JP5090008B2 - Semiconductor device and shift register circuit - Google Patents

Abstract

A dual-gate transistor formed of two transistors connected in series between a first power terminal and a first node is used as a charging circuit for charging a gate node (first node) of a transistor intended to pull up an output terminal of a unit shift register. The dual-gate transistor is configured such that the connection node (second node) between the two transistors constituting the dual-gate transistor is pulled down to the L level by the capacitive coupling between the gate and second node in accordance with the change of the gate from the H level to the L level.

Description

  The present invention relates to a technique for preventing malfunction due to deterioration of electric characteristics of a transistor, and in particular, a semiconductor capable of suppressing a negative shift of a threshold voltage of an amorphous silicon thin film transistor or an organic transistor. It relates to the device.

  In an image display device such as a liquid crystal display device, as a gate line driver circuit (scan line driver circuit) for scanning a display panel, a shift register that performs a shift operation that makes a round in one frame period of a display signal can be used. . The shift register is preferably composed only of field effect transistors of the same conductivity type in order to reduce the number of steps in the manufacturing process of the display device.

  A display device in which a shift register of a gate line driving circuit is configured by an amorphous silicon thin film transistor (hereinafter referred to as “a-Si transistor”) is easy to increase in area and has high productivity. Widely used in screen display devices.

  On the other hand, it is known that the a-Si transistor has a characteristic that the threshold voltage shifts when the gate electrode is continuously (DC) biased. In addition, as shown in Non-Patent Document 1 below, an a-Si transistor generally has progressive deterioration.

  Further, since the shift of the threshold voltage of the transistor (Vth shift) causes a malfunction of the circuit, various shift registers in which countermeasures are taken have been proposed (for example, Patent Document 1). This Vth shift problem has been found to occur not only in a-Si transistors but also in organic transistors.

JP 2006-107692 A R. B. Wehrspohn etc, “Relative importance of the Si-Si bond and Si-H bond for the stability of amorphous silicon thin film transistors” Journal of applied physics vol.87 pp. 144-154

  The threshold voltage of the a-Si transistor shifts in the negative (minus) direction with the passage of time when the gate is kept at a low potential state with respect to both the drain and the source. When the threshold voltage of the a-Si transistor shifts in the negative direction, even if an attempt is made to turn the a-Si transistor off (non-conducting state) by reducing its gate-source potential, it is not completely turned off. . That is, the a-Si transistor cannot completely cut off the current, thereby causing a malfunction of the circuit.

  The shift register includes an output pull-up transistor (transistor Q1 in FIG. 1) that supplies a clock signal to an output terminal to pull up the potential of the output terminal, and a gate node (node N1 in the figure) of the output pull-up transistor. And a charging transistor (transistor Q3 in the figure). Although details will be described later, in the normal operation of the shift register, the charging transistor is in the above-described potential state (state in which the gate potential of the transistor is lower than both the drain potential and the source potential) for a certain period. There has been a problem that the threshold voltage is shifted in the negative direction to cause a malfunction.

  The present invention has been made to solve the above problems, and provides a transistor capable of suppressing a negative shift of a threshold voltage (minus shift), and a semiconductor including a shift register. The purpose is to prevent malfunction of the device.

A semiconductor device according to the present invention is a semiconductor device including a plurality of transistors that are connected in series between a predetermined first node and a second node, and whose control electrodes are connected to each other. Each of the connection nodes is a third node, and each of the first to third nodes and the control electrode is in an H (High) level that is higher than a threshold voltage of the plurality of transistors. When the potential of the control electrode changes to an L (Low) level lower than the threshold voltage while the second node remains at the H level, the level of the third node is also lowered to the L level accordingly. In each of the third nodes, a capacitance component between the control electrode and the third node is C1, and a parasitic capacitance associated with the third node not included in the C1 is C1. 2. When the difference between the H level and the L level at the control electrode is Vd and the threshold voltage of the first transistor is Vth, the relationship of C1 ≧ C2 × (Vd−Vth) / Vth is satisfied .

According to the first aspect of the shift register circuit of the present invention, the shift register circuit receives an input terminal, an output terminal, a first clock terminal, a reset terminal, and a first clock signal input to the first clock terminal. A first transistor supplied to the output terminal, a second transistor that discharges the output terminal, and a first node connected to the control electrode of the first transistor are charged according to an input signal input to the input terminal. A charging circuit; and a discharging circuit that discharges the first node in response to a reset signal input to the reset terminal, wherein the charging circuit is connected in series between the first node and a power supply terminal, look including a plurality of third transistors which control electrodes are both connected to the input terminal, the high H (high) level than the input signal is the threshold voltage of the third transistor Then, when the plurality of third transistors are turned on, the first node is charged, and then when the input signal changes to L (Low) level lower than the threshold voltage, each of the plurality of third transistors is changed. The connection node is configured to be pulled down to an L level, and in each of the connection nodes between the plurality of third transistors, a capacitance component between the input terminal and the connection node is not included in C1 and C1 Assuming that the parasitic capacitance associated with the connection node is C2, the difference between the H level and the L level of the input signal is Vd, and the threshold voltage of the third transistor is Vth, C1 ≧ C2 × (Vd−Vth) / The relationship of Vth is satisfied .

  According to the semiconductor device of the first aspect of the present invention, in each of the transistors connected in series, a potential state in which both the source and the drain are at the H level and the control electrode is at the L level is prevented. Is prevented from shifting in the negative direction. Thereby, when the control electrode is set to the L level, each transistor can be surely turned off (blocked state), so that malfunction of the semiconductor device is prevented.

  According to the first aspect of the shift register circuit of the present invention, in each of the transistors constituting the charging circuit for the control electrode of the first transistor, the potential at which both the source and the drain are at the H level and the control electrode is at the L level. Since the state is prevented, the threshold voltage of the transistor is prevented from shifting in the negative direction. Accordingly, when the control electrode of the charging circuit is set to the L level, the charging circuit can be reliably turned off (shut off state), so that the first transistor can be prevented from being turned on unnecessarily, and the shift register circuit Malfunction is prevented.

  Embodiments of the present invention will be described below with reference to the drawings. In addition, in order to avoid duplication and redundant description, elements having the same or corresponding functions are denoted by the same reference symbols in the respective drawings.

<Embodiment 1>
First, in order to facilitate the description of the present invention, a conventional shift register will be described. Usually, the shift register has a multi-stage structure in which a plurality of shift register circuits are connected in cascade (cascade connection). In this specification, each of the shift register circuits in each stage constituting the multi-stage shift register is referred to as a “unit shift register”.

  As described above, the shift register can be used as a gate line driver circuit of a display device. A specific configuration example of a display device using a shift register as a gate line driver circuit is disclosed in patent documents (for example, FIGS. 1 and 2 of JP-A-2006-277860) by the present inventors. In the following description, the low-potential power supply potential (VSS), which is the reference voltage of the circuit, is described as 0 V. However, in an actual display device, the reference potential is set based on the voltage of data written to the pixel. The low potential side power supply potential (VSS) is set to -12V, the high potential side power supply potential (VDD) is set to 17V, and the like.

FIG. 1 is a circuit diagram showing a configuration of a conventional unit shift register. FIG. 2 is a diagram showing the configuration of a multistage shift register. The shift register of FIG. 2 includes n unit shift registers SR ₁ , SR ₂ , SR ₃ ,..., SR _n connected in cascade, and dummy units provided further downstream of the last unit shift register SR _n . Unit shift register SRD (hereinafter, unit shift registers SR ₁ , SR ₂ ... SR _n , SRD are collectively referred to as “unit shift register SR”). In the conventional shift register, each unit shift register SR is the circuit of FIG.

  Further, the clock generator 31 shown in FIG. 2 supplies two-phase clock signals CLKA and CLKB having opposite phases (the active periods do not overlap) to the plurality of unit shift registers SR. In the gate line driving circuit, the clock signals CLKA and CLKB are controlled so as to be sequentially activated at a timing synchronized with the scanning period of the display device.

  As shown in FIGS. 1 and 2, each unit shift register SR has an input terminal IN1, an output terminal OUT, a clock terminal CK1, and a reset terminal RST. Each unit shift register SR is supplied with the low-potential-side power supply potential VSS (= 0 V) via the first power-supply terminal S1, and is supplied with the high-potential-side power supply potential VDD via the second power-supply terminal S2. (Not shown in FIG. 2).

  As shown in FIG. 1, the output stage of the unit shift register SR includes a transistor Q1 connected between the output terminal OUT and the clock terminal CK1, and a transistor Q2 connected between the output terminal OUT and the first power supply terminal S1. It is configured. That is, the transistor Q1 is a transistor (first transistor) that supplies the clock signal CLKA input to the clock terminal CK1 to the output terminal OUT, and the transistor Q2 is a transistor that discharges the output terminal OUT (second transistor). . Hereinafter, a node to which the gate (control electrode) of the transistor Q1 is connected is defined as “node N1”, and a node to which the gate of the transistor Q2 is connected is defined as “node N2”.

  A capacitive element C1 is provided between the gate and source of the transistor Q1 (that is, between the node N1 and the output terminal OUT). The capacitive element C1 is an element (bootstrap capacitance) that capacitively couples the output terminal OUT and the node N1 and boosts the node N1 in response to a rise in the level of the output terminal OUT. However, the capacitor C1 can be replaced if the gate-channel capacitance of the transistor Q1 is sufficiently large, and may be omitted in such a case.

  A transistor Q3 whose gate is connected to the input terminal IN1 is connected between the node N1 and the second power supply terminal S2. A transistor Q4 whose gate is connected to the reset terminal RST is connected between the node N1 and the first power supply terminal S1. That is, the transistor Q3 constitutes a charging circuit that charges the node N1 according to the signal input to the input terminal IN1, and the transistor Q4 discharges the node N1 according to the signal input to the reset terminal RST. Is configured. In this conventional example, the gate (node N2) of the transistor Q2 is also connected to the reset terminal RST.

  As shown in FIG. 2, the output terminal OUT of the preceding unit shift register SR is connected to the input terminal IN1 of each unit shift register SR. However, a predetermined start pulse ST is input to the input terminal IN1 of the unit shift register SR1, which is the first stage. In addition, one of the clock signals CLKA and CLKB is input to the clock terminal CK1 of each unit shift register SR so that clock signals having different phases are input to the unit shift registers SR adjacent to each other.

The output terminal OUT of the next unit shift register SR is connected to the reset terminal RST of each unit shift register SR. However, the reset terminal RST of the next stage provided dummy unit shift register SRD of the unit shift register SR _n of the last stage, a predetermined end pulse EN is input. In the gate line driving circuit, the start pulse ST and the end pulse EN are input at timings corresponding to the beginning and end of each frame period of the image signal, respectively.

Next, the operation of each conventional unit shift register SR shown in FIG. 1 will be described. Basically, all the unit shift registers SR of each stage operate in the same manner, so here, the operation of the k-th unit shift register SR _k among the multi-stage shift registers will be representatively described. The clock terminal CK1 of the unit shift register SR _k shall clock signal CLKA is input (for example, the unit shift register SR _1, SR ₃ in FIG. 2 corresponds to this).

  Here, it is assumed that the H level potential of the clock signals CLKA and CLKB is VDD (high potential side power source potential) and the L level potential is VSS (low potential side power source potential). The threshold voltage of each transistor Qx constituting the unit shift register SR is represented as Vth (Qx).

FIG. 3 is a timing chart showing the operation of the conventional unit shift register SR _k (FIG. 1). First, as an initial state of the unit shift register SR _k , it is assumed that the node N1 is at the L level (hereinafter, the state where the node N1 is at the L level is referred to as a “reset state”). The input terminal IN1 (previous output signal G _k-1 ), the reset terminal RST (next output signal G _{k + 1} ), and the clock terminal CK1 (clock signal CLKA) are all at L level. At this time, since the transistors Q1 and Q2 are both off, the output terminal OUT is in a high impedance state (floating state), but in the initial state, the output terminal OUT (output signal G _k ) is also at L level.

From that state, at time t ₁ , the clock signal CLKA changes to L level and the clock signal CLKB changes to H level, and the output signal G _{k-1 of} the previous stage (start pulse ST in the case of the first stage) changes to H level. comes to the transistor Q3 of the unit shift register SR _k is turned on, the node N1 is charged to H level (hereinafter, the node N1 is called a state of an H level and "set state"). At this time, the potential level of the node N1 (hereinafter simply referred to as “level”) rises to VDD−Vth (Q3). In response, transistor Q1 is turned on.

At time t ₂ , the clock signal CLKB changes to the L level and the clock signal CLKA changes to the H level, and the output signal G _{k−1 of the} previous stage becomes the L level. Then, the transistor Q3 is turned off, and the node N1 is in the floating state with the H level. Since the transistor Q1 is on, the level of the output terminal OUT rises following the clock signal CLKA.

  When the levels of the clock terminal CK1 and the output terminal OUT rise, the level of the node N1 is boosted as shown in FIG. 3 by the coupling through the capacitance element C1 and the gate-channel capacitance of the transistor Q1. Since the boost amount at this time substantially corresponds to the amplitude (VDD) of the clock signal CLKA, the node N1 is boosted to approximately 2 × VDD−Vth (Q3).

As a result, the voltage between the gate (node N1) and source (output terminal OUT) of the transistor Q1 is kept large even while the output signal _Gk is at the H level. That the on-resistance of the transistor Q1 is kept low, the output signal G _k becomes H level rises quickly following the clock signal CLKA. Since operating at this time, the transistor Q1 is a linear region (non-saturation region), the level of the output signal G _k rises to the same VDD and the amplitude of the clock signal CLKA.

Further, when the clock signal CLKB changes to the H level and the clock signal CLKA changes to the L level at time t ₃ , the on-resistance of the transistor Q1 is kept low, and the output signal G _k follows the clock signal CLKA and falls quickly. To return to the L level.

At time t ₃ , the output signal G _{k + 1} at the next stage becomes H level, so that the transistors Q2 and Q4 of the unit shift register SR _k are turned on. As a result, the output terminal OUT is sufficiently discharged through the transistor Q2, and is surely set to the L level (VSS). Node N1 is discharged to low level by transistor Q4. That is, the unit shift register SR _k returns to the reset state.

After the next stage output signal G _{k + 1} returns to the L level at time t ₄ , the unit shift register SR _k is maintained in the reset state until the next stage output signal G _k-1 is next input. The output signal G _k is kept at the L level.

To summarize the above operations, the unit shift register SR _k is in a reset state during the period when the signal (start pulse SP or the previous stage output signal G _k-1 ) is not input to the input terminal IN1, and the transistor Q1 is kept off. Therefore, the output signal G _k is maintained at the L level (VSS). When the signal is input to the input terminal IN1, the unit shift register SR _k is switched to the set state. Since the transistor Q1 is turned on in the set state, while the signal of the clock terminal CK1 (clock signal CLKA) becomes the H level, the output signal G _k becomes the H level. Thereafter, when a signal (next-stage output signal G _{k + 1} or end pulse EN) is input to the reset terminal RST, the original reset state is restored.

According to the multi-stage shift register composed of the plurality of unit shift registers SR operating in this way, when the start pulse ST is input to the first-stage unit shift register SR ₁ , the output signal G Are transmitted in order with unit shift registers SR ₁ , SR ₂ , SR ₃ ... While being shifted at a timing synchronized with the clock signals CLKA, CLKB. In the gate line driving circuit, the output signal G output in this order is used as a horizontal (or vertical) scanning signal of the display panel.

  Hereinafter, a period during which a specific unit shift register SR outputs the output signal G is referred to as a “selection period” of the unit shift register SR.

The dummy unit shift register SRD is provided to reset the unit shift register SR _n by the output signal GD immediately after the last unit shift register SR _n outputs the output signal G _n . . For example, in the case of a gate line driving circuit, unless the last unit shift register SR _n is reset immediately after the output signal G _n is output, the corresponding gate line (scanning line) is activated unnecessarily. A malfunction will occur.

  Note that the dummy unit shift register SRD is reset by the end pulse EN input at a timing after the output signal GD is output. When the signal shift operation is repeatedly performed as in the gate line driving circuit, the start pulse ST of the next frame period may be used instead of the end pulse EN.

  In the case of driving using a two-phase clock as shown in FIG. 2, each of the unit shift registers SR is reset by the output signal G of its next stage, so that the unit shift register SR of the next stage is at least The normal operation as shown in FIGS. 3 and 4 cannot be performed until after the operation once. Therefore, prior to the normal operation, it is necessary to perform a dummy operation for transmitting a dummy signal from the first stage to the last stage. Alternatively, a reset transistor is separately provided between the reset terminal RST (node N2) and the second power supply terminal S2 (high potential side power supply) of each unit shift register SR, and the node N2 is forcibly set before the normal operation. You may perform the reset operation which makes it H level. In this case, however, a reset signal line is required separately.

  Here, the problem of the negative shift of the threshold voltage of the a-Si transistor in the above-described conventional unit shift register SR will be described in detail.

As can be seen from the timing chart of FIG. 3, the node N1 of the unit shift register SR _k is charged to the H level (VDD−Vth (Q3)) when the output signal G _k−1 of the previous stage becomes the H level (time t _1). Thereafter, even if the output signal G _k-1 of the previous stage returns to the L level (time t ₂ ), it is maintained at the H level in the floating state. Moreover, while the output signal G is at the H level (selection period: time t ₂ to time t ₃ ), the node N1 is boosted to the level of 2 × VDD−Vth (Q3).

  That is, in each unit shift register SR, the drain (second power supply terminal S2) of the transistor Q3 is VDD, the source (node N1) is 2 × VDD−Vth (Q3), and the gate (input terminal IN1) is VSS during the selection period. Level and the gate is negatively biased with respect to both the source and drain. For example, assuming that VSS = 0V and VDD = 30V, as shown in FIG. 5, the gate-drain voltage Vgd of the transistor Q3 is about -30V and the gate-source voltage Vgs is about -57V.

  FIG. 6 is a diagram showing experimental results showing the relationship between the potential state of the a-Si transistor and the threshold voltage shift. As indicated by a broken line in the figure, when the gate of the a-Si transistor is in a low potential state with respect to both the drain and the source, the threshold voltage shifts in the negative (minus) direction with time. Therefore, in the conventional unit shift register SR, a negative shift of the threshold voltage occurs in the transistor Q3 during the selection period.

  In the unit shift register SR, when the threshold voltage of the transistor Q3 is shifted in the negative direction, a current flows through the transistor Q3 even when the input terminal IN1 is at L level, and charge is supplied to the node N1 during the non-selection period. , That level rises. In this case, there is a problem in that a malfunction occurs in which the transistor Q1 of the unit shift register SR is turned on and the output signal G as an error signal is output from the output terminal OUT in spite of the non-selection period.

  On the other hand, even if the gate of the a-Si transistor is at a lower potential than the drain, the shift of the threshold voltage is reduced if the gate and the source are substantially at the same potential. For example, as shown by the solid line in FIG. 6, even if the gate is at a low potential with respect to the drain, if the gate-source voltage is set to 0 V, the threshold voltage hardly shifts.

  A shift register circuit according to the present invention that can solve the above problem will be described below. FIG. 7 is a circuit diagram of the unit shift register SR according to the first embodiment of the present invention. The unit shift register SR is obtained by replacing the transistor Q3 with a dual gate transistor Q3D in the circuit of FIG. Since the other configuration is the same as that of FIG. 1, detailed description thereof is omitted here.

  Note that the “dual gate transistor” in this specification refers to two transistors connected in series and having their gates connected to each other. That is, the dual gate transistor Q3D includes transistors Q3a and Q3b connected in series between the node N1 and the second power supply terminal S2, and the gates of the transistors Q3a and Q3b are both connected to the input terminal IN1. . Here, a connection node between the transistor Q3a and the transistor Q3b is defined as “node N3”.

FIG. 8 is a timing chart showing the operation of the unit shift register SR of FIG. Here again, the k-th unit shift register SR _k will be described representatively. Further, as an initial state of the unit shift register SR _k , the node N1 is assumed to be in a reset state of L level, the clock terminal CK1 (clock signal CLKA), the reset terminal RST (next stage output signal G _{k + 1} ), and the input terminal It is assumed that IN1 (previous output signal G _k-1 ) and output terminal OUT (output signal G _k ) are at L level.

From that state, at time t ₁ , the clock signal CLKA changes to L level and the clock signal CLKB changes to H level, and the output signal G _{k-1 of} the previous stage (start pulse ST in the case of the first stage) changes to H level. becomes, the transistors Q3a constituting the dual-gate transistor Q3D of the unit shift register SR _k, Q3b is turned on both the node N1 is charged in the H level. That is, the unit shift register SR _k is set. At this time, the levels of the nodes N1 and N3 both rise to VDD-Vth (Q3a). In response, transistor Q1 is turned on.

The clock signal CLKB is at the L level at time t _2, the the clock signal CLKA is changed to H level, the previous stage of the output signal G _k-1 becomes L level, the transistors Q3a, the node N1 Q3b is turned off floating state At H level. Therefore the transistor Q1 is kept ON state, the H-level output signal G _k is following the clock signal CLKA. At this time, the level of the node N1 is boosted to about 2 × VDD−Vth (Q3a).

In the present embodiment, each of transistors Q3a and Q3b constituting dual gate transistor Q3D has a large gate-source overlap capacity (the gate-source overlap capacity of transistors Q3a and Q3b is increased). The method will be described in Embodiment 9). Parasitic capacitance is large between the first input terminal IN1 and a node N3 Therefore, when the clock signal CLKB is at the L level at time t _2, the by coupling through the parasitic capacitance, the level of the node N3 is at the L level (i.e. To a level lower than the threshold voltage of the transistors Q3a and Q3b). If the capacitance component between the input terminal IN1 and a node N3 is sufficiently large, the level of the node N3 as the time t ₂ in FIG. 8 is pulled down to approximately VSS.

  At this time, from the potential relationship, in the transistor Q3a, the second power supply terminal S2 side is the drain and the node N3 side is the source, and in the transistor Q3b, the node N1 side is the drain and the node N3 side is the source.

Subsequently, when the clock signal CLKB changes to H level and the clock signal CLKA changes to L level at time t ₃ , the output signal G _k returns to L level. Since with the next-stage output signal G _{k + 1} it becomes H level, the transistors Q2, Q4 of the unit shift register SR _k is turned on, the unit shift register SR _k returns to the reset state.

After the next stage output signal G _{k + 1} becomes L level at time t ₄ , the unit shift register SR _k is maintained in the reset state until the next stage output signal G _k-1 is next input. The output signal G _k is kept at the L level.

  As described above, the signal shift operation of the unit shift register SR according to the present embodiment is almost the same as that of the conventional one (FIG. 1), and the multistage shift register configured therewith is as described with reference to FIG. Is possible.

In the present embodiment, as described above, at time t ₂ the preceding stage output signal G _k-1 nodes N3 in response to the falls is pulled to L level. Therefore, between time t ₂ and time t ₃ (selection period), the transistor Q3a has a potential state in which the gate (input terminal IN1) and the source (node N3) are VSS, and the drain (second power supply terminal S2) is VDD. The transistor Q3a has a potential state in which the gate (input terminal IN1) and the source (node N3) are VSS, and the drain (node N1) is 2 × VDD−Vth (Q3a).

  That is, in the unit shift register SR according to the present embodiment, the voltage between the gates and the sources of the transistors Q3a and Q3b is substantially 0 V during the selection period. For example, assuming that VSS = 0V and VDD = 30V, as shown in FIG. 9, the gate-drain voltage Vgd of the transistor Q3a is -30V, the gate-source voltage Vgs is about 0V, and the gate-drain voltage of the transistor Q3b The voltage Vgd is about -57V, and the gate-source voltage Vgs is about 0V.

  As shown by the solid line in FIG. 6, the a-Si transistor can be in a state where the gate is at the same potential as the source (the gate-source voltage is about 0 V) even if the gate is at a lower potential than the drain. For example, the threshold voltage shift hardly occurs. Therefore, in the unit shift register SR of the present embodiment, a negative shift of the threshold voltage of the transistors Q3a and Q3b, that is, the dual gate transistor Q3D does not occur. Therefore, it is possible to prevent a current from flowing through the dual gate transistor Q3D during the non-selection period and an increase in the level of the node N1, thereby preventing a malfunction.

  Here, a condition for the voltage between the gate and the source of the transistors Q3a and Q3b to be 0 V or less during the selection period of the unit shift register SR (that is, a condition that the node N3 is lowered to VSS or less) will be described. Again, VSS = 0V. That is, the amplitude (the difference (Vd) between the H level and the L level) of the output signal G of each unit shift register SR is VDD.

In this case the capacitance component between the input terminal IN1 and a node N3 to (C1) and Cgs, the parasitic capacitance associated with the node N3 which is not included in the Cgs (C2) and Cstr, front at time t ₂ in FIG. 8 When the output signal G _k−1 of the output signal changes from the H level (VDD) to the L level (VSS = 0 V), the amount of change in the level of the node N3 of the unit shift register SR _k is VDD × Cgs / (Cgs + Cstr) It is represented by Since just before the time t ₂ the level of the node N3 has a VDD-Vth (Q3a), to the node N3 is pulled down below the VSS, should satisfy the following formula (1).

  In the unit shift register SR of the present embodiment, the capacitance component Cgs is a parasitic capacitance between the input terminal IN1 and the node N3, most of which is the gate-source overlap capacitance of the transistors Q3a and Q3b. Therefore, as shown in FIG. 10, if the gate-source overlap capacitances of the transistors Q3a and Q3b are Cgso (Q3a) and Cgso (Q3b), Cgs≈Cgso (Q3a) + Cgso (Q3b) can be obtained. 1) can be transformed into the following equation (2).

  Further, assuming that the gate-source overlap capacitances of the transistors Q3a and Q3b are equal to each other, and assuming that the value is Cgso, Cgs≈2 × Cgso, the above equation (1) is expressed by the following equation (3): Can be transformed.

  In the present embodiment, a configuration is shown in which two transistors whose gates are connected to the first input terminal IN1 are connected in series between the node N1 and the second power supply terminal S2, but three or more transistors are connected. May be connected in series. In that case, if the condition of the expression (1) is satisfied at each connection node between the transistors, each connection node becomes equal to or lower than VSS during the non-selection period, thereby preventing a negative shift of the threshold voltage of each transistor. be able to.

  In this embodiment mode, an example in which the dual gate transistor according to the present invention is applied to a shift register circuit is shown. However, the dual gate transistor is widely applied to transistors that operate so that the gate is negatively biased with respect to both the source and the drain. Can do. The present invention can be applied not only to an a-Si transistor but also to various transistors having a problem of negative shift of the threshold voltage, such as an organic transistor.

<Embodiment 2>
In the following embodiments, specific examples of shift register circuits to which the dual gate transistor according to the present invention can be applied will be shown.

  FIG. 11 is a circuit diagram of the unit shift register SR according to the second embodiment. The unit shift register SR is provided with an inverter having the node N1 (the gate of the transistor Q1) as an input terminal and the node N2 (the gate of the transistor Q2) as an output terminal with respect to the circuit of FIG. That is, unlike FIG. 7, the gate (node N2) of the transistor Q2 is not connected to the reset terminal RST.

  The inverter includes a transistor Q5 that is diode-connected between the node N2 and the second power supply terminal S2, and a transistor Q6 that is connected between the node N2 and the first power supply terminal S1 and has a gate connected to the node N1. Yes. The on-resistance of the transistor Q6 is set to be sufficiently smaller than that of the transistor Q5.

  When the node N1 is at L level, the transistor Q6 is turned off, so that the node N2 is at H level (VDD-Vth (Q5)). Conversely, when the node N1 is at the H level, the transistors Q5 and Q6 are both turned on, but the node N2 is at the L level of the potential (≈0 V) determined by the ratio of the on resistances of the transistors Q5 and Q6. That is, the inverter is a so-called “ratio inverter”.

In the unit shift register SR _k of FIG. 7, the node N2 becomes H level only when the output signal G _{k + 1} of the next stage becomes H level (that is, the selection period of the next stage). The output terminal OUT is set to a low impedance L level by turning on only for a period. During the other non-selection period, the transistor Q2 is off, and the output terminal OUT is at a high impedance (floating state) L level. Therefore, the output signal G _k is easily affected by noise and leakage current, and the operation is likely to be unstable.

On the other hand, in the unit shift register SR _k of FIG. 11, while the node N1 is at the L level, the inverter composed of the transistors Q5 and Q6 maintains the node N2 at the H level, so that the transistor Q2 remains on during the non-selection period. Kept in a state. That is, since the output terminal OUT (output signal G) in the non-selection period is maintained at the L level with low impedance, the operation is stabilized.

  Of course, also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. Therefore, it is possible to prevent the level of the node N1 from increasing during the non-selection period, and it is possible to prevent malfunction.

<Embodiment 3>
FIG. 12 is a circuit diagram of the unit shift register SR according to the third embodiment. The unit shift register SR is provided with a transistor Q7 connected between the node N1 and the first power supply terminal S1 and having a gate connected to the node N2 in the circuit of FIG. That is, the transistor Q7 has a gate electrode connected to the node N2, and discharges the node N1.

In the unit shift register SR _k of FIG. 11, when the output signal G _{k + 1} of the next stage becomes H level (next stage selection period), the transistor Q4 is turned on to discharge the node N1. In a non-selection period other than the above, the node N1 is at the L level with high impedance (floating state). Therefore, when charge is supplied to the node N1 due to noise or leakage current during the non-selection period, the level of the node N1 increases. Then, a malfunction occurs in which the transistor Q1 is turned on and the output signal G is output as an error signal.

  On the other hand, in the unit shift register SR of FIG. 12, when the node N1 becomes L level, the inverter consisting of the transistors Q5 and Q6 turns the node N2 to H level and the transistor Q7 is turned on accordingly, so that the node N1 is not selected. It becomes L level with low impedance throughout the period. Accordingly, an increase in the level of the node N1 during the non-selection period is suppressed, and the occurrence of the malfunction is prevented.

  Transistor Q7 has a sufficiently higher on-resistance than dual gate transistor Q3D so that dual gate transistor Q3D can raise the level of node N1.

  Also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. In the present embodiment, even if a current flows through the dual gate transistor Q3D and charges are supplied to the node N1 during the non-selection period, the charges are discharged to the first power supply terminal S1 by the transistor Q7. Malfunctions are less likely to occur. However, the current causes an increase in power consumption of the unit shift register SR and a decrease in the high potential side power supply potential VDD. Therefore, preventing a negative shift of the threshold voltage of the dual gate transistor Q3D is also very effective in this embodiment.

<Embodiment 4>
As described in the second and third embodiments, in the unit shift register SR of FIGS. 11 and 12, the gate (node N2) of the transistor Q2 is continuously at the H level during the non-selection period, so that the output The terminal OUT can be set to a low impedance L level. However, when the gate of the a-Si transistor is continuously positively biased with respect to the source, the threshold voltage shifts in the positive direction. When the threshold voltage of the transistor Q2 shifts in the positive direction, the on-resistance of the transistor Q2 becomes high, causing a problem that the output terminal OUT cannot be made sufficiently low impedance.

  In the unit shift register SR of FIG. 12, since the gate of the transistor Q7 is continuously at the H level during the non-selection period, the threshold voltage of the transistor Q7 is also shifted in the positive direction, and the node N1 is There also arises a problem that the output terminal OUT cannot be made sufficiently low impedance.

  FIG. 13 is a circuit diagram of the unit shift register SR according to the fourth embodiment, in which measures against the problem are taken. In the unit shift register SR of FIG. 13, two transistors (corresponding to the transistor Q2 of FIGS. 11 and 12) that discharge the output terminal OUT are provided in parallel (transistors Q2A and Q2B). Here, nodes to which the gates of the transistors Q2A and Q2B are connected are defined as “node N2A” and “node N2B”, respectively.

  In the unit shift register SR, the one corresponding to the transistor Q7 in FIG. 12 is provided in each of the nodes N2A and N2B (transistors Q7A and Q7B). That is, the transistor Q7A has a gate electrode connected to the node N2A and discharges the node N1, and the transistor Q7B has a gate electrode connected to the node N2B and discharges the node N1.

  The unit shift register SR according to the present embodiment has a first control terminal TA to which a predetermined control signal VFRA is input and a second control terminal TB to which a control signal VFRB is input. The control signals VFRA and VFRB are complementary signals and are generated by a control device (not shown) outside the shift register. The control signals VFRA and VFRB are switched in level at a constant cycle. In the gate line driving circuit, it is desirable to control the level to switch (alternate) during the blanking period between frames of the display image. For example, the level is controlled to switch for each frame of the display image.

  The transistor Q8A is connected between the first control terminal TA and the node N2A, and the transistor Q8B is connected between the second control terminal TB and the node N2B. Transistor Q8A has its gate connected to node N2B, and transistor Q8B has its gate connected to node N2A. That is, the transistor Q8A and the transistor Q8B have one main electrode (drain in this case) connected to each other's control electrode (gate) by dragging to constitute a so-called flip-flop circuit.

  The unit shift register SR further includes a transistor Q9A connected between the output terminal of the inverter composed of the transistors Q5 and Q6 and the node N2A, and a transistor Q9B connected between the output terminal of the inverter and the node N2B. ing. The gate of the transistor Q9A is connected to the first control terminal TA, and the gate of the transistor Q9B is connected to the second control terminal TB.

  Since the transistor Q9A is on and the transistor Q9B is off while the control signal VFRA is at the H level and the control signal VFRB is at the L level, the output terminal of the inverter composed of the transistors Q5 and Q6 is connected to the node N2A. At this time, the transistor Q8B is turned on, and the node N2A becomes L level. That is, during that period, the transistor Q2A is driven, and the transistor Q2B is in a dormant state.

  Conversely, during the period when the control signal VFRA is at L level and the control signal VFRB is at H level, the transistor Q9A is turned off and the transistor Q9B is turned on, so that the output terminal of the inverter composed of the transistors Q5 and Q6 is connected to the node N2B. . At this time, the transistor Q8A is turned on, and the node N2B becomes L level. That is, during that period, the transistor Q2B is driven, and the transistor Q2A is in a resting state.

  Thus, the transistors Q9A and Q9B function as a switching circuit that alternately connects the output terminals of the inverters composed of the transistors Q5 and Q6 to the nodes N2A and N2B based on the control signals VFRA and VFRB.

  In the present embodiment, every time the control signals VFRA and VFRB are inverted, the pair of transistors Q2A and Q5A and the pair of transistors Q2B and Q5B are alternately in a resting state, so that their gates are continuously biased. Can be prevented. Therefore, malfunction due to a positive shift of the threshold value of the a-Si transistor can be prevented, and the operation reliability is improved.

  Also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. Accordingly, it is possible to prevent the malfunction of the unit shift register SR, the increase in power consumption, and the decrease in the power supply voltage.

<Embodiment 5>
In each of the above embodiments, the dual gate transistor Q3D operates in the source follower mode when charging the node N1 of the unit shift register SR. That is, as the charging of the node N1 proceeds, the voltage between the gate (input terminal IN1) and the source (node N1) of the transistor Q3b decreases and the driving capability (ability to flow current) decreases, so that the node N1 is sufficiently high. It takes some time to charge to the level. This hinders the speeding up of the shift register operation.

  FIG. 14 is a circuit diagram of the unit shift register SR according to the fifth embodiment, in which measures against the problem are taken. In the case of cascade connection, the unit shift register SR is driven using three-phase clocks CLKA, CLKB, and CLKC having different phases as shown in FIG.

  Each unit shift register SR has two input terminals, a first input terminal IN1 and a second input terminal IN2, and the first input terminal IN1 has an output terminal at the previous stage (two stages before). The second input terminal IN2 is connected to the output terminal OUT of the previous stage (one stage before). Further, start pulses ST1 and ST2 are input to the first input terminal IN1 and the second input terminal IN2, respectively, of the first stage unit shift register SR1. The start pulses ST1 and ST2 are activated (become H level) at different timings, and the start pulse ST2 is activated after the start pulse ST1.

  In the unit shift register SR of the present embodiment, as means for charging the node N1, a dual gate transistor Q3D, a transistor Q10 charging its gate node (defined as “node N4”), and a capacitor for boosting the node N4 It includes element C2 and transistor Q4 that discharges node N4. As shown in FIG. 14, the transistor Q10 is connected between the node N4 and the second power supply terminal S2, and the gate is connected to the first input terminal IN1. The capacitive element C2 is connected between the node N4 and the second input terminal IN2. The transistor Q4 is connected between the node N4 and the first power supply terminal S1, and has a gate connected to the reset terminal RST.

  The unit shift register SR includes inverters (transistors Q5 and Q6) having the node N4 as an input terminal, and both the output terminal OUT and the gates (nodes N2) of the transistors Q2 and Q7 that discharge the node N1, respectively. Connected to the output terminal of the inverter. A transistor Q11 is connected in parallel with the transistor Q4 between the node N4 and the first power supply terminal S1, and its gate is connected to the node N2.

  The basic operation theory of the unit shift register SR of FIG. 14 is almost the same as that described in the first embodiment, but the gate of the dual gate transistor Q3D that charges the node N1 is different between the previous stage and the previous stage. It is characterized in that it is charged and boosted using two signals called output signals.

That is, in the unit shift register SR _k , the gate (node N4) of the dual gate transistor Q3D is first set to the level of VDD−Vth (Q10) by the transistor Q10 when the output signal G _{k−2 of the} preceding stage becomes H level. Is precharged. Next, when the output signal G _k-1 at the previous stage becomes the H level, the node N4 is boosted to about 2 × VDD−Vth (10) by the capacitive element C2. That is, the gate potential of the dual gate transistor Q3D is about VDD higher than that in the case of the circuit of FIG. 1, and the dual gate transistor Q3D can charge the node N1 not by the source follower mode but by the operation in the non-saturated region. . Therefore, the node N1 is charged at high speed and becomes H level (VDD), so that the above problem is solved.

In the present embodiment, since the gate (node N4) of the dual gate transistor Q3D is in a floating state during the selection period, the transistor Q4 controlled by the output signal G _{k + 1 at the} next stage does not discharge the node N4. (This is different from the transistor Q4 in FIG. 7). When the transistor Q4 brings the node N4 to L level, the inverter consisting of the transistors Q5 and Q6 brings the node N2 to H level, and accordingly the transistor Q7 is turned on to discharge the node N1. That is, in this embodiment, the transistor Q7 plays a role of discharging the node N1 in accordance with a signal input to the reset terminal RST (that is, the role of the transistor Q4 in FIG. 7).

  The transistor Q11 operates to maintain the node N4 at the low impedance L level while the node N2 is at the H level (non-selection period), thereby causing the unit shift register SR to malfunction. It is prevented.

  Also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. Accordingly, it is possible to prevent the malfunction of the unit shift register SR, the increase in power consumption, and the decrease in the power supply voltage.

Further, in the unit shift register SR of FIG. 14, when the output signal G _k-2 at the preceding stage becomes L level after the transistor Q10 charges the node N4, the potential of the gate of the transistor Q10 is lower than that of the source and drain. become. Therefore, the transistor Q10 may have a problem of negative shift of the threshold voltage similar to the transistor Q3 in FIG.

  In order to avoid this problem, the dual gate transistor of the present invention may be applied to the transistor Q10 of FIG. That is, the transistor Q10 may be replaced with a dual gate transistor Q10D comprising transistors Q10a and Q10b as shown in FIG. Similarly to the dual gate transistor Q3D, when the gate electrode changes from H level to L level, the dual gate transistor Q10D is defined as a connection node between the transistors Q10a and Q10b (defined as “node N10”). ) Is lowered to the L level, and the parasitic capacitance between the gate and the node N6 is increased.

  According to this configuration, in the dual gate transistor Q10D, that is, the transistors Q10a and Q10b, the effect of suppressing the negative shift of the threshold voltage can be obtained, the occurrence of malfunction of the unit shift register SR according to the present embodiment, and the power consumption Increase and decrease in power supply voltage can be prevented.

<Embodiment 6>
In the fourth embodiment, one of the methods for solving the problem of the positive shift of the threshold voltage in the transistors Q2 and Q7 for setting the output terminal OUT and the node N1 to the low impedance L level in the non-selection period, respectively. Explained. This embodiment also shows a method for solving the problem.

  FIG. 17 is a circuit diagram of the unit shift register SR according to the sixth embodiment. The unit shift register SR has two clock terminals. That is, in addition to the first clock terminal CK1 to which the drain of the transistor Q1 is connected, a second clock terminal CK2 to which a clock signal having a phase different from that input thereto is input.

  In addition, a transistor Q12 having a gate connected to the first clock terminal CK1 is provided between the node N1 and the output terminal OUT, and an output terminal (defined as “node N5”) of the inverter including the transistors Q5 and Q6 and the first terminal. A transistor Q13 having a gate connected to the first clock terminal CK1 is provided between the one power supply terminal S1. Further, in the present embodiment, the gate of the transistor Q2 connected between the output terminal OUT and the first power supply terminal S1 is connected to the second clock terminal CK2.

  The inverter composed of the transistors Q5 and Q6 has the node N1 as an input end as in the second and third embodiments, but differs in that the gate and drain of the transistor Q5 are connected to the second clock terminal CK2. That is, the clock signal input to the second clock terminal CK2 serves as a power source for the inverter.

  The basic operation theory of the unit shift register SR in FIG. 17 is almost the same as that described in the first embodiment, but the clock signal input to the second clock terminal CK2 by the inverter composed of the transistors Q5 and Q6. It is characterized in that it is activated when power is supplied by the transistor Q13, and its output is forcibly set to L level by the transistor Q13.

Here, the operation of the k-th stage unit shift register SR _k will also be described as a representative. For simplicity, in the the unit shift register SR _k, the clock signal CLKA is input to the first clock terminal CK1, and those clock signal CLKB is input to the second clock terminal CK2.

First, the operation during the non-selection period of the unit shift register SR _k will be described. Since the node N1 is at the L level during the non-selection period, the node N5 becomes the H level when the inverter formed of the transistors Q5 and Q6 is activated by the clock signal CLKB. When the inverter is inactivated, the transistor Q13 is turned on by the clock signal CLKA, so that the node N5 becomes L level. That is, in the non-selection period, the level of the node N5 changes almost in the same manner as the clock signal CLKB. Therefore, the transistor Q7 sets the node N1 to the low impedance L level at the timing when the clock signal CLKB becomes the H level.

  The transistor Q7 is turned off when the clock signal CLKB is at the L level. During this period, the clock signal CLKA turns on the transistor Q12, so that the charge at the node N1 is discharged to the output terminal OUT by the transistor Q12. Usually, since a capacitive load (in the case of a gate line driving circuit, a gate line of a display panel) is connected to the output terminal OUT, with the charge that is discharged to the output terminal OUT at this time, the output terminal OUT Never goes high.

In this way, non-selection period of the unit shift register SR _k, elevated levels of the node N1 by the transistor Q7 and the transistor Q12 operates to discharge the node N1 alternately is prevented. Since the gate electrodes of the transistors Q7 and Q12 are not continuously positively biased, the positive shift of their threshold voltages is suppressed.

  Further, the transistor Q2 is turned on when the clock signal CLKB becomes H level, and sets the output terminal OUT to L level with low impedance. That is, since the gate of the transistor Q2 is not continuously positively biased, the positive shift of the threshold voltage is also suppressed.

Further, when the output signal G _k-1 in the previous stage becomes H level and the selection period of the unit shift register SR _k is reached, the node N1 becomes H level. In the meantime, even if the inverter composed of the transistors Q5 and Q6 is activated by the clock signal CLKB, the node N5 is at the L level, so the transistor Q7 is turned off and the H level of the node N1 is maintained. When the clock signal CLKA becomes H level, the gate of the transistor Q12 becomes H level. At the same time, the output terminal OUT (output signal G _k ) also becomes H level, so that the transistor Q12 is not turned on and the node N1 is floating. In this state, it is maintained at the H level (stepped up by the clock signal CLKA). Therefore, the unit shift register SR _k can normally output the output signal G _k .

  As described above, also in the unit shift register SR of the present embodiment, the level of the node N1 changes as in the first embodiment. That is, the unit shift register SR operates so as to be maintained in the reset state during the non-selection period and to be set during the selection period. Therefore, a signal shift operation similar to that in Embodiment 1 can be performed.

  In the above description, the source of the transistor Q2 is connected to the first power supply terminal S1, but it may be connected to the first clock terminal CK1. In that case, when the clock signal CLKB input to the gate of the transistor Q2 becomes L level and the transistor Q2 is turned off, the clock signal CLKA input to the source becomes H level, so that the gate of the transistor Q2 Is equivalent to being negatively biased with respect to the source. As a result, the threshold voltage shifted in the positive direction returns and recovers in the negative direction, so that the reduction in the driving capability of the transistor Q2 is reduced, and the operation life of the circuit is extended.

  Also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. Accordingly, it is possible to prevent the malfunction of the unit shift register SR, the increase in power consumption, and the decrease in the power supply voltage.

<Embodiment 7>
Also in the seventh embodiment, a technique for solving the problem of the positive shift of the threshold voltage in the transistors Q2 and Q7 for setting the output terminal OUT and the node N1 to the low impedance L level in the non-selection period will be described.

  FIG. 18 is a circuit diagram of the unit shift register SR according to the seventh embodiment. The unit shift register SR also includes a first clock terminal CK1 to which the drain of the transistor Q1 is connected, and a second clock terminal CK2 to which a clock signal having a phase different from that input thereto is input.

  The circuit in FIG. 18 has a configuration similar to that in FIG. 12, except that an inverter having the gate node (node N1) of the transistor Q1 as an input terminal and the gate node (node N2) of the transistor Q2 as an output terminal is provided. It is different in that it is a capacitive load type inverter. That is, the inverter uses the load capacitance C3 as a load element. The inverter is different from a normal inverter in that the clock signal input to the first clock terminal CK1 is a power source. That is, the load capacitor C3 is connected between the node N2 that is the output terminal of the inverter and the first clock terminal CK1. The load capacitor C3 is a load element of the inverter and also functions as a coupling capacitor between the first clock terminal CK1 and the node N2.

  Further, in the circuit of FIG. 18, the transistor Q14 is connected in parallel with the transistor Q2 whose gate is connected to the output terminal of the inverter. The gate of the transistor Q14 is connected to the second clock terminal CK2.

  The basic operation theory of the unit shift register SR of FIG. 18 is almost the same as that described in the first embodiment, but an inverter composed of a load capacitor C3 and a transistor Q6 is input to the first clock terminal CK1. It is characterized in that it is activated when power is supplied by the clock signal.

Here, the operation of the k-th stage unit shift register SR _k will also be described as a representative. For simplicity, in the the unit shift register SR _k, the clock signal CLKA is input to the first clock terminal CK1, and those clock signal CLKB is input to the second clock terminal CK2.

First, the operation during the non-selection period of the unit shift register SR _k will be described. Since the node N1 is at the L level during the non-selection period, the node N2 becomes the H level when the inverter composed of the load capacitor C3 and the transistor Q6 is activated by the clock signal CLKA. When the inverter becomes inactive, the node N2 becomes L level in response to the fall of the clock signal CLKA because of coupling through the load capacitor C3. That is, in the non-selection period, the level of the node N2 changes almost in the same manner as the clock signal CLKA. Therefore, the transistor Q7 sets the node N1 to L level with low impedance at the timing when the clock signal CLKA becomes H level.

  Similarly to the transistor Q7, the transistor Q2 is turned on at a timing synchronized with the clock signal CLKA, thereby setting the output terminal OUT to the L level of low impedance. When the clock signal CLKA is at L level, the transistor Q2 is turned off. At this time, the transistor Q14 is turned on by the clock signal CLKB, and the output terminal OUT is set to L level with low impedance.

In this way, non-selection period of the unit shift register SR _k, is prevented elevated levels of the node N1 by operating as the transistor Q7 to discharge the node N1 at a timing synchronized with the clock signal CLKA. Also by the transistor Q2 and the transistor Q14 to discharge the output terminal OUT alternately, so as to prevent the output signal G _k as a false signal occurs. Since the gate electrodes of these transistors Q2, Q7, and Q14 are not continuously positively biased, the positive shift of their threshold voltages is suppressed.

Further, when the output signal G _{k-1 of} the previous stage becomes H level and the selection period of the unit shift register SR _k is reached, the dual gate transistor Q3D of the unit shift register SR _k is turned on, so that the node N1 becomes H level. At that time, the inverter composed of the load capacitor C3 and the transistor Q6 is inactive, and the node N2 is at the L level. When the clock signal CLKA becomes H level, the inverter is activated. However, since the transistor Q6 is turned on, the node N2 maintains L level. Therefore, the transistor Q7 is kept off during the selection period, and the node N1 is kept at the H level in a floating state (boosted by the clock signal CLKA). Therefore, the unit shift register SR _k can normally output the output signal G _k .

  As described above, also in the unit shift register SR of the present embodiment, the level of the node N1 changes as in the first embodiment. That is, the unit shift register SR operates so as to be maintained in the reset state during the non-selection period and to be set during the selection period. Therefore, a signal shift operation similar to that in Embodiment 1 can be performed.

  Also in the present embodiment, an effect of suppressing the negative shift of the threshold voltages of the transistors Q3a and Q3b constituting the dual gate transistor Q3D can be obtained. Accordingly, it is possible to prevent the malfunction of the unit shift register SR, the increase in power consumption, and the decrease in the power supply voltage.

<Eighth embodiment>
In the eighth embodiment, an example in which the dual gate transistor Q3D according to the present invention is applied to a shift register (bidirectional shift register) capable of switching the signal shift direction will be described.

  FIG. 19 is a circuit diagram of the unit shift register SR according to the eighth embodiment. The unit shift register SR is a bidirectional shift register. The circuit configuration itself of the unit shift register SR is substantially the same as that of FIG. 11, but the transistor Q4 is replaced with the dual gate transistor Q4D (transistors Q4a and Q4b) according to the present invention. Similar to the above-described dual gate transistor Q3D, when the gate electrode changes from H level to L level, the dual gate transistor Q4D is defined as a connection node (defined as “node N6”) between the transistors Q4a and Q4b accordingly. ) Is lowered to the L level, and the parasitic capacitance between the gate and the node N6 is increased.

  A unit shift register that performs a shift in only one direction basically enters a set state when a signal is input to the input terminal, and enters a reset state when a signal is input to the reset terminal. Since the input terminal and the reset terminal are functionally interchanged according to the direction, there is no distinction. For convenience of explanation, a terminal to which the gate of the dual gate transistor Q3D is connected is referred to as a “first input terminal IN1”, and a terminal to which the gate of the dual gate transistor Q4D is connected is referred to as a “second input terminal IN2.”

  Further, the first voltage signal VN and the second voltage signal VR which are control signals for determining the signal shift direction are input to the unit shift register SR which is a bidirectional shift register. The dual gate transistor Q3D is connected between the first voltage signal terminal TN to which the first voltage signal VN is input and the node N1, and the dual gate transistor Q4D is the second voltage signal to which the second voltage signal VR is input. Connect to terminal TR. The first voltage signal VN and the second voltage signal VR are complementary signals.

  For example, when the first voltage signal VN is H level (VDD) and the second voltage signal VR is L level (VSS), the first voltage signal terminal TN is VDD and the second voltage signal terminal TR is VSS in FIG. Therefore, the dual gate transistor Q3D serves as a charging circuit for the node N1, and the dual gate transistor Q4D serves as a discharging circuit for the node N1. That is, in this state, the first input terminal IN1 functions as the input terminal IN1 in FIG. 11, and the second input terminal IN2 functions as the reset terminal RST in FIG.

  On the other hand, when the first voltage signal VN is at L level (VSS) and the second voltage signal VR is at H level (VDD), the dual gate transistor Q3D becomes a discharge circuit of the node N1, and the dual gate transistor Q4D is at the node N1. It becomes a charging circuit. That is, in this state, the first input terminal IN1 functions as the reset terminal RST in FIG. 11, and the second input terminal IN2 functions as the input terminal IN1 in FIG.

That is, according to the multistage shift register in which the unit shift registers SR are cascaded as shown in FIG. 2, when the first voltage signal VN is at the H level and the second voltage signal VR is at the L level, the output signal G1 , G2, G3,... (Forward shift). On the contrary, when the first voltage signal VN is L level and the second voltage signal VR is H level, the output signals G _n , G _n−1 , G _n−2. shift).

  Therefore, in the present embodiment, the effect of suppressing the negative shift of the threshold voltage can be obtained in the transistors Q3a and Q3b constituting the dual gate transistor Q3D during the forward shift operation. Further, in the reverse shift operation, an effect of suppressing the negative shift of the threshold voltage can be obtained in the transistors Q4a and Q4b constituting the dual gate transistor Q4D. Therefore, also in this embodiment, it is possible to prevent the malfunction of the unit shift register SR, increase in power consumption, and decrease in power supply voltage.

  FIG. 19 shows an example in which the dual gate transistors Q3D and Q4D according to the present invention are applied to the bidirectional shift register based on the configuration of the unit shift register SR of the second embodiment (FIG. 11). The application of the present invention to the bidirectional shift register is not limited thereto. Hereinafter, modifications of the present embodiment will be shown.

  For example, FIG. 20 shows an example in which dual gate transistors Q3D and Q4D are applied to the bidirectional shift register based on the unit shift register SR of the third embodiment (FIG. 12). That is, a transistor Q7 for discharging the node N1 during the non-selection period is provided for the circuit of FIG. Further, for example, FIG. 21 and FIG. 22 show dual gate transistors Q3D, Q2D, QB, D, R, R, and R respectively for a bidirectional shift register based on the unit shift register SR of the seventh embodiment (FIG. 18) and the fourth embodiment (FIG. 13). This is an example in which Q4D is applied. In these modified examples, the same effect as described above can be obtained.

<Embodiment 9>
As described above, in the dual gate transistor Q3D, the transistor Q3a when the gates of the transistors Q3a and Q3b (the input terminal IN1 of the unit shift register SR) change from the H level (VDD) to the L level (VSS = 0V). , Q3b, the amount of change in the level of the node N3 is VDD × Cgs, where Cgs is a capacitance component between the input terminal IN1 and the node N3, and Cstr is a parasitic capacitance associated with the node N3 not included in the Cgs. / (Cgs + Cstr). That is, as the capacitance component Cgs between the input terminal IN1 and the node N3 is larger than the parasitic capacitance Cstr (that is, the proportion of the capacitance component Cgs in the total parasitic capacitance associated with the node N3 is larger), the node N3 is changed. Can be lowered to a lower level.

  Therefore, in the present embodiment, a method for increasing the capacitance component between the input terminal IN1 and the node N3 in the dual gate transistor Q3D will be described. In general, a field effect transistor has an overlap capacitance between a gate and a source / drain as a parasitic capacitance. Therefore, as shown in FIG. 23, the dual gate transistor Q3D includes, as parasitic capacitances, a gate-drain overlap capacitance Cgdo (Q3a) and a gate-source overlap capacitance Cgso (Q3a) of the transistor Q3a, and a gate-drain overlap of the transistor Q3b. It has a wrap capacitance Cgdo (Q3b) and a gate-source overlap capacitance Cgso (Q3b).

  Of these, the gate-source overlap capacitances Cgso (Q3a) and Cgso (Q3b) of the transistors Q3a and Q3b contribute to the capacitance component (Cgs) between the input terminal IN1 and the node N3. It is desirable that their capacitance values be sufficiently large.

  FIGS. 24A and 24B are diagrams showing the configuration of the dual gate transistor Q3D according to the present embodiment. FIG. 24A is a cross-sectional view of the dual gate transistor Q3D, and FIG. 24B is a top view thereof. Note that FIG. 24A corresponds to the cross section along the line AA shown in FIG.

  The dual gate transistor Q3D is a so-called “bottom gate transistor” in which source / drain regions are disposed on a gate electrode. That is, the dual gate transistor Q3D includes a gate electrode 11 formed on the glass substrate 10, a gate insulating film 12 formed on the gate electrode 11, and an active region 13 (intrinsic silicon formed on the gate insulating film 12). ), And a contact layer 14 (N + type silicon) formed on the active region 13. The contact layer 14 becomes the source / drain of the transistors Q3a and Q3b, and the drain wiring 15 of the transistor Q3a, the drain wiring 16 of the transistor Q3b, and the source wiring 17 of the transistors Q3a and Q3b are formed thereon, respectively.

  For example, in the unit shift register SR of FIG. 7, the gate electrode 11 is connected to the input terminal IN1 of the unit shift register SR, the drain wiring 15 of the transistor Q3a is connected to the second power supply terminal S2, and the drain wiring 16 of the transistor Q3b is connected to the node. Connected to N1. The source wiring 17 of the transistors Q3a and Q3b becomes the node N3.

  As shown in FIG. 24B, in the present embodiment, the pattern of the wiring 17 serving as the node N3 is made larger (wider) than the other wirings 15 and 16. By doing so, the area where the gate electrode 11 and the wiring 17 face each other is increased, and the overlap capacitances Cgso (Q3a) and Cgso (Q3b) can be increased. That is, the capacitance component (Cgs≈Cgso (Q3a) + Cgso (Q3b)) between the input terminal IN1 and the node N3 can be increased.

  As a result, when the gate of the dual gate transistor Q3D is changed from the H level to the L level, the level of the node N3 between the transistors Q3a and Q3b can be lowered to a sufficiently low level. The effect of the present invention of suppressing the negative voltage shift is improved.

  At this time, the pattern of the drain wiring 15 of the transistor Q3a and the drain wiring 16 of the transistor Q3b may be enlarged, and the above-described effect is obtained. However, since the formation area of the dual gate transistor Q3D increases remarkably, it is desirable to enlarge only the pattern of the source wiring 17 serving as the node N3 as shown in FIG. That is, in FIG. 23, it is desirable to increase only the values of Cgso (Q3a) and Cgso (Q3b) while maintaining the values of Cgdo (Q3a) and Cgdo (Q3b). As a result, the relationship of Cgso (Q3a)> Cgdo (Q3a) and Cgso (Q3b)> Cgdo (Q3b) is established.

  By the way, when the width of the wiring 17 serving as the node N3 is increased as in the present embodiment, it seems that not only the capacitance component Cgs but also the parasitic capacitance Cstr of the node N3 not included in the Cgs is increased. The parasitic capacitance Cstr hardly increases.

  The parasitic capacitance Cstr is a capacitance of the wiring 17 such as a grounding capacitance in the wiring 17 or a parasitic capacitance between the counter electrode (common electrode) disposed above the glass substrate 10 and the wiring 17 in the case of a liquid crystal display device, for example. “Fringe capacity”. Since the ground electrode and the common electrode are far from the wiring 17, even if the width of the wiring 17 changes, the value of the fringe capacitance hardly changes. This is why the parasitic capacitance Cstr is hardly increased even if the width of the wiring 17 is increased in this embodiment.

  On the other hand, the gate-source overlap capacitors Cgso (Q3a) and Cgso (Q3b) can be regarded as parallel plate type capacitors in which the wiring 17 and the gate electrode 11 face each other. Therefore, if the width of the wiring 17 is widened, the values of Cgso (Q3a) and Cgso (Q3b) increase in proportion to it.

  Therefore, according to the present embodiment, it is possible to increase the value of the capacitance component Cgs while maintaining the value of the parasitic capacitance Cstr. In other words, the ratio of the capacitive component Cgs to the parasitic capacitance associated with the node N3 can be increased. As a result, when the gates of the transistors Q3a and Q3b (input terminal IN1 of the unit shift register SR) change from the H level to the L level, the node N3 is lowered to a lower level, and the above effect is obtained. can get.

  In the above description, an example of a bottom gate type transistor has been described. However, the present embodiment can also be applied to, for example, a “top gate type transistor” in which the gate electrode 11 is disposed on the active region 13. It is. FIGS. 25A and 25B show an example where the dual gate transistor Q3D is a top gate transistor.

  In the case of a top gate type transistor, the gate electrode 11 is formed in a line pattern crossing the active region 13 as shown in FIG. Since the drain region 151 of the transistor Q3a, the drain region 161 of the transistor Q3b, and the source region 171 of the transistors Q3a and Q3b are formed in the active region 13 below the gate electrode 11, they are connected to an upper layer wiring. A contact 18 is formed for this purpose. FIG. 25B shows the wiring pattern.

  Also in this case, the pattern of the wiring 17 serving as the node N3 is made larger than those of the other wirings 15 and 16. As a result, the area where the gate electrode 11 and the wiring 17 face each other is increased, the overlap capacitances Cgso (Q3a) and Cgso (Q3b) can be increased, and the same effect as described above can be obtained.

  This embodiment is applicable to all the dual gate transistors Q3D in the first to eighth embodiments. Although only the dual gate transistor Q3D has been described above, the present invention can also be applied to the dual gate transistor Q10D in the modification of the fifth embodiment shown in FIG. 16 and the dual gate transistor Q4D in the eighth embodiment. It is clear.

<Embodiment 10>
In the above embodiment, the means for lowering the level of the node N3 of the dual gate transistor Q3D is the parasitic capacitance between the gate (input terminal IN1) and the node N3 (the gate-source overlap capacitance of the transistors Q3a and Q3b). )Met.

  As described above, in order to lower the level of the node N3 to a lower level, the capacitance component between the gate and the node N3 may be increased. Therefore, in the present embodiment, as shown in FIG. 26A, the capacitive element C4 is connected between the gate of the dual gate transistor Q3D and the node N3. Thus, the capacitance component between the gate and the node N3 can be increased without using the technique as in the ninth embodiment, and the node N3 can be more reliably lowered to the L level. That is, the negative shift of the threshold voltage of the dual gate transistor Q3D can be prevented more reliably.

  This embodiment is applicable to all the dual gate transistors Q3D in the first to eighth embodiments. The present invention is also applicable to the dual gate transistor Q4D in the eighth embodiment. In that case, a capacitive element C5 may be connected between the gate of the dual gate transistor Q4D and the node N6 as shown in FIG. Also, the present invention can be applied to the dual gate transistor Q10D in the modification of the fifth embodiment shown in FIG. 16, and in that case, as shown in FIG. 26C, the gate of the dual gate transistor Q10D, the node N10, The capacitor C6 may be connected between the two.

<Embodiment 11>
In the above embodiment, the means for lowering the level of the node N3 of the dual gate transistor Q3D is the capacitive component between the gate (input terminal IN1) and the node N3, but a diode can also be used.

  FIG. 27A shows a dual gate transistor Q3D according to the present embodiment. As shown in the figure, a diode D1 is connected between the gate of the dual gate transistor Q3D and the node N3 so that the gate side is the cathode and the node N3 is the anode. Also in this case, when the gate of the dual gate transistor Q3D changes from the H level to the L level, the node N3 is pulled down to the L level following the change. Therefore, the dual gate transistor Q3D shown in FIG. 27A can achieve the same effect as that of the first embodiment in which the negative shift of the threshold voltage is suppressed.

  In each of the above-described embodiments, when the level of the node N3 is lowered, the node N3 becomes L level in a floating state. For this reason, when a leak current is generated in the transistors Q3a and Q3b, the level of the node N3 rises, causing a problem that the effect of the present invention is reduced.

  On the other hand, in the dual gate transistor Q3D of FIG. 27A, even if the level of the node N3 increases due to the leakage current, the charge is discharged through the diode D1. That is, even when a leak current occurs, the level of the node N3 does not become higher than the threshold voltage of the diode D1, so that the above problem can be solved.

  This embodiment is applicable to the dual gate transistors Q3D of all the unit shift registers SR in the first to eighth embodiments. The present invention can also be applied to the dual gate transistor Q4D of the unit shift register SR in the eighth embodiment. In this case, as shown in FIG. 27B, a diode D2 is connected between the gate of the dual gate transistor Q4D and the node N6 so that the gate side is the cathode and the node N6 side is the anode. Also, the present invention can be applied to the dual gate transistor Q10D in the modification of the fifth embodiment shown in FIG. 16, in which case the gate of the dual gate transistor Q10D, the node N10, and the like as shown in FIG. Between them, the diode D3 may be connected so that the gate side becomes the cathode and the node N10 side becomes the anode.

<Embodiment 12>
In the present embodiment, a transistor is used as means for lowering the level of node N3 of dual gate transistor Q3D.

FIG. 28A shows a dual gate transistor Q3D according to the present embodiment. As shown in the figure, between the gate of the dual gate transistor Q3D and the node N3, a transistor that is turned on when a signal (previous output signal G _k-1 ) input to the gate changes from H level to L level. Connect Q3c. For example, FIG. 29 shows an example in which the dual gate transistor Q3D of FIG. 28A is applied to the unit shift register SRk of FIG.

When the unit shift registers SR are connected in cascade, the output signal G _k-1 of the preceding stage is input to the input terminal IN1 of the _k-th unit shift register SR _k and is input to the gate of the dual gate transistor Q3D. In order to turn on the transistor Q3c when the signal (the previous stage output signal G _k-1 ) changes from the H level to the L level, the gate of the transistor Q3c may be connected to the clock terminal CK1.

For example, when it is assumed that the clock signal CLKA is input to the clock terminal CK1 as the unit shift register SR _k , the output signal G _{k−1 at the} preceding stage becomes H level at the rising timing of the clock signal CLKB. When the output signal _{Gk-1 at the} previous stage becomes H level, the transistors Q3a and Q3b are turned on, and the nodes N1 and N3 become H level. At this time, since the clock signal CLKA input to the clock terminal CK1 is at L level, the transistor Q3c is off.

Thereafter, when the output signal G _{k-1 at} the previous stage changes to L level, the transistors Q3a and Q3b are turned off. At this time, since the clock signal CLKA becomes H level, the transistor Q3c is turned on. As a result, the node N3 is discharged by the transistor Q3c and pulled down to the L level. That is, the transistors Q3a and Q3b are prevented from having the gate at a low potential with respect to both the source and the drain. Therefore, an effect similar to that of the first embodiment in which the negative shift of the threshold voltage in the dual gate transistor Q3D is suppressed can be obtained.

  This embodiment is applicable to the dual gate transistors Q3D of all the unit shift registers SR in the first to eighth embodiments. The present invention can also be applied to the dual gate transistor Q4D of the unit shift register SR in the eighth embodiment. In that case, as shown in FIG. 28B, the transistor Q4c whose gate is connected to the clock terminal CK1 is connected between the gate of the dual gate transistor Q4D and the node N6.

The present invention can also be applied to the dual gate transistor Q10D in the modification of the fifth embodiment shown in FIG. In this case, as shown in FIG. 28C, the transistor Q10c is connected between the gate of the dual gate transistor Q10D and the node N10. However, the output signal G _k-2 of the previous stage is input to the gate of the dual gate transistor Q10D, and it is necessary to turn on the transistor Q10c when it changes from the H level to the L level. Therefore, as shown in FIG. 28C, the gate of the transistor Q10c is connected to the clock terminal CK2.

It is a circuit diagram which shows the structure of the conventional unit shift register. It is a figure which shows the structure of a multistage shift register. It is a timing diagram which shows operation | movement of the conventional unit shift register. It is a figure which shows operation | movement of a multistage shift register. It is a figure for demonstrating the problem of the conventional unit shift register. It is a figure which shows the experimental result showing the relationship between the electric potential state of an a-Si transistor, and the shift of a threshold voltage. FIG. 3 is a circuit diagram of a unit shift register according to the first embodiment. FIG. 6 is a timing diagram illustrating an operation of the unit shift register according to the first embodiment. FIG. 6 is a diagram for explaining an effect of the first embodiment. FIG. 6 is a diagram for explaining an effect of the first embodiment. 6 is a circuit diagram of a unit shift register according to Embodiment 2. FIG. FIG. 6 is a circuit diagram of a unit shift register according to a third embodiment. FIG. 6 is a circuit diagram of a unit shift register according to a fourth embodiment. FIG. 10 is a circuit diagram of a unit shift register according to a fifth embodiment. FIG. 10 is a diagram illustrating a configuration of a multistage shift register including unit shift registers according to a fifth embodiment. FIG. 10 is a circuit diagram of a unit shift register that is a modification of the fifth embodiment. FIG. 10 is a circuit diagram of a unit shift register according to a sixth embodiment. FIG. 10 is a circuit diagram of a unit shift register according to a seventh embodiment. FIG. 10 is a circuit diagram of a unit shift register according to an eighth embodiment. FIG. 25 is a circuit diagram of a unit shift register that is a modification of the eighth embodiment. FIG. 25 is a circuit diagram of a unit shift register that is a modification of the eighth embodiment. FIG. 25 is a circuit diagram of a unit shift register that is a modification of the eighth embodiment. It is a figure which shows distribution of the overlap capacity | capacitance in a dual gate transistor. FIG. 10 is a diagram illustrating a configuration of a dual gate transistor according to a ninth embodiment. FIG. 10 is a diagram illustrating a configuration of a dual gate transistor according to a ninth embodiment. It is a figure which shows the structure of the dual gate transistor concerning Embodiment 10. FIG. FIG. 38 shows a structure of a dual gate transistor according to the eleventh embodiment. FIG. 20 is a diagram showing a configuration of a dual gate transistor according to a twelfth embodiment. FIG. 20 is a circuit diagram of a unit shift register according to a twelfth embodiment.

Explanation of symbols

  Q1-Q14 transistor, Q3D, Q4D, Q10D dual gate transistor, CK1, CK2 clock terminal, RST reset terminal, S1 first power supply terminal, S2 second power supply terminal, IN1, IN2 input terminal, OUT output terminal, TA first control Terminal, TB second control terminal, TN first voltage signal terminal, TR second voltage signal terminal, C1 to C6 capacitive element, D1, D2 diode.

Claims (44)

A semiconductor device comprising a plurality of first transistors connected in series between a predetermined first node and a second node and having control electrodes connected to each other,
Each of connection nodes between the plurality of first transistors is a third node,
From the state where each of the first to third nodes and the control electrode is at an H (High) level higher than the threshold voltage of the plurality of first transistors, the first and second nodes are at an H level. When the potential of the control electrode changes to an L (Low) level lower than the threshold voltage, the level of the third node is also lowered to the L level accordingly .
In each of the third nodes,
The capacitance component between the control electrode and the third node is C1, the parasitic capacitance associated with the third node not included in the C1 is C2, the difference between the H level and the L level at the control electrode is Vd, When the threshold voltage of the first transistor is Vth,
C1 ≧ C2 × (Vd−Vth) / Vth
A semiconductor device characterized by satisfying the relationship: The semiconductor device according to claim 1,
The means for lowering the level of the third node is a parasitic capacitance between the control electrode and the third node. The semiconductor device according to claim 2,
The semiconductor device according to claim 1, wherein a width of the electrode to be the third node is wider than a width of the electrode to be the first and second nodes. A semiconductor device according to any one of claims 1 to 3,
A semiconductor device, further comprising a capacitive element connected between the control electrode and the third node as means for lowering the level of the third node. The semiconductor device according to claim 1,
As means for lowering the level of the third node, a diode connected between the control electrode and the third node and having the control electrode side as a cathode and the third node side as an anode is further provided.
A semiconductor device. The semiconductor device according to claim 1,
The device further includes a second transistor connected between the control electrode of the first transistor and the third node as means for lowering the level of the third node.
A semiconductor device. A semiconductor device according to any one of claims 1 to 6,
The plurality of first transistors are amorphous silicon thin film transistors.
A semiconductor device. A semiconductor device according to any one of claims 1 to 6,
The plurality of first transistors are organic transistors
A semiconductor device. An input terminal, an output terminal, a first clock terminal and a reset terminal;
A first transistor for supplying a first clock signal input to the first clock terminal to the output terminal;
A second transistor for discharging the output terminal;
A charging circuit that charges a first node to which the control electrode of the first transistor is connected in response to an input signal input to the input terminal;
A discharge circuit for discharging the first node in response to a reset signal input to the reset terminal;
The charging circuit is
A plurality of third transistors connected in series between the first node and a power supply terminal, each having a control electrode connected to the input terminal;
When the input signal becomes H (High) level higher than the threshold voltage of the third transistor, the plurality of third transistors are turned on to charge the first node, and then the input signal is the threshold value. Each connection node between the plurality of third transistors is configured to be pulled down to L level when the level changes to L (Low) level lower than the voltage.
In each of connection nodes between the plurality of third transistors,
The capacitance component between the input terminal and the connection node is C1, the parasitic capacitance associated with the connection node not included in the C1 is C2, the difference between the H level and the L level of the input signal is Vd, and the first When the threshold voltage of the three transistors is Vth,
C1 ≧ C2 × (Vd−Vth) / Vth
Satisfy the relationship
A shift register circuit. A shift register circuit according to claim 9,
The control electrode of the second transistor is connected to the reset terminal
A shift register circuit. A shift register circuit according to claim 9,
An inverter having the first node as an input terminal and a second node connected to a control electrode of the second transistor as an output terminal;
A shift register circuit. A shift register circuit according to claim 11,
A fourth transistor having a control electrode connected to the second node and discharging the first node;
A shift register circuit. A shift register circuit according to claim 9,
Two second transistors,
Nodes to which the control electrodes of each of the two second transistors are connected are second and third nodes, respectively.
The two second transistors are alternately driven based on a predetermined control signal.
A shift register circuit. A shift register circuit according to claim 13,
The control signal is
The first and second control signals are complementary to each other;
The shift register circuit is
First and second control terminals to which the first and second control signals are respectively input;
A fourth transistor connected between the first control terminal and the second node;
A fifth transistor connected between the second control terminal and the third node;
The fourth and fifth transistors are:
One of the main electrodes is connected to each other's control electrode.
A shift register circuit. The shift register circuit according to claim 13 or 14,
The means for alternately driving the two second transistors comprises:
An inverter having the first node as an input end;
And a switching circuit for alternately connecting the output terminal of the inverter to the second and third nodes based on the control signal.
A shift register circuit. The shift register circuit according to any one of claims 13 to 15,
A sixth transistor having a control electrode connected to the second node and discharging the first node;
A seventh transistor having a control electrode connected to the third node and discharging the first node;
A shift register circuit. A shift register circuit according to claim 9,
A fourth transistor having a control electrode connected to the first clock terminal and connected between the first node and the output terminal;
A shift register circuit. A shift register circuit according to claim 17,
A second clock terminal to which a second clock signal having a phase different from that of the first clock signal is input;
An inverter having the first node as an input terminal and activated by the second clock signal;
A fifth transistor having a control electrode connected to the first clock terminal and discharging an output terminal of the inverter;
A control transistor connected to the output terminal of the inverter; and a sixth transistor for discharging the first node.
A shift register circuit. The shift register circuit according to claim 18,
The control electrode of the second transistor is connected to the second clock terminal.
A shift register circuit. The shift register circuit according to claim 19,
The second transistor is connected between the output terminal and the first clock terminal.
A shift register circuit. A shift register circuit according to claim 9,
An inverter having the first node as an input terminal and activated by the first clock signal;
A control electrode connected to the output terminal of the inverter, and a fourth transistor for discharging the first node;
The inverter is
A first capacitive element connected between the output terminal of the inverter and the first clock terminal is provided as a load element.
A shift register circuit. The shift register circuit according to claim 21,
The control electrode of the second transistor is connected to the output terminal of the inverter
A shift register circuit. The shift register circuit according to claim 22,
A second clock terminal to which a second clock signal having a phase different from that of the first clock signal is input;
And a fifth transistor having a control electrode connected to the second clock terminal and discharging the output terminal.
A shift register circuit. A shift register circuit according to any one of claims 9 to 23, wherein
A second capacitive element connected between each connection node between the plurality of third transistors and the input terminal;
A shift register circuit. A shift register circuit according to any one of claims 9 to 23, wherein
A diode further connected between each connection node between the plurality of third transistors and the input terminal with the input terminal side as a cathode and the connection node side as an anode is further provided.
A shift register circuit. A shift register circuit according to any one of claims 9 to 23, wherein
A sixth transistor having a control electrode connected to the first clock terminal is further provided between each input node between the plurality of third transistors and the input terminal.
A shift register circuit. First and second input terminals, an output terminal, a first clock terminal and a reset terminal;
A first transistor for supplying a first clock signal input to the first clock terminal to the output terminal;
A second transistor for discharging the output terminal;
A first charging circuit for charging a first node to which the control electrode of the first transistor is connected;
A first discharge circuit for discharging the first node in response to a reset signal input to the reset terminal;
The first charging circuit includes:
A plurality of third transistors connected in series between the first node and a power supply terminal, the control electrodes of which are both connected to a predetermined second node;
A second charging circuit that charges the second node in response to a first input signal input to the first input terminal;
A booster circuit that boosts the second node in response to a second input signal input to the second input signal;
A second discharge circuit for discharging the second node in response to the reset signal;
In each of connection nodes between the plurality of third transistors,
The capacitance component between the second node and the connection node is C1, the parasitic capacitance associated with the connection node not included in the C1 is C2, and the difference between the H level and the L level at the second node is Vd, When the threshold voltage of the third transistor is Vth,
C1 ≧ C2 × (Vd−Vth) / Vth
Satisfy the relationship
A shift register circuit. The shift register circuit according to claim 27, wherein
The first charging circuit includes:
When the second node becomes H (High) level higher than the threshold voltage of the third transistor, the plurality of third transistors are turned on to charge the first node, and then the second node is Each connection node between the plurality of third transistors is lowered to L level when the level changes to L (Low) level lower than the threshold voltage.
A shift register circuit. A shift register circuit according to claim 27 or claim 28,
An inverter having the second node as an input end;
The first discharge circuit includes:
A fourth transistor having a control electrode connected to an output terminal of the inverter and discharging the first node;
A shift register circuit. A shift register circuit according to claim 29,
A control transistor connected to an output terminal of the inverter; and a fifth transistor for discharging the second node.
A shift register circuit. A shift register circuit according to claim 29 or claim 30, wherein
The control electrode of the second transistor is connected to the output terminal of the inverter
A shift register circuit. A shift register circuit according to any one of claims 27 to 31,
And a capacitive element connected between each connection node between the plurality of third transistors and the second node.
A shift register circuit. A shift register circuit according to any one of claims 27 to 31,
And a diode connected between each connection node between the plurality of third transistors and the second node with the second node side serving as a cathode and the connection node side serving as an anode.
A shift register circuit. A shift register circuit according to any one of claims 27 to 31,
A sixth transistor having a control electrode connected to the first clock terminal is further provided between each connection node between the plurality of third transistors and the second node.
A shift register circuit. A shift register circuit according to any one of claims 27 to 34, wherein
The second charging circuit includes:
A plurality of seventh transistors are connected in series between the second node and the power supply terminal, and the control electrodes are both connected to the first input terminal.
A shift register circuit. The shift register circuit according to claim 35, wherein
The second charging circuit includes:
When the first input terminal becomes H (High) level higher than the threshold voltage of the seventh transistor, the plurality of seventh transistors are turned on to charge the second node, and then the first input terminal Is changed to L (Low) level lower than the threshold voltage, each connection node between the plurality of seventh transistors is pulled down to L level.
A shift register circuit. A shift register circuit according to claim 35 or claim 36,
And a capacitive element connected between each connection node between the plurality of seventh transistors and the first input terminal.
A shift register circuit. A shift register circuit according to claim 35 or claim 36,
A diode connected between each connection node between the plurality of seventh transistors and the first input terminal with the first input terminal side as a cathode and the connection node side as an anode is further provided.
A shift register circuit. A shift register circuit according to claim 35 or claim 36,
The second clock terminal is connected between each connection node between the plurality of seventh transistors and the first input terminal, and the control electrode is input to a second clock signal having a phase different from that of the first clock signal. It further has a connected eighth transistor
A shift register circuit. First and second input terminals, output terminals and clock terminals;
First and second voltage signal terminals to which first and second voltage signals complementary to each other are respectively input;
A first transistor for supplying a clock signal input to the clock terminal to an output terminal;
A second transistor for discharging the output terminal;
A first drive circuit for supplying the first voltage signal to a first node to which a control electrode of the first transistor is connected based on a first input signal input to the first input terminal;
A second driving circuit for supplying the second voltage signal to the first node based on a second input signal input to the second input terminal;
An inverter having the first node as an input terminal and a second node connected to a control electrode of the second transistor as an output terminal;
The first drive circuit includes:
A plurality of third transistors connected in series between the first node and the first voltage signal terminal, each having a control electrode connected to the first input terminal;
The second driving circuit includes:
A plurality of fourth transistors connected in series between the first node and the second voltage signal terminal, each having a control electrode connected to the second input terminal;
In a first connection node that is each of connection nodes between the plurality of third transistors,
The capacitance component between the first input terminal and the first connection node is C1, the parasitic capacitance associated with the first connection node not included in C1 is C2, and the H level and L level of the first input signal. Is Vd, and the threshold voltage of the third transistor is Vth,
C1 ≧ C2 × (Vd−Vth) / Vth
Satisfy the relationship
In a second connection node that is each of connection nodes between the plurality of fourth transistors,
The capacitance component between the second input terminal and the second connection node is C1a, the parasitic capacitance associated with the second connection node not included in the C1a is C2a, and the H level and L level of the second input signal. And Vda, and the threshold voltage of the fourth transistor is Vtha,
C1a ≧ C2a × (Vda−Vtha) / Vtha
Satisfy the relationship
A shift register circuit. The shift register circuit according to claim 40, wherein
The first drive circuit includes:
The first voltage signal is at an H (High) level that is higher than the threshold voltage of the third and fourth transistors, and the second voltage signal is lower than the threshold voltage of the third and fourth transistors. When it is L (Low) level,
When the first input signal becomes H level, the plurality of third transistors are turned on to charge the first node, and when the first input signal subsequently changes to L level, each of the plurality of third transistors is The connection node is configured to be pulled down to L level,
The second driving circuit includes:
When the first voltage signal is at L level and the second voltage signal is at H level,
When the second input signal becomes H level, the plurality of fourth transistors are turned on to charge the first node, and when the second input signal subsequently changes to L level, each of the plurality of fourth transistors is changed. The connection node is configured to be pulled down to L level
A shift register circuit. A shift register circuit according to claim 40 or claim 41,
A first capacitive element connected between each connection node between the plurality of third transistors and the first input terminal;
A second capacitive element connected between each connection node between the plurality of fourth transistors and the second input terminal;
A shift register circuit. A shift register circuit according to claim 40 or claim 41,
A first node connected between each connection node between the plurality of third transistors and the first input terminal with the first input terminal side as a cathode and a connection node side between the plurality of third transistors as an anode. A diode,
A second node connected between each connection node between the plurality of fourth transistors and the second input terminal with the second input terminal side as a cathode and a connection node side between the plurality of fourth transistors as an anode. And further comprising a diode
A shift register circuit. A shift register circuit according to claim 40 or claim 41,
A fifth transistor having a control electrode connected to the clock terminal is further provided between each connection node between the plurality of third transistors and the first input terminal.
A sixth transistor having a control electrode connected to the clock terminal is further provided between each connection node between the plurality of fourth transistors and the second input terminal.
A shift register circuit.

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