CN103532493B - A kind of Low-power-consumptiohigh-gain high-gain broadband frequency mixer - Google Patents

Abstract

A low-power high-gain broadband mixer, including the basic structure of a Gilbert mixer, with a radio frequency balun, a local oscillator balun, a transconductance unit, a switch unit, and a parallel connection with a differential inductance and a parasitic capacitance at the load end The RF balun converts the input RF single-ended signal into a differential signal and then outputs it to the transconductance unit. The transconductance unit converts the RF voltage signal into a RF current and inputs it to the switch unit; the local oscillator balun converts the input local After the single-ended signal is converted into a differential signal, it is also output to the switch unit. The switch unit multiplies the input RF differential signal and the input local oscillator differential signal to output an intermediate frequency differential signal and passes through a load containing a differential inductance and a parallel connection with the parasitic capacitance of the load end. The unit output is characterized in that a positive feedback unit, an LCR resonant unit and a buffer unit are added after the switch unit to form a new load unit together with a load unit containing differential inductance and parallel parasitic capacitance, and the buffer unit outputs the final intermediate frequency differential signal .

Description

A Low Power High Gain Wideband Mixer

technical field

The invention relates to a mixer in a millimeter-wave circuit, especially a low-power high-gain broadband mixer, which adopts MOS technology and has great advantages in a millimeter-wave circuit. The design structure is simple, and the mixer can be simultaneously Obtain high gain and broadband characteristics, and can greatly reduce the power consumption of the mixer while having the same gain and bandwidth performance as the traditional mixer.

Background technique

In 1968, BarrieGilbert first proposed the Gilbert double-balanced multiplier structure, and it was widely used in mixers (hereinafter referred to as "Gilbert mixer"). Its circuit block diagram and circuit principle are shown in Figure 1 and Figure 2 respectively. As shown, the radio frequency signal RF converts the single-ended signal into a differential signal RF+ and RF- through the balun, and connects the gates of the common source structure MOS transistors M1 and M2 of the transconductance unit respectively, and the MOS transistors M1 and M2 convert the input radio frequency voltage signal into RF current signal. The local oscillator signal LO converts the single-ended signal into differential signals LO+ and LO- through the balun, which are respectively connected to the gates of the switching unit MOS transistors. The switch unit is composed of MOS tubes M3, M4, M5, and M6. The drains of the MOS tubes of the switch unit output intermediate frequency differential signals. Output the inverted intermediate frequency signal IF-. Due to its own structure, the Gilbert mixer has the following disadvantages:

On the one hand, the gain is low, and the voltage on the load is determined by the product of the intermediate frequency output current and the load resistance. When the intermediate frequency current remains unchanged, the load resistance must be increased to increase the load voltage. If a resistive load is used, the use of a large resistor will inevitably consume too much voltage, which will reduce the voltage margin of the circuit, and due to the influence of parasitic capacitance, it is difficult for a resistive load to obtain high gain at a higher intermediate frequency. If a tuned load is used, the load resistance depends on the quality factor Q of the inductor, and generally the Q of the on-chip inductor is very low, making the gain of the mixer very low.

On the other hand, the bandwidth is narrow. If a resistive load is used, the gain will become a low-pass characteristic due to the influence of parasitic capacitance, and the gain of the higher intermediate frequency frequency is extremely low. Therefore, a tuned load must be used, and the 3dB tuning bandwidth of a typical LC resonant network and resonance The frequency ratio is 10% to 15%, that is to say, the ratio of the IF bandwidth of the mixer to the IF frequency is generally 10% to 15%, which is obviously not enough for applications where the channel bandwidth is relatively large compared to the IF frequency, such as in In a 60GHz millimeter wave system, the IF frequency is 12GHz, and the IF bandwidth of 2.5GHz must be supported, and the ratio of the bandwidth to the carrier frequency reaches 20.8%.

It is worth pointing out that sometimes gain can be exchanged for bandwidth, such as intentionally connecting a resistor in series with an inductor, which can reduce the Q of the inductor so that the bandwidth becomes larger and the gain becomes smaller; sometimes bandwidth can also be exchanged for gain, such as using a positive feedback unit , by introducing a parallel negative resistance on the load to increase the total resistance, so that the gain becomes higher and the bandwidth narrows. However, these methods all sacrifice one of gain and bandwidth in exchange for the improvement of the other, and cannot optimize these two indicators at the same time.

After analyzing the above-mentioned traditional Gilbert mixer, the following conclusions can be drawn: ①One reason for the low gain of the traditional Gilbert mixer is that the load conductance component seen after the intermediate frequency current flows out of the switch unit is relatively large, thus The generated intermediate frequency voltage is small. When using a tuned load, the limited Q value of the inductance unit and the limited output resistance of the switch stage restrict the reduction of the load conductance, which is an important reason why the traditional structure cannot increase the gain; ②The bandwidth of the traditional Gilbert mixer is narrow The main reason is that the susceptance component of the tuned load increases too fast at the frequency that deviates from the resonance point, so that the voltage generated by the current on the load drops rapidly as the frequency deviates from the resonance point; ③ By increasing the Q value of the load inductance, the frequency mixing The gain and bandwidth of the device cannot be improved at the same time, and the gain and bandwidth are often in conflict with each other, and a trade-off is required.

Contents of the invention

The purpose of the present invention is to provide a kind of low power consumption high gain broadband mixer for overcoming the deficiencies in the prior art, and the technical scheme adopted is:

A low-power high-gain broadband mixer, including the basic structure of a Gilbert mixer, is provided with a balun unit, a transconductance unit, a switch unit, and a load unit containing a differential inductance and a parallel connection of a parasitic capacitance at the load end, wherein , the balun unit includes a radio frequency balun and a local oscillator balun. The radio frequency balun converts the input radio frequency single-ended signal into a differential voltage signal V _RF + and V _RF - and then outputs it to the transconductance unit, and the transconductance unit converts the radio frequency voltage signal Converted into radio frequency current signals RF+ and RF- and input to the switch unit; the local oscillator balun converts the input local oscillator single-ended signal into differential voltage signals V _LO + and V _LO - and then outputs to the switch unit, the switch unit will input The radio frequency differential current signals RF+ and RF- are multiplied by the input local oscillator differential voltage signals V _LO + and V _LO - to output the intermediate frequency differential current signals IF+ and IF- and pass through the load unit containing the differential inductance and the parasitic capacitance of the load terminal in parallel Output intermediate frequency differential voltage signals V _OUT+ and V _OUT- , the center tap of the differential inductor is connected to the power supply Vdd, and the two ends of the differential inductor are respectively connected to the intermediate frequency differential current signals IF+ and IF- output by the switch unit, which is characterized in that: after the switch unit, an additional The positive feedback unit, the LCR resonant unit and the buffer unit, together with the load unit containing differential inductance and parallel parasitic capacitance, constitute a new load unit, and the buffer unit outputs the final intermediate frequency differential voltage signals V _IF+ and V _IF- ;

The positive feedback unit is used to reduce the conductance component of the new load unit to increase the gain of the mixer, including NMOS transistors M7, M8 and capacitors C3, C4, the sources of the NMOS transistor M7 and the NMOS transistor M8 are interconnected and grounded, The gates of the NMOS transistor M7 and the NMOS transistor M8 are respectively connected to the bias voltage V _bias1 , the drain of the NMOS transistor M7 is connected to one end of the capacitor C4 and the differential signal output positive terminal IF+ of the switch unit, and the other end of the capacitor C4 is connected to the NMOS transistor M8 The gate of the NMOS transistor M8 is connected to one end of the capacitor C3 and the differential signal output reverse terminal IF- of the switch unit, and the other end of the capacitor C3 is connected to the gate of the NMOS transistor M7;

The said LCR resonant unit is used to reverse the equivalent sign of the susceptance component of the LCR resonant unit and the susceptance component of the new load unit itself, offset the susceptance component of the new load unit, and expand the bandwidth of the mixer, including inductance L1, L2, resistors R1, R2 and capacitors C1, C2, one end of resistor R1 is connected to the drain of NMOS transistor M7 in the positive feedback unit and the connection end of capacitor C4, the other end of resistor R1 is connected in series with capacitor C1 and inductor L1, and then connected to inductor L2 One end, the other end of the inductor L2 is connected in series with the capacitors C2 and R2, and then connected to the drain of the NMOS transistor M8 in the positive feedback unit and the connection end of the capacitor C3;

The buffer unit includes NMOS transistors M9, M10, M11, M12, the drains of the NMOS transistors M11, M12 are interconnected and connected to the power supply Vdd, the gate of the NMOS transistor M11 is connected to the resistor R2 in the LCR resonant unit and the NMOS transistor in the positive feedback unit The drain of M8 is connected to the connection end of capacitor C3, the gate of NMOS transistor M12 is connected to the connection end of resistor R1 in the LCR resonance unit and the drain of NMOS transistor M7 in the positive feedback unit and capacitor C4, and the sources of NMOS transistors M9 and M10 interconnected and grounded, the gates of NMOS transistors M9 and M10 are respectively connected to the bias voltage V _bias2 , the drain of NMOS transistor M9 is interconnected with the source of M11 and serves as the final intermediate frequency differential signal output terminal V _IF+ , the NMOS transistor M10 The drain is interconnected with the source of M12 and serves as the final intermediate frequency differential signal output terminal V _IF- .

Two PMOS transistors M13 and M14 can be added in the switch unit, the sources of the PMOS transistors M13 and M14 are connected to Vdd, the drains and gates of the PMOS transistors M13 and M14 are respectively connected to the drains of the two MOS transistors in the transconductance unit. It is connected with the gate to form a current injection structure to reduce the current flowing through the switching unit.

The balun unit includes on-chip or off-chip baluns, and the types of baluns include lumped element baluns, transformer baluns, and transmission line baluns;

The buffer unit only needs to have a capacitive reactance input component and a low-impedance output, for example, a common-source structure in which the drain is connected to the tuning inductance can be used;

The MOS tubes in the circuit can be replaced by bipolar transistors or MOS tubes and bipolar transistors can be used in combination;

The parasitic capacitance at the load end connected in parallel with the differential inductance in the load unit can be an external capacitor.

Advantage of the present invention and remarkable effect:

(1) In the design of millimeter-wave mixers, how to increase the gain and bandwidth at low power consumption has always been a design problem. In existing designs, the improvement of gain and bandwidth is mainly exchanged for power consumption, and the improvement of gain and bandwidth is often contradictory, and a trade-off is required. The invention adopts a positive feedback unit to reduce the load conductance, and uses an additional series resonant circuit in parallel with the load unit to offset the change of the power-down susceptance part with frequency, so that the gain can be increased and the bandwidth can be expanded at the same time. Compared with the traditional structure, the gain and bandwidth have been greatly improved, as shown in Figure 5.

(2) When using the mixer of the present invention, the power consumption of the mixer can be greatly reduced while obtaining the performance of gain and bandwidth equivalent to that of the traditional structure. For example, when the power consumption is the same, the gain of the mixer with this structure is four times that of the traditional structure, and the bandwidth is twice that of the traditional structure. When the gain and bandwidth are equivalent to the traditional structure, the power consumption can be reduced to eight times that of the traditional structure. one.

Description of drawings

Fig. 1 is the circuit block diagram of traditional Gilbert mixer;

Fig. 2 is the circuit schematic diagram of traditional Gilbert mixer;

Fig. 3 is the circuit block diagram of mixer of the present invention;

Fig. 4 is the circuit schematic diagram of mixer of the present invention;

Figure 5 is a comparison of the frequency conversion gain curves of the traditional Gilbert mixer, the load is added to the positive feedback unit, and the load is simultaneously added to the positive feedback unit and the LCR resonant unit;

FIG. 6 is another implementation circuit of the switching unit in FIG. 4 .

detailed description

Referring to Figure 3, the radio frequency signal is converted into differential voltage signals V _RF + and V _RF - by the radio frequency balun unit 1 and sent to the transconductance unit 2, and the transconductance unit 2 converts the input voltage signal into a current signal and sends it to the switch unit 3 At the same time, the local oscillator signal is converted into differential voltage signals V _LO + and V _LO − by the balun unit 4 and then added to the switch unit 3 to make it switch the radio frequency current at the local oscillator frequency. The switch unit 3 outputs the intermediate frequency differential current signals IF+ and IF- to be connected to the load unit 5 . The above part is exactly the same as the circuit block diagram 1 of the traditional Gilbert structure mixer. The present invention adds a positive feedback unit 6, an LCR resonant unit 7 and a buffer unit 8 after the switch unit 3, and together with the load unit 5 they form a new load unit as the intermediate frequency output load of the switch unit 3, and the intermediate frequency difference before the buffer 8 The positive feedback unit 6 and the LCR resonance unit 7 are connected in parallel on the signal line, and the buffer unit 8 outputs the final intermediate frequency differential voltage signals V _IF+ and V _IF- .

Referring to Figure 4, the single-ended input radio frequency signal V _RF is connected to the single-ended port of the radio frequency balun 1, and the differential output of the radio frequency balun 1 is respectively connected to the gates of M1 and M2 (M1 and M2 constitute the transconductance unit 2), M1 and M2 The source of M1 and M2 is connected to the source of M3, M4, M5, and M6 respectively (M3, M4, M5, and M6 form a switching unit); the single-ended input local oscillator signal V _LO is connected to the local oscillator balun The single-ended port of 2, the differential output of local oscillator balun 2 is respectively connected to the gates of M3, M6, M4, and M5, the drain of M3 is connected to M5 and outputs the positive port intermediate frequency differential signal, and the drain of M4 is connected to M6 And output the reverse port IF differential signal. After the intermediate frequency differential signal is output from the switch unit 3, it is connected to the differential inductance L _diff in the load unit. The center tap of the differential inductance L _diff is connected to the power supply VDD, and is connected in parallel with the differential inductance L _diff . The capacitance connected with a dotted line represents the parasitic capacitance or The sum of the capacitances added intentionally. The above part of the circuit is no different from the traditional Gilbert structure mixer. The present invention adds positive feedback unit 6, LCR resonant unit 7 and buffer unit 8 on the intermediate frequency output load of above-mentioned switch unit 3, differential inductance L _diff and the total capacitance (comprising the parasitic capacitance of switch tube drain, positive feedback The capacitance introduced by the unit and the input capacitance of the buffer unit, etc.) resonate at the IF frequency.

The positive feedback unit 6 is used to reduce the conductance component of the new load unit to increase the gain of the mixer, including NMOS transistors M7, M8 and capacitors C3, C4, the sources of the NMOS transistor M7 and the NMOS transistor M8 are interconnected and grounded, and the NMOS transistor M8 The gates of the transistor M7 and the NMOS transistor M8 are respectively connected to the bias voltage V _bias1 , the drain of the NMOS transistor M7 is connected to one end of the capacitor C4 and the differential signal output positive terminal IF+ of the switch unit, and the other end of the capacitor C4 is connected to the NMOS transistor M8 The gate, the drain of the NMOS transistor M8 is connected to one end of the capacitor C3 and the differential signal output reverse terminal IF- of the switch unit, and the other end of the capacitor C3 is connected to the gate of the NMOS transistor M7;

The LCR resonant unit 7 is used to reverse the equivalent sign of the susceptance component of the LCR resonant unit and the susceptance component of the new load unit itself, offset the susceptance component of the new load unit, and expand the bandwidth of the mixer, including inductance L1, L2 , resistors R1, R2 and capacitors C1, C2, one end of the resistor R1 is connected to the drain of the NMOS transistor M7 in the positive feedback unit and the connection end of the capacitor C4 (that is, the IF+ end), and the other end of the resistor R1 is connected in series with the capacitor C1 and the inductor L1. One end of the inductor L2 is connected, and the other end of the inductor L2 is connected in series with the capacitors C2 and R2, and then connected to the drain of the NMOS transistor M8 in the positive feedback unit and the connection end of the capacitor C3 (ie, the IF-end). Among them, L1, C1 and R1 form a series resonant network, L2, C2 and R2 form a series resonant network, and the resonant frequencies are all intermediate frequency frequencies.

The buffer unit 8 includes NMOS transistors M9, M10, M11, and M12. The drains of the NMOS transistors M11 and M12 are interconnected and connected to the power supply Vdd. The gate of the NMOS transistor M11 is connected to the resistor R2 in the LCR resonance unit and the NMOS transistor M8 in the positive feedback unit. The drain of the NMOS transistor M12 is connected to the connection end of the capacitor C3, the gate of the NMOS transistor M12 is connected to the connection end of the resistor R1 in the LCR resonance unit and the drain of the NMOS transistor M7 in the positive feedback unit and the capacitor C4, and the sources of the NMOS transistors M9 and M10 are connected to each other. connected and grounded, the gates of NMOS transistors M9 and M10 are respectively connected to the bias voltage V _bias2 , the drain of NMOS transistor M9 is interconnected with the source of M11 and serves as the final intermediate frequency differential signal output terminal V _IF+ , and the drain of NMOS transistor M10 The pole is interconnected with the source of M12 and serves as the final intermediate frequency differential signal output terminal V _IF- .

Compared with the traditional Gilbert mixer, the low-power high-gain broadband mixer composed of the above-mentioned circuit of the present invention has increased: ① Positive feedback unit: a cross-coupling structure is used to generate negative conductance, which is different from the positive load of the switch unit itself. The conductance is connected in parallel, thereby reducing the load conductance and increasing the voltage generated by the intermediate frequency current on the load, thereby increasing the gain. But it should also be seen that when only the positive feedback unit is added, the load conductance component is reduced, but the load susceptance component does not change, so when the frequency shifts from the resonant frequency, the susceptance component will divert more current, making the mixing Therefore, the positive feedback can only increase the gain, but cannot improve the overall performance of gain and bandwidth, and there is still a trade-off between gain and bandwidth. ②LCR resonant unit: By connecting an LCR network in parallel to the intermediate frequency load, its resonant frequency is the same as that of the intermediate frequency tuning network, and this network generates a susceptance component at a frequency point away from the resonant frequency. The sign of this susceptance component is the same as that of the intermediate frequency resonant load The susceptance generated by itself counteracts the change of susceptance with frequency offset, making the load appear as a pure conductance in a wide frequency band, which greatly expands the bandwidth of the mixer. Among them, the value of resistance, inductance, and capacitance should be selected appropriately, and the selection of resistance should be moderate. If the resistance value is too small, the gain will be very low, and if it is too large, self-excitation will occur. The inductance and capacitance should be tuned at the intermediate frequency frequency, and they deviate from the resonance frequency The magnitude of the susceptance component generated at the time should be equal to the magnitude of the susceptance generated by the load network itself, so as to completely cancel it. The specific instructions are as follows:

When there is no parallel LCR network, the resistance, inductance and capacitance of the load network are respectively R ₀ , L ₀ and C ₀ , and the resistance, inductance and capacitance of the parallel LCR network are R, L and C respectively. After derivation, the conductance of the new load network is obtained Na:

$\mathrm{<span\; class="notranslate">\; Y</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&Delta;</span>}\mathrm{<span\; class="notranslate">\; \&omega;</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; R</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&Delta;</span>}\mathrm{<span\; class="notranslate">\; \&omega;</span>}\mathrm{<span\; class="notranslate">\; L</span>}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&Delta;</span>}\mathrm{<span\; class="notranslate">\; \&omega;</span>}<span\; class="notranslate">\; \&lsqb;</span>{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; L</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&Delta;</span>}\mathrm{<span\; class="notranslate">\; \&omega;</span>}\mathrm{<span\; class="notranslate">\; L</span>}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; \&rsqb;</span>$

where Δω is the frequency offset from the resonance point _ω0 . It can be seen from the above formula that in order to make the load behave as pure conductance in a wide frequency band, the imaginary part of the above formula should be 0 in a wide range, and L and C must also satisfy the resonance at ω ₀ , thus The values of L and C can be determined. Due to the effect of positive feedback, R ₀ in the above formula is a negative value, if the value of R is too small, when Δω→0, the real part of the above formula will be very large, so the load conductance is large and the gain is very low, but if If the value of R is too large, the absolute value of the second term of the above formula will be smaller than the first term (negative value), then the entire conductance will be negative, resulting in self-excitation, so the value of the resistance R should be selected appropriately.

The advantages of adding positive feedback unit 6 and LCR resonant unit 7 at the same time are: ① The gain and bandwidth are improved at the same time. Compared with the traditional method of increasing gain or expanding bandwidth, this solution controls the conductance and susceptance components of the load at the same time , so that the gain and bandwidth are greatly improved on the basis of the original structure; ②The gain can be adjusted by changing the bias voltage Vbias1 of the positive feedback. In use, the gain of the mixer can be adjusted according to the actual needs. When the noise is the primary constraint factor of the system, the gain of the mixer can be increased to suppress the noise of the subsequent circuit; when the linearity is the primary constraint factor of the system, it can be appropriately reduced The gain of the mixer to relieve the pressure on the linearity of the system. It can be seen from the previous formula that when the parameters of all components are determined, there is still R ₀ that can change with the strength of positive feedback. R ₀ itself is negative. The stronger the positive feedback is, the smaller the absolute value of R ₀ is, so the load conductance is higher. Smaller but higher gain, the strength of positive feedback can be easily adjusted through the bias voltage Vbias1, the higher the Vbias1, the stronger the positive feedback, so changing the bias voltage Vbias1 can play a role in adjusting the gain. In use, the gain of the mixer can be adjusted according to the actual needs. When the noise is the primary constraint factor of the system, the gain of the mixer can be increased to suppress the noise of the subsequent circuit; when the linearity is the primary constraint factor of the system, it can be appropriately reduced The gain of the mixer to relieve the pressure on the linearity of the system.

Fig. 5 compares the change of the mixer gain with the frequency when the traditional Gilbert structure, only the positive feedback unit is added, and the positive feedback unit and the LCR resonant unit are added at the same time. It can be seen that compared with the traditional Gilbert structure, the gain is significantly increased after adding positive feedback, but the bandwidth is also greatly reduced, but after adding positive feedback and LCR resonant network at the same time, the gain and bandwidth are greatly improved at the same time.

FIG. 6 is an embodiment in which PMOS transistors M13 and M14 are added to the circuit of switching power supply 3 in the circuit of FIG. 4 . The sources of M13 and M14 are connected to the power supply Vdd, the drains are respectively connected to the drains of M1 and M2, and the gates are respectively connected to the gates of M1 and M2 to form a current injection structure to reduce the current flowing through the switching unit.

Some units in the present invention can be replaced with other structures, which does not affect the use of the present invention. The balun unit can be implemented on-chip or off-chip, and can be various types of baluns, such as lumped element baluns, transformer baluns, transmission line baluns, etc., or any form of passive or active . The function of the buffer unit is to isolate the circuits before and after the buffer. Since the conductance before and after the buffer is greatly different after using the positive feedback technology, the buffer should have better isolation. In addition to the source-follower structure used in this solution, the buffer can also be selected as a common-source structure, as long as it has a capacitive reactance input component and a low-impedance output, such as a common-source structure in which the drain is connected to a tuning inductor. The MOS tubes used in the circuit can be replaced by bipolar transistors, or MOS tubes and bipolar transistors can be used in combination, and the circuit realizes the same function.

Claims (3)

1. A low-power high-gain broadband mixer, including the basic structure of a Gilbert mixer, with a balun unit, a transconductance unit, a switch unit, and a load unit containing a differential inductance and a parasitic capacitance at the load end in parallel , wherein, the balun unit includes a radio frequency balun and a local oscillator balun, and the radio frequency balun converts the input radio frequency single-ended signal into a differential voltage signal V _RF + and V _RF - and outputs it to the transconductance unit, and the transconductance unit converts the radio frequency The voltage signal is converted into radio frequency current signals RF+ and RF- and input to the switch unit; the local oscillator balun converts the input local oscillator single-ended signal into a differential voltage signal V _LO + and V _LO - and then outputs to the switch unit, the switch unit Multiply the input radio frequency differential current signals RF+ and RF- with the input local oscillator differential voltage signals V _LO + and V _LO - and then output the intermediate frequency differential current signals IF+ and IF-, and pass through the differential inductance and the parasitic capacitance of the load terminal in parallel The load unit outputs intermediate frequency differential voltage signals V _OUT+ and V _OUT- , the center tap of the differential inductor is connected to the power supply Vdd, and the two ends of the differential inductor are respectively connected to the intermediate frequency differential current signals IF+ and IF- output by the switch unit, which is characterized in that: in the switch unit Then add positive feedback unit, LCR resonant unit and buffer unit, together with the load unit containing differential inductance and parallel parasitic capacitance to form a new load unit, the buffer unit outputs the final intermediate frequency differential voltage signal V _IF+ and V _IF- ; The positive feedback unit is used to reduce the conductance component of the new load unit to increase the gain of the mixer, including NMOS transistors M7, M8 and capacitors C3, C4, the sources of the NMOS transistor M7 and the NMOS transistor M8 are interconnected and grounded, The gates of the NMOS transistor M7 and the NMOS transistor M8 are respectively connected to the bias voltage V _bias1 , the drain of the NMOS transistor M7 is connected to one end of the capacitor C4 and the differential signal output positive terminal IF+ of the switch unit, and the other end of the capacitor C4 is connected to the NMOS transistor M8 The gate of the NMOS transistor M8 is connected to one end of the capacitor C3 and the differential signal output reverse terminal IF- of the switch unit, and the other end of the capacitor C3 is connected to the gate of the NMOS transistor M7; The said LCR resonant unit is used to reverse the equivalent sign of the susceptance component of the LCR resonant unit and the susceptance component of the new load unit itself, offset the susceptance component of the new load unit, and expand the bandwidth of the mixer, including inductance L1, L2, resistors R1, R2 and capacitors C1, C2, one end of resistor R1 is connected to the drain of NMOS transistor M7 in the positive feedback unit and the connection end of capacitor C4, the other end of resistor R1 is connected in series with capacitor C1 and inductor L1, and then connected to inductor L2 One end, the other end of the inductor L2 is connected in series with the capacitors C2 and R2, and then connected to the drain of the NMOS transistor M8 in the positive feedback unit and the connection end of the capacitor C3; The buffer unit includes NMOS transistors M9, M10, M11, M12, the drains of the NMOS transistors M11, M12 are interconnected and connected to the power supply Vdd, the gate of the NMOS transistor M11 is connected to the resistor R2 in the LCR resonant unit and the NMOS transistor in the positive feedback unit The drain of M8 is connected to the connection end of capacitor C3, the gate of NMOS transistor M12 is connected to the connection end of resistor R1 in the LCR resonance unit and the drain of NMOS transistor M7 in the positive feedback unit and capacitor C4, and the sources of NMOS transistors M9 and M10 interconnected and grounded, the gates of NMOS transistors M9 and M10 are respectively connected to the bias voltage V _bias2 , the drain of NMOS transistor M9 is interconnected with the source of M11 and serves as the final intermediate frequency differential signal output terminal V _IF+ , the NMOS transistor M10 The drain is interconnected with the source of M12 and serves as the final intermediate frequency differential signal output terminal V _IF- . 2. A kind of low-power high-gain broadband mixer according to claim 1, characterized in that: two PMOS transistors M13 and M14 are added in the said switch unit, and the sources of the PMOS transistors M13 and M14 are connected to the power supply Vdd , the drains and gates of the PMOS transistors M13 and M14 are respectively connected to the drains and gates of the two MOS transistors in the transconductance unit to form a current injection structure to reduce the current flowing through the switching unit. 3. a kind of low power consumption high gain broadband mixer according to claim 1 or 2, is characterized in that: The balun unit includes on-chip or off-chip baluns, and the types of baluns include lumped element baluns, transformer baluns, and transmission line baluns; The buffer unit needs to have a capacitive reactance input component and a low-impedance output, including a common-source structure in which the drain is connected to the tuning inductance; The positive feedback unit, the MOS tube in the buffer unit and the two PMOS tubes added in the switch unit are replaced with bipolar transistors or MOS tubes and bipolar transistors are used in combination; The parasitic capacitance at the load end connected in parallel with the differential inductance in the load unit is an external capacitance.