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WO2021042773A1 - 一种llc谐振变换器及控制方法 - Google Patents

一种llc谐振变换器及控制方法 Download PDF

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Publication number
WO2021042773A1
WO2021042773A1 PCT/CN2020/092918 CN2020092918W WO2021042773A1 WO 2021042773 A1 WO2021042773 A1 WO 2021042773A1 CN 2020092918 W CN2020092918 W CN 2020092918W WO 2021042773 A1 WO2021042773 A1 WO 2021042773A1
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Prior art keywords
switching tube
switch
resonant
inverter circuit
transformer
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Application number
PCT/CN2020/092918
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English (en)
French (fr)
Inventor
余逸群
李思远
王志燊
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广州金升阳科技有限公司
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Publication of WO2021042773A1 publication Critical patent/WO2021042773A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to the technical field of switching converters, in particular to an LLC resonant converter and a control method thereof.
  • LLC resonant converter has many advantages, such as low noise, low stress, and low switching loss.
  • traditional LLC resonant converters generally need to adjust the output voltage by changing the switching frequency.
  • the switching frequency needs to be changed in a wide range, which brings great influence to the design, analysis, and control of the converter. It was extremely difficult.
  • the efficiency of the traditional variable-frequency control LLC resonant converter drops significantly.
  • the traditional control method is to use phase shift control, but the current flowing when the switch tube on the leading and lagging bridge arms of the primary side is turned on is different, making the lagging bridge It is difficult for the arm to achieve soft switching, which is an inherent shortcoming of traditional phase-shifting control.
  • this control mode it is necessary to consider that the lagging bridge arm can still achieve soft switching under the maximum phase shift angle. The larger the phase shift angle, the wider the gain range. Due to the aforementioned concerns, the phase shift angle is limited. Therefore, the conventional fixed frequency phase shift control LLC resonant converter cannot have a wider gain range.
  • the application number is 201810587249.5, the invention application named "Full-bridge resonant DC-DC converter with wide voltage output range and modulation method", hereinafter referred to as background document 2, shows an LLC resonant converter, as shown in Figure 2. It shows that by adding a bidirectional switch in the primary side resonant cavity, building a boost circuit with the primary side bridge arm and resonant inductor, and realizing output voltage stabilization by controlling the bidirectional switch and the phase shift angle of each switch tube of the primary side.
  • the side switches S1 and S4 are turned on and turned off at the same time, and S2 and S3 are turned on and turned off at the same time.
  • the first technical problem to be solved by the present invention is to provide an LLC resonant converter with lower design difficulty.
  • the second technical problem to be solved by the present invention is to provide a control method for the above LLC resonant converter.
  • an LLC resonant converter including an inverter circuit, LLC resonant cavity, transformer and rectifier network connected in sequence from input to output;
  • the LLC resonant cavity includes a resonant inductance Lr, an excitation inductance Lm, and a resonant capacitor Cr, and a two-way switch is additionally provided; the resonant inductor Lr and the resonant capacitor Cr are connected in series between the first output end of the inverter circuit and the first output end of the transformer primary coil.
  • the second output end of the inverter circuit is connected to the second end of the primary coil of the transformer, the magnetizing inductance Lm is connected in parallel with the primary coil of the transformer, and the first end of the two-way switch is connected to the resonant inductor Lr and the resonant capacitor Cr through Are connected to the first end of the primary coil of the transformer, and the second end of the bidirectional switch is connected to the second end of the primary coil of the transformer;
  • the resonant inductor Lr is connected between the first end of the bidirectional switch and the first end of the transformer primary coil
  • the resonant capacitor Cr is connected between the first end of the bidirectional switch and the first output end of the inverter circuit.
  • the two-way switch of the present invention is located on the primary side of the transformer, and there is no need to consider the problem of isolation drive. Moreover, the resonance capacitor Cr is placed before the two-way switch, and the resonance inductance Lr is placed after the two-way switch, which destroys the boost circuit in the background document 2, thereby solving This solves the problem of greater device stress.
  • first end, the first output end, the second end, and the second output end stated above are only codes given for ease of description. If it corresponds to FIG. 3, the first end or the first output end Corresponding to the input/output terminals of the inverter circuit, the bidirectional switch and the primary winding of the transformer respectively, and the second terminal or the second output terminal respectively corresponds to the input/output terminals of the inverter circuit, the bidirectional switch and the primary winding of the transformer.
  • the inverter circuit is a full-bridge inverter circuit composed of four switching tubes S1, S2, S3, and S4, wherein the switching tubes S1 and S2 are respectively used to control the positive pole of the input power source Vin and the first inverter circuit. , Whether the second output terminal is connected or not, the switch tubes S3 and S4 are respectively used to control whether the first and second output terminals of the inverter circuit are connected with the negative electrode of the input power source Vin.
  • the inverter circuit may also be a half-bridge inverter circuit.
  • the number of switch tubes on the primary side is reduced from 6 to 4, which saves the number of components and is suitable for small and medium power occasions.
  • the bidirectional switch is composed of two switch tubes S5 and S6 connected in reverse series, wherein the parasitic diode points to the connection end of the bidirectional switch and the resonant inductor Lr and the resonant capacitor Cr is the switch tube S5.
  • the rectification network adopts a full-wave rectification structure, such as a full-bridge rectification structure, and the full-bridge rectification structure is composed of rectifier diodes or switch tubes.
  • the second technical problem of the present invention is solved by the following solution: a control method of the above LLC resonant converter, characterized in that the LLC resonant converter adopts fixed frequency PWM (short for Pulse Width Modulation) control.
  • PWM pulse for Pulse Width Modulation
  • the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, the switching tube S1 and the switching tube S5 are complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, and the switching tube S1 and the switching tube S4 are turned on and turned off at the same time ,
  • the switching tube S2 and the switching tube S3 are turned on and turned off at the same time, the duty cycle of the switching tube S1 is equal to that of the switching tube S2, and both are not greater than 0.5 and the phase difference between the two is 180°.
  • the switching tube S5 accounts for The duty cycle is equal to the duty cycle of the switch tube S6, both are not less than 0.5 and the phase difference between the two is 180°.
  • the output voltage is controlled by adjusting the duty cycle of the switch tube S1. The greater the duty cycle of the switch tube S1, The greater the output voltage gain.
  • the present invention has the following beneficial effects:
  • the two-way switch of the present invention is located on the primary side of the transformer, and there is no need to consider the isolation drive problem, which reduces the difficulty of circuit drive design; compared with background literature 2, the present invention changes the resonant inductor Lr and the resonant capacitor in Figure 2
  • the position of Cr turns the original step-up circuit into a step-down circuit, which solves the problem of greater device stress.
  • the change trend (gradual change, non-jumping) of the output current I 0 in Figure 4 can also be seen.
  • the LLC resonant converter with the structure of the present invention reduces the design difficulty of the LLC resonant converter, and can adopt constant frequency PWM control.
  • the LLC resonant converter with the structure of the present invention when the bidirectional switch is turned on, the energy of the resonant current is stored in the loop composed of the transformer magnetizing inductance Lm, the resonant inductance Lr and the bidirectional switch during the circulating phase, and does not flow through the resonant capacitor Cr , Due to the parasitic resistance of the resonant capacitor Cr, this is beneficial to reduce the energy loss on the parasitic resistance of the resonant capacitor Cr, so compared to the reference document 2, the circuit structure of the present invention can improve the working efficiency;
  • the present invention has excellent comprehensive performance: the present invention adopts fixed frequency PWM control, the frequency conversion range is small, the requirements for magnetic components such as transformers and inductors are low, there are no leading and lagging bridge arms, and the voltage gain range is wide and efficiency High, low component stress, low circuit control and drive design difficulty;
  • the present invention can achieve a wider gain range than the background documents 1 and 2, because the fixed frequency phase shift control cannot be inverted with the half bridge
  • the gain range of the half-bridge inverter circuit is half that of the full bridge, so the circuit of the present invention can achieve a wider gain range.
  • Fig. 1 is a schematic diagram of the LLC resonant converter shown in Background Document 1;
  • FIG. 2 is a schematic diagram of the LLC resonant converter shown in background document 2;
  • Fig. 3 is a schematic diagram of an LLC resonant converter according to a preferred embodiment of the present invention.
  • Fig. 4 is the main working waveform of the LLC resonant converter of the preferred embodiment of the present invention when the fixed frequency control is adopted;
  • 5 to 10 are equivalent circuit diagrams of each switch mode when the LLC resonant converter of the preferred embodiment of the present invention adopts constant frequency control.
  • the LLC resonant converter of this embodiment includes an inverter circuit 10, an LLC resonant cavity 20, a transformer T, and a rectifier network 30 that are sequentially connected from input to output.
  • Vin is the input power of the converter
  • Ro is the output load R 0 of the converter.
  • the inverter circuit 10 is a full-bridge inverter circuit composed of a switching tube S1, a switching tube S2, a switching tube S3, and a switching tube S4.
  • the LLC resonant cavity 20 includes a resonant inductance Lr, an excitation inductance Lm and a resonant capacitor Cr, and is additionally provided with a bidirectional switch composed of a switch tube S5 and a switch tube S6.
  • the rectifier network 30 is composed of a full-bridge rectifier circuit composed of four diodes D1-D4 in parallel with an output filter capacitor C 0 .
  • the drain of the switch S1 is connected to the drain of the switch S2 and the positive terminal of the input power Vin
  • the source of the switch S1 is connected to the drain of the switch S3 and one end of the resonant capacitor Cr
  • the other end of the resonant capacitor Cr is connected
  • One end of the resonant inductor Lr and the drain of the switch S5 the other end of the resonant inductor Lr is connected to one end of the magnetizing inductance Lm and the first end of the transformer T primary winding Np, and the second end of the transformer T primary winding Np is connected
  • the source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power supply Vin.
  • the source of the tube S6 is connected to the source of the switching tube S5; the first end of the secondary winding Ns of the transformer T is connected to the anode of the secondary rectifier diode D1 and the cathode of the secondary rectifier diode D3, and the cathode of the secondary rectifier diode D1 Connected to the cathode of the secondary side rectifier diode D2, one end of the secondary side output filter capacitor Co and one end of the output load Ro, the other end of the output load Ro is connected to the other end of the secondary side output filter capacitor Co, and the secondary side rectifier diode D3
  • the anode of the secondary rectifier diode D4 and the anode of the secondary rectifier diode D4, the cathode of the secondary rectifier diode D4 is connected to the anode of the secondary rectifier diode D2 and the second end
  • the first ends of the transformer primary winding and the secondary winding have the same name each other, and the second ends of the transformer primary winding and the secondary winding have the same name each other.
  • the above LLC resonant converter can adopt the following fixed-frequency PWM control method: the switching frequencies of the switching tubes S1 to S6 are equal and fixed, the switching tubes S1 and S5 are complementarily turned on, and the switching tubes S2 and S6 are complementarily turned on.
  • the switching tube S1 and the switching tube S4 are turned on and off at the same time, the switching tube S2 and the switching tube S3 are turned on and off at the same time, the duty cycle of the switching tube S1 is equal to the duty cycle of the switching tube S2, and both are not greater than 0.5 and The phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is achieved by adjusting the duty cycle of the switching tube S1 For the control, the greater the duty cycle of the switch S1, the greater the output voltage gain.
  • a reasonable dead time must be set between the switching signals of the switching tube S1 and the switching tube S5 to realize the soft switching of the switching tube S1, the switching tube S4, and the switching tube S5; the switching of the switching tube S2 and the switching tube S6
  • a reasonable dead time must be set between the signals to realize the soft switching of the switching tube S2, the switching tube S3, and the switching tube S6.
  • Coss1 to Coss6 respectively represent the output capacitances to the sixth switching tubes S1 to S6.
  • the turns ratio of the primary and secondary sides of the transformer is 2:1.
  • Figure 4 shows the main operating waveforms of this resonant converter when fixed-frequency PWM control is used.
  • Vgs1/4 is the driving signal of the switching tubes S1 and S4
  • Vgs2/3 is the driving signal of the switching tubes S2 and S3
  • Vgs5 is the switching tube.
  • the driving signal of S5, Vgs6 is the driving signal of the switch tube S6, and Vc, iLr, iLm, and i 0 respectively represent the voltage across Cr, the current through Lr, the current through Lm, and the current through resistor R 0 .
  • the LLC resonant converter has six switching modes in a half cycle, as shown in Figure 5-10 (the working modes of the second half cycle and the first half cycle of the LLC resonant converter are symmetrical. It can also be seen from the waveform diagram. Generally, The description of LLC resonant converter only describes half a cycle).
  • Switch mode 1 [t 0 , t 1 ]: As shown in Figure 5, before time t 0 , the switch S6 has been turned on, the switch S5 is turned off, and its body diode bears a reverse voltage and reversely stops; At t0, the switching tube S1 and the switching tube S4 are turned on at zero voltage; the secondary side rectifier diode D1 and the secondary side rectifier diode D4 are turned on, and the current flowing through the diode is proportional to the difference between the resonance current and the excitation current; both ends of the excitation inductance Lm
  • the output voltage is clamped to nVO (n is the transformer turns ratio); the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is negative, and the magnetizing inductor current iLm increases linearly, but is less than the resonant current iLr;
  • Switching mode 2 [t 1 , t 2 ]: As shown in Figure 6, at t 1 , the resonance current iLr crosses the zero point; the secondary side rectifier diode D1 and the secondary side rectifier diode D4 continue to conduct; both ends of the magnetizing inductance Lm The voltage is output clamped to nV O ; the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is positive, and the magnetizing inductor current iLm increases linearly, but is less than the resonant current iLr;
  • Switch mode 3 [t 2 , t 3 ]: As shown in Figure 7, at t 2 the switch S1 and the switch S4 are turned off, the resonance current iLr is still greater than the excitation inductance current iLm, the secondary side rectifier diode D1 and the secondary side rectifying diode D4 is still turned on; resonant current iLr to switch Sl, the switch output capacitance C oss1 S4 of, C oss4 charge, to the switch S2, the switch S3 output capacitor C oss2, C oss3 discharge, to switch S5 discharge of the output capacitor C oss5; C oss5 when the voltage across the capacitor drops to zero, the body diode of the switch S5 is turned tube, switch S5 is implemented to provide zero-voltage conditions.
  • Switch Mode 4 [t 3, t 4] : As shown in the drawings, the switch 8 S5 ZVS at time t 3, the switch S6 and the secondary side rectifying diode D1, a secondary rectifier diode D4 is still turned on; Excitation The inductance Lm is still clamped by the output voltage, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;
  • Switching mode 5 [t 4 , t 5 ]: As shown in Figure 9, at t 4 , the resonant current iLr is equal to the excitation current iLm, the current flowing through the secondary rectifier diode D1 naturally crosses 0, and the secondary side is rectified The diode D1 and the secondary side rectifier diode D4 are turned off at zero current to avoid the diode reverse recovery problem; the switch tube S5 and the switch tube S6 continue to conduct, and the excitation current and the resonance current iLr are equal and remain unchanged;
  • Switching mode 6 [t 5 , t 6 ]: As shown in Figure 10, at t 5 , the switch S6 is turned off while the switch S5 continues to conduct; the resonant current iLr is equal to the excitation current iLm, and the secondary rectifier diode is still in reverse off-state; resonant current iLr to switch Sl, switch the output capacitor C oss1 S4, C oss4 charge, to the switch S2, the switch of the output capacitor C oss2 S3, C oss3 discharge, to the output of switch S6 The capacitor C oss6 is charged; when the voltage across the capacitors C oss2 and C oss3 drops to 0, the body diode of the switching tube S2 and the switching tube S3 is turned on, providing conditions for the switching tube S2 and the switching tube S3 to achieve zero voltage turn-on; t6 At the moment, the switch tube S2 and the switch tube S3 realize ZVS, and the circuit enters the second
  • all switching devices of the converter can achieve zero voltage turn-on, and the rectifier devices on the secondary side can also achieve zero current turn-off. There is no diode reverse recovery problem, and all switching devices are Can realize soft switching.
  • the two-way switch of the present invention is located on the primary side of the transformer, without considering the problem of isolation driving, and reducing the difficulty of circuit driving design.
  • the present invention changes the positions of the resonant inductor Lr and the resonant capacitor Cr in Fig. 2, so that the original booster circuit becomes a bucker circuit, which solves the problem of greater device stress. Therefore, overall, the LLC resonant converter with the structure of the present invention reduces the design difficulty of the LLC resonant converter.
  • the energy of the resonant current is stored in the loop composed of the transformer magnetizing inductance Lm, the resonant inductance Lr and the bidirectional switch during the circulating phase, and will not flow through the resonant capacitor Cr.
  • the resonant capacitor Cr has a parasitic resistance, which is beneficial to reduce the energy loss on the parasitic resistance of the resonant capacitor Cr.
  • the circuit structure of the present invention can further improve the working efficiency of the circuit.
  • the invention realizes output voltage stabilization by controlling the duty cycle, realizing fixed-frequency PWM control, facilitating the design of magnetic components such as transformers, reducing the difficulty of circuit drive design and device stress, and realizing the soft switching of all switching devices, and there is no Leading bridge arm and lagging bridge arm, wide voltage gain range, high efficiency, high power density, the inverter circuit can use either full bridge or half bridge, which can achieve a wider voltage gain range than fixed frequency phase shift, satisfying wide The needs of voltage gain range conversion occasions. In short, the converter of the present invention has excellent overall performance.
  • the switch tubes S5 and S6 of the present invention can also be connected in reverse series with common drain (the respective control timings remain unchanged), and the rectifier network Other full-wave rectifier circuits can also be used, and the above rectifier diodes can also be replaced with switching tubes, etc.
  • any modification or equivalent made without departing from the principle of the present invention Replacement, improvement, etc. should all be included in the protection scope of the present invention.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

本发明公开了一种LLC谐振变换器及控制方法，所述LLC谐振变换器设计难度较低，包括从输入到输出依次连接的逆变电路、LLC谐振腔、变压器和整流网络；所述LLC谐振腔包括谐振电感Lr、励磁电感Lm和谐振电容Cr，还增设有双向开关；所述谐振电感Lr、谐振电容Cr串联在逆变电路的第一输出端与变压器原边线圈的第一端之间，逆变电路的第二输出端与变压器原边线圈的第二端连接，励磁电感Lm与变压器原边线圈并联，双向开关的第一端通过连在谐振电感Lr、谐振电容Cr之间与变压器原边线圈的第一端连接，双向开关的第二端与变压器原边线圈的第二端连接；谐振电感Lr连于双向开关的第一端与变压器原边线圈的第一端之间，谐振电容Cr连于双向开关的第一端与逆变电路的第一输出端之间。

Description

一种LLC谐振变换器及控制方法 技术领域
本发明涉及开关变换器技术领域，具体涉及一种LLC谐振变换器及其控制方法。
背景技术
随着电力电子领域迅猛发展，开关变换器应用越来越广泛。人们对开关变换器提出更多要求：高功率密度、高可靠性和小体积。LLC谐振变换器作为一种谐振变换器，具有诸多优点，比如低噪声、低应力、开关损耗低等。然而传统LLC谐振变换器一般需要通过改变开关频率实现输出电压的调节，当负载或输入电压波动时，开关频率需要在很宽的范围内变化，这给变换器的设计、分析、控制都带来了极大的困难。当电压增益较宽时，传统变频控制LLC谐振变换器的效率明显下降。
对于定频控制的LLC谐振变换器，传统的控制方法是采用移相控制，但由于其原边的超前桥臂和滞后桥臂上的开关管开通的时候流过的电流不一样，使得滞后桥臂较难实现软开关，这是传统移相控制的固有缺点。在这种控制方式下，就需要考虑最大移相角下滞后桥臂仍能实现软开关。移相角越大，增益范围越宽，由于有前述顾虑，所以移相角受到限制，因此，传统定频移相控制的LLC谐振变换器的增益范围无法做宽。
为了解决传统定频移相控制LLC谐振变换器存在的滞后桥臂难以实现软开关的问题，申请号为201410777221.X，名为《一种谐振变换器及其控制方法》的发明申请，以下简称背景文献1，示出了一种可以实现定频控制的LLC谐振变换器的结构，如图1所示，该种LLC谐振变换器在副边增加双向开关，通过控制原副边开关管的移相角来实现输出稳压，由于其原边开关S1、S4同时开通、同时关断，S2、S3同时开通、同时关断，所以其不存在超前桥臂和滞后桥臂，所以其移相角可以做得较大，相应的增益范围就可以做的较宽。但由于其在副边的双向开关需要隔离驱动，这给电路的驱动控制设计带来了较大的不便。
申请号为201810587249.5，名为《具有宽电压输出范围的全桥谐振DC-DC变换器及调制方法》的发明申请，以下简称背景文献2，示出了一种LLC谐振 变换器，如图2所示，其通过在原边谐振腔内增加双向开关，与原边桥臂与谐振电感构建一个升压电路，通过控制双向开关和原边各开关管的移相角来实现输出稳压，由于其原边开关S1、S4同时开通、同时关断，S2、S3同时开通、同时关断，所以其不存在超前桥臂和滞后桥臂，所以其移相角可以做得较大，相应的增益范围就可以做的较宽。但由于存在升压电路，其器件应力会较大，这也给变换器的设计带来一些不便。
发明内容
本发明所要解决的第一个技术问题是，提供一种设计难度较低的LLC谐振变换器。
本发明所要解决的第二个技术问题是，提供一种针对上述LLC谐振变换器的控制方法。
本发明第一个技术问题通过如下方案解决：一种LLC谐振变换器，包括从输入到输出依次连接的逆变电路、LLC谐振腔、变压器和整流网络；
所述LLC谐振腔包括谐振电感Lr、励磁电感Lm和谐振电容Cr，还增设有双向开关；所述谐振电感Lr、谐振电容Cr串联在逆变电路的第一输出端与变压器原边线圈的第一端之间，逆变电路的第二输出端与变压器原边线圈的第二端连接，励磁电感Lm与变压器原边线圈并联，双向开关的第一端通过连在谐振电感Lr、谐振电容Cr之间与变压器原边线圈的第一端连接，双向开关的第二端与变压器原边线圈的第二端连接；
其特征在于，谐振电感Lr连于双向开关的第一端与变压器原边线圈的第一端之间，谐振电容Cr连于双向开关的第一端与逆变电路的第一输出端之间。
本发明双向开关位于变压器原边，无需考虑隔离驱动问题，而且将谐振电容Cr放到了双向开关前，将谐振电感Lr放在了双向开关后，破坏了背景文献2中的升压电路，从而解决了器件应力较大的问题。
需要说明的是，上面陈述的第一端、第一输出端以及第二端、第二输出端只是为了便于描述而给予的代号，若对应到图3中，则第一端或第一输出端分别对应逆变电路、双向开关和变压器原边绕组在上的输入/输出端，第二端或第二输出端分别对应逆变电路、双向开关和变压器原边绕组在下的输入/输出端。
所述逆变电路为由四个开关管S1、S2、S3、S4组成的全桥逆变电路，其中，开关管S1、S2分别用于控制输入电源Vin正极与所述逆变电路的第一、第二输出端连通与否，开关管S3、S4分别用于控制所述逆变电路的第一、第二输出端与输入电源Vin负极连通与否。
所述逆变电路也可以是半桥逆变电路。当原边采用半桥逆变电路时，原边的开关管数量由6个减为4个，节省了器件数量，宜适用在中小功率场合。
所述双向开关由两个反向串联的开关管S5、S6构成，其中，寄生二极管指向所述双向开关与所述谐振电感Lr、谐振电容Cr连接端的为开关管S5。
所述整流网络采用全波整流结构，如全桥整流结构，所述全桥整流结构由整流二极管或开关管构成。
本发明第二个技术问题通过如下方案解决：一种上述LLC谐振变换器的控制方法，其特征在于，所述LLC谐振变换器采用定频PWM(Pulse Width Modulation的简称)控制。
具体：开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小来实现输出电压的控制，开关管S1占空比越大，输出电压增益越大。
相比于现有技术，本发明具有如下有益效果：
1)相比背景文献1，本发明双向开关位于变压器原边，无需考虑隔离驱动问题，降低了电路驱动设计的难度；相比背景文献2，本发明改变了图2中谐振电感Lr、谐振电容Cr的位置，使原本的升压电路变成了降压电路，解决了器件应力较大的问题，从图4中输出电流I 0的变化趋势(逐渐变化、非跳变)也可以看出。总之，本发明结构的LLC谐振变换器降低了LLC谐振变换器的设计难度，而且可以采用定频PWM控制。
2)本发明结构的LLC谐振变换器，在双向开关导通时，谐振电流在环流阶段能量存储在由变压器励磁电感Lm、谐振电感Lr及双向开关组成的回路中， 不会流经谐振电容Cr，由于谐振电容Cr存在寄生电阻，这样有利于降低能量在谐振电容Cr的寄生电阻上的损耗，所以相比于对比文件2，本发明电路结构可提升工作效率；
3)本发明具有优秀的综合性能：本发明采用定频PWM控制，频率变换范围小，对变压器、电感等磁性元件的要求低，不存在超前桥臂和滞后桥臂，电压增益范围宽、效率高，器件应力小，电路控制及驱动设计难度低；另外，在增益范围方面，本发明能实现相比背景文献1、2更宽的增益范围，由于定频移相控制不能与半桥逆变电路结合，采用半桥逆变电路增益范围是全桥的一半，所以本发明电路能做到更宽的增益范围。
附图说明
图1为背景文献1示出的LLC谐振变换器的原理图；
图2为背景文献2示出的LLC谐振变换器的原理图；
图3为本发明较佳实施例的LLC谐振变换器的原理图；
图4为本发明较佳实施例的LLC谐振变换器采用定频控制时的主要工作波形；
图5～10为本发明较佳实施例的LLC谐振变换器采用定频控制时各开关模态的等效电路图。
具体实施方式
如图3所示，本实施例的LLC谐振变换器，包括从输入到输出依次连接的逆变电路10、LLC谐振腔20、变压器T和整流网络30。图中Vin为变换器的输入电源，Ro为变换器的输出负载R 0。
逆变电路10为由开关管S1、开关管S2、开关管S3、开关管S4组成的全桥逆变电路。LLC谐振腔20包括谐振电感Lr、励磁电感Lm和谐振电容Cr，还增设有由开关管S5、开关管S6构成的双向开关。整流网络30由4个二极管D1-D4构成的全桥整流电路并联输出滤波电容C 0构成。
开关管S1的漏极连于开关管S2的漏极和输入电源Vin的正端，开关管S1的源极连于开关管S3的漏极和谐振电容Cr的一端，谐振电容Cr的另一端连于谐振电感Lr的一端和开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器T原边绕组Np的第一端，变压器T原边绕组Np的第二端连 接励磁电感Lm的另一端、开关管S2的源极、开关管S4的漏极、开关管S6的漏极，开关管S4的源极连于开关管S3的源极和输入电源Vin的负极，开关管S6的源极连于开关管S5的源极；变压器T的副边绕组Ns的第一端连于副边整流二极管D1的阳极和副边整流二极管D3的阴极，副边整流二极管D1的阴极连于副边整流二极管D2的阴极、副边输出滤波电容Co的一端和输出负载Ro的一端，输出负载Ro的另一端连于副边输出滤波电容Co的另一端、副边整流二级管D3的阳极和副边整流二级管D4的阳极，副边整流二级管D4的阴极连于副边整流二级管D2的阳极和变压器T的副边绕组Ns的第二端。
变压器原边绕组与副边绕组的第一端互为同名端，变压器原边绕组与副边绕组的第二端互为同名端。
上述LLC谐振变换器可以采取如下定频PWM控制方法：开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小来实现输出电压的控制，开关管S1占空比越大，输出电压增益越大。
在具体实施时，开关管S1和开关管S5的开关信号之间必须设置合理的死区时间以实现开关管S1、开关管S4、开关管S5的软开关；开关管S2和开关管S6的开关信号之间必须设置合理的死区时间以实现开关管S2、开关管S3、开关管S6的软开关。Coss1～Coss6分别表示至第六开关管S1～S6的输出电容。
下面结合图3，具体说明下LLC谐振变换器采用定频PWM控制时的工作过程。
在本实施例中，参数选取如下：Lr＝220nH，Lm＝800nH，Cr＝82nF，输入电压范围为36～75VDC。变压器原副边匝比为2:1。图4为这种谐振变换器采用定频PWM控制时的主要工作波形图，Vgs1/4为开关管S1、S4的驱动信号，Vgs2/3为开关管S2、S3的驱动信号，Vgs5为开关管S5的驱动信号，Vgs6为开关管S6的驱动信号，Vc、iLr、iLm、i 0分别表示Cr两端电压、通过Lr的电流、通过Lm的电流和通过电阻R 0的电流。从图4可以看出，本发明输出电流I 0变化 平缓，器件应力小。LLC谐振变换器在半个周期内共有六种开关模态，分别如图5-10所示(LLC谐振变换器后半个周期与前半个周期工作模态对称从波形图也可以看出，一般对LLC谐振变换器的描述只描述半个周期即可)。
开关模态1[t 0，t 1]：如附图5所示，在t 0时刻前，开关管S6已导通，开关管S5关断，其体二极管承受反向电压而反向截至；t0时刻，开关管S1、开关管S4零电压开通；副边整流二极管D1、副边整流二极管D4导通，流经二极管的电流与谐振电流和励磁电流的差值成正比；励磁电感Lm两端电压被输出钳位至nVO(n为变压器匝比)；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为负值，励磁电感电流iLm线性增加，但小于谐振电流iLr；
开关模态2[t 1，t 2]：如附图6所示，在t 1时刻，谐振电流iLr过零点；副边整流二极管D1、副边整流二极管D4继续导通；励磁电感Lm两端电压被输出钳位至nV O；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为正值，励磁电感电流iLm线性增加，但小于谐振电流iLr；
开关模态3[t 2，t 3]：如附图7所示，在t 2时刻开关管S1、开关管S4关断，谐振电流iLr仍大于励磁电感电流iLm，副边整流二极管D1、副边整流二极管D4继续导通；谐振电流iLr给开关管S1、开关管S4的输出电容C oss1、C oss4充电，给开关管S2、开关管S3输出电容C oss2、C oss3放电，给开关管S5的输出电容C oss5放电；当电容C oss5两端电压降到零时，开关管S5的体二极管管导通，为开关管S5实现零电压开通提供条件。
开关模态4[t 3，t 4]：如附图8所示，在t 3时刻开关管S5零电压开通，开关管S6和副边整流二极管D1、副边整流二极管D4继续导通；励磁电感Lm依然被输出电压钳位，励磁电流iLm继续线性增加，谐振电流iLr线性下降；
开关模态5[t 4，t 5]：如附图9所示，在t 4时刻，谐振电流iLr等于励磁电流iLm，流过副边整流二极管D1的电流自然过0，副边副边整流二极管D1、副边整流二极管D4零电流关断，避免二极管反向恢复问题；开关管S5和开关管S6继续导通，励磁电流和谐振电流iLr相等且保持不变；
开关模态6[t 5，t 6]：如附图10所示，t 5时刻，开关管S6关断而开关管S5继续导通；谐振电流iLr等于励磁电流iLm，副边整流二极管仍处于反向截止 状态；谐振电流iLr给开关管S1、开关管S4的输出电容C oss1、C oss4充电，给开关管S2、开关管S3的输出电容C oss2、C oss3放电，给开关管S6的输出电容C oss6充电；当电容C oss2、C oss3两端电压降到0时，开关管S2、开关管S3的体二极管管导通，为开关管S2、开关管S3实现零电压开通提供条件；t6时刻，开关管S2、开关管S3实现ZVS，电路进入后半个周期。
根据上述变换器的工作过程描述可知，该变换器各开关器件均可以实现零电压开通，副边的整流器件也都可以实现零电流关断，不存在二极管反向恢复的问题，所有开关器件都能实现软开关。
本发明双向开关位于变压器原边，无需考虑隔离驱动问题，降低了电路驱动设计的难度。本发明改变了图2中谐振电感Lr、谐振电容Cr的位置，使原本的升压电路变成了降压电路，解决了器件应力较大的问题。所以，整体而言，本发明结构的LLC谐振变换器降低了LLC谐振变换器的设计难度。
本发明结构的LLC谐振变换器，在双向开关导通时，谐振电流在环流阶段能量存储在由变压器励磁电感Lm、谐振电感Lr及双向开关组成的回路中，不会流经谐振电容Cr，由于谐振电容Cr存在寄生电阻，这样有利于降低能量在谐振电容Cr的寄生电阻上的损耗，本发明电路结构可可进一步提升电路的工作效率。
本发明通过控制占空比来实现输出稳压，实现定频PWM控制，便于变压器等磁性元件的设计，还降低了电路驱动设计难度和器件应力，实现了所有开关器件的软开关，而且不存在超前桥臂和滞后桥臂，电压增益范围宽、效率、功率密度高，逆变电路既可以采用全桥也可以采用半桥，可实现比定频移相更为宽的电压增益范围，满足宽电压增益范围变换场合的需要。总之，本发明变换器具有优秀的综合性能。
以上实施例的说明只是用于帮助理解本申请的发明构思，并不用以限制本发明，如本发明开关管S5、S6还可以采用共漏极反向串联(各自控制时序不变)，整流网络还可以采用其他全波整流电路，上面整流二极管也可以替换为开关管等，总之，对于本技术领域的普通技术人员来说，凡在不脱离本发明原理的前提下，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (9)

  1. 一种LLC谐振变换器，包括从输入到输出依次连接的逆变电路、LLC谐振腔、变压器和整流网络；
    所述LLC谐振腔包括谐振电感Lr、励磁电感Lm和谐振电容Cr，还增设有双向开关；所述谐振电感Lr、谐振电容Cr串联在逆变电路的第一输出端与变压器原边线圈的第一端之间，逆变电路的第二输出端与变压器原边线圈的第二端连接，励磁电感Lm与变压器原边线圈并联，双向开关的第一端通过连在谐振电感Lr、谐振电容Cr之间与变压器原边线圈的第一端连接，双向开关的第二端与变压器原边线圈的第二端连接；
    其特征在于，谐振电感Lr连于双向开关的第一端与变压器原边线圈的第一端之间，谐振电容Cr连于双向开关的第一端与逆变电路的第一输出端之间。
  2. 根据权利要求1所述的LLC谐振变换器，其特征在于，所述逆变电路是半桥逆变电路。
  3. 根据权利要求1所述的LLC谐振变换器，其特征在于，所述逆变电路为由四个开关管S1、S2、S3、S4组成的全桥逆变电路，其中，开关管S1、S2分别用于控制输入电源Vin正极与所述逆变电路的第一、第二输出端连通与否，开关管S3、S4分别用于控制所述逆变电路的第一、第二输出端与输入电源Vin负极连通与否。
  4. 根据权利要求3所述的LLC谐振变换器，其特征在于，所述双向开关由两个反向串联的开关管S5、S6构成，其中，寄生二极管指向所述双向开关与所述谐振电感Lr、谐振电容Cr连接端的为开关管S5。
  5. 根据权利要求4所述的LLC谐振变换器，其特征在于，所述整流网络采用全波整流结构。
  6. 根据权利要求5所述的LLC谐振变换器，其特征在于，所述整流网络采用全桥整流结构，所述全桥整流结构由整流二极管或开关管构成。
  7. 一种权利要求1-6任一权利要求所述LLC谐振变换器的控制方法，其特征在于，所述LLC谐振变换器采用定频PWM控制。
  8. 一种权利要求4-6任一权利要求所述LLC谐振变换器的控制方法，其特征在于，所述LLC谐振变换器采用定频PWM控制。
  9. 根据权利要求8所述的控制方法，其特征在于，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小来实现输出电压的控制，开关管S1占空比越大，输出电压增益越大。
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