JP7563190B2 - Bidirectional isolated DC-DC converter and control method - Google Patents

Description

The present invention relates to a bidirectional isolated DC-DC converter and a control method, and relates to technology that can contribute to the conversion efficiency of, for example, a bidirectional isolated DC-DC converter.

One example of a DC power supply device that is used in various fields (for example, in industrial equipment such as battery simulators) is a configuration in which power is transmitted (power is transmitted in both directions) between two DC power sources (first and second DC power sources 1 and 2 in FIG. 1 described below) using a bidirectional isolated DC-DC converter (hereinafter simply referred to as a bidirectional converter).

In power transmission, for example, the primary side DC power of a DC power supply device is converted to AC power using a switching element or the like and sent to the primary side of a transformer, and the AC power transmitted to the secondary side of the transformer is rectified using a switching element and converted back to DC power.

Since the volume of a transformer depends on the frequency of the applied voltage, it is possible to make it smaller if the frequency is high. However, as mentioned above, the switching loss that can occur when converting power using a switching element is roughly proportional to the frequency. For this reason, if a bidirectional converter is simply configured to have a high frequency, there is a risk that the losses in the bidirectional converter will increase.

In recent years, bidirectional converters using the Dual Active Bridge method have been researched as a configuration that reduces the switching loss per switching and achieves both miniaturization and low loss in the bidirectional converter (see, for example, Non-Patent Document 1).

This Dual Active Bridge bidirectional converter is equipped with a coupling circuit (hereinafter simply referred to as a DAB circuit) in which a pair of converter parts (first and second converter parts 4 and 10 in FIG. 1 described later) capable of bidirectional voltage conversion are coupled by a transformer or the like. The DAB circuit is configured so that the desired power transmission can be performed between the converter parts by outputting gate signals to the switching elements of each converter part for appropriate switching control. There are also configurations in which a capacitor is connected in parallel to the switching elements, and a reactor is connected in series to the transformer.

In addition, there is also a configuration that utilizes the resonance phenomenon caused by a capacitor and a reactor during the dead time of the switching element to reduce the applied voltage to zero and achieve zero voltage switching, making it possible to reduce the turn-on loss of the switching element to zero.

Incidentally, in the control configuration of the bidirectional converter, for example, there is a configuration in which the DC voltage of the DC output section (such as the second smoothing capacitor 11 in FIG. 1 described later) on the secondary side of the DAB circuit is detected and the voltage is controlled based on the detection result. In the case of this voltage control, the bidirectional converter is operated so that the DC voltage can be output according to the voltage command from the outside.

However, the transformers and switching elements that are components of bidirectional converters have limited allowable currents, so in the case of a control configuration that simply controls voltage as described above, there is a possibility that the components may burn out or be destroyed due to, for example, an inrush current at startup or an increase in current during a sudden load change.

One way to prevent this is to use a control configuration that applies current control. For example, Patent Document 1 proposes a technology that detects the current (load current) of the DC output section on the secondary side of the DAB circuit and controls the current based on the detection results.

In power transmission using a bidirectional converter (hereinafter simply referred to as conventional converter J1) as shown in Non-Patent Document 1 and Patent Document 1, for example, when a voltage is applied to the primary side of a transformer by switching control of a switching element on the primary side of the DAB circuit, a voltage is induced on the secondary side of the transformer, and the induced voltage is rectified by a switching element on the secondary side of the DAB circuit. If a capacitor (second smoothing capacitor 11 in FIG. 1 described later) is connected to the DC output section on the secondary side of the DAB circuit, the capacitor is charged by the rectification.

Here, in the conventional converter J1, if there is inductance due to reactors on the primary and secondary sides of the transformer, or capacitance of the capacitor, a response delay may occur in the power transmission. After the response delay, a current will start to flow from the secondary side of the DAB circuit to the load (connected equipment, etc.) connected to the secondary side of the DAB circuit due to the voltage difference between the load and the secondary side of the DAB circuit.

In power transmission with this kind of response delay, it takes time from when the primary side of the DAB circuit is switched to when the current in the DC output section of the secondary side of the DAB circuit increases. In other words, time is wasted in the control, and the control response is slow.

In addition, the conventional converter J1 is configured not to directly detect the input/output current (input current or output current) of the transformer, so if an overcurrent occurs in the transformer due to, for example, a short circuit on the secondary side of the DAB circuit, it may be difficult to protect the transformer.

First of all, the current passing through the transformer in a DAB circuit is a high-frequency AC current at about the switching frequency of the DAB circuit (for example, about 10 kHz to 100 kHz). In order to detect such a high-frequency AC current, it is necessary to detect it at a sampling frequency about 10 times the detection frequency, which means that very expensive hardware may be required.

To solve the problems of the conventional converter J1 described above, Patent Document 2 considers a configuration that directly detects the input/output currents of the transformer of the DAB circuit (hereinafter, simply referred to as conventional converter J2).

Shigenori Inoue and Hirofumi Akagi, "Operating Voltage and Loss Analysis of Bidirectional Isolated DC/DC Converters," IEEJ Transactions on Power Systems, Vol. 127, No. 2, 2007, pp. 189-197. Higa, Jun, Nagano, Tsuyoshi, and Ito, Junichi, "A Study on Reducing Transformer Losses According to the Control Method of Dual Active Bridge Converters," Institute of Electrical Engineers of Japan National Convention 2015, No. 4-077, pp. 130-131.

JP 2014-087134 A JP 2020-150574 A

In the case of the conventional converter J2, when the difference in DC voltage between the primary and secondary sides of the DAB circuit becomes large, switching loss, conduction loss, copper loss, etc. also become large, which may slow down the control response of the bidirectional converter.

The present invention was made in consideration of these technical issues, and aims to provide technology that can contribute to improving the control response of bidirectional converters and protecting their components.

One aspect of the present invention is a power supply comprising: first and second converter units coupled via a transformer and disposed between first and second DC power sources;
and a control circuit that controls the output power of power transmission by the first and second converter sections.

In this bidirectional isolated DC-DC converter, the DC voltage of the first DC power supply is lower than or equal to the DC voltage of the second DC power supply, the first converter unit is connected between the first DC power supply and the primary side of the transformer, and has a single-phase full bridge circuit in which a first switching arm in which first and second semiconductor switching elements are connected in series and a second switching arm in which third and fourth semiconductor switching elements are connected in series are connected in parallel, and the second converter unit is connected to the second DC power supply and the secondary side of the transformer, and has a single-phase full bridge circuit in which a third switching arm in which fifth and sixth semiconductor switching elements are connected in series and a fourth switching arm in which seventh and eighth semiconductor switching elements are connected in series are connected in parallel.

The control circuit also includes a gate signal generating unit that generates gate signals for each semiconductor switching element of the first and second converter units, and calculates a primary side pulse width command value that determines the pulse width of the output voltage of the first converter unit, a secondary side pulse width command value that determines the pulse width of the output voltage of the second converter unit, and a phase difference command value that determines the phase difference of the output voltage between the primary side pulse width command value and the secondary side pulse width command value.

The gate signal generating unit generates gate signals for each switching element of the second converter unit based on the current detection value detected from the input or output current of the transformer, the primary side pulse width command value, the secondary side pulse width command value, the primary side voltage detection value detected from the DC side of the first converter unit, the secondary side voltage detection value detected from the DC side of the second converter unit, and the triangular wave carrier signal so that the input/output power factor of the first converter unit becomes 1.

The current detection value is sampled at either the peak or the valley of the triangular wave carrier signal at a timing delayed from the peak of the other peak, and a low-pass filter removes high-frequency components greater than a switching frequency, which is the frequency of the triangular wave carrier signal , from the current detection value , and the control circuit controls the current detection value to become a current command value .

The control circuit may also be characterized in that it generates a phase difference command value based on the current detection value sampled at the delayed timing, the secondary side voltage detection value, and the voltage command value set on the DC side of the second converter unit.

In addition, it may be characterized in that the following equation (1) is satisfied, where the primary side pulse width command value is _W1 , the secondary side pulse width command value is _W2 , the primary side voltage detection value is _Vdc1 , the secondary side voltage detection value is _Vdc2 , and the phase difference command value is θ.

It may also be characterized in that the following formula (2) is satisfied:

Another aspect is a method for controlling a bidirectional isolated DC-DC converter that includes first and second converter units that are coupled via a transformer and disposed between first and second DC power sources, and a control circuit that controls the output power of the power transmission by the first and second converter units.

In this other aspect of the control method for a bidirectional isolated DC-DC converter, the DC voltage of the first DC power supply is lower than or equal to the DC voltage of the second DC power supply, the first converter unit is connected between the first DC power supply and the primary side of the transformer and has a single-phase full bridge circuit in which a first switching arm in which first and second semiconductor switching elements are connected in series and a second switching arm in which third and fourth semiconductor switching elements are connected in series are connected in parallel, and the second converter unit is connected to the second DC power supply and the secondary side of the transformer and has a single-phase full bridge circuit in which a third switching arm in which fifth and sixth semiconductor switching elements are connected in series and a fourth switching arm in which seventh and eighth semiconductor switching elements are connected in series are connected in parallel.

The control circuit also includes a gate signal generating unit that generates gate signals for each semiconductor switching element of the first and second converter units, and calculates a primary side pulse width command value that determines the pulse width of the output voltage of the first converter unit, a secondary side pulse width command value that determines the pulse width of the output voltage of the second converter unit, and a phase difference command value that determines the phase difference of the output voltage between the primary side pulse width command value and the secondary side pulse width command value.

The gate signal generating unit generates gate signals for each switching element of the second converter unit so that the input/output power factor of the first converter unit is 1 based on the current detection value detected from the input or output current of the transformer, the primary side pulse width command value, the secondary side pulse width command value, the primary side voltage detection value detected from the DC side of the first converter unit, the secondary side voltage detection value detected from the DC side of the second converter unit, and the triangular wave carrier signal.

The current detection value is sampled at either the peak or the valley of the triangular wave carrier signal at a timing delayed from the peak of the other peak, and a low-pass filter removes high-frequency components greater than a switching frequency, which is the frequency of the triangular wave carrier signal , from the current detection value , and the control circuit controls the current detection value to become a current command value .

As described above, the present invention can contribute to improving the control response of bidirectional converters and protecting their components.

FIG. 1 is a schematic configuration diagram of a DC power supply device for explaining a bidirectional converter S according to an example of the present embodiment (a diagram for explaining a main circuit configuration of the bidirectional converter S). Control configuration diagram of the control circuit 200 FIG. 2 is a diagram showing an example of the internal configuration of a primary side gate signal generating unit 281 in the control circuit 200. 4 is a waveform diagram illustrating an example of the control operation of the control circuit 200 (triangular wave carrier signal A, compared wave, primary side switching signal, and AC voltage V1). FIG. 2 is a diagram showing an example of the internal configuration of a secondary side gate signal generating unit 282 in the control circuit 200. 4 is a waveform diagram illustrating an example of the control operation of the control circuit 200 (triangular wave carrier signal B, compared wave, secondary side switching signal, AC voltage V2). 1 is a waveform diagram illustrating an example of sampling of current detection values in the control circuit 200 (triangular wave carrier signals A and B, compared wave, AC voltages V1 and V2, and current detection values I1 and Ifil).

The control configuration of the bidirectional converter and control method in the embodiment of the present invention is completely different from, for example, a configuration in which current is controlled simply based on the current detection value of the DC output section on the secondary side of the DAB circuit, as in the conventional converter J1, or a configuration in which current is controlled by simply directly detecting the input/output current of the transformer of the DAB circuit, as in the conventional converter J2.

That is, in the control configuration according to this embodiment, a control circuit that controls the output power of the power transmission of the DAB circuit between the first and second DC power sources calculates a primary side pulse width command value that determines the pulse width of the output voltage of the first converter unit arranged on the primary side of the DAB circuit, a secondary side pulse width command value that determines the pulse width of the output voltage of the second converter unit arranged on the secondary side of the DAB circuit, and a phase difference command value that determines the phase difference of the output voltage between the primary side pulse width command value and the secondary side pulse width command value. The DC voltage of the first DC power source is lower than the DC voltage of the second DC power source or is equal to the DC voltage of the second DC power source.

The gate signal of each switching element of the second converter unit is generated so that the input/output power factor of the first converter unit is 1 based on the current detection value detected from the input or output current of the transformer, the primary side pulse width command value, the secondary side pulse width command value, the primary side voltage detection value detected from the DC side of the first converter unit, the secondary side voltage detection value detected from the DC side of the second converter unit, and the triangular wave carrier signal.

The current detection value is sampled at one of the peaks and valleys of the triangular wave carrier signal at a sampling point delayed from the peak of the other peak, and a low-pass filter removes high-frequency components greater than the switching frequency, which is the frequency of the triangular wave carrier signal , from the current detection value , and the control circuit controls the current detection value to become a current command value .

According to the control configuration of this embodiment, the detected value of the input current or output current of the transformer is applied to the current control, so that it is possible to suppress the control response delay that may occur in the conventional converter J1, for example. In addition, the current detection value makes it easy to appropriately grasp the presence or absence of an overcurrent that may occur in the transformer.

In addition, the gate signals of each switching element of the second converter are generated so that the input/output power factor of the first converter is 1, so that, for example, the passing current of each switching element, reactor, and transformer can be reduced. This makes it possible to sufficiently suppress switching loss, conduction loss, and copper loss (sufficiently suppressed, for example, compared to the conventional converter J2) even when, for example, there is a large difference in the DC voltage between the primary and secondary sides of the DAB circuit.

This can contribute to improving the control response of bidirectional converters and protecting their components. Such contributions can enable highly accurate load operation of a load (connected equipment, etc.) connected to a DC power supply, and can potentially improve the reliability of the DC power supply.

As described above, the bidirectional converter and control method of this embodiment can be modified in design by appropriately applying common technical knowledge in various fields (e.g., power conversion technology, etc.) and referring to prior art documents as necessary, as long as the gate signals of the switching elements of the second converter section can be appropriately generated so that the input/output power factor of the first converter section arranged on the primary side of the DAB circuit is 1. One example of this is shown in the following example.

In the following examples, detailed explanations will be omitted as appropriate, for example by referring to the same reference numerals for similar contents. In addition, the following explanations will be mainly based on the assumption that the DC voltage of DC power source 1 described below is lower than the DC voltage of DC power source 2 described below, or that the DC voltages of DC power sources 1 and 2 are equal.

Example
The DC power supply device shown in Fig. 1 is for explaining a bidirectional converter S, which is an example of this embodiment. The bidirectional converter S shown in Fig. 1 mainly comprises a DAB circuit 100 and a control circuit 200, and is configured to be interposed between two DC power sources 1 and 2 so as to transmit power (transmit power in both directions).

The DAB circuit 100 mainly comprises a first smoothing capacitor 3 connected in parallel to the DC power source 1, a first converter section 4 having a switching circuit, a high-frequency transformer 8 as an isolated transformer, a second converter section 10 having a switching circuit, and a second smoothing capacitor 11 connected in parallel to the DC power source 2.

The switching circuit of the first converter section 4 is configured (full bridge circuit configuration) with multiple semiconductor switching elements 5a to 5d (hereinafter simply referred to as switches 5a to 5d) made of, for example, IGBTs, MOSFETs, etc.

The switching circuit of the first converter section 4 in FIG. 1 is configured as a single-phase full-bridge circuit in which a first switching arm consisting of switches 5a and 5b connected in series and a second switching arm consisting of switches 5c and 5d connected in series are connected in parallel.

The DC side of the first converter unit 4 is connected to the first smoothing capacitor 3, and the AC side of the first converter unit 4 is connected to the first winding (primary side) 8a of the high-frequency transformer 8, making it possible to perform bidirectional power conversion between DC and AC.

In the case of each of the switches 5a to 5d in FIG. 1, a diode is connected in anti-parallel. In addition, each of the switches 5a to 5d is connected in parallel to a capacitor 6a to 6d, respectively, forming a zero-voltage switching circuit configuration. This zero-voltage switching circuit configuration makes it possible to make the voltage across each element almost zero when each of the switches 5a to 5d is turned on.

In addition, a first reactor 7 is connected to the AC input/output lines between each switch 5a to 5d and the transformer 8, and the first reactor 7 and the first winding 8a are connected in series.

The switching circuit of the second converter section 10, like that of the first converter section 4, is configured with multiple switches 12a to 12d (full bridge circuit configuration) that are made up of, for example, IGBTs, MOSFETs, etc.

In the case of the switching circuit of the second converter section 10 in FIG. 1, it is configured as a single-phase full bridge circuit in which a third switching arm consisting of switches 12a and 12b connected in series and a fourth switching arm consisting of switches 12c and 12d connected in series are connected in parallel.

The DC side of the second converter unit 10 is connected to the second smoothing capacitor 11, and the AC side of the second converter unit 10 is connected to the second winding (secondary side) 8b of the transformer 8, making it possible to perform bidirectional power conversion between DC and AC.

In the case of each of the switches 12a to 12d in FIG. 1, a diode is connected in anti-parallel. In addition, each of the switches 12a to 12d is connected in parallel to a capacitor 13a to 13d, respectively, forming a zero-voltage switching circuit configuration. This zero-voltage switching circuit configuration makes it possible to make the voltage across each element almost zero when each of the switches 12a to 12d is turned on.

In addition, a second reactor 9 is connected to the AC input/output lines between each switch 12a to 12d and the transformer 8, and the second reactor 9 and the second winding 8b are connected in series.

In the DAB circuit 100, a current detector 20 is installed to detect the AC current I. As long as the current detector 20 can detect the AC current I as described above, various configurations can be applied, and the installation position can also be changed as appropriate. A specific example of the current detector 20 is one that uses a sensor such as a Hall CT. In addition, the installation position of the current detector 20 can be a position where the AC current I can be detected on the primary side or secondary side of the transformer 8.

In the case of the current detector 20 in FIG. 1, the AC current I is detected on the primary side of the transformer 8 (between the first converter section 4 and the first winding 8a of the transformer 8), but it may also be detected, for example, between the common connection point of the switches 5a and 5b and the first reactor 7, or between the common connection point of the switches 5c and 5d and the first winding 8a. The current detection value detected in this way (current detection value I1 in FIG. 7 described later) is input to the control circuit 200.

In addition, a primary side voltage detector 31 is installed on the DC side of the first converter unit 4 of the DAB circuit 100 to detect the DC voltage on the DC side. In addition, a secondary side voltage detector 32 is installed on the DC side of the second converter unit 10 to detect the DC voltage on the DC side. As long as the primary side voltage detector 31 and secondary side voltage detector 32 can detect DC voltage as described above, various configurations can be applied, and their installation positions can also be changed as appropriate.

In the case of FIG. 1, the primary side voltage detector 31 is installed between the first smoothing capacitor 3 and the DC power supply 1, and the secondary side voltage detector 32 is installed between the second smoothing capacitor 11 and the DC power supply 2. As a result, the primary side voltage detector 31 and the secondary side voltage detector 32 are configured to detect the DC voltages of the first and second smoothing capacitors 3 and 11, respectively. The voltage detection values detected in this way (primary side voltage detection value, secondary side voltage detection value) are also input to the control circuit 200.

The control circuit 200 receives the current detection value detected by the current detector 20 and the voltage detection values detected by the primary side voltage detector 31 and the secondary side voltage detector 32. In addition, a desired voltage command value is set on the DC side of the second converter unit 10, so that a desired triangular wave carrier signal can be generated.

The control circuit 200 is also capable of generating gate signals G-5 (specifically, gate signals G-5a to G-5d for each switch 5a to 5d; on/off command signals) and G-12 (specifically, gate signals G-12a to G-12d for each switch 12a to 12d; on/off command signals) that control the switching of each switch 5a to 5d, 12a to 12d of the first and second converter units 4 and 10, respectively, based on the current detection value, each voltage detection value, voltage command value, and triangular wave carrier signal (for example, triangular wave carrier signal A and triangular wave carrier signal B described below).

The control circuit 200 is configured to transmit the generated gate signals G-5 and G-12 to each switch 5a to 5d and 12a to 12d, respectively, to drive and control the switching circuits of the first and second converter units 4 and 10.

In addition, a first reactor 7 is connected to the AC input/output lines between each switch 5a to 5d and the transformer 8, and the first reactor 7 and the first winding 8a are connected in series.

The bidirectional converter S described above can be modified in design as needed depending on the purpose. For example, the capacitors 6a-6d, 13a-13d and the first and second reactors 7, 9 may be omitted as needed.

In addition, as for the control circuit 200, as long as the control configuration is such that it can drive and control the switching circuits of the first and second converter units 4 and 10 as described above, various aspects can be applied, and specific examples include the configuration example shown below.

<Configuration example of control circuit 200>
Fig. 2 shows an example of the configuration of the control circuit 200. The control circuit in Fig. 2 mainly includes voltage detection low-pass filter units (LPF: Low-Pass Filter) 210, 220, a current detection low-pass filter unit 230, a voltage control unit (AVR: Automatic Voltage Regulator) 240, a current limiting unit 250, a current control unit (ACR: Automatic Current Regulator) 260, a pulse width calculation unit 270, a primary side gate signal generation unit 281, a secondary side gate signal generation unit 282, a triangular wave generation unit 300, and an ADC unit (ADC: Analog-to-Digital Converter) 310.

The primary side voltage detection value detected by the primary side voltage detector 31 is input to the voltage detection low pass filter unit 210, and the secondary side voltage detection value detected by the secondary side voltage detector 32 is input to the voltage detection low pass filter unit 220. These voltage detection low pass filter units 210, 220 are configured to remove high frequency noise components from the input primary side voltage detection value and secondary side voltage detection value, respectively.

The subtractor 24a derives the difference between the output of the voltage detection low-pass filter unit 220 and the voltage command value, and is configured to input this difference to the voltage control unit 240.

The voltage control unit 240 is configured to be able to control the secondary side voltage detection value to become a voltage command value based on the output of the subtractor 24a. The control configuration of this voltage control unit 240 can be, for example, a configuration using a PID compensator or the like. The output of the voltage control unit 240 is then applied as a current command value.

The current limiting unit 250 has a preset allowable current value related to the current command value, and is configured to limit the current command value input from the voltage control unit 240 so that it is equal to or less than the allowable current value.

The triangular wave generating unit 300 is configured to generate a triangular wave carrier signal A that serves as a reference for switching the switches 5a to 5d, 12a to 12d and for sampling detection of the AC current I. The triangular wave carrier signal A has a frequency equal to the switching frequency (fc) and is a signal with positive and negative symmetry (maximum value 1, minimum value -1) such that the average value of the triangular wave carrier signal A is zero. With such a triangular wave carrier signal A, the positive parts of the triangular wave carrier signal A appear as peaks and the negative parts appear as valleys, for example, as shown in FIG. 4 described later.

The current detection value detected by the current detector 20 is input to the current detection low-pass filter unit 230, which removes high-frequency noise components. Then, the output of the current detection low-pass filter unit 230 and the triangular wave carrier signal generated by the triangular wave generating unit 300 are each input to the ADC unit 310.

The ADC section 310 is configured to convert the current detection value, from which noise has been removed by the current detection low-pass filter section 230, into a digital value and sample it. The timing of the conversion and sampling of the current detection value is either one of the peaks or valleys of the triangular wave carrier signal A, and the sampling point is delayed from the peak of that one (for example, the sample point indicated by the symbol "▽" in Figure 7 described below).

The subtractor 26a derives the difference between the output of the current limiting section 250 and the output of the ADC section 310, and is configured to input this difference to the current control section 260.

The current control unit 260 is configured to be able to control the current detection value to become the current command value based on the output of the subtractor 26a. The control configuration of this current control unit 260 can be, for example, a configuration using a PID compensator. The output of the current control unit 260 is applied as the phase difference command value θ.

The phase difference command value θ is input to the pulse width calculation unit 270 together with each voltage detection value (filtered primary side voltage detection value, secondary side voltage detection value) output from the voltage detection low pass filter units 210, 220 and the primary side pulse width command value _W1 of the first converter unit 4. The primary side pulse width command value _W1 is, for example, a pulse width set in advance in the control circuit 200.

The pulse width calculation section 270 is configured to be able to calculate a secondary side pulse width command value _W2 of the second converter section 10, for example, as shown in a calculation example described later.

The primary side gate signal generating unit 281 is configured to receive the primary side pulse width command value _W1 and the triangular wave carrier signal A generated by the triangular wave generating unit 300, and to generate a gate signal G-5.

The secondary side gate signal generating unit 282 is configured to receive the secondary side pulse width command value _W2 , the triangular wave carrier signal A generated by the triangular wave generating unit 300, and the phase difference command value θ, and to generate the secondary side gate signal G-12.

<Example of Calculation by Pulse Width Calculation Unit 270>
For example, according to the contents shown in Non-Patent Document 2, it can be read that copper loss and conduction loss can be suppressed by reducing the current I (i.e., the current detection value detected by the current detector 20) flowing through the first reactor 7 of the bidirectional converter S shown in FIG. 1.

In addition, one of the conditions for minimizing the current flowing through the first reactor 7 can be seen as the inverter with the higher DC voltage (the switching circuit of the second converter unit 10 in FIG. 1) bearing all of the reactive power supplied to the high-frequency transformer and reactor, and the inverter with the lower DC voltage (the switching circuit of the first converter unit 4 in this embodiment) operating at a power factor of 1.

Therefore, in the pulse width calculation unit 270, the secondary side pulse width command value _W2 is calculated so as to satisfy the following formula (1) according to the contents of the above-mentioned non-patent document 2. In the following formula (1), _Vdc1 is a primary side voltage detection value that determines the pulse width of the output voltage of the first converter unit 4, _Vdc2 is a secondary side voltage detection value that determines the pulse width of the output voltage of the second converter unit 10, and θ is a phase difference command value (rad) that determines the phase difference of the output voltage between the primary side pulse width command value W1 and the secondary side pulse width command value W2. In addition, the primary side pulse width command value _W1 is an arbitrary fixed value, _W1 satisfies 0< _W1 ≦1, and _W2 satisfies 0< _W2 ≦1.

If _Vdc1 in this formula (1) is too small, it may be impossible to operate with a power factor of 1 even if _W1 = 1. In such a case, the pulse width calculation unit 270 performs calculations within a range that satisfies the following formula (2).

By operating the pulse width calculation unit 270 as described above, the current I flowing through the first reactor 7 of the bidirectional converter S shown in Figure 1 can be reduced, and copper loss and conduction loss can be suppressed.

<Example of generation of gate signal G-5 by primary side gate signal generating unit 281>
In the primary side gate signal generating section 281 of the control circuit 200, for example, a configuration as shown in FIG. 3 is applied, and the control operation is performed as follows.

The primary side gate signal generating unit 281 shown in FIG. 3 mainly comprises a calculation unit 400, a rectangular wave generating unit 410, multipliers 421 and 422, triangular wave comparing units 431 and 432, and dead time generating units 441, 442, 443, and 444.

A primary side pulse width command value _W1 set in advance in the control circuit 200 is input to this primary side gate signal generation unit 281. The primary side pulse width command value _W1 is input to the calculation unit 400 and converted to a value of 1- _W1 (that is, when _W1 satisfies 0< _W1 ≦1, 0≦(1- _W1 )<1).

The square wave generating unit 410 is configured to generate two types of square wave signals synchronized with the triangular wave carrier signal A, and input each of the generated square wave signals to the multipliers 421 and 422. Specifically, a square wave signal that is -1 when the triangular wave carrier signal A rises (when the slope is positive) and +1 when it falls (when the slope is negative) is input to the multiplier 421. In addition, a square wave signal that is +1 when the triangular wave carrier signal A rises (when the slope is positive) and -1 when it falls (when the slope is negative) is input to the multiplier 422.

The multiplier 421 multiplies the rectangular wave signal generated by the rectangular wave generating unit 410 by (1-W ₁ ) calculated by the calculating unit 400 to generate a compared wave for the switching signals G-5a, G-5b (hereinafter simply referred to as the compared wave for G-5a, G-5b) related to the switches 5a, 5b of the first converter unit 4. This compared wave for G-5a, G-5b is input to the triangular wave comparing unit 431 together with the triangular wave carrier signal A.

The triangular wave comparator 431 generates switching signals G-5a and G-5b based on the results of the comparison between the G-5a and G-5b comparison waves and the triangular wave carrier signal A, as shown in FIG. 4, for example.

In the case of the switching signals G-5a and G-5b shown in FIG. 4, in the region where the compared wave for G-5a and G-5b is less than or equal to the triangular wave carrier signal A, switch 5a switches on and switch 5b switches off, and in the region where the compared wave for G-5a and G-5b is greater than or equal to the triangular wave carrier signal A, switch 5a switches off and switch 5b switches on.

The switching signals G-5a and G-5b generated in this way are input to the dead time generating units 441 and 442, respectively. Then, the rising parts of the switching signals G-5a and G-5b are deleted by the dead time generating units 441 and 442 for the dead time (for example, the on signal is corrected to an off signal for the dead time), and become the gate signals G-5a and G-5b for the switches 5a and 5b, respectively.

The multiplier 422 multiplies the rectangular wave signal generated by the rectangular wave generating unit 410 by (1-W ₁ ) calculated by the calculating unit 400 to generate a compared wave for the switching signals G-5c, G-5d (hereinafter simply referred to as the compared wave for G-5c, G-5d) related to the switches 5c, 5d of the first converter unit 4. This compared wave for G-5c, G-5d is input to the triangular wave comparing unit 432 together with the triangular wave carrier signal A.

The triangular wave comparator 432 generates switching signals G-5c and G-5d based on the results of the comparison between the compared waves for G-5c and G-5d and the triangular wave carrier signal A, as shown in FIG. 4, for example.

In the case of the switching signals G-5c and G-5d shown in FIG. 4, in the region where the compared wave for G-5c and G-5d is less than or equal to the triangular wave carrier signal A, switch 5c is switched off and switch 5d is switched on, and in the region where the compared wave for G-5c and G-5d is greater than or equal to the triangular wave carrier signal A, switch 5c is switched on and switch 5d is switched off.

The switching signals G-5c and G-5d thus generated are input to the dead time generating units 443 and 444, respectively. Then, the rising edge of the switching signals G-5c and G-5d is deleted by the dead time generating units 443 and 444 for the dead time (for example, the on signal is corrected to an off signal for the dead time), and the resulting signals become the gate signals G-5c and G-5d for the switches 5c and 5d, respectively.

<Example of generation of gate signal G-12 by secondary side gate signal generating unit 282>
In the secondary side gate signal generating section 282 of the control circuit 200, for example, a configuration as shown in FIG. 5 is applied, and the control operation is performed as follows.

The secondary gate signal generating unit 282 shown in FIG. 5 mainly comprises a calculation unit 500, a phase shift unit 501, a rectangular wave generating unit 510, multipliers 521 and 522, triangular wave comparing units 531 and 532, and dead time generating units 541, 542, 543, and 544.

The secondary gate signal generating section 282 receives the secondary pulse width command value W ₂ calculated by the pulse width calculating section 270 and the phase difference command value θ generated by the current control section 260 .

The calculation unit 500 is configured to calculate {(2θ/π)+(W ₂ /2)} and {(2θ/π)-(W ₂ /2)} from the secondary side pulse width command value W ₂ and the phase difference command value θ.

The phase shift unit 501 is configured to convert the triangular wave carrier signal A input from the triangular wave generating unit 300 into a triangular wave carrier signal B with a 90 degree phase lead.

The square wave generating unit 510 is configured to receive the triangular wave carrier signal B output from the phase shift unit 501, generate a square wave signal synchronized with this triangular wave carrier signal B, and input the generated square wave signal to the multipliers 521 and 522. Specifically, a square wave signal that is +1 when the triangular wave carrier signal B rises (when the slope is positive) and -1 when it falls (when the slope is negative) is input to the multipliers 521 and 522.

In the multiplier 521, the square wave signal generated in the square wave generating section 510 is multiplied by {(2θ/π)+( _W2 /2)} calculated in the calculating section 500 to generate a compared wave for the switching signals G-12a, G-12b (hereinafter simply referred to as the compared wave for G-12a, G-12b) related to the switches 12a, 12b of the second converter section 10. This compared wave for G-12a, G-12b is input to the triangular wave comparing section 531 together with the triangular wave carrier signal B.

The triangular wave comparator 531 generates switching signals G-12a and G-12b based on the results of the comparison between the compared waves for G-12a and G-12b and the triangular wave carrier signal B, as shown in FIG. 6 for example.

In the case of the switching signals G-12a and G-12b shown in FIG. 6, in the region where the compared wave for G-12a and G-12b is less than or equal to the triangular wave carrier signal B, switch 12a switches on and switch 12b switches off, and in the region where the compared wave for G-12a and G-12b is greater than or equal to the triangular wave carrier signal B, switch 12a switches off and switch 12b switches on.

The switching signals G-12a and G-12b generated in this way are input to the dead time generating units 541 and 542, respectively. Then, the rising parts of the switching signals G-12a and G-12b are deleted by the dead time generating units 541 and 542 for the dead time (for example, the on signal is corrected to an off signal for the dead time), and become the gate signals G-12a and G-12b for the switches 12a and 12b, respectively.

In the multiplier 522, the rectangular wave signal generated in the rectangular wave generating section 510 is multiplied by {(2θ/π)-( _W2 /2)} calculated in the calculating section 500 to generate a compared wave for the switching signals G-12c, G-12d (hereinafter simply referred to as the compared wave for G-12c, G-12d) related to the switches 12c, 12d of the second converter section 10. This compared wave for G-12c, G-12d is input to the triangular wave comparing section 532 together with the triangular wave carrier signal B.

The triangular wave comparator 532 generates switching signals G-12c and G-12d based on the results of the comparison between the comparison waves for G-12c and G-12d and the triangular wave carrier signal B, as shown in FIG. 6, for example.

In the case of the switching signals G-12c and G-12d shown in FIG. 6, in the region where the compared wave for G-12c and G-12d is less than or equal to the triangular wave carrier signal B, switch 12c switches on and switch 12d switches off, and in the region where the compared wave for G-12c and G-12d is greater than or equal to the triangular wave carrier signal B, switch 12c switches off and switch 12d switches on.

The switching signals G-12c and G-12d thus generated are input to the dead time generating units 543 and 544, respectively. Then, the rising parts of the switching signals G-12c and G-12d are deleted by the dead time generating units 543 and 544 for the dead time (for example, the on signal is corrected to an off signal for the dead time), and become the gate signals G-12c and G-12d for the switches 12c and 12d, respectively.

Here, if the output voltage (AC voltage) of the second converter unit 10 is voltage V2, as shown in FIG. 6, voltage V2 becomes secondary-side pulse width command value _W2 , and the time from the zero-cross point of triangular wave carrier signal B to the center of the voltage pulse is θ/(2πfc).

<An example of sampling in the control circuit 200>
Next, an example of sampling of the current detection value in the control circuit 200 will be described with reference to Fig. 7. In Fig. 7, the voltage V1 which is the output voltage (AC voltage) of the first converter unit 4, the voltage V2 which is the output voltage (AC voltage) of the second converter unit 10, and the current detection value I1 detected by the current detector 20 are depicted on the same time axis.

As shown in FIG. 7, the voltage V1 and the current detection value I1 are in a power factor 1 relationship (the input/output power factor of the first converter unit 4 is 1) and in phase due to the calculations of the pulse width calculation unit 270. In addition, the zero crossing point of the triangular wave carrier signal B is the apex of the peak or valley of the triangular wave carrier signal A, and is the center of the voltage pulse of voltage V1. Therefore, the time between the centers of the voltage pulses of voltages V1 and V2 is θ/(2πfc), and the phase difference command value between voltages V1 and V2 is θ [rad].

The current detection value I1 contains a fair amount of high frequency components, but the peak of the fundamental wave component of the current detection value I1 is at the center of the voltage pulse of the voltage V1, that is, on the apex side of the crest or valley of the triangular wave carrier signal A. In the case of the control circuit 200, the current detection value I1 is input to the current detection low-pass filter unit 230, so that the high frequency components described above are removed.

In this current detection low-pass filter unit 230, for example, by setting the time constant to about three times the switching frequency fc, the fundamental wave component of frequency fc is not attenuated, and high-frequency components of the third harmonic or higher are reduced. In this way, the signal waveform of the current detection value I1 after passing through the current detection low-pass filter unit 230 becomes approximately a sine wave, like the current signal Ifil shown in Figure 7.

This current signal Ifil is delayed in phase from the current detection value I1 by the current detection low-pass filter unit 230, and the peak of the current signal Ifil is a delayed point (a point delayed by a certain time) from the apex of the crest or valley of the triangular wave carrier signal A. Such a delay time (the time from the apex of the crest or valley to the delayed point) is determined by the delay by the current detection low-pass filter unit 230 and a detection circuit such as the current detector 20, and does not depend on the amplitude of the current I, etc.

In the case of the control circuit 200, the ADC unit 310 is configured to compare the triangular wave carrier signal A with an arbitrary sample timing reference value and determine the current sample timing. This makes it possible to generate sample timing that always has a constant delay time from the apex of the peak or valley of the triangular wave carrier signal A, such as the sample point indicated by the symbol "▽" in Figure 7.

In addition, such a sample timing point is the point where dI/dt (slope) is smallest in the current signal Ifil, and compared to sampling at other points, it is easier to avoid the effects of jitter in the sample timing (such as large fluctuations in the detected value).

By sampling the current detection value in the control circuit 200 as described above, high frequency components higher than the switching frequency can be removed from the current detection value, and the current detection value can be suitably used as a detection value for feedback control, which also results in good stability in current control.

Although the present invention has been described in detail above only with respect to the specific examples, it will be clear to those skilled in the art that various modifications are possible within the scope of the technical concept of the present invention, and it goes without saying that such modifications fall within the scope of the claims.

For example, in the embodiment, the current detection value detected from the AC current I between the primary side of the DAB circuit 100 (first converter unit 4) and the primary side of the transformer 8 is applied, but even if the current detection value detected between the secondary side of the DAB circuit (second converter unit 10) and the secondary side of the transformer 8 is applied, the same effect as that shown in the embodiment is achieved.

In addition, since the input current or output current of the transformer 8 can be constantly detected, the detection results can be appropriately reused. For example, a function can be added (e.g., added separately from the configuration in FIG. 1) that switches off the gate signals of all the switches (switches 5a-5d, 12a-12d) of the DAB circuit 100 when the detected value of the input current or output current of the transformer 8 exceeds a predetermined value (such as a preset threshold value). This makes it possible to protect the transformer 8 from destruction, for example, when an overcurrent occurs. The detection point of the AC current I in such a protection function is not particularly limited (it is not limited to the valleys or peaks of the triangular wave carrier signal).

S... Bidirectional converter 100... DAB circuit 200... Control circuit 1, 2... DC power supply 4, 10... First and second converter units 5a to 5d... Switches (first to fourth semiconductor switching elements)
12a to 12d: Switches (fifth to eighth semiconductor switching elements)
8. Transformer

Claims (5)

a first converter unit and a second converter unit coupled via a transformer and disposed between the first and second DC power sources;
A control circuit for controlling output power of power transmission by the first and second converter units;
Equipped with
a DC voltage of the first DC power source is lower than or equal to a DC voltage of the second DC power source;
the first converter unit is connected between the first DC power supply and a primary side of the transformer,
a single-phase full bridge circuit in which a first switching arm in which first and second semiconductor switching elements are connected in series and a second switching arm in which third and fourth semiconductor switching elements are connected in series are connected in parallel;
the second converter unit is connected to the second DC power supply and a secondary side of the transformer,
a single-phase full bridge circuit in which a third switching arm in which fifth and sixth semiconductor switching elements are connected in series and a fourth switching arm in which seventh and eighth semiconductor switching elements are connected in series are connected in parallel,
the control circuit includes a gate signal generating unit that generates gate signals for the semiconductor switching elements of the first and second converter units,
A primary-side pulse width command value that determines a pulse width of an output voltage of the first converter unit;
A secondary-side pulse width command value that determines a pulse width of an output voltage of the second converter unit;
a phase difference command value that determines a phase difference between an output voltage of the primary pulse width command value and the secondary pulse width command value;
Calculate
The gate signal generation unit generates a gate signal for each switching element of the second converter unit.
a current detection value detected from an input current or an output current of the transformer ;
The primary side pulse width command value;
The secondary pulse width command value;
A primary side voltage detection value detected from a DC side of the first converter unit;
A secondary side voltage detection value detected from a DC side of the second converter unit;
A triangular wave carrier signal;
based on the above , the input/output power factor of the first converter unit is generated to be 1;
The detected current value is
Sampling is performed at a timing delayed from one of the crest and the trough of the triangular wave carrier signal,
A low-pass filter removes high-frequency components higher than a switching frequency, which is the frequency of the triangular wave carrier signal , from the current detection value ,
The control circuit controls the current detection value to become a current command value .
A bidirectional isolated DC-DC converter comprising: 2. The bidirectional isolated DC-DC converter according to claim 1, wherein the control circuit generates the phase difference command value based on the current detection value sampled at the delayed timing, the secondary side voltage detection value, and a voltage command value set on the DC side of the second converter unit. The bidirectional isolated DC- DC converter according to claim ₁ or 2, characterized in that the following formula (1) is satisfied, where the primary side pulse width command value is _W1 , the secondary side pulse width command value is _W2 , the primary side voltage detection value is _Vdc1 , the secondary side voltage detection value is Vdc2, and the phase difference command value is θ.

4. The bidirectional isolated DC-DC converter according to claim 3, wherein the following formula (2) is satisfied:

a first converter unit and a second converter unit coupled via a transformer and disposed between the first and second DC power sources;
A control circuit for controlling output power of power transmission by the first and second converter units;
A control method for a bidirectional isolated DC-DC converter comprising:
a DC voltage of the first DC power source is lower than or equal to a DC voltage of the second DC power source;
the first converter unit is connected between the first DC power supply and a primary side of the transformer,
a single-phase full bridge circuit in which a first switching arm in which first and second semiconductor switching elements are connected in series and a second switching arm in which third and fourth semiconductor switching elements are connected in series are connected in parallel;
the second converter unit is connected to the second DC power source and a secondary side of a transformer,
a single-phase full bridge circuit in which a third switching arm in which fifth and sixth semiconductor switching elements are connected in series and a fourth switching arm in which seventh and eighth semiconductor switching elements are connected in series are connected in parallel,
the control circuit includes a gate signal generating unit that generates gate signals for the semiconductor switching elements of the first and second converter units,
A primary-side pulse width command value that determines a pulse width of an output voltage of the first converter unit;
A secondary-side pulse width command value that determines a pulse width of an output voltage of the second converter unit;
a phase difference command value that determines a phase difference between an output voltage of the primary pulse width command value and the secondary pulse width command value;
Calculate
The gate signal generation unit generates a gate signal for each switching element of the second converter unit.
a current detection value detected from an input current or an output current of the transformer ;
The primary side pulse width command value;
The secondary pulse width command value;
A primary side voltage detection value detected from a DC side of the first converter unit;
A secondary side voltage detection value detected from a DC side of the second converter unit;
A triangular wave carrier signal;
based on the above , the input/output power factor of the first converter unit is generated to be 1;
The detected current value is
Sampling is performed at a timing delayed from one of the crest and the trough of the triangular wave carrier signal,
A low-pass filter removes high-frequency components higher than a switching frequency, which is the frequency of the triangular wave carrier signal , from the current detection value ,
The control circuit controls the current detection value to become a current command value .
A control method for a bidirectional isolated DC-DC converter.

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