JP7251424B2 - INVERTER DEVICE AND INVERTER DEVICE CONTROL METHOD - Google Patents

Description

The present invention relates to an inverter device and an inverter device control method.

In general, PM (Permanent Magnet) motors use permanent magnets in their rotors, so they do not generate excitation current and secondary current, and can operate more efficiently than induction motors.
In particular, PMSMs (Permanent Magnet Synchronous Motors), which are permanent magnet synchronous motors, are widely used.
Vector control and V/f control can be typically exemplified as the inverter device that controls the PMSM.
Here, vector control requires rotor position information obtained by a rotor position sensor, but the rotor position sensor is susceptible to heat and vibration and has low environmental resistance.
Also, the installation of rotor position sensors leads to increased costs.

On the other hand, V/f control is known as a simple PMSM control method that does not require rotor position information.
V/f control is a control method that keeps the ratio between the amplitude and frequency of the output voltage of the inverter device constant.
V/f control requires less computation than vector control.
Therefore, the control section of the inverter device can be realized by a low-cost microcomputer.

However, in the V/f control, since the rotor position is not detected, it is difficult to appropriately control the current phase during load operation, and there is a problem that reactive power is generated and efficiency is lowered.

Non-Patent Document 1, which is an example of conventional technology, describes a technology that can suppress the generation of reactive power by appropriately controlling the voltage correction amount to match the inverter output voltage with the q-axis, which is the motor control axis. disclosed.

Junichi Ito, Jiro Toyosaki, Hiroshi Osawa, "High Performance V/f Control of Permanent Magnet Synchronous Motor", The Institute of Electrical Engineers of Japan Transaction D, 2003, Vol. 122, No. 3, p. 253-259

However, the conventional technology described above has a problem that the q-axis inductance fluctuates due to the influence of magnetic saturation during load operation, and the accuracy of estimating reactive power decreases.

The present invention has been made in view of the above, and provides a technology that enables an inverter device that drives a permanent magnet synchronous motor by V/f control to be robust against magnetic saturation and to operate with high efficiency. intended to

The present invention, which solves the above-described problems and achieves the object, is an inverter device for driving a permanent magnet synchronous motor by V/f control, wherein the γ-axis voltage command and the δ-axis voltage are controlled based on a rotational angular frequency command. A command is generated, and ΔQ calculated by the following equation (6) is calculated by any one of P calculation, PI calculation, and PID calculation to calculate a voltage compensation amount command, and the voltage command for the γ axis, The corrected δ-axis voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command is coordinate-transformed into the α-axis voltage command vα and the β-axis voltage command vβ, and the α-axis voltage command A gate signal for turning on and off the switching element is generated based on vα and the β-axis voltage command vβ. is a value corrected by multiplying by the variation rate of the δ-axis inductance from the time of no load, which is calculated by the above calculation.

where ω: rotation angular frequency command, iγ: γ-axis current estimate, iδ: δ-axis current estimate, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , L.delta.: the .delta.-axis inductance value of the permanent magnet synchronous motor, and R: the winding resistance value.

Alternatively, the present invention is a control method for an inverter device that drives a permanent magnet synchronous motor by V/f control, wherein a γ-axis voltage command and a δ-axis voltage command are generated based on a rotational angular frequency command; calculating a voltage compensation amount command by calculating ΔQ calculated by the following equation (6) by any one of P calculation, PI calculation, and PID calculation; coordinate conversion of the δ-axis corrected voltage command obtained by adding the voltage compensation amount command to the voltage command into the α-axis voltage command vα and the β-axis voltage command vβ, and the α-axis voltage command vα and The q-axis inductance value Lq in the above equation (6) is the δ-axis inductance calculated by the following equation (10). is a control method of an inverter device, which is a value corrected by multiplying the variation rate of the δ-axis inductance from no load calculated using

where ω: rotation angular frequency command, iγ: γ-axis current estimate, iδ: δ-axis current estimate, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , L.delta.: the .delta.-axis inductance value of the permanent magnet synchronous motor, and R: the winding resistance value.

ADVANTAGE OF THE INVENTION According to this invention, the effect that the inverter apparatus which drives a permanent-magnet synchronous motor by V/f control can be made robust with respect to magnetic saturation, and can be operated with high efficiency is produced.

FIG. 1 is a diagram showing the coordinate system of PMSM. FIG. 2 is a diagram showing the relationship between the output voltage output by the inverter device in V/f control and the dq axis direction of the motor. FIG. 3 is a diagram showing the relationship between the output voltage output by the inverter device in V/f control, the δ-axis, the γ-axis, and the dq-axis of the motor. FIG. 4 is a diagram showing the relationship between the corrected output voltage output by the inverter device in V/f control, the δ-axis, the γ-axis, and the dq-axis of the motor. FIG. 5 is a diagram showing a system configuration including a conventional inverter device. FIG. 6 is a diagram showing a system configuration including a block diagram showing details of the inverter device according to the technique described above. FIG. 7 is a diagram showing a voltage compensation amount command calculator, a reactive power estimator, and an inductance value compensator included in the inverter device according to the present embodiment.

BEST MODE FOR CARRYING OUT THE INVENTION Hereinafter, embodiments for carrying out the present invention will be described with reference to the accompanying drawings.
However, the present invention is not to be construed as limited by the description of the following embodiments.

<Embodiment 1>
FIG. 1 is a diagram showing the coordinate system of PMSM.
The coordinate system shown in FIG. 1 is set so that the space vector of the phase voltage is oriented in the δ-axis direction during operation of the PMSM.

In PMSM control, the direction of magnetic flux generated by the magnets provided on the rotor is defined as the d-axis direction, and the direction of induced voltage generation perpendicular to the d-axis is defined as the q-axis direction.
Here, the q-axis direction current iq flowing in the q-axis direction contributes to torque and thus generates active power, and the d-axis direction current id flowing in the q-axis direction perpendicular to the q-axis direction current iq generates reactive power.
Therefore, by controlling the d-axis direction current id=0, the maximum torque and maximum current can be obtained in the V/f control, and the copper loss of the motor can be minimized.

FIG. 2 is a diagram showing the relationship between the output voltage V1 output by the inverter device in V/f control and the dq axis direction of the motor.
In FIG. 2, the voltage phase output by the inverter is used as a reference, the axis of the voltage output by the inverter is the δ axis, the axis perpendicular to this is the γ axis, and hereinafter the γδ axis is the V/f control axis. called.
At this time, as shown in FIG. 2, the dq axis, which is the motor control axis based on the rotor position of the motor, and the V/f control axis are different.
That is, the direction of the output voltage V1 output by the inverter device does not match the q-axis direction of the motor.
Therefore, since the d-axis direction current id≠0, the current flows in the d-axis direction, increasing the copper loss of the motor, generating reactive power, and reducing the power efficiency of the inverter device.
Here, the phase difference between the dq axis and the V/f control axis increases according to the magnitude of the load.
Therefore, the decrease in power efficiency during high load is a particular problem.

FIG. 3 is a diagram showing the relationship between the output voltage V1 output by the inverter device in V/f control, the δ axis, the γ axis, and the dq axes of the motor.
At the predetermined load angle shown in FIG. 3, the output current I1 flows.
When the output current I1 shown in FIG. 3 is resolved into a dq-axis coordinate system, a d-axis direction current id flows in the d-axis direction, so reactive power is generated.

FIG. 4 is a diagram showing the relationship between the corrected output voltage output by the inverter device in V/f control, the δ-axis, the γ-axis, and the dq-axis of the motor.
Here, in order to suppress the reactive power due to the d-axis direction current id, the voltage compensation amount ΔV is added to the output voltage V1, and the output voltage V2 after addition of the compensation amount is V2=V1+ΔV.
Then, the current that flows due to the output voltage V2 after addition of the compensation amount is assumed to be the output current I2.
The output current I2 coincides with the q-axis, which is the control axis of the motor.

However, in the V/f control, since the rotor position is not detected, it is difficult to appropriately control the current phase during load operation, and there is a problem that reactive power is generated and efficiency is lowered.

Therefore, in Non-Patent Document 1, which is an example of the above-mentioned conventional technology, in a system in which the PMSM is operated at a desired rotational angular frequency command ω, the δ-axis voltage command Vδ is corrected according to the reactive power estimated values Qdq and Qγδ. thereby improving efficiency.

FIG. 5 is a diagram showing a system configuration including a conventional inverter device.
FIG. 5 shows voltage command calculator 11, inverter 12, PMSM 13, voltage compensation amount command calculator 14, reactive power estimator 15, and adder 16. FIG.

A rotation angular frequency command ω from a control unit (not shown) is input to the voltage command calculation unit 11, and the voltage command calculation unit 11 outputs a δ-axis voltage command vδ and a γ-axis voltage command vγ.
In general, in V/f control, the γ-axis voltage command vγ is set to vγ=0, and the δ-axis voltage command vδ is set to a value proportional to the rotational angular frequency command ω.

The corrected voltage command vδ′ for the δ axis and the voltage command vγ for the γ axis are input to the inverter unit 12 , and the inverter unit 12 outputs a three-phase voltage to the PMSM 13 .
Here, the post-correction voltage command vδ′ for the δ-axis is obtained by adding the voltage compensation amount command Vcmd to the voltage command vδ for the δ-axis by the adder 16 .
The inverter unit 12 generates a three-phase voltage command vuvw based on the corrected voltage command vδ′ of the δ-axis and the voltage command vγ of the γ-axis, compares the three-phase voltage command vuvw with the triangular carrier signal, A three-phase voltage having a predetermined amplitude and a predetermined frequency is output to the PMSM 13 by generating a gate signal for turning on and off the switching element in the inverter unit 12 based on the result of this comparison.

In general, motor control uses a two-phase coordinate system, which is a rotating coordinate system in which a symmetrical three-phase alternating current is converted into an equivalent two-phase alternating current for simplification of control.

Here, using the winding resistance value R, the d-axis inductance value Ld of the motor, the q-axis inductance value Lq of the motor, the differential operator p, the induced voltage constant Ke, and the rotational angular frequency command ω, the PMSM 13 is expressed by the following equation ( 1) is represented by the voltage equation.

Reactive power Qdq on the dq axis, which is the motor control axis, is expressed by the following equation (2).

By substituting the above formula (1) into the above formula (2), the following formula (3) is obtained.

Here, if the d-axis direction current id is controlled to be 0, the maximum torque and the maximum current can be obtained. Equation (4) is obtained.

Here, assuming that the output current of the inverter section 12 is I1, I1=(iq) ² during control with the d-axis direction current id=0.

Next, the reactive power Qγδ on the V/f control axis is calculated.
Similarly, if iδ=0 on the V/f control axis, the reactive power Qγδ on the V/f control axis is expressed by the following equation (5).

In order to obtain the maximum torque and maximum current on the V/f control shaft, the above equations (4) and (5) should be matched.
Here, when the output current I1 of the inverter unit 12 is decomposed into the V/f control axis, that is, the γδ axis, (I1) ² =(iγ) ² +(iδ) ² as shown in FIG. The output voltage V1 on the V/f control axis is corrected by performing PI (Proportional-Integral) control so that Expression (6) is satisfied, that is, ΔQ=0.

FIG. 6 is a diagram showing a system configuration including a block diagram showing details of the inverter device according to the technique described above.
FIG. 6 shows the inverter section 12, the PMSM 13, the voltage compensation amount command calculation section 14, the reactive power estimation section 15, and the adder 16. As shown in FIG.
The inverter section 12 shown in FIG. 6 includes a coordinate converter 120, a two-to-three phase converter 122, a PWM circuit and inverter 123, and a current detector .

Here, the output voltage phase θvf shown in FIG. 1 is estimated by an output voltage phase estimator (not shown) by integrating the frequency command fcmd as shown in the following equation (7).
Note that the output voltage phase θvf is the phase difference between the αβ axis and the γδ axis.

where fcmd=ω/2π is the frequency command and s is the Laplace operator.

Next, the α-axis voltage command vα and the β-axis voltage command vβ on the fixed coordinate axes are calculated from the output voltage phase θvf and the γ-axis voltage command vγ of the rotating coordinates on the V/f control axis and the voltage command vδ of the δ-axis, Based on this, a three-phase voltage command vuvw is generated, and gate signals for the PWM circuit and the inverter 123 are generated.

The γ-δ axis voltage command, that is, the corrected δ-axis voltage command vδ′ and the γ-axis voltage command vγ are input to the coordinate converter 120, and the coordinate converter 120 performs coordinate conversion to convert α- It outputs the β-axis voltage command, that is, the α-β-axis voltage command vαβ.
The α-β axis voltage command vαβ includes an α-axis voltage command vα and a β-axis voltage command vβ.

The α-β axis voltage command vαβ is input to the two-to-three-phase converter 122, and the two-to-three-phase converter 122 converts the α-β-axis voltage command vαβ into a three-phase coordinate system, Outputs the phase voltage command vuvw.

A three-phase voltage command vuvw is input to the PWM circuit and inverter 123, and the PWM circuit and inverter 123 compares the three-phase voltage command vuvw with a triangular carrier signal, and based on the result of this comparison, the PWM circuit And by generating a gate signal for turning on and off the switching element in the inverter 123 , a three-phase voltage with a predetermined amplitude and a predetermined frequency is output to the PMSM 13 .

The current detector 124 is arranged between the output side of the PWM circuit/inverter 123 and the PMSM 13, detects a three-phase current, and outputs a three-phase current detection value iuvw.

The reactive power estimator 15 includes a three-phase to two-phase converter 150 , multipliers 151 , 152 and 155 , an adder 153 , a subtractor 156 and a multiplier 154 .
The output voltage phase θvf and the three-phase current detection value iuvw are input to the three-phase-to-two-phase converter 150, and the three-phase-to-two-phase converter 150 performs coordinate transformation of the three-phase current detection value iuvw on the γδ axis. , to output a γ-axis current estimate value iγ and a δ-axis current estimate value iδ.
Here, coordinate transformation is performed by the following equations (8) and (9).

The multiplier 151 squares the input δ-axis current estimated value iδ and outputs it.
The multiplier 152 squares the input γ-axis current estimated value iγ and outputs it.
Adder 153 adds the output of multiplier 152 to the output of multiplier 151 .
A multiplier 154 multiplies the output of the adder 153 by the rotational angular frequency command ω and the q-axis inductance value Lq of the motor.
A multiplier 155 multiplies the corrected δ-axis voltage command vδ′ by the γ-axis current estimated value iγ.
Subtractor 156 subtracts the output of multiplier 155 from the output of multiplier 154 and outputs the result.
That is, the reactive power estimator 15 calculates and outputs ΔQ of the above equation (6).

The voltage compensation amount command calculator 14 shown in FIG. 6 includes a PI controller 141 .

The output of the subtractor 156 is input to the PI controller 141, and the dq-axis reactive current target value Qdq*=0 is input to the PI controller 141 so that ΔQ in the above equation (6) becomes 0. , to output a voltage compensation amount command Vcmd.
Then, the adder 16 adds the voltage compensation amount command Vcmd to the δ-axis voltage command vδ and outputs the δ-axis post-correction voltage command vδ′, thereby realizing feedback control.
Instead of the PI controller 141, a P controller that performs P (Proportional) calculation or a PID controller that performs PID (Proportional-Integral-Differential) calculation may be provided.

As described above, it is said that the maximum torque and the maximum current can be obtained in the configuration shown in FIG. 6 by controlling the d-axis direction current id=0.
In the configuration described above, the reactive power Qdq on the control axis of the motor is expressed using the q-axis inductance value Lq.
However, the q-axis inductance value Lq is susceptible to magnetic saturation.
In particular, in an IPMSM (Interior Permanent Magnet Synchronous Motor), which is an embedded synchronous motor in which magnets are embedded in the rotor, the q-axis inductance value Lq fluctuates significantly depending on the load current.
As a result, an error occurs between the set value of the q-axis inductance value Lq in the above equation (4) or the reactive power estimator 15 shown in FIG. There is a possibility that the estimation accuracy will be lowered and the accuracy of high-efficiency control of the inverter device will be lowered accordingly.

Therefore, in the present embodiment, by estimating the fluctuation of the q-axis inductance value Lq from the fluctuation of the δ-axis inductance on the V/f control axis and correcting the q-axis inductance value Lq, the q-axis inductance Bring the value Lq closer to the true value.

FIG. 7 is a diagram showing the voltage compensation amount command calculator 14, the reactive power estimator 15a, and the inductance value compensator 20 provided in the inverter device according to the present embodiment.
Among the configurations shown in FIG. 7, the same components as those shown in FIG. 6 are denoted by the same reference numerals as in FIG.
The reactive power estimation unit 15a shown in FIG. 7 differs from the reactive power estimation unit 15 only in that a multiplier 211 is added between the output of the adder 153 and the input of the multiplier 154, and the rest of the configuration is the same. be.
The inductance value compensator 20 shown in FIG. 7 includes a differentiator 201 , multipliers 202 and 208 , multipliers 203 and 206 , subtractors 204 and 207 , and dividers 205 , 209 and 210 .

First, during pre-operation with no load, the no-load δ-axis inductance value Lδ0 is calculated by the following configuration and operation.
The output voltage phase θvf is input to the differentiator 201, and the differentiator 201 differentiates the output voltage phase θvf to output the rotational angular frequency ωf.
Note that the rotational angular frequency ωf may be replaced with the rotational angular frequency command ω.
The rotation angular frequency ωf and the no-load γ-axis current estimated value iγ0 are input to the multiplier 202, and the multiplier 202 outputs a value obtained by multiplying the rotation angular frequency ωf by the no-load γ-axis current estimated value iγ0. .
The no-load γ-axis current estimated value iγ0 is input to the multiplier 203, and the multiplier 203 outputs a value obtained by multiplying the no-load γ-axis current estimated value iγ0 by the winding resistance value R.
A subtractor 204 subtracts the no-load γ-axis voltage command vγ0 from the output of the multiplier 203 and outputs the result.
The output of the multiplier 202 and the output of the subtractor 204 are input to the divider 205, and the divider 205 divides the output of the subtractor 204 by the output of the multiplier 202, and outputs the no-load δ-axis inductance value Lδ0. do.

Here, the no-load δ-axis inductance value Lδ0 is stored in advance in a storage unit (not shown) as a parameter for preliminary operation performed with no load.
Next, during load operation, the δ-axis inductance value Lδ is calculated by the following configuration and operation.
The γ-axis current estimate value iγ is input to the multiplier 206, and the multiplier 206 outputs a value obtained by multiplying the γ-axis current estimate value iγ by the winding resistance value R.
Subtractor 207 outputs a value obtained by subtracting γ-axis voltage command vγ from the output of multiplier 206 .
The rotation angular frequency ωf and the δ-axis current estimated value iδ are input to the multiplier 208, and the multiplier 208 outputs a value obtained by multiplying the rotation angular frequency ωf by the δ-axis current estimated value iδ.
The output of the subtractor 207 and the output of the multiplier 208 are input to the divider 209, and the divider 209 divides the output of the subtractor 207 by the output of the multiplier 208 to output the δ-axis inductance value Lδ.
Further, the divider 210 receives the δ-axis inductance value Lδ output from the divider 209 and the no-load δ-axis inductance value Lδ0 output from the divider 205 (the value stored in the storage unit). A divider 210 divides the output of the divider 209 by the output of the divider 205 to output the rate of change of the δ-axis inductance from no load.
The output of the adder 153 and the output of the divider 210 are input to the multiplier 211 , and the multiplier 211 outputs a value obtained by multiplying the output of the adder 153 by the output of the divider 210 .

Next, when the δ-axis current estimated value iδ under a predetermined load is set as the δ-axis current estimated value iδ1 under a predetermined load, and the δ-axis inductance value Lδ at this time is set as the δ-axis inductance value Lδ1 under a predetermined load, The time .delta.-axis inductance value L.delta.1 is obtained using the parameters during operation under this predetermined load and the following equation (10).

Here, the variation rate of the δ-axis inductance value Lδ is Lδ1/Lδ0.
Then, in the above equation (6), the no-load q-axis inductance value Lq0 is set to a nominal value, and q The shaft inductance value Lq is corrected.

Here, the reason why the variation rate of the δ-axis inductance value Lδ is used to correct the q-axis inductance value Lq as described above will be explained.
First, the q-axis inductance value Lq is represented by the following equation (11).

A roughly proportional relationship is established between the magnitude of the γ-axis current estimated value iγ in the above equation (10) and the magnitude of the d-axis current estimated value id in the above equation (11).
Similarly, a roughly proportional relationship is established between the magnitude of the γ-axis voltage command vγ and the magnitude of the d-axis voltage command vd.
Similarly, a roughly proportional relationship is established between the magnitude of the δ-axis current estimated value iδ and the magnitude of the q-axis current estimated value iq.
Therefore, a substantially proportional relationship is established between the magnitude of the δ-axis inductance value Lδ and the magnitude of the q-axis inductance value Lq with respect to changes in the load.
Therefore, the variation rate of the δ-axis inductance value Lδ can be used to correct the q-axis inductance value Lq.

As described above, according to the inverter device according to the present embodiment, it is possible to perform correction according to fluctuations in the q-axis inductance value Lq.
Therefore, according to the inverter device according to the present embodiment, it is possible to be robust against magnetic saturation and operate with high efficiency.

In addition, the present invention is not limited to the above-described embodiment, and includes various modifications in which components are added, deleted, or converted to the above-described configuration.

11 voltage command calculation unit 12 inverter unit 120 coordinate converter 122 two-phase three-phase converter 123 PWM circuit and inverter 124 current detector 13 PMSM
14 voltage compensation amount command calculator 141 PI controllers 15 and 15a reactive power estimator 150 three-phase two-phase converter 151, 152, 155 multiplier 153 adder 154 multiplier 156 subtractor 16 adder 20 inductance value compensator 201 Differentiator 202, 203, 206, 208 Multiplier 204, 207 Subtractor 205, 209, 210 Divider 211 Multiplier

Claims (2)

An inverter device that drives a permanent magnet synchronous motor by V/f control,
generating a γ-axis voltage command and a δ-axis voltage command based on the rotation angular frequency command;
Calculate the voltage compensation amount command by calculating ΔQ calculated by the following formula (6) by any one of P calculation, PI calculation and PID calculation,
The γ-axis voltage command and the corrected δ-axis voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command are coordinate-transformed into an α-axis voltage command vα and a β-axis voltage command vβ. death,
generating a gate signal for turning on and off a switching element based on the α-axis voltage command vα and the β-axis voltage command vβ;
The q-axis inductance value Lq in the above equation ( 6) is calculated using the δ-axis inductance calculated by the following equation (10). An inverter device that is a corrected value by multiplying the inductance value Lq0 .

where ω: rotation angular frequency command, iγ: γ-axis current estimate, iδ: δ-axis current estimate, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , L.delta.: the .delta.-axis inductance value of the permanent magnet synchronous motor, and R: the winding resistance value. A control method for an inverter device that drives a permanent magnet synchronous motor by V/f control,
generating a γ-axis voltage command and a δ-axis voltage command based on the rotation angular frequency command;
calculating a voltage compensation amount command by calculating ΔQ calculated by the following formula (6) by any one of P calculation, PI calculation and PID calculation;
The γ-axis voltage command and the corrected δ-axis voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command are coordinate-transformed into an α-axis voltage command vα and a β-axis voltage command vβ. and generating a gate signal for turning on and off a switching element based on the α-axis voltage command vα and the β-axis voltage command vβ,
The q-axis inductance value Lq in the above equation ( 6) is calculated using the δ-axis inductance calculated by the following equation (10). A control method for an inverter device that is a corrected value by multiplying an inductance value Lq0 .

where ω: rotation angular frequency command, iγ: γ-axis current estimate, iδ: δ-axis current estimate, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , L.delta.: the .delta.-axis inductance value of the permanent magnet synchronous motor, and R: the winding resistance value.

Images