JP2021048740A - Inverter device and method for controlling inverter device - Google Patents

Abstract

To provide a technique capable of making an inverter device for driving a permanent magnet synchronous motor by V/f control robust for magnetic saturation and operating the inverter device with high efficiency.SOLUTION: An inverter device 12 for driving a permanent magnet synchronous motor 13 by V/f control: generates a γ-axis voltage command and a δ-axis voltage command on the basis of a rotational angle frequency command; calculates a voltage compensation amount command by calculating ΔQ by any one of P operation, PI operation, and PID operation; coordinate-converts the γ-axis voltage command and a δ-axis post-correction voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command to a α-axis voltage command vα and a β-axis voltage command vβ; and generates a gate signal which turns on/off a switching element on the basis of the α-axis voltage command vα and the β-axis voltage command vβ. A q-axis inductance value Lq is a value corrected by multiplying a fluctuation rate of the δ-axis inductance from a no-load time.SELECTED DRAWING: Figure 6

Description

The present invention relates to an inverter device and a control method for the inverter device.

In general, a PM (Permanent Magnet) motor uses a permanent magnet as a rotor, so that an exciting current and a secondary current are not generated, and it is possible to operate with higher efficiency than an induction motor.
In particular, PMSM (Permanent Magnet Synchronous Motor), which is a permanent magnet synchronous motor, is widely used.
Typical examples of the inverter device for controlling the PMSM include vector control and V / f control.
Here, vector control requires rotor position information obtained by the rotor position sensor, but the rotor position sensor is easily affected by heat and vibration and has low environmental resistance.
In addition, the installation of the rotor position sensor leads to an increase in cost.

On the other hand, V / f control is known as a simple control method for PMSM that does not require rotor position information.
V / f control is a control method in which the ratio of the amplitude of the output voltage of the inverter device to the frequency is constant.
In V / f control, the amount of calculation is smaller than in vector control.
Therefore, the control unit of the inverter device can be realized by a low-cost microcomputer.

However, in V / f control, since the rotor position is not detected, it is difficult to properly control the current phase during load operation, and there is a problem that reactive power is generated and efficiency is lowered.

Non-Patent Document 1, which is an example of the prior art, provides a technique capable of matching the inverter output voltage with the q-axis, which is the motor control axis, and suppressing the generation of invalid power by appropriately controlling the voltage correction amount. It is disclosed.

Junichi Ito, Jiro Toyosaki, Hiroshi Osawa, "High Performance V / f Control of Permanent Magnet Synchronous Motors", IEEJ Journal D, 2003, Vol. 122, No. 3, p. 253-259

However, in the above-mentioned conventional technique, there is a problem that the inductance of the q-axis fluctuates due to the influence of magnetic saturation during load operation, and the estimation accuracy of the reactive power is lowered.

The present invention has been made in view of the above, and provides a technique for making an inverter device for driving a permanent magnet synchronous motor by V / f control robust against magnetic saturation and capable of operating with high efficiency. The purpose is to do.

The present invention that solves the above-mentioned problems and achieves the object is an inverter device that drives a permanent magnet synchronous electric motor by V / f control, and has a γ-axis voltage command and a δ-axis voltage based on a rotation angle frequency command. A command is generated, and the ΔQ calculated by the following equation (6) is calculated by any one of P calculation, PI calculation, and PID calculation to calculate the voltage compensation amount command. The corrected voltage command of the δ axis obtained by adding the voltage compensation amount command to the voltage command of the δ axis is coordinate-converted into the voltage command vα of the α axis and the voltage command vβ of the β axis, and the voltage command of the α axis is converted. A gate signal for turning on and off the switching element is generated based on vα and the voltage command vβ of the β-axis, and the q-axis inductance value Lq of the above equation (6) uses the δ-axis inductance calculated by the following equation (10). It is an inverter device which is a value corrected by multiplying the fluctuation rate of the δ-axis voltage from the time of no load calculated by the above.

However, ω: rotation angle frequency command, iγ: γ-axis current estimated value, iδ: δ-axis current estimated value, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , Lδ: δ-axis inductance value of the permanent magnet synchronous motor, R: winding resistance value.

Alternatively, the present invention is a control method for an inverter device that drives a permanent magnet synchronous electric motor by V / f control, and generates a γ-axis voltage command and a δ-axis voltage command based on a rotation angle frequency command. To calculate the voltage compensation amount command by calculating ΔQ calculated by the following equation (6) by any one of P calculation, PI calculation and PID calculation, the voltage command of the γ axis and the voltage command of the δ axis. The δ-axis corrected voltage command obtained by adding the voltage compensation amount command to the voltage command is coordinate-converted into the α-axis voltage command vα and the β-axis voltage command vβ, and the α-axis voltage command vα and The q-axis inductance value Lq of the above equation (6) includes the generation of a gate signal for turning on / off the switching element based on the β-axis voltage command vβ, and the q-axis inductance value Lq of the equation (6) is the δ-axis inductance calculated by the following equation (10). This is a control method for an inverter device, which is a value corrected by multiplying the fluctuation rate of the δ-axis voltage from no load, which is calculated using the above.

However, ω: rotation angle frequency command, iγ: γ-axis current estimated value, iδ: δ-axis current estimated value, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , Lδ: δ-axis inductance value of the permanent magnet synchronous motor, R: winding resistance value.

According to the present invention, the inverter device that drives the permanent magnet synchronous motor by V / f control can be operated with high efficiency while being robust against magnetic saturation.

FIG. 1 is a diagram showing a coordinate system of PMSM. FIG. 2 is a diagram showing the relationship between the output voltage output by the inverter device in V / f control and the dq-axis direction of the motor. FIG. 3 is a diagram showing the relationship between the output voltage output by the inverter device in V / f control, the δ axis, the γ axis, and the dq axis of the motor. FIG. 4 is a diagram showing the relationship between the corrected output voltage output by the inverter device in V / f control, the δ axis, the γ axis, and the dq axis of the motor. FIG. 5 is a diagram showing a system configuration including an inverter device of the prior art. FIG. 6 is a diagram showing a system configuration including a block diagram showing details of the inverter device according to the technique described above. FIG. 7 is a diagram showing a voltage compensation amount command calculation unit, a reactive power estimation unit, and an inductance value compensation unit included in the inverter device according to the present embodiment.

Hereinafter, embodiments for carrying out the present invention will be described with reference to the accompanying drawings.
However, the present invention is not limited to the description of the following embodiments.

<Embodiment 1>
FIG. 1 is a diagram showing a coordinate system of PMSM.
In the coordinate system shown in FIG. 1, the space vector of the phase voltage is set to face in the δ-axis direction during the operation of the PMSM.

In the control of PMSM, the magnetic flux generation direction of the magnet provided in the rotor is defined as the d-axis direction, and the generation direction of the induced voltage orthogonal to the d-axis is defined as the q-axis direction.
Here, the q-axis direction current iq flowing in the q-axis direction contributes to torque, so that active power is generated, and the d-axis direction current id flowing in the q-axis direction orthogonal to the q-axis direction current iq generates ineffective power.
Therefore, if the control is performed so that the d-axis direction current id = 0, the maximum torque and the maximum current in the V / f control can be obtained, and the copper loss of the motor can be minimized.

FIG. 2 is a diagram showing the relationship between the output voltage V1 output by the inverter device in V / f control and the dq-axis direction of the motor.
In FIG. 2, the voltage phase output by the inverter is used as a reference, the axis of the voltage output by the inverter is the δ axis, the axis in the direction orthogonal to this is the γ axis, and thereafter, this γδ axis is referred to as the V / f control axis. Refer to.
At this time, as shown in FIG. 2, the dq axis, which is the motor control axis based on the rotor position of the motor, and the V / f control axis are different.
That is, the direction of the output voltage V1 output by the inverter device does not match the direction of the q-axis of the motor.
Therefore, since the d-axis direction current id ≠ 0, the current flows in the d-axis direction, which increases the copper loss of the motor, which generates ineffective power and lowers the power efficiency of the inverter device.
Here, the phase difference between the dq axis and the V / f control axis increases according to the magnitude of the load.
Therefore, the decrease in power efficiency at the time of high load becomes a particular problem.

FIG. 3 is a diagram showing the relationship between the output voltage V1 output by the inverter device in V / f control, the δ axis, the γ axis, and the dq axis of the motor.
The output current I1 flows at a predetermined load angle shown in FIG.
When the output current I1 shown in FIG. 3 is decomposed into the coordinate system of the dq axis, the current id in the d-axis direction flows in the d-axis direction, so that invalid power is generated.

FIG. 4 is a diagram showing the relationship between the corrected output voltage output by the inverter device in V / f control, the δ axis, the γ axis, and the dq axis of the motor.
Here, in order to suppress the reactive power due to the current id in the d-axis direction, the voltage compensation amount ΔV is added to the output voltage V1, and the output voltage V2 = V1 + ΔV after the compensation amount is added.
Then, the current flowing by the output voltage V2 after the compensation amount is added is defined as the output current I2.
The output current I2 coincides with the q-axis, which is the control axis of the motor.

However, in V / f control, since the rotor position is not detected, it is difficult to properly control the current phase during load operation, and there is a problem that reactive power is generated and efficiency is lowered.

Therefore, in Non-Patent Document 1, which is an example of the above-mentioned prior art, in a system in which the PMSM is operated with the desired rotation angular frequency command ω, the voltage command Vδ on the δ axis is corrected according to the estimated reactive power values Qdq and Qγδ. By doing so, the efficiency is improved.

FIG. 5 is a diagram showing a system configuration including an inverter device of the prior art.
FIG. 5 shows a voltage command calculation unit 11, an inverter unit 12, a PMSM 13, a voltage compensation amount command calculation unit 14, a reactive power estimation unit 15, and an adder 16.

A rotation angular frequency command ω from a control unit (not shown) is input to the voltage command calculation unit 11, and the voltage command calculation unit 11 outputs a voltage command vδ on the δ axis and a voltage command vγ on the γ axis.
Here, in general, in V / f control, the voltage command vγ on the γ axis is set to vγ = 0, and the voltage command vδ on the δ axis is set to a value proportional to the rotation angular frequency command ω.

The corrected voltage command vδ'of the δ axis and the voltage command vγ of the γ axis are input to the inverter unit 12, and the inverter unit 12 outputs a three-phase voltage to the PMSM 13.
Here, the corrected voltage command vδ'on the δ axis is the sum of the voltage command vδ on the δ axis and the voltage compensation amount command Vcmd by the adder 16.
The inverter unit 12 generates a three-phase voltage command vuvw based on the corrected voltage command vδ'on the δ axis and the voltage command vγ on the γ axis, compares the three-phase voltage command vuvw with the triangular wave-shaped carrier signal, and compares the three-phase voltage command vuvw with the triangular wave-shaped carrier signal. By generating a gate signal for turning on and off the switching element in the inverter unit 12 based on the result of this comparison, a three-phase voltage having a predetermined amplitude and a predetermined frequency is output to the PMSM 13.

Generally, in motor control, a two-phase coordinate system, which is a rotating coordinate system in which a symmetric three-phase AC is converted into an equivalent two-phase AC, is used for simplification of control.

Here, using the winding resistance value R, the d-axis inductance value Ld of the motor, the q-axis inductance value Lq of the motor, the differential operator p, the induced voltage constant Ke, and the rotation angular frequency command ω, PMSM13 has the following equation ( It is represented by the voltage equation of 1).

The ineffective power Qdq on the dq axis, which is the motor control axis, is represented by the following equation (2).

Substituting the above equation (1) into the above equation (2) gives the following equation (3).

Here, if the current in the d-axis direction is controlled to be id = 0, the maximum torque and the maximum current can be obtained. Therefore, if id = 0 and the differential operator p = 0 in consideration of the operation in the steady state, the following Equation (4) is obtained.

Here, assuming that the output current of the inverter unit 12 is I1, I1 = (iq) ² when the current is controlled in the d-axis direction id = 0.

Next, the ineffective power Qγδ in the V / f control axis is calculated.
Similarly, assuming that iδ = 0 in the V / f control axis, the ineffective power Qγδ of the V / f control axis is expressed by the following equation (5).

Then, in order to obtain the maximum torque and the maximum current in the V / f control shaft, the above equations (4) and (5) may be matched.
Here, when the output current I1 of the inverter unit 12 is decomposed into the V / f control axis, that is, the γδ axis, (I1) ² = (iγ) ² + (iδ) ² as shown in FIG. The output voltage V1 on the V / f control axis is corrected by performing PI (Proportional-Integral) control so as to satisfy the equation (6), that is, so that ΔQ = 0.

FIG. 6 is a diagram showing a system configuration including a block diagram showing details of the inverter device according to the technique described above.
FIG. 6 shows an inverter unit 12, a PMSM 13, a voltage compensation amount command calculation unit 14, an reactive power estimation unit 15, and an adder 16.
The inverter unit 12 shown in FIG. 6 includes a coordinate converter 120, a two-phase three-phase converter 122, a PWM circuit, an inverter 123, and a current detector 124.

Here, as shown in the following equation (7), the output voltage phase θvf shown in FIG. 1 is estimated by an output voltage phase estimator (not shown) by integrating the frequency command fcmd.
The output voltage phase θvf is the phase difference between the αβ axis and the γδ axis.

Here, fcmd = ω / 2π is a frequency command, and s is a Laplace operator.

Next, the α-axis voltage command vα and the β-axis voltage command vβ on the fixed coordinate axes are calculated from the output voltage phase θvf and the γ-axis voltage command vγ of the rotation coordinates on the V / f control axis and the voltage command vδ of the δ-axis. Based on this, the three-phase voltage command vuvw is generated, and the gate signal of the PWM circuit and the inverter 123 is generated.

A voltage command on the γ-δ axis, that is, a corrected voltage command vδ'on the δ axis and a voltage command vγ on the γ axis are input to the coordinate converter 120, and the coordinate converter 120 performs coordinate conversion to perform α-. The β-axis voltage command, that is, the α-β-axis voltage command vαβ is output.
The α-β-axis voltage command vαβ includes the α-axis voltage command vα and the β-axis voltage command vβ.

The α-β-axis voltage command vαβ is input to the two-phase three-phase converter 122, and the two-phase three-phase converter 122 converts the α-β-axis voltage command vαβ into a three-phase coordinate system. The phase voltage command vuvw is output.

A three-phase voltage command vuvw is input to the PWM circuit and the inverter 123, and the PWM circuit and the inverter 123 compare the three-phase voltage command vuvw with the triangular wave-shaped carrier signal, and the PWM circuit is based on the result of the comparison. By generating a gate signal for turning on and off the switching element in the inverter 123, a three-phase voltage having a predetermined amplitude and a predetermined frequency is output to the PMSM 13.

The current detector 124 is arranged between the output side of the PWM circuit and the inverter 123 and the PMSM 13, detects the three-phase current, and outputs the three-phase current detection value iuvw.

The ineffective power estimation unit 15 includes a three-phase two-phase converter 150, a multiplier 151, 152, 155, an adder 153, a subtractor 156, and a multiplier 154.
The output voltage phase θvf and the three-phase current detection value iuvw are input to the three-phase two-phase converter 150, and the three-phase two-phase converter 150 converts the three-phase current detection value iuvw into coordinates on the γδ axis. , Γ-axis current estimated value iγ and δ-axis current estimated value iδ are output.
Here, the coordinate transformation is performed by the following equations (8) and (9).

The multiplier 151 squares the input δ-axis current estimated value iδ and outputs it.
The multiplier 152 squares the input γ-axis current estimated value iγ and outputs it.
The adder 153 adds the output of the multiplier 152 to the output of the multiplier 151.
The multiplier 154 multiplies the output of the adder 153 by the rotation angular frequency command ω and the q-axis inductance value Lq of the motor.
The multiplier 155 multiplies the δ-axis corrected voltage command vδ'by the γ-axis current estimate iγ.
The subtractor 156 subtracts the output of the multiplier 155 from the output of the multiplier 154 and outputs it.
That is, the ineffective power estimation unit 15 calculates and outputs ΔQ of the above equation (6).

The voltage compensation amount command calculation unit 14 shown in FIG. 6 includes a PI controller 141.

The output of the subtractor 156 is input to the PI controller 141, and the dq-axis reactive current target value Qdq * = 0 is input to the PI controller 141 so that ΔQ in the above equation (6) becomes 0. , Outputs the voltage compensation amount command Vcmd.
Then, the adder 16 adds the voltage compensation amount command Vcmd to the voltage command vδ on the δ axis and outputs the corrected voltage command vδ'on the δ axis, thereby realizing feedback control.
In addition, instead of the PI controller 141, a P controller that performs P (Proportional) calculation or a PID controller that performs PID (Proportional-Integral-Differential) calculation may be provided.

As described above, it is said that the maximum torque and the maximum current can be obtained in the configuration shown in FIG. 6 by controlling the current id in the d-axis direction to be 0.
In the above configuration, the reactive power Qdq on the control shaft of the motor is expressed using the q-axis inductance value Lq.
However, the q-axis inductance value Lq is susceptible to magnetic saturation.
In particular, in the IPMSM (Interior Permanent Magnet Synchronous Motor), which is an embedded synchronous motor in which a magnet is embedded in a rotor, the q-axis inductance value Lq fluctuates remarkably depending on the load current.
As a result, an error occurs between the set value and the true value of the q-axis inductance value Lq in the reactive power estimation unit 15 shown in the above equation (4) or FIG. There is a risk that the estimation accuracy will decrease and the accuracy of high-efficiency control of the inverter device will decrease accordingly.

Therefore, in the present embodiment, the fluctuation of the q-axis inductance value Lq is estimated from the fluctuation of the δ-axis inductance in the V / f control axis, and the q-axis inductance value Lq is corrected to correct the q-axis inductance used for estimating the reactive power. Brings the value Lq closer to the true value.

FIG. 7 is a diagram showing a voltage compensation amount command calculation unit 14, an ineffective power estimation unit 15a, and an inductance value compensation unit 20 included in the inverter device according to the present embodiment.
Among the configurations shown in FIG. 7, the same configurations as those shown in FIG. 6 are designated by the same reference numerals as those in FIG.
The negative power estimation unit 15a shown in FIG. 7 is different from the negative power estimation unit 15 only in that the multiplier 211 is added between the output of the adder 153 and the input of the multiplier 154, and the other configurations are the same. is there.
The inductance value compensating unit 20 shown in FIG. 7 includes a differentiator 201, multipliers 202 and 208, multipliers 203 and 206, subtractors 204 and 207, and dividers 205, 209 and 210.

First, in the pre-operation performed with no load in advance, the δ-axis inductance value Lδ0 with no load is calculated by the configuration and operation shown below.
The output voltage phase θvf is input to the differentiator 201, and the differentiator 201 differentiates the output voltage phase θvf and outputs the rotation angular frequency ωf.
The rotation angular frequency ωf may be replaced with the rotation angular frequency command ω.
The rotation angular frequency ωf and the unloaded γ-axis current estimated value iγ0 are input to the multiplier 202, and the multiplier 202 outputs the value obtained by multiplying the rotational angular frequency ωf by the unloaded γ-axis current estimated value iγ0. ..
The unloaded γ-axis current estimated value iγ0 is input to the multiplier 203, and the multiplier 203 outputs a value obtained by multiplying the unloaded γ-axis current estimated value iγ0 by the winding resistance value R.
The subtractor 204 subtracts the γ-axis voltage command vγ0 at no load from the output of the multiplier 203 and outputs it.
The output of the multiplier 202 and the output of the subtractor 204 are input to the divider 205, and the divider 205 outputs the unloaded δ-axis inductance value Lδ0 obtained by dividing the output of the subtractor 204 by the output of the multiplier 202. To do.

Here, the δ-axis inductance value Lδ0 under no load is stored in a storage unit (not shown) as a parameter during pre-operation performed with no load in advance.
Next, during load operation, the δ-axis inductance value Lδ is calculated according to the configuration and operation shown below.
The γ-axis current estimated value iγ is input to the multiplier 206, and the multiplier 206 outputs a value obtained by multiplying the γ-axis current estimated value iγ by the winding resistance value R.
The subtractor 207 outputs a value obtained by subtracting the γ-axis voltage command vγ from the output of the multiplier 206.
The rotation angular frequency ωf and the δ-axis current estimated value iδ are input to the multiplier 208, and the multiplier 208 outputs a value obtained by multiplying the rotation angular frequency ωf by the δ-axis current estimated value iδ.
The output of the subtractor 207 and the output of the multiplier 208 are input to the divider 209, and the divider 209 outputs the δ-axis inductance value Lδ obtained by dividing the output of the subtractor 207 by the output of the multiplier 208.
Further, the δ-axis inductance value Lδ which is the output of the divider 209 and the δ-axis inductance value Lδ0 (value stored in the storage unit) at no load which is the output of the divider 205 are input to the divider 210. The divider 210 divides the output of the divider 209 by the output of the divider 205 and outputs the fluctuation rate of the δ-axis inductance from the time of no load.
The output of the adder 153 and the output of the divider 210 are input to the multiplier 211, and the multiplier 211 outputs a value obtained by multiplying the output of the adder 153 by the output of the divider 210.

Next, if the δ-axis current estimated value iδ under a predetermined load is set as the δ-axis current estimated value iδ1 under a predetermined load and the δ-axis inductance value Lδ at this time is set as the δ-axis inductance value Lδ1 under a predetermined load, the predetermined load is set. The hour δ-axis inductance value Lδ1 is acquired by using the parameters during operation under this predetermined load and the following equation (10).

Here, the volatility of the δ-axis inductance value Lδ is Lδ1 / Lδ0.
Then, in the above equation (6), the q-axis inductance value Lq0 under no load is set as a nominal value, and the fluctuation rate of the δ-axis inductance value Lδ is multiplied according to the change in the δ-axis current estimated value iδ to q. The shaft inductance value Lq is corrected.

Here, as described above, the reason why the volatility of the δ-axis inductance value Lδ is used for the correction of the q-axis inductance value Lq will be described.
First, the q-axis inductance value Lq is expressed by the following equation (11).

A proportional relationship is generally established between the magnitude of the γ-axis current estimated value iγ in the above equation (10) and the magnitude of the d-axis current estimated value id in the above equation (11).
Similarly, a proportional relationship is generally established between the magnitude of the γ-axis voltage command vγ and the magnitude of the d-axis voltage command vd.
Similarly, a proportional relationship is generally established between the magnitude of the δ-axis current estimated value iδ and the magnitude of the q-axis current estimated value iq.
Therefore, a proportional relationship is generally established between the magnitude of the δ-axis inductance value Lδ and the magnitude of the q-axis inductance value Lq with respect to the change in load.
Therefore, the volatility of the δ-axis inductance value Lδ can be used to correct the q-axis inductance value Lq.

As described above, according to the inverter device according to the present embodiment, it is possible to make a correction according to the fluctuation of the q-axis inductance value Lq.
Therefore, according to the inverter device according to the present embodiment, it is possible to operate robustly against magnetic saturation and with high efficiency.

The present invention is not limited to the above-described embodiment, and includes various modifications in which components are added, deleted, or converted from the above-described configuration.

11 Voltage command calculation unit 12 Inverter unit 120 Coordinate converter 122 Two-phase three-phase converter 123 PWM circuit and inverter 124 Current detector 13 PMSM
14 Voltage Compensation Amount Command Calculation Unit 141 PI Controller 15, 15a Invalid Power Estimator 150 Three-Phase Two-Phase Converter 151, 152, 155 Multiplier 153 Adder 154 Multiplier 156 Adder 16 Adder 20 Inductivity Value Compensator 201 Differentiator 202,203,206,208 Multiplier 204,207 Subtractor 205,209,210 Divider 211 Multiplier

Claims (2)

An inverter device that drives a permanent magnet synchronous motor by V / f control.
Generates a γ-axis voltage command and a δ-axis voltage command based on the rotation angular frequency command.
The voltage compensation amount command is calculated by calculating ΔQ calculated by the following equation (6) by any one of P calculation, PI calculation and PID calculation.
Coordinate conversion of the γ-axis voltage command and the δ-axis corrected voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command into an α-axis voltage command vα and a β-axis voltage command vβ. And
A gate signal for turning on / off the switching element is generated based on the α-axis voltage command vα and the β-axis voltage command vβ.
The q-axis inductance value Lq of the above equation (6) was corrected by multiplying the volatility of the δ-axis inductance from no load, which was calculated using the δ-axis inductance calculated by the following equation (10). Inductance device that is the value.

However, ω: rotation angle frequency command, iγ: γ-axis current estimated value, iδ: δ-axis current estimated value, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , Lδ: δ-axis inductance value of the permanent magnet synchronous motor, R: winding resistance value. It is a control method of an inverter device that drives a permanent magnet synchronous motor by V / f control.
Generating a γ-axis voltage command and a δ-axis voltage command based on the rotation angular frequency command,
To calculate the voltage compensation amount command by calculating ΔQ calculated by the following formula (6) by any one of P calculation, PI calculation and PID calculation.
Coordinate conversion of the γ-axis voltage command and the δ-axis corrected voltage command obtained by adding the voltage compensation amount command to the δ-axis voltage command into an α-axis voltage command vα and a β-axis voltage command vβ. And to generate a gate signal to turn the switching element on and off based on the α-axis voltage command vα and the β-axis voltage command vβ.
The q-axis inductance value Lq of the above equation (6) was corrected by multiplying the volatility of the δ-axis inductance from no load, which was calculated using the δ-axis inductance calculated by the following equation (10). The control method of the inverter device which is the value.

However, ω: rotation angle frequency command, iγ: γ-axis current estimated value, iδ: δ-axis current estimated value, vγ: γ-axis voltage command, vδ: δ-axis voltage command, Lq: q-axis inductance value of permanent magnet synchronous motor , Lδ: δ-axis inductance value of the permanent magnet synchronous motor, R: winding resistance value.

Images