JP7012931B2 - Inverter controller and motor drive system - Google Patents

Description

Embodiments of the present invention relate to an inverter control device and a motor drive system.

A method of estimating the angle of rotation is used in a rotation angle sensorless control device of a permanent magnet synchronous motor (PMSM) or a synchronous reluctance motor (SynRM). In the speed estimation, a method called PLL (Phase Locked Loop) is known in which the speed is estimated by performing PI control so that the evaluation index corresponding to the axis error becomes 0.

Further, in a speed sensorless control device of an induction motor (IM), a primary frequency estimator that calculates an estimated primary frequency from a q-axis secondary magnetic flux induced voltage and feedforwards it to a PI controller has been proposed.

Japanese Patent No. 3692085

"Sensorless vector control of AC drive system", Institute of Electrical Engineers of Japan, edited by the research committee on the arrangement of sensorless vector control, Ohmsha, released in September 2016

When a high-speed speed estimation response is required, there is a possibility of step-out unless the gain of the PI controller is increased in the PLL control. On the other hand, if the gain of the PI controller is increased, it becomes over-controlled and the control becomes oscillating.

On the other hand, in the permanent magnet synchronous motor (PMSM) and the synchronous reluctance motor (SynRM) as well as the induction motor (IM), the estimated speed calculated from the q-axis induced voltage is given by feed forward to the PI controller. It is possible to realize a high-speed speed response without increasing the gain of. However, in permanent magnet synchronous motors (PMSM) and synchronous reluctance motors (SynRM) that mainly utilize reluctance torque, it is difficult to calculate the estimated speed from the q-axis induced voltage because the q-axis induced voltage changes according to the load. Met.

An embodiment of the present invention has been made in view of the above circumstances, and an object of the present invention is to provide an inverter control device and a motor drive system that realize a high-speed speed response.

The inverter control device according to the embodiment is described so that the rotation speed and the rotation speed command value of the inverter, the current detector that detects the current response value output from the inverter, and the motor connected to the inverter are equal to each other. A speed control circuit including a first PI controller that calculates a torque command value of a motor, a current command value generation circuit that calculates a current command value corresponding to the current response value based on the torque command value, and a current command. A current control circuit that calculates the output voltage output from the inverter so that the value and the current response value are equal, and the difference between the rotation speed command value and the rotation speed approximate value is input to output the torque command estimated value. A second PI controller and a motor machine model circuit that calculates the rotational speed approximate value using the torque command estimated value are provided, and the gain used in the second PI controller is used in the first PI controller. A schematic speed estimation circuit equivalent to a gain and a speed estimation circuit for calculating a speed estimation value of the motor using the rotation speed approximate value are provided.

FIG. 1 is a block diagram schematically showing a configuration example of the inverter control device and the motor drive system of the first embodiment. FIG. 2 is a block diagram schematically showing a configuration example of a speed control circuit of the inverter control device shown in FIG. 1. FIG. 3 is a block diagram schematically showing a configuration example of the schematic speed estimation circuit shown in FIG. FIG. 4 is a block diagram schematically showing a configuration example of the speed / rotation phase angle estimation circuit shown in FIG. 1. FIG. 5 is a block diagram schematically showing a configuration example of the rotation phase angle error calculation circuit shown in FIG. FIG. 6 is a block diagram schematically showing another configuration example of the schematic speed estimation circuit shown in FIG. 1. FIG. 7 is a block diagram schematically showing another configuration example of the motor machine model shown in FIG. FIG. 8 is a block diagram schematically showing a configuration example of the inverter control device and the motor drive system of the second embodiment. FIG. 9 is a block diagram schematically showing a configuration example of the speed estimation circuit shown in FIG.

Hereinafter, the inverter control device and the motor drive system of a plurality of embodiments will be described with reference to the drawings.
FIG. 1 is a block diagram schematically showing a configuration example of the inverter control device and the motor drive system of the first embodiment.

The motor drive system of the present embodiment includes an inverter control device, an inverter 1, and a synchronous machine 2.
The synchronous machine 2 is, for example, a synchronous reluctance motor (SynRM) including a stator and a rotor. The synchronous machine 2 is an AC motor driven by a three-phase alternating current supplied from the inverter 1, generates a magnetic field by the three-phase alternating current flowing in each exciting phase, and generates torque by magnetic interaction with the rotor. be.

The inverter 1 includes, for example, an inverter main circuit (none of which is shown) including a DC power supply (DC load), two power semiconductor elements for each of U-phase, V-phase, and W-phase. .. The two power semiconductor elements in each phase are connected in series between a DC line (DC link) connected to the positive electrode of the DC power supply and a DC line (DC link) connected to the negative electrode of the DC power supply. The operation of the power semiconductor element of the inverter 1 is controlled by a gate command received from an inverter control device described later. The inverter 1 is a three-phase AC inverter that outputs U-phase current iu, V-phase current iv, and W-phase current iwa to the synchronous machine 2 which is an AC load. Further, the inverter 1 can also charge the electric power generated by the synchronous machine 2 to the DC power source.

The inverter control device of this embodiment may be configured by hardware, may be configured by software, or may be configured by combining hardware and software. The inverter controller may include, for example, at least one processor and a memory in which a program executed by the processor is recorded.

The inverter control device of this embodiment includes a current detector 3, a PWM modulation circuit 22, a coordinate conversion (UVW / dcqc) circuit 23, a coordinate conversion (dcqc / UVW) circuit 24, a current control circuit 25, and a speed. A rotation phase angle estimation circuit 26, a current command generation circuit 27, a speed control circuit 28, and an approximate speed estimation circuit 29 are provided.

The current detector 3 has a current response value of at least two phases (for example, U phase) among the three-phase AC currents (U-phase current value iu, V-phase current value iv, W-phase current value iwa) output from the inverter 1. The current value iu and the W phase current value iw) are detected. The current response value detected by the current detector 3 is input to the coordinate conversion (UVW / dcqc) circuit 23.

The coordinate conversion (UVW / dcqc) circuit 23 is a vector conversion circuit that performs coordinate conversion from a three-phase fixed coordinate system to a dcqc axis rotating coordinate system using the rotation phase angle estimated value θest. The coordinate conversion (UVW / dcqc) circuit 23 converts the U-phase current value iu and the W-phase current value iw of the three-phase fixed coordinate system into the dc-axis current value idc and the qc-axis current value iqc of the dcqc-axis rotating coordinate system. And output.

The dcqc axis rotating coordinate system is an estimated rotating coordinate system. The d-axis is a vector axis having the smallest static inductance in the rotor of the synchronous machine 2, and the q-axis is a vector axis orthogonal to the d-axis in terms of electrical angle. On the other hand, the estimated rotating coordinate system corresponds to the d-axis and the q-axis at the estimated position of the rotor. That is, the vector axis rotated by the rotation angle error Δθ from the d-axis is the dc axis, and the vector axis rotated by the rotation angle error Δθ from the q-axis is the qc axis.

The speed control circuit 28 is provided with, for example, a PI controller, and the torque command T is provided so that the speed command ω ^* supplied from the outside and the speed estimated value ωest supplied from the speed / rotation phase angle estimation circuit 26 are equal to each other. ^* Is calculated and output.

FIG. 2 is a block diagram schematically showing a configuration example of a speed control circuit of the inverter control device shown in FIG. 1.
The speed control circuit 28 includes a subtractor 31, a PI controller (first PI controller) 32, and a torque limiting circuit 33.

The subtractor 31 subtracts the speed estimation value ωest from the speed command ω ^* , and outputs the speed error Δω (= ω ^* −ωest) which is the calculation result.
The PI controller 32 receives the speed error Δω from the subtractor 31, and calculates the torque command calculation value T ^* cal so that the speed error Δω becomes zero.

The torque limit circuit 33 limits the torque command calculation value T ^* cal with the torque limit value Slim, and outputs the torque command value T ^* which is the value after the limit. The torque limit value Slim may be a preset value or a value input from the outside.
The current command generation circuit 27 generates and outputs dq-axis current commands idc ^* and iqc ^* using the torque command T ^* and the estimated speed ωest.

The current control circuit 25 includes current response values iuc and iqc obtained by vector-converting the current response values iu and iw detected by the current detector 3, and current command values idc ^* and iqc ^* generated by the current command generation circuit 27. , And determine the fundamental wave voltage command value (modulation rate command value) vdc ^* and vqc ^* so as to realize the current command values idc ^* and iqc ^* .

The coordinate conversion (dcqc / UVW) circuit 24 is a vector conversion circuit that performs coordinate conversion from the dcqc axis rotating coordinate system to the three-phase fixed coordinate system using the rotation phase angle estimated value θest. The coordinate conversion (dcqc / UVW) circuit 24 transfers the voltage command values vdc ^* and vqc ^* of the dcqc axis rotating coordinate system to the voltage command values (modulation rate command values) vu ^* , vv ^* , vw ^* of the three-phase fixed coordinate system. Convert and output.

The PWM modulation circuit 22 modulates the voltage command values vu ^* , vv ^* , and vw ^* for driving the synchronous machine 2 by triangular wave PWM, and turns on (conducts) / off (non-conducts) each phase power semiconductor element of the inverter 1. Outputs the gate signal, which is a command (conduction).
The approximate speed estimation circuit 29 calculates the speed estimation FF (feed forward) value ωest FF using the speed command ω ^* .

FIG. 3 is a block diagram schematically showing a configuration example of the schematic speed estimation circuit shown in FIG.
The approximate speed estimation circuit 29 includes a subtractor 41, a PI controller (second PI controller) 42, a torque limiting circuit 43, and a motor machine model circuit 44.

The subtractor 41 subtracts the speed estimation FF value ω ^* estFF from the speed command ω ^* and calculates the speed error Δω ^* .
The PI controller 42 calculates the torque command calculation estimated value T ^* estcal so that the speed error Δω ^* becomes 0. Here, the proportional gain and the integrated gain used in the PI controller 42 are the same as the proportional gain and the integrated gain used in the PI controller 32 of the speed control circuit 28.

The torque limit circuit 43 limits the torque command calculation estimated value T ^* est cal by the torque limit value Slim and outputs the torque command estimated value T ^* est. Here, the torque limit value Slim is the same as the torque limit value Slim used in the torque limit circuit 33 used in the speed control circuit 28.

The motor machine model circuit 44 includes a divider 48 and an integrator 49.
The divider 48 divides the torque command estimated value T ^* est by the inertia estimated value Jest and outputs it.
The integrator 49 integrates the value T ^* est / Jest output from the divider 48, calculates the speed estimation FF value ω ^* est FF, and outputs it. The inertia estimate Jest can use a fixed value adjusted by speed control. That is, the speed estimation FF value ω ^* estFF can be expressed by the following equation.

上記速度推定ＦＦ値ω^＊estFFは、モータ機械モデル回路４４により演算された同期機２の回転速度概略値である。
速度・回転位相角推定回路２６は、速度推定ＦＦ値ω^＊estFF、電圧指令ｖｄｃ^＊、ｖｑｃ^＊および電流応答値ｉｄｃ、ｉｑｃを用いて速度推定値ωestおよび回転位相角推定値θestを演算する。

The speed estimation FF value ω ^* estFF is an approximate rotation speed value of the synchronous machine 2 calculated by the motor machine model circuit 44.
The velocity / rotation phase angle estimation circuit 26 calculates the velocity estimation value ωest and the rotation phase angle estimation value θest using the velocity estimation FF value ω ^* estFF, the voltage command vdc ^* , vqc ^* , and the current response values idc and iqc.

FIG. 4 is a block diagram schematically showing a configuration example of the speed / rotation phase angle estimation circuit shown in FIG. 1.
FIG. 5 is a block diagram schematically showing a configuration example of the rotation phase angle error calculation circuit shown in FIG.
The speed / rotation phase angle estimation circuit 26 includes a rotation phase angle error calculation circuit 51, a PI controller 52, an adder 53, and an integrator 54.

The rotation phase angle error calculation circuit 51 includes a phase difference δ setting circuit 60, a γ voltage calculation circuit 61, a γ voltage estimation circuit 62, and a subtractor 63.
The phase difference δ setting circuit 60 takes the torque command value T ^* as an input and sets the phase angle δ near the maximum voltage change when an error occurs between the rotor phase estimated value and the rotor phase. .. Even if the phase difference δ setting circuit 60 includes a table of set values of the phase angle δ that stores the set values of the phase angle δ obtained in advance analytically or experimentally according to the torque command value T ^* . good.

The γ voltage calculation circuit 61 calculates the γ voltage Vγ by the following equation with the voltage commands vdc ^* and vqc ^* and the phase difference δ as inputs.
Vγ = vdc ^* × cos (δ) + vqc ^* × sin (δ)
The γ voltage estimation circuit 62 calculates the γ voltage estimation value VγH by inputting the current response values idc and iqc of the dcqc axis rotating coordinate system, the velocity estimation value ωest, and the phase difference δ.

The γ voltage estimation circuit 62 calculates the dcqc axis voltage estimated values VdcH and VqcH by the following equation.

In the above formula, R is the winding resistance, Ldc is the dc-axis inductance, Lqc is the qc-axis inductance, φPM is the permanent magnet magnetic flux, and s is the Laplace operator.

The γ voltage estimation circuit 62 calculates the γ voltage estimated value VγH by the following formula using the dcqd axis voltage estimated values VdcH and VqcH obtained by the above formula.
VγH = VdcH × cos (δ) + VqcH × sin (δ)
The subtractor 63 subtracts the estimated γ voltage value VγH from the γ voltage Vγ and outputs the voltage error ΔVγ. The voltage error ΔVγ is a voltage error in the γ-axis direction deviated from the dc axis by the phase difference δ. When the rotation phase angle error Δθ is zero, the rotation phase angle estimated value θest follows the actual rotation phase angle θ. That is, the rotation phase angle error Δθ is an error included in the rotation phase angle estimated value θest, and when the voltage error ΔVγ is zero, the rotation phase angle error Δθ becomes zero.

The rotation phase angle error calculation circuit 51 has, for example, a formula or table representing the relationship between the voltage error ΔVγ and the rotation phase angle error Δθ in advance, and calculates and outputs the rotation phase angle error Δθ based on the voltage error ΔVγ. be able to.

The relationship between the rotation phase angle error Δθ and the voltage error ΔVγ differs depending on the inductance characteristics of the synchronous machine 2. For example, when the synchronous machine 2 is a synchronous relaxation motor (SinRM), the relationship between the rotation phase angle error Δθ and the voltage error ΔVγ is such that when the rotation phase angle error Δθ is zero, the voltage error ΔVγ becomes zero and the rotation phase. This is a sine wave having the maximum voltage error ΔVγ when the angle error Δθ is ± 90 °. Further, for example, when the synchronous machine 2 is a permanent magnet synchronous motor (PMSM), the relationship between the rotational phase angle error Δθ and the voltage error ΔVγ is such that the voltage error ΔVγ becomes zero when the rotational phase angle error Δθ is zero. This is a sine wave having the maximum voltage error ΔVγ when the rotation phase angle error Δθ is ± 180 °. The rotation phase angle error calculation circuit 51 is not limited to the above method, and may calculate a value corresponding to the rotation phase angle error Δθ.

In the PI controller 52, the rotation phase angle error calculation circuit 51 receives the rotation phase angle error Δθ, performs PI control so that the rotation phase angle error Δθ becomes zero, and performs speed estimation FB (feedback) value ω ^* est FB. Is calculated. The speed estimation FB value ω ^* estFB corresponds to the error of the speed estimation FF value ω ^* estFF with respect to the speed estimation value ωest.

The adder 53 adds the speed estimated FB value ω ^* estFB and the speed estimated FF value ω ^* estFF to calculate the speed estimated value ωest. The integrator 54 integrates the velocity estimated value ωest to calculate the rotation angle estimated value θest.

In the inverter control device and the motor drive system, when the speed command ω ^* changes, the approximate speed estimation circuit 29 calculates and outputs the speed estimation FF value ω ^* est FF corresponding to the change. At this time, since the proportional gain and the integrated gain used in the PI controller 42 are the same as the proportional gain and the integrated gain used in the PI controller 32 of the speed control circuit 28, the motor machine model circuit 44 is the actual synchronous machine 2. If it matches the response of the mechanical system, the estimated speed FF value ω ^* estFF matches the actual speed. The velocity estimation FF value ω ^* estFF is used in the calculation of the velocity estimation value ω est by the adder 53 of the velocity / rotation phase angle estimation circuit 26.

When the value obtained by feedback is used when calculating the speed estimation value ωest, the rotation angle error Δθ which is the input of the PI controller 52 becomes large when the speed command ω ^* changes. On the other hand, in the present embodiment, the speed estimation FF value ω ^* estFF acts as a feed forward when the speed estimation value ω est is calculated, so that the rotation phase angle error Δθ becomes large even when the speed command ω ^* changes. Never.
Therefore, in the present embodiment, high-speed speed response can be realized without step-out without increasing the gain of the PI controller 52 that receives the rotation phase angle error Δθ.

It is difficult for the motor machine model circuit 44 to completely match the response of the mechanical system of the actual synchronous machine 2. However, even if the response of the motor machine model circuit 44 and the mechanical system of the actual synchronous machine 2 do not completely match, it is possible to reduce the error between the speed estimation value ωest and the actual speed ω. In particular, when a high-speed speed response is required, step-out can be prevented.

As described above, according to the present embodiment, by using the same gain and the same torque limit as the speed control circuit 28 in the approximate speed estimation circuit 29, the primary delay of the speed command ω ^* and the speed command ω ^* is simply fed. It is possible to estimate the actual velocity ω and the error is smaller than when adding by forward.

That is, according to the present embodiment, it is possible to provide an inverter control device and a motor drive system that realize a high-speed speed response.
In the above embodiment, an example in which a synchronous reluctance motor is used as the synchronous machine 2 is shown, but the same effect can be obtained with a permanent magnet synchronous motor using a reluctance torque.

Next, a modification of the inverter control device and the motor drive system of the first embodiment will be described. The modification described below is different from the inverter control device of the first embodiment described above in the configuration of the approximate speed estimation circuit 29.

FIG. 6 is a block diagram schematically showing another configuration example of the schematic speed estimation circuit shown in FIG. 1.
In this example, the schematic speed estimation circuit 29 includes a torque-limited PI controller 45 in place of the PI controller 42 and the torque limiting circuit 43 in the example shown in FIG. Other than this point, it is the same as the configuration example shown in FIG.

The PI controller 45 calculates the torque command estimated value T ^* est by PI control so that the speed estimation error Δω ^* est becomes zero.
The PI controller 45 includes a proportional gain multiplier 451, an integrator gain multiplier 452, an integrator 453, a limiter 454, an adder 455, and a subtractor 456, 457.

The proportional gain multiplier 451 multiplies the velocity estimation error Δω ^* est by the proportional gain P and outputs the result.
The integral gain multiplier 452 outputs the velocity estimation error Δω ^* est multiplied by the integral gain I.
The subtractor 457 (first subtractor) calculates and outputs the difference (ΔT ^* est) between the input value and the output value of the limiter 454.
The subtractor (second subtractor) 456 outputs a value obtained by subtracting the calculation result (ΔT ^* est) of the subtractor 457 from the value (I × Δω ^* est) output from the integral gain multiplier 452.

The integrator 453 integrates and outputs the value (I × Δω ^* est−ΔT ^* est) output from the subtractor 456.
The adder 455 adds and outputs the output value (P × Δω ^* est) from the proportional gain multiplier 451 and the output value (∫ (I × Δω ^* est−ΔT ^* est) dt) of the integrator 453. ..
The limiter 454 receives the value output from the adder 455, and outputs the torque command estimated value T ^* est limited so as not to exceed the output limit of the inverter 1.

For example, when there is no subtractor 456 in front of the integrator 453, the control error caused by limiting the torque command estimated value cannot be made zero, and the integrator 453 continues to add the error. Therefore, as shown in FIG. 6, by subtracting the difference between the input value and the output value of the limiter 454 from the input of the integrator 453 in advance, it is possible to prevent the integrator 453 from continuously adding the error. can.

For example, the motor machine model circuit 44 does not match the mechanical system of the actual synchronous machine 2, and in particular, when the actual synchronous machine 2 has a load torque, the speed estimated FF value ω ^* estFF is higher than the actual speed ω. Will grow. In the example shown in FIG. 5, when there is a load torque and the torque is limited by the torque limiting circuit 33 of the speed control circuit 28, the error between the actual speed ω and the speed estimated FF value ω ^* estFF may increase. It disappears. This makes it possible to prevent the control from becoming unstable (broken) according to the amount of continuous error addition when the limit state is changed to the non-limit state (non-limit state).

Next, another modification of the inverter control device and the motor drive system of the first embodiment will be described. The modifications described below differ in the configuration of the motor machine model circuit 44 of the inverter control device of the first embodiment described above.

FIG. 7 is a block diagram schematically showing another configuration example of the motor machine model shown in FIG.
In this example, in the motor machine model circuit 44, the speed estimation FF value ω ^* estFF considering the friction of bearings and the like is calculated, and the actual mechanical system of the synchronous machine 2 is modeled more accurately.
The motor machine model circuit 44 includes a divider 48, an integrator 49, a subtractor 441, and a multiplier 442.

The divider 48 divides the torque command estimated value T ^* est by the inertia estimated value Jest and outputs it.
The multiplier 442 outputs the product of the velocity estimation FF value ω ^* est FF output from the integrator 49 and the friction coefficient Dest.
The subtractor 441 outputs the difference obtained by subtracting the output value (Dest × ω ^* estFF) of the multiplier 442 from the output value (T ^* est / Jest) of the divider 48.

The integrator 49 integrates the value (T ^* est / Jest-Dest × ω ^* estFF) output from the divider 48, calculates the speed estimation FF value ω ^* estFF, and outputs it. The inertia estimate Jest can use a fixed value adjusted by speed control. That is, the speed estimation FF value ω ^* estFF can be expressed by the following equation.

As described above, by calculating the speed estimation FF value ω ^* estFF in consideration of the friction in the synchronous machine 2, the error between the speed estimation FF value ω ^* estFF and the actual speed value can be reduced.
That is, according to the above-described modification, it is possible to provide an inverter control device and a motor drive system that can obtain the same effects as those of the above-mentioned first embodiment and perform more reliable sensorless control. Is. Furthermore, if the characteristics of the load coupled to the motor are known, the characteristics can be further reduced from the value output from the divider 48 (T ^* est / Jest-Dest × ω ^* estFF) to achieve a more accurate speed FF. The value can be calculated and the speed response can be improved. For example, when a fan pump is joined to the motor, it can be regarded as a load that changes with the cube of the motor rotation speed ω (ω ³ ), and the value considering this characteristic is subtracted from the output value of the divider 48. By setting it, a more accurate speed FF value can be calculated and the speed response can be improved.

Next, the inverter control device and the motor drive system of the second embodiment will be described in detail with reference to the drawings. In the following description, the same components as those in the above-described first embodiment are designated by the same reference numerals and the description thereof will be omitted.

FIG. 8 is a block diagram schematically showing a configuration example of the inverter control device and the motor drive system of the second embodiment.
The electric motor drive system of the present embodiment includes an inverter control device, an inverter 1, and an induction motor 4.

The induction motor 4 is, for example, an alternating current motor including a stator and a rotor and driven by a three-phase alternating current supplied from the inverter 1.
The inverter control device of this embodiment includes a current detector 3, a PWM modulation circuit 22, a coordinate conversion (UVW / dcqc) circuit 23, a coordinate conversion (dcqc / UVW) circuit 24, a current control circuit 25, and a speed. The estimation circuit 102, the current command generation circuit 27, the speed control circuit 28, the approximate speed estimation circuit 29, the slip frequency calculation circuit 101, the speed estimation circuit 102, the adder 103, and the integrator 104 are provided. ing.

ここで、Ｒ_２は二次抵抗、Ｌ_２は二次自己インダクタンスをそれぞれ表す。 The slip frequency calculation circuit 101 calculates the slip frequency command ωs ^* from the current commands idc ^* and iqc ^* . For example, the following formula is used for calculation.

Here, R ₂ represents a secondary resistance and L ₂ represents a secondary self-inductance.

The speed estimation circuit 102 receives the current command iqc ^* , the current iq, and the speed estimation FF value ω ^* estFF, and outputs the speed estimation value ωest.

FIG. 9 is a block diagram schematically showing a configuration example of the speed estimation circuit shown in FIG.
The speed estimation circuit 102 includes a subtractor 71, a PI controller 72, and an adder 73.
The subtractor 71 subtracts the current iq from the current command iq ^* , calculates the q-axis current deviation Δiq, and outputs it.
The PI controller 72 calculates and outputs the speed estimation FB value ω ^* est FB so that the q-axis current deviation Δiq becomes zero.

The adder 73 adds the speed estimation FF value ω ^* estFF and the speed estimation FB value ω ^* estFB, calculates the speed estimation value ωest, and outputs the calculation.
The adder 103 adds the speed estimation value ωest and the slip frequency command ωs ^* to calculate and output the primary frequency command ω1 ^* .
The integrator 104 integrates the primary frequency command ω1 ^* , calculates the d-axis phase θ, and outputs it. The d-axis phase θ output from the integrator 104 is input to the coordinate conversion (UVW / dcqc) circuit 23 and the coordinate conversion (dcqc / UVW) circuit 24, and is used for vector conversion.

In the present embodiment as well, as in the first embodiment described above, in the approximate speed estimation circuit 29, when the speed command ω ^* changes, the speed estimation FF value ω ^* est FF is calculated accordingly. At this time, since the same gain and torque limit as the speed control circuit 28 are used, if the motor machine model circuit 44 matches the response of the actual mechanical system of the induction motor 4, the speed estimation FF value ω ^* est FF is Matches the actual speed. The speed estimation FF value ω ^* estFF is used in the calculation of the speed estimation value ω est in the adder 73 of the speed estimation circuit 102.

When the speed command ω ^* changes, the q-axis current deviation Δiq, which is the input of the PI controller 72, usually increases, but the speed estimation FF value ω ^* estFF acts as a feed forward, so that the speed command ω ^* becomes The q-axis current deviation Δiq does not change as it did when it did not change. Therefore, step-out does not occur without increasing the gain of the PI controller 72, and a high-speed speed response can be realized.

For example, in the method proposed conventionally, the one corresponding to the velocity estimation FF value ω ^* estFF using the q-axis secondary magnetic flux induced voltage is calculated, but in this method, it is estimated correctly when the velocity ω is small. Difficult to do. On the other hand, in the inverter control device and the motor drive system of the present embodiment, feedforward can be added regardless of the speed, and a high-speed speed response can be realized.
That is, according to this embodiment, the same effect as that of the above-mentioned first embodiment can be obtained.

In the inverter control device and the motor drive system of the present embodiment, the schematic speed estimation circuit 29 described in the first embodiment may be applied.
Further, in the present embodiment, an example of an inverter control device and an electric motor drive system including an induction motor 4 is shown, but similarly, high-speed response in a low-speed range is realized even in sensorless control of PMSM without using reluctance torque. can do.

Although some embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other embodiments, and various omissions, replacements, and changes can be made without departing from the gist of the invention. These embodiments and variations thereof are included in the scope and gist of the invention, and are also included in the scope of the invention described in the claims and the equivalent scope thereof.

1 ... Inverter, 2 ... Synchronizer, 3 ... Current detector, 4 ... Induction motor, 22 ... PWM modulation circuit, 23 ... Coordinate conversion circuit, 24 ... Coordinate conversion circuit, 25 ... Current control circuit, 26 ... Speed / rotation phase Angle estimation circuit, 27 ... current command generation circuit, 28 ... speed control circuit, 29 ... approximate speed estimation circuit, 31 ... subtractor, 32 ... PI controller (first PI controller), 33 ... torque limiting circuit (first torque) Limit circuit), 41 ... subtractor, 42 ... PI controller (second PI controller), 43 ... torque limit circuit (second torque limit circuit), 44 ... motor machine model circuit, 45 ... PI controller with torque limit, 48 ... Divider, 49 ... Integrator, 51 ... Rotational phase angle error calculation circuit, 52 ... PI controller, 53 ... Adder, 54 ... Integrator, 60 ... Phase difference δ setting circuit, 61 ... γ voltage calculation circuit, 62 ... γ voltage estimation circuit, 63 ... subtractor, 71 ... subtractor, 72 ... PI controller, 73 ... adder, 101 ... frequency calculation circuit, 102 ... speed estimation circuit, 103 ... adder, 104 ... adder, 441 ... subtractor, 442 ... multiplier, 451 ... proportional gain multiplier, 452 ... integral gain multiplier, 453 ... adder, 454 ... limiter, 455 ... adder, 456 ... subtractor, 457 ... subtractor.

Claims (5)

With an inverter
A current detector that detects the current response value output from the inverter, and
A speed control circuit including a first PI controller that calculates a torque command value of the motor so that the rotation speed of the motor connected to the inverter and the rotation speed command value are equal to each other.
A current command value generation circuit that calculates a current command value corresponding to the current response value using the torque command value, and a current command value generation circuit.
A current control circuit that calculates the output voltage output from the inverter so that the current command value and the current response value are equal to each other.
A second PI controller that outputs a torque command estimated value by inputting a difference between the rotational speed command value and the rotational speed approximate value, and a motor machine model circuit that calculates the rotational speed approximate value using the torque command estimated value. , And the gain used in the second PI controller is the same as the gain used in the first PI controller.
An inverter control device including a speed estimation circuit that calculates a speed estimation value of the electric motor using the rotation speed approximate value. The speed control circuit includes a first torque limiting circuit that limits the torque command value.
The inverter control device according to claim 1, wherein the schematic speed estimation circuit includes a second torque limiting circuit that limits the torque command estimated value with the same value as the first torque limiting circuit. The speed control circuit includes a torque limiting circuit that limits the torque command value.
The second PI controller outputs a product of a limiter that limits the torque command estimated value with the same value as the torque limiting circuit, and the difference between the rotation speed command value and the rotation speed approximate value multiplied by a proportional gain. A proportional gain multiplier, an integrated gain multiplier that outputs the product of the difference between the rotational speed command value and the rotational speed approximate value multiplied by the integrated gain, and the torque command estimated value before being limited by the limiter. The first subtractor that outputs the difference from the torque command estimated value after being limited by the limiter, and the first subtractor that outputs the difference obtained by subtracting the output value of the first subtractor from the output value of the adder gain multiplier. Adder that supplies the sum of the two subtractors, the adder that integrates and outputs the output values of the second subtractor, and the output values of the proportional gain multiplier and the output values of the integrator to the limiter. The inverter control device according to claim 1, further comprising a device. The inverter control device according to any one of claims 1 to 3,
An electric motor drive system including the above-mentioned electric motor. The motor drive system according to claim 4, wherein the motor is a synchronous reluctance motor.

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