JP6820047B2 - Contactless power transfer system - Google Patents

Description

The present invention relates to a non-contact power transmission system that transmits power in a non-contact manner.

Technology has been developed to transmit the power output by the power supply to the load in a non-contact manner without directly connecting the load and the power supply. The technology is commonly referred to as contactless power transfer or wireless power transfer. The technology is applied to power transmission to mobile phones, home appliances, electric vehicles, automatic guided vehicles (AGVs), and the like.

In non-contact power transmission, power is transmitted non-contactly from a power transmission device connected to a high-frequency power supply device to a power receiving device connected to a load. The power transmission device is equipped with a power transmission coil, and the power reception device is equipped with a power reception coil. Non-contact power transmission is performed by magnetically coupling the power transmission coil and the power reception coil. For example, Patent Document 1 describes a magnetic field resonance type non-contact power transmission device in which both a power transmission coil and a power reception coil form a series resonance circuit.

The inventor of the present application uses a high-frequency power supply device as a constant current source, a transmitting coil constitutes a series resonance circuit, a power receiving coil constitutes a parallel resonance circuit, and non-contact power transmission is performed by a magnetic field resonance method. We have developed a non-contact power transmission system that charges a power storage device with a constant current with the output current from the parallel resonant circuit set to a constant magnitude (Japanese Patent Application No. 2015-221504).

Japanese Unexamined Patent Publication No. 2015-231287

However, when the power transmitting coil and the power receiving coil have the same specifications, there is a problem that the output current from the parallel resonant circuit on the power receiving side is smaller than the output current of the high frequency power supply device.

The present invention has been conceived under the above circumstances, and even when a high-frequency power supply device is used as a constant current source, the output current from the resonance circuit on the power receiving side is compared with the output current of the high-frequency power supply device. The purpose is to provide a non-contact power transmission system that can be increased in size.

In order to solve the above problems, the following technical measures are taken in the present invention.

The non-contact power transmission system provided by the present invention includes a high-frequency power supply device that outputs a constant high-frequency current, a power transmission device that transmits power output by the high-frequency power supply device, and the power transmission device that transmits power. The power transmission device includes a power transmission device and a power reception device that receives power in a non-contact manner, and the power transmission device includes a conversion means for converting a current output from the high frequency power supply device into a voltage output. According to this configuration, the constant high frequency current output by the high frequency power supply device is converted into a constant high frequency voltage by the conversion means. Therefore, by adjusting the mutual inductance due to the magnetic coupling between the power transmission coil of the power transmission device and the power reception coil of the power reception device, the power reception device can output a larger current than the output current of the high frequency power supply device.

In a preferred embodiment of the present invention, the conversion means is a current-voltage conversion circuit. According to this configuration, the conversion means can be configured as hardware.

In a preferred embodiment of the present invention, the power transmission device comprises a power transmission unit having a power transmission coil for transmitting power and a resonance capacitor connected in series to the power transmission coil to form a resonance circuit, and the power reception device. The device includes a power receiving unit having a power receiving coil that receives power and a resonance capacitor that is connected in series to the power receiving coil to form a resonance circuit. According to this configuration, both the power transmission unit and the power reception unit are series resonance circuits, and perform power transmission by a magnetic field resonance method. Therefore, the power receiving unit can output a constant high frequency current.

In a preferred embodiment of the present invention, the impedance of the mutual inductance due to the magnetic coupling between the power transmitting coil and the power receiving coil is smaller than the characteristic impedance of the current-voltage conversion circuit. According to this configuration, the power receiving unit can output a current larger than the output current of the high frequency power supply device.

In a preferred embodiment of the present invention, the current-voltage conversion circuit comprises an inductor and a capacitor designed so that the magnitudes of the respective impedances are equal at the frequency of the high frequency current output by the high frequency power supply device. Either a filter circuit arranged in a T-type or a π-type, or a transmission line whose length is approximately one-fourth of the transmission wavelength of the transmission line at the said frequency. It has one. According to this configuration, the current-voltage conversion circuit can be configured only by the filter circuit or the transmission line.

In a preferred embodiment of the present invention, the current-voltage conversion circuit includes a filter circuit in which one inductor and two capacitors are arranged in a π shape, and a transmission line, and the length of the transmission line is increased. When the frequency f ₀ output by the high-frequency power supply device is x times the transmission wavelength λ in the transmission line, the self-inductance L of the inductor and the capacitance C of the capacitor are designed based on the following equations. The characteristic impedance of the transmission line is Z _0, and β is a phase constant (2π / λ). According to this configuration, the filter circuit can be appropriately designed according to the length of the transmission line.

In a preferred embodiment of the present invention, the current-voltage conversion circuit includes a filter circuit in which two inductors and one capacitor are arranged in a T shape and a transmission line, and the length of the transmission line is increased. When the frequency f ₀ output by the high-frequency power supply device is x times the transmission wavelength λ in the transmission line, the self-inductance L of the inductor and the capacitance C of the capacitor are designed based on the following equations. The characteristic impedance of the transmission line is Z _0, and β is a phase constant (2π / λ). According to this configuration, the filter circuit can be appropriately designed according to the length of the transmission line.

According to the present invention, a constant high frequency current output by a high frequency power supply device is converted into a constant high frequency voltage by a conversion means. Therefore, by adjusting the mutual inductance due to the magnetic coupling between the power transmission coil of the power transmission device and the power reception coil of the power reception device, the power reception device can output a larger current than the output current of the high frequency power supply device.

Other features and advantages of the present invention will become more apparent with the detailed description given below with reference to the accompanying drawings.

It is a figure which shows the outline of the whole structure of the non-contact power transmission system which concerns on 1st Embodiment. It is a circuit diagram which shows the whole structure of the non-contact power transmission system which concerns on 1st Embodiment. It is a circuit diagram which shows the detail of the internal structure of a high frequency power supply device. It is a circuit diagram which shows the whole structure of the non-contact power transmission system for comparison. It is a figure for demonstrating the circuit of the main part of the non-contact power transmission system for comparison by the equivalent circuit. It is a figure which shows each waveform at the time of performing the simulation based on the circuit shown in FIG. It is a figure for demonstrating the circuit of the main part of the non-contact power transmission system which concerns on 1st Embodiment by the equivalent circuit. It is a figure which shows each waveform at the time of performing the simulation based on the circuit shown in FIG. It is a figure which shows the modification of the current-voltage conversion circuit which concerns on 1st Embodiment. (A) is a circuit diagram showing the configuration of the non-contact power transmission system according to the second embodiment, and (b) is a diagram showing a main part of the non-contact power transmission system according to the second embodiment. It is a figure which shows each waveform at the time of performing the simulation based on the circuit shown in FIG. 10A. It is a circuit diagram which shows the structure of the non-contact power transmission system which concerns on 3rd Embodiment. It is a figure which shows each waveform at the time of performing the simulation based on the circuit shown in FIG. It is a circuit diagram which shows the structure of the non-contact power transmission system which concerns on 4th Embodiment. It is a figure which shows the modification of the current-voltage conversion circuit which concerns on 4th Embodiment. It is a figure which shows each waveform when the simulation was performed based on the circuit shown in FIG. It is a figure which shows the other embodiment of the non-contact power supply system which concerns on 1st to 4th Embodiment.

Hereinafter, embodiments of the present invention will be specifically described with reference to the drawings. It should be noted that the dimensions, shapes, materials, various values, etc. described below are examples for explanation and are not limited to those described. It can be changed as appropriate according to the specifications. Further, in each drawing, the same or similar elements are designated by the same reference numerals, and duplicate description is omitted.

1 to 3 are diagrams for explaining the non-contact power transmission system S1 according to the first embodiment. FIG. 1 shows an outline of the overall configuration of the non-contact power transmission system S1. FIG. 2 is a circuit diagram showing the overall configuration of the non-contact power transmission system S1. FIG. 3 is a circuit diagram showing details of the internal configuration of the high frequency power supply device 1.

As shown in FIG. 1, the non-contact power transmission system S1 includes a power receiving device B provided in a vehicle body of an electric vehicle or the like, and a power transmission device A1 embedded in a floor surface of a parking lot or the like. The power transmission device A1 includes a power transmission coil arranged on the floor surface, and the power reception device B includes a power reception coil arranged on the bottom surface of the vehicle body. By magnetically coupling the power transmitting coil and the power receiving coil, the power receiving device B receives the high frequency power transmitted from the power transmitting device A1. That is, the magnetic flux changes when a high-frequency current flows through the power transmission coil, and the high-frequency current flows through the power receiving coil interlinking with this magnetic flux. As a result, electric power can be transmitted from the power transmitting device A1 to the power receiving device B in a non-contact manner. The power receiving device B rectifies the high frequency current with a rectifying smoothing circuit and supplies it to the power storage device D.

The power transmission coil and the power reception coil are planar coils wound in a spiral shape, and the coil surfaces are arranged so as to be substantially parallel to the floor surface. When power is supplied, as shown in FIG. 1, the vehicle body is arranged so that the power receiving device B comes directly above the power transmission device A1 and the power receiving coil overlaps the power transmission coil when viewed from above. FIG. 2 shows a state in which the power receiving coil is magnetically coupled to the power transmission coil.

As shown in FIG. 2, the power transmission device A1 includes a high-frequency power supply device 1, a current-voltage conversion circuit 51, and a power transmission unit 2.

The high frequency power supply device 1 outputs high frequency power. The high-frequency power supply device 1 is a so-called constant current source and outputs a high-frequency current of a constant magnitude. As shown in FIG. 3, the high-frequency power supply device 1 includes a DC power supply device 11, a control device 12, a switching element Qs, a diode D1, inductors L1, L2, L3, and capacitors C1, C2, C3, C4, C10. ing. The high-frequency power supply device 1 converts the DC power generated by the DC power supply device 11 into high-frequency power by the switching operation of the switching element Qs.

The DC power supply device 11 generates and outputs DC power. The DC power supply device 11 converts an AC voltage input from a commercial power source (for example, a commercial voltage 200 [V], etc.) into a DC voltage by rectifying it with a rectifying circuit (not shown) and smoothing it with a smoothing circuit (not shown). Then, it is converted into a DC voltage of a predetermined level (target voltage) by a DC-DC converter (not shown). The DC power supply device 11 converts the rectified and smoothed DC voltage into a predetermined level DC voltage by controlling the conversion operation of the DC-DC converter by the voltage control signal CS1 input from the control device 12. The configuration of the DC power supply device 11 is not limited as long as it outputs a predetermined DC voltage.

The switching element Qs, the diode D1, the inductors L1 and L3, and the capacitors C1, C3, C4 and C10 form a so-called class E amplifier. The class E amplifier receives DC power from the DC power supply device 11 to generate and output high-frequency power.

The capacitor C10 is connected in parallel to the DC power supply device 11 and smoothes the DC voltage input from the DC power supply device 11.

The inductor L1 is connected in series between the output terminal on the high potential side of the DC power supply device 11 and the switching element Qs. When the DC power supply device 11 outputs a constant DC voltage, the inductor L1 supplies a constant DC current to the switching element Qs.

The switching element Qs switches between an on state and an off state according to the high frequency control signal CS2 input from the control device 12. In this embodiment, a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) is used as the switching element Qs. The switching element Qs is not limited to the MOSFET, and may be a bipolar transistor, an IGBT (Insulated Gate Bipolar Transistor: Insulated Gate Bipolar Transistor), or the like. The drain terminal of the switching element Qs is connected to one terminal of the inductor L1 (the terminal different from the terminal connected to the output terminal of the DC power supply device 11). The source terminal of the switching element Qs is connected to the output terminal on the low potential side of the DC power supply device 11. A high frequency control signal CS2 is input from the control device 12 to the gate terminal of the switching element Qs. The high frequency control signal CS2 is a pulse signal that repeats high level and low level at a predetermined frequency f ₀ (for example, 85 [kHz], 13.56 [MHz], etc.). Since the frequency f ₀ is the frequency at which the switching element Qs is switched, it may be described as "switching frequency f ₀ " below. The switching element Qs is turned off when the high frequency control signal CS2 is at a low level, and is turned on when the high frequency control signal CS2 is at a high level.

The diode D1 is a so-called flywheel diode, and is connected in antiparallel between the drain terminal and the source terminal of the switching element Qs. That is, the anode terminal of the diode D1 is connected to the source terminal of the switching element Qs, and the cathode terminal of the diode D1 is connected to the drain terminal of the switching element Qs. The diode D1 is for preventing a high voltage in the reverse direction due to the counter electromotive force generated by switching the switching element Qs from being applied to the switching element Qs. If the switching element Qs has a function of operating a diode inside, the diode D1 may not be provided.

The capacitor C1 is connected in parallel to the switching element Qs, and when the switching element Qs is in the off state, a current flows and electric energy is stored. Then, after the voltage across the capacitor C1 reaches its peak, discharge is performed to release electrical energy. Then, at the timing when the voltage across the capacitor C1 becomes zero, the switching element Qs switches from the off state to the on state.

The inductor L3 and the capacitor C3 are connected in series to form a resonant circuit LC3. The inductor L3 and the capacitor C3 are designed so that the resonance frequency coincides with the switching frequency f ₀ . The resonance circuit LC3 is connected in series between the connection point between the drain terminal of the switching element Qs and one terminal of the inductor L1 and the current-voltage conversion circuit 51. Due to the resonance characteristics of the resonance circuit LC3, the output current becomes a sinusoidal shape with a resonance frequency (switching frequency f ₀ ). The resonant circuit LC3 corresponds to the "first resonant circuit" of the present invention.

The capacitor C4 is connected to the output side of the resonance circuit LC3 so as to be in parallel with the DC power supply device 11. The capacitor C4, the inductor L3 and the capacitor C3 function as an impedance matching circuit. Further, the capacitor C3 cuts a DC component from the high frequency current output from the high frequency power supply device 1.

From the above configuration, the class E amplifier including the switching element Qs, the diode D1, the inductors L1 and L3, and the capacitors C1, C3, C4 and C10 switches according to the high frequency control signal CS2 input from the control device 12. By switching the element Qs, a high frequency current having a switching frequency f ₀ is generated and output.

Further, in the present embodiment, in the high frequency power supply device 1, the resonance circuit LC2 in which the inductor L2 and the capacitor C2 are connected in series is connected in parallel to the switching element Qs. The inductor L2 and the capacitor C2 are designed so that the resonance frequency coincides with a frequency twice the switching frequency f ₀ . Resonant circuit LC2, to the double frequency component of the switching frequency f ₀ (2-order harmonic component) becomes a low impedance, the components of the switching frequency f ₀ (fundamental wave component) and 3 times the frequency component (3 High impedance with respect to the next harmonic component). The resonant circuit LC2 corresponds to the "second resonant circuit" of the present invention.

Further, the filter LC1 composed of the inductor L1 and the capacitor C1 also has an impedance combined with the resonance circuit LC2 with respect to a component (fundamental wave component) having a switching frequency f ₀ and a frequency component (third harmonic component) three times as high as that. It is designed to have high impedance and low impedance with respect to a frequency component (second harmonic component) that is twice the switching frequency f ₀ . The capacitance of the capacitor C1 is designed in consideration of the capacitance component inside the switching element Qs.

From the above configuration, the second harmonic component of the generated high frequency current flows through the resonant circuit LC2, and the voltage generated by the second harmonic component current between the drain and the source of the switching element Qs can be suppressed.

The control device 12 controls the high-frequency power supply device 1, and is a microcomputer or FPGA (Field-Programmable Gate Array) equipped with a CPU (Central Processing Unit), a ROM (Read Only Memory), and a RAM (Random Access Memory). It is composed.

The control device 12 controls the level of the DC voltage output from the DC power supply device 11 by feedback control. Specifically, the control device 12 generates a control pulse signal for making the deviation between the output voltage of the DC power supply device 11 and the set target voltage zero. Then, the control pulse signal is amplified to a level at which the DC-DC converter can be driven by a drive circuit (not shown), and output to the DC power supply device 11 as a voltage control signal CS1. As a result, the control device 12 can control the DC voltage output from the DC power supply device 11 to the target voltage and output a constant DC voltage from the DC power supply device 11. Further, the control device 12 changes the level of the output voltage of the DC power supply device 11 by changing the target voltage.

Further, the control device 12 generates a pulse signal having a switching frequency f ₀ (note that a sine wave signal or the like) is generated based on the reference clock, and the switching element Qs can be driven by a drive circuit (not shown) of the pulse signal. Is amplified to and output as a high frequency control signal CS2 to the gate terminal of the switching element Qs.

The current-voltage conversion circuit 51 converts the current output into a voltage output. The current-voltage conversion circuit 51 is a filter circuit in which the inductor L11 and the two capacitors C11 and C12 are arranged in a π shape. The inductor L11 is provided on one of the power lines of the power line pair between the high frequency power supply device 1 and the power transmission unit 2. The capacitor C11 is connected to the upstream side of the inductor L11, and the capacitor C12 is connected to the downstream side of the inductor L11 in parallel between the power line pairs. Each inductance and capacitance are determined so that the magnitudes of the impedances of the inductor L11 and the capacitors C11 and C12 at the switching frequency f ₀ are equal. The current-voltage conversion circuit 51 converts the current output from the high-frequency power supply device 1 into a voltage output and outputs it to the power transmission unit 2.

The power transmission unit 2 includes a power transmission coil Lt and a resonance capacitor Ct. The power transmission coil Lt transmits high-frequency power supplied from the high-frequency power supply device 1 to the power receiving device B via the current-voltage conversion circuit 51. The resonance capacitor Ct is connected in series to the power transmission coil Lt to form a series resonance circuit.

The transmission coil Lt and the resonance capacitor Ct are designed so that the resonance frequency coincides with the frequency f ₀ (switching frequency f ₀ ) of the high frequency power supplied from the high frequency power supply device 1. That is, the self-inductance L _t of the power transmission coil Lt and the capacitance C _t of the resonance capacitor Ct are designed to have the relationship of the following equation (1). When the switching frequency f ₀ is high, the stray capacitance between the windings of the power transmission coil Lt may be used as the resonance capacitor Ct.

Further, as shown in FIG. 2, the power receiving device B includes a power receiving unit 3 and a rectifying smoothing circuit 4.

The power receiving unit 3 includes a power receiving coil Lr and a resonance capacitor Cr. The power receiving coil Lr is magnetically coupled with the power transmission coil Lt to receive power in a non-contact manner. The resonance capacitor Cr is connected in series to the power receiving coil Lr to form a series resonance circuit.

The power receiving coil Lr and the resonance capacitor Cr are designed so that the resonance frequency matches the frequency f ₀ (switching frequency f ₀ ) of the high frequency power supplied from the high frequency power supply device 1, similarly to the transmission coil Lt and the resonance capacitor Ct. To. When the switching frequency f ₀ is high, the stray capacitance between the windings of the power receiving coil Lr may be used as the resonance capacitor Cr.

Both the power transmission unit 2 and the power reception unit 3 are resonance circuits, and power is transmitted from the power transmission unit 2 to the power reception unit 3 in a non-contact manner by a magnetic field resonance method. The electric power received by the power receiving unit 3 is output to the rectifying smoothing circuit 4.

The rectifying / smoothing circuit 4 rectifies the high-frequency current output from the power receiving unit 3 and converts it into a direct current. The rectifying / smoothing circuit 4 includes a full-wave rectifying circuit in which four diodes are bridge-connected. The rectifying smoothing circuit 4 also includes a smoothing circuit for smoothing the output after rectification. The configuration of the rectifying / smoothing circuit 4 is not limited as long as it converts a high-frequency current into a direct current. The direct current output from the rectifying / smoothing circuit 4 is supplied to the power storage device D.

The power storage device D is a secondary battery such as a lithium ion battery. The power storage device D is charged by the DC power output from the rectifying / smoothing circuit 4, and supplies power to a motor or the like (not shown). A constant magnitude of current is input to the power storage device D regardless of the charging state of the power storage device D. The type of secondary battery is not limited, and may be a lead storage battery or the like. Further, instead of the power storage device D, an electric double layer capacitor or a lithium ion capacitor may be used.

Next, it will be described that the current supplied to the power storage device D becomes constant and the current supplied to the power storage device D becomes a larger current than the output current of the high frequency power supply device 1.

First, for comparison, in the non-contact power transmission system S'shown in FIG. 4, the current supplied to the power storage device D becomes constant, and the current supplied to the power storage device D is the output current of the high frequency power supply device 1. It will be explained that the current becomes smaller than that of. In the non-contact power transmission system S', the high-frequency power supply device 1 is used as a constant current source, the transmission unit 2 constitutes a series resonance circuit, and the power reception unit 3'consists of a parallel resonance circuit. By performing the above, the power storage device D is constantly charged with the output current from the power receiving unit 3 having a constant magnitude. The non-contact power transmission system S'is non-contact except that the power transmission device A'does not have a current-voltage conversion circuit 51 and the power receiving unit 3'of the power receiving device B'consists of a parallel resonant circuit. It has the same configuration as the power transmission system S1.

FIG. 5A is an extraction of the main part of the non-contact power transmission system S'shown in FIG.

Let the output voltage of the high frequency power supply device 1 be V ₁ and the output current be I ₁ . Further, let V ₂ be the voltage applied to the rectifying and smoothing circuit 4, and I ₂ be the current input to the rectifying and smoothing circuit 4. The voltages V ₁ and V ₂ and the currents I ₁ and I ₂ are all vectors.

In general, an equivalent circuit of a non-contact power transmission system can be represented by replacing a magnetically coupled power transmission coil and a power reception coil with a T-type circuit composed of three coils. When the circuit shown in FIG. 5 (a) is converted into an equivalent circuit represented by using the T-type circuit, the circuit shown in FIG. 5 (b) is obtained. In FIG. 5B, as shown in the figure, the impedance of the capacitor or the coil is represented as Z1 to Z4. It should be noted that each impedance Z1 to Z4 is a vector. The inductance of the coil on the power transmission unit side (coil included in impedance Z1) of the coils of the T-type circuit is the mutual inductance due to magnetic coupling between the power transmission coil Lt and the power reception coil Lr from the self-inductance of the power transmission coil Lt. It will be reduced. The inductance of the coil on the power receiving unit side (coil included in impedance Z3) of the coils of the T-type circuit is the mutual inductance due to magnetic coupling between the power transmitting coil Lt and the power receiving coil Lr from the self-inductance of the power receiving coil Lr. It will be reduced. Further, the inductance of the coils (coils included in the impedance Z2) connected in parallel among the coils of the T-type circuit is mutual inductance due to magnetic coupling between the power transmission coil Lt and the power reception coil Lr. Therefore, each impedance Z1 to Z4 can be represented by the following equations (2) to (5). The self-inductances of the power transmission coil Lt and the power reception coil Lr are L _t and L _r , respectively, and the capacitances of the resonance capacitor Ct and the resonance capacitor Cr are C _t and C _r , respectively. Further, the coupling coefficient of the power transmission coil Lt and the power reception coil Lr is k, and the angular frequency corresponding to the frequency f ₀ is ω ₀ (= 2πf ₀ ).

FIG. 5 (c) is a diagram showing an equivalent circuit showing the circuit shown in FIG. 5 (b) using the F parameter. Each element A, B, C, D of the F parameter is a vector, and the F parameter is as shown in the following equation (6).

Substituting Z1 + Z2 = Z2 + Z3 + Z4 = 0, which is a conditional expression of magnetic field resonance, into the above equation (6) gives the following equation (7). From this, the following equation (9) can be obtained from the following equation (8) and the above equations (3) and (4).

If the distance between the power transmission coil Lt and the power reception coil Lr does not change, the coupling coefficient k does not change. Therefore, from the above equation (9), the magnitude of the current I ₂ output from the power receiving unit 3 is proportional to the magnitude of the current I ₁ input to the power transmission unit 2. Further, the current I ₁ is input to the power transmission unit 2, an output current I ₁ of the high-frequency power supply device 1. When the DC voltage output by the DC power supply device 11 is constant, the magnitude of the output current I ₁ of the high frequency power supply device 1 is constant. Therefore, when the DC voltage output by the DC power supply device 11 is constant, the magnitude of the output current I ₂ of the power receiving unit 3 is constant regardless of the impedance of the connected load or the like. That is, the output of the power receiving unit 3 can be considered as a constant current source that outputs a current I ₂ having a constant magnitude. Since the magnitude of the output current I ₂ of the power receiving unit 3 is constant, the current rectified by the rectifying smoothing circuit 4 becomes constant. Therefore, the current supplied to the power storage device D becomes substantially constant regardless of the charging state of the power storage device D.

Further, when the power transmission coil Lt and the power reception coil Lr have the same specifications, the self-inductances L _t and L _r are substantially equal to each other, and L _t / L _r ≈ 1. Since the coupling coefficient k <1, the magnitude of the current I ₂ is smaller than the magnitude of the current I ₁ from the above equation (9). Further, the smaller the coupling coefficient k, the smaller the magnitude of the current I ₂ .

FIG. 6 shows each waveform when a simulation is performed in the circuit shown in FIG. The DC voltage output by the DC power supply device 11 is 40 [V], the high frequency control signal CS2 is a rectangular wave signal with a frequency f = 13.56 [MHz], and the coupling coefficient of the transmission coil Lt and the power receiving coil Lr is k = 0. The simulation was performed with the capacitance of the capacitor of the rectifying smoothing circuit 4 set to 2 and 0.4 and set to 1 [μF] and 100 [μF]. The self-inductances L _t and L _r are both 0.6 [μH], and the capacitances C _t and C _r are both 230 [pF]. FIG. 6A shows the waveform of the drain-source voltage V _ds of the switching element Qs, and FIG. 6B shows the waveform of the output current I ₁ of the high frequency power supply device 1. Further, FIG. 6 (c) shows the waveform of the output current I ₂ of the power receiving unit 3, and FIG. 6 (d) shows the waveform of the current I _LOAD flowing through the capacitor of the rectifying smoothing circuit 4. In FIGS. 6C and 6D, the one having the larger amplitude is the one when the coupling coefficient is k = 0.4. Since there was no significant change in each waveform between the case where the capacitance of the capacitor of the rectifying smoothing circuit 4 was 1 [μF] and the case where the capacitance was 100 [μF], they are displayed overlapping in FIGS. 6 (a) to 6 (d). ing.

Where the amplitude of the current I ₁ is about 1 [A] as shown in FIG. 6 (b), the amplitude of the current I ₂ is k = as shown in FIG. 6 (c). When it is 0.4, it is about 0.4 [A], and when k = 0.2, it is about 0.2 [A]. That is, the magnitude of the current I ₂ is smaller than the magnitude of the current I _1, as the coupling coefficient k is small, it was confirmed that the magnitude of the current I ₂ is reduced.

Next, in the non-contact power transmission system S1 shown in FIG. 2, the current supplied to the power storage device D becomes constant, and the current supplied to the power storage device D is larger than the output current of the high frequency power supply device 1. Explain that

FIG. 7A is an extraction of the main part of the non-contact power transmission system S1 shown in FIG.

Let the output voltage of the high frequency power supply device 1 be V ₁ and the output current be I ₁ . Further, the output voltage of the current-voltage conversion circuit 51 is V ₂ , and the output current is I ₂ . Further, the output voltage of the power receiving unit 3 is V ₃ , and the output current is I ₃ . That is, the voltage applied to the rectifying / smoothing circuit 4 is V ₃ , and the current input to the rectifying / smoothing circuit 4 is I ₃ . The voltages V ₁ , V ₂ , V ₃ and the currents I ₁ , I ₂ , and I ₃ are all vectors.

FIG. 7B shows the circuit of the current-voltage conversion circuit 51. In FIG. 7B, as shown in the figure, the impedance of the capacitor or coil is represented as Z1 to Z3. It should be noted that each impedance Z1 to Z3 is a vector. When the circuit shown in FIG. 7B is represented by using the F parameter, the F parameter is as shown in the following equation (10). Each element A, B, C, D of the F parameter is a vector.

The inductance and each capacitance are determined so that the magnitudes of the impedances of the inductor L11 and the capacitors C11 and C12 at the frequency f ₀ (switching frequency f ₀ ) of the high frequency power supplied from the high frequency power supply device 1 are equal. Assuming that the impedances of the inductor L11 and the capacitors C11 and C12 are the characteristic impedance Z ₀ and the angular frequency corresponding to the frequency f ₀ is ω ₀ (= 2πf ₀ ), the impedances Z1 to Z3 are the following (11) and (12). ) Can be expressed by the formula.

Substituting the above equations (11) and (12) into the above equation (10) gives the following equation (13). From this, the following equation (14) is obtained.

FIG. 7C shows an equivalent circuit of the power transmission unit 2 and the power reception unit 3, in which the magnetically coupled power transmission coil Lt and the power reception coil Lr are replaced with a T-type circuit composed of three coils. It is a thing. In FIG. 7C, as shown in the figure, the impedance of the capacitor or the coil is represented as Z4 to Z6. Each impedance Z4 to Z6 is a vector and can be expressed by the following equations (15) to (17). The inductances of the transmission coil Lt and the power receiving coil Lr are L _t and L _r , respectively, and the capacitances of the resonance capacitor Ct and the resonance capacitor Cr are C _t and C _r , respectively. Further, the coupling coefficient of the power transmission coil Lt and the power reception coil Lr is k, and the angular frequency corresponding to the frequency f ₀ is ω ₀ (= 2πf ₀ ).

When the circuit shown in FIG. 7C is represented by using the F parameter, the F parameter is as shown in the following equation (18). Each element A, B, C, D of the F parameter is a vector.

Substituting Z4 + Z5 = Z5 + Z6 = 0, which is a conditional expression of magnetic field resonance, into the above equation (18) gives the following equation (19). From this, the following equation (20) can be obtained by using the above equation (16).

From the above equations (14) and (20), the following equation (21) is obtained, and the following equation (22) is obtained.

If the distance between the power transmission coil Lt and the power reception coil Lr does not change, the coupling coefficient k does not change. Therefore, from the above equation (22), the magnitude of the current I ₃ output from the power receiving unit 3 is proportional to the magnitude of the output current I ₁ of the high frequency power supply device 1. Further, when the DC voltage output by the DC power supply device 11 is constant, the magnitude of the output current I ₁ of the high frequency power supply device 1 is constant, so that the magnitude of the output current I ₃ of the power receiving unit 3 is the magnitude of the connected load. It is constant regardless of impedance. That is, the output of the power receiving unit 3 can be considered as a constant current source that outputs a current I ₃ having a constant magnitude. Since the magnitude of the output current I ₃ of the power receiving unit 3 is constant, the current rectified by the rectifying smoothing circuit 4 becomes constant. Therefore, the current supplied to the power storage device D becomes substantially constant regardless of the charging state of the power storage device D.

Since the impedance Z _M (= jω ₀ M) of the mutual inductance M due to the magnetic coupling between the power transmission coil Lt and the power reception coil Lr is the following equation (23), it is changed from the above equation (22) to the following equation (24). Become. Therefore, by setting Z _M <Z ₀ , the magnitude of the current I ₃ can be made larger than the magnitude of the current I ₁ .

FIG. 8 shows each waveform when a simulation is performed in the circuit shown in FIG. The DC voltage output by the DC power supply device 11 is 40 [V], the high frequency control signal CS2 is a rectangular wave signal with a frequency f = 13.56 [MHz], and the coupling coefficient of the transmission coil Lt and the power receiving coil Lr is k = 0. The simulation was performed with the capacitance of the capacitor of the rectifying smoothing circuit 4 set to 2 and 0.4 and set to 1 [μF] and 100 [μF]. The self-inductances L _t and L _r are both 0.6 [μH], and the capacitances C _t and C _r are both 230 [pF]. FIG. 8A shows the waveform of the drain-source voltage V _ds of the switching element Qs, and FIG. 8B shows the waveform of the output current I ₁ of the high frequency power supply device 1. Further, FIG. 8 (c) shows the waveform of the output current I ₃ of the power receiving unit 3, and FIG. 8 (d) shows the waveform of the current I _LOAD flowing through the capacitor of the rectifying smoothing circuit 4. In FIGS. 8C and 8D, the one having the larger amplitude is the one when the coupling coefficient is k = 0.2. In FIGS. 8A to 8D, there is a slight phase shift between the case where the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and the case where the capacitance is 100 [μF], but the same waveform is obtained. ing.

Where the amplitude of the current I ₁ is about 1.2 [A] as shown in FIG. 8 (b), the amplitude of the current I ₃ is as shown in FIG. 8 (c). When k = 0.4, it is about 2.5 [A], and when k = 0.2, it is about 5 [A]. That is, it was confirmed that the magnitude of the current I ₃ is larger than the magnitude of the current I ₁ , and the smaller the coupling coefficient k, the larger the magnitude of the current I ₂ .

Next, the operation and effect of the non-contact power transmission system S1 according to the first embodiment will be described.

According to this embodiment, the constant high frequency current output by the high frequency power supply device 1 is converted into a constant high frequency voltage by the current-voltage conversion circuit 51. The power transmission unit 2 and the power reception unit 3 are both series resonance circuits, and perform power transmission by a magnetic field resonance method. Therefore, the output of the power receiving unit 3 becomes equivalent to the output of the constant current source. Therefore, the current output to the power storage device D is substantially constant regardless of the charging state of the power storage device D. That is, the power storage device D can be charged with a constant current. Further, the output current I ₃ of the output current I ₁ and the power receiving unit 3 of the high frequency power supply device 1, the relation of the expression (24). Therefore, the output current I ₃ can be made larger than the output current I ₁ by adjusting the mutual inductance M (impedance Z _M ).

Further, according to the present embodiment, the high frequency power supply device 1 can output a high frequency current having a constant magnitude by inputting the high frequency control signal CS2 to one switching element Qs. Since there is only one switching element, the drive circuit can have a simple configuration. Further, a resonance circuit LC2 composed of an inductor L2 and a capacitor C2 is connected in parallel to the switching element Qs of the high frequency power supply device 1. The resonant circuit LC2, to the double frequency component of the switching frequency f ₀ (2-order harmonic component) becomes a low impedance, the components of the switching frequency f ₀ (fundamental wave component) and 3 times the frequency components ( High impedance with respect to the third harmonic component). Therefore, the second harmonic component of the high frequency current generated by the high frequency power supply device 1 flows into the resonance circuit LC2, and the voltage generated by the second harmonic component current between the drain and the source of the switching element Qs can be suppressed. it can. Therefore, it is not necessary for the switching element Qs to have a high withstand voltage.

In the first embodiment, the case where the current-voltage conversion circuit 51 is a filter circuit in which the inductor L11 and the two capacitors C11 and C12 are arranged in a π shape has been described, but the circuit of the current-voltage conversion circuit 51 The configuration is not limited to those described above. For example, it may be a filter circuit in which two inductors and one capacitor are arranged in a π type (see FIG. 9A), or a filter circuit in which one inductor and two capacitors are arranged in a T type (FIG. 9). (See (b)) may be used, or a filter circuit (see FIG. 9 (c)) in which two inductors and one capacitor are arranged in a T shape may be used.

Further, the current-voltage conversion circuit 51 is not limited to a filter circuit in which an inductor and a capacitor are combined. The current-voltage conversion circuit 51 may be any one that converts the current output from the high-frequency power supply device 1 into a voltage output.

FIG. 10A is a circuit diagram showing the configuration of the non-contact power transmission system S2 according to the second embodiment. Since the configuration of the power receiving device B is the same as that of the power receiving device B according to the first embodiment, only the power transmitting device A2 is described. The non-contact power transmission system S2 differs from the current-voltage conversion circuit 51 of the non-contact power transmission system S1 according to the first embodiment in the configuration of the current-voltage conversion circuit 52.

The current-voltage conversion circuit 52 converts the current output into a voltage output. The current-voltage conversion circuit 52 includes a transmission line TL. The transmission line TL is connected in series between the high frequency power supply device 1 and the power transmission unit 2. In the present embodiment, the transmission line TL is a coaxial cable. The transmission line TL is not limited to the coaxial cable, and may be, for example, a coaxial tube, a line formed on a substrate, or the like.

The length of the transmission line TL is approximately one-fourth of the transmission wavelength of the high-frequency fundamental wave output by the high-frequency power supply device 1 in the transmission line TL. The high-frequency wavelength λ output by the high-frequency power supply device 1 is represented by λ [m] = ν [m / s] / f [Hz], where f is the frequency and ν is the speed of the radio wave in the transmission line TL. .. The radio wave velocity ν of the coaxial cable (polyethylene), so approximately 66% of the speed of a radio wave in vacuum ^{(3.0 × 10 8 [m /} s]), for example, the switching frequency f ₀ = 13.56 When [MHz], the frequency of the wavelength lambda of the high-frequency power supply device 1 ^{outputs, λ = (3.0 × 10 8} ) × (66/100) / (13.56 × 10 6) ≒ 14.60 [m ]. Since the length of the transmission line TL is approximately 1/4 of this wavelength λ, it is 14.60 × (1/4) ≈3.65 [m]. The speed ν of the radio wave on the coaxial cable is set to about 66% of the speed of the radio wave in vacuum, but the speed of the radio wave on the coaxial cable is the wavelength shortening rate of the coaxial cable used (specifically, that of the coaxial cable. Insulation material). Therefore, the length of the transmission line TL may be appropriately changed according to the type of coaxial cable used. Further, as can be seen from the above calculation formula, the lower the frequency of the high frequency output by the high frequency power supply device 1, the longer the wavelength λ. Therefore, when the frequency is low, it is necessary to use a long transmission line TL, and the size of the power transmission device A2 must be increased in order to accommodate the transmission line TL in the housing of the power transmission device A2. Therefore, it is desirable that the high frequency frequency output by the high frequency power supply device 1 is 6.78 MHz or more.

FIG. 10B is an extraction of the main part of the non-contact power transmission system S2 shown in FIG. 10A.

Let the output voltage of the high frequency power supply device 1 be V ₁ and the output current be I ₁ . Further, the output voltage of the current-voltage conversion circuit 52 is V ₂ , and the output current is I ₂ . Further, the output voltage of the power receiving unit 3 is V ₃ , and the output current is I ₃ . That is, the voltage applied to the rectifying / smoothing circuit 4 is V ₃ , and the current input to the rectifying / smoothing circuit 4 is I ₃ . The voltages V ₁ , V ₂ , V ₃ and the currents I ₁ , I ₂ , and I ₃ are all vectors.

When the circuit of the current-voltage conversion circuit 52 is represented by using the F parameter, the F parameter is as shown in the following equation (25). Each element A, B, C, D of the F parameter is a vector. Z ₀ is the characteristic impedance of the transmission line TL, β is the phase constant (2π / λ) (λ is the transmission wavelength in the transmission line TL), and l is the line length. Since the line length l of the transmission line TL is one-fourth of the transmission wavelength λ, β · l = π / 2. Therefore, the F parameter is as shown in the following equation (26). From this, the following equation (27) is obtained.

That is, when the circuit of the current-voltage conversion circuit 52 is represented by using the F parameter, it is the same as when the circuit of the current-voltage conversion circuit 51 is represented by using the F parameter ((13), (14) above). See formula). Further, since the power transmission unit 2 and the power reception unit 3 are common to the first embodiment, the equivalent circuits of the power transmission unit 2 and the power reception unit 3 are the same as those in FIG. 7C, and the above equation (20) is obtained. Be done. Therefore, in the case of the non-contact power transmission system S2, the above equation (22) is required as in the case of the non-contact power transmission system S1.

From the above equation (22), the magnitude of the current I ₃ output from the power receiving unit 3 is proportional to the magnitude of the output current I ₁ of the high frequency power supply device 1. Further, when the DC voltage output by the DC power supply device 11 is constant, the magnitude of the output current I ₁ of the high frequency power supply device 1 is constant, so that the magnitude of the output current I ₃ of the power receiving unit 3 is the magnitude of the connected load. It is constant regardless of impedance. That is, the output of the power receiving unit 3 can be considered as a constant current source that outputs a current I ₃ having a constant magnitude. Since the magnitude of the output current I ₃ of the power receiving unit 3 is constant, the current rectified by the rectifying smoothing circuit 4 becomes constant. Therefore, the current supplied to the power storage device D becomes substantially constant regardless of the charging state of the power storage device D.

Further, also in the case of the non-contact power transmission system S2, the above equation (24) is required as in the case of the non-contact power transmission system S1. Therefore, by setting Z _M <Z ₀ , the magnitude of the current I ₃ can be made larger than the magnitude of the current I ₁ .

FIG. 11 shows each waveform when a simulation is performed in the circuit shown in FIG. 10 (a). In FIG. 10A, the description of the power storage device D and the power receiving device B is omitted. Each condition of the simulation is the same as the simulation shown in FIG. That is, the DC voltage output by the DC power supply device 11 is 40 [V], the high frequency control signal CS2 is a rectangular wave signal with a frequency f = 13.56 [MHz], and the coupling coefficient of the transmission coil Lt and the power receiving coil Lr is k. = 0.2, 0.4, and the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and 100 [μF]. The self-inductances L _t and L _r are both 0.6 [μH], and the capacitances C _t and C _r are both 230 [pF]. Further, in this simulation, the characteristic impedance Z ₀ = 50 [Ω] of the transmission line TL is set. The line length of the transmission line TL is one-fourth (0.25λ) of the transmission wavelength λ. FIG. 11A shows the waveform of the drain-source voltage V _ds of the switching element Qs, and FIG. 11B shows the waveform of the output current I ₁ of the high frequency power supply device 1. Further, FIG. 11 (c) shows the waveform of the output current I ₃ of the power receiving unit 3, and FIG. 11 (d) shows the waveform of the current I _LOAD flowing through the capacitor of the rectifying smoothing circuit 4. In FIGS. 11C and 11D, the one having the larger amplitude is the one when the coupling coefficient is k = 0.2. In FIGS. 11A to 11D, the same waveform is obtained although there is a slight phase shift between the case where the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and the case where the capacitance is 100 [μF]. ing.

Where the amplitude of the current I ₁ is about 1.2 [A] as shown in FIG. 11 (b), the amplitude of the current I ₃ is as shown in FIG. 11 (c). When k = 0.4, it is about 2.5 [A], and when k = 0.2, it is about 5 [A]. That is, it was confirmed that the magnitude of the current I ₃ is larger than the magnitude of the current I ₁ , and the smaller the coupling coefficient k, the larger the magnitude of the current I ₂ .

Also in the second embodiment, the same effect as that of the first embodiment can be obtained. Further, in the case of the second embodiment, since the high frequency power supply device 1 and the power transmission unit 2 may be connected by a transmission line TL such as a coaxial cable having a predetermined length, a filter circuit combining an inductor and a capacitor is required. Do not.

However, in the case of the second embodiment, since the length of the transmission line TL is limited, it is necessary to use an unnecessarily long transmission line TL. Hereinafter, the case where the transmission line TL has a desired length will be described as the third and fourth embodiments.

FIG. 12 is a circuit diagram showing the configuration of the non-contact power transmission system S3 according to the third embodiment. Since the configuration of the power receiving device B is the same as that of the power receiving device B according to the first embodiment, only the power transmitting device A3 is described. In the non-contact power transmission system S3, the configuration of the current-voltage conversion circuit 53 is different from that of the current-voltage conversion circuit 51 of the non-contact power transmission system S1 according to the first embodiment.

The current-voltage conversion circuit 53 converts the current output into a voltage output. The current-voltage conversion circuit 53 includes a filter circuit in which an inductor L11'and two capacitors C11'and C12' are arranged in a π shape, and a transmission line TL'. The transmission line TL'has the same configuration as the transmission line TL of the current-voltage conversion circuit 52 according to the second embodiment, and its length is the transmission line TL of the high-frequency fundamental wave output by the high-frequency power supply device 1. It is x times (xλ) the transmission wavelength λ at ′. Further, the filter circuit has the same configuration as the filter circuit of the current-voltage conversion circuit 51 according to the first embodiment, and the self-inductance of the inductor L11'and the capacitance of the two capacitors C11'and C12' are the transmission line TL. It is designed according to the length xλ of'. In the current-voltage conversion circuit 53, the filter circuit is connected to the high frequency power supply device 1 side, and the transmission line TL'is connected to the power transmission unit 2 side.

Next, the design of the filter circuit will be described.

The filter circuit is designed to be equivalent to a transmission line with a line length l = (0.25-x) λ. That is, the filter circuit and the transmission line TL'are combined to be 0.25λ (a quarter of the transmission wavelength λ).

When the impedance of the capacitor C11'is Z1, the impedance of the inductor L11' is Z2, and the impedance of the capacitor C12' is Z3, the F parameter of the filter circuit is given by the following equation (28).

On the other hand, the F parameter of the transmission line having the characteristic impedance Z ₀ and the line length l is given by the following equation (29). Note that β is a phase constant (2π / λ).

The self-inductance of the inductor L11'and the capacitances of the two capacitors C11' and C12'are designed so that the F parameters of the above equations (28) and (29) match. The capacitance of the capacitor C11'and the capacitance of the capacitor C12'are the same. From the above equations (28) and (29), the following equations (30a) to (30d) can be obtained.

From the above equations (30b) and (30d) and Z1 = Z3, the following equation (31) is obtained.

Assuming that the self-inductance of the inductor L11'is L and the capacitances of the capacitors C11' and C12'are C, Z1 = Z3 = 1 / (j2πf ₀ C) and Z2 = j2πf ₀ L, so equations (30b) and (31) Therefore, the following equations (32) and (33) are obtained.

By designing the self-inductance L of the inductor L11'and the capacitance C of the capacitors C11' and C12' to be the above equations (32) and (33), the current-voltage conversion circuit 53 has a line length of 0.25λ. It is equivalent to a transmission line (a quarter of the transmission wavelength λ), and the current output can be converted into a voltage output.

When the circuit of the current-voltage conversion circuit 53 is represented by using the F parameter, it is the same as when the circuit of the current-voltage conversion circuits 51 and 52 is represented by using the F parameter ((13), (14) above). See formula). Further, since the power transmission unit 2 and the power reception unit 3 are common to the first embodiment, the equivalent circuits of the power transmission unit 2 and the power reception unit 3 are the same as those in FIG. 7C, and the above equation (20) is obtained. Be done. Therefore, in the case of the non-contact power transmission system S3, the above equation (22) is required as in the case of the non-contact power transmission system S1.

From the above equation (22), the magnitude of the current I ₃ output from the power receiving unit 3 is proportional to the magnitude of the output current I ₁ of the high frequency power supply device 1. Further, when the DC voltage output by the DC power supply device 11 is constant, the magnitude of the output current I ₁ of the high frequency power supply device 1 is constant, so that the magnitude of the output current I ₃ of the power receiving unit 3 is the magnitude of the connected load. It is constant regardless of impedance. That is, the output of the power receiving unit 3 can be considered as a constant current source that outputs a current I ₃ having a constant magnitude. Since the magnitude of the output current I ₃ of the power receiving unit 3 is constant, the current rectified by the rectifying smoothing circuit 4 becomes constant. Therefore, the current supplied to the power storage device D becomes substantially constant regardless of the charging state of the power storage device D.

Further, also in the case of the non-contact power transmission system S3, the above equation (24) is required as in the case of the non-contact power transmission system S1. Therefore, by setting Z _M <Z ₀ , the magnitude of the current I ₃ can be made larger than the magnitude of the current I ₁ .

FIG. 13 shows each waveform when a simulation is performed in the circuit shown in FIG. In FIG. 12, the description of the power storage device D and the power receiving device B is omitted. Each condition of the simulation is the same as the simulation shown in FIG. That is, the DC voltage output by the DC power supply device 11 is 40 [V], the high frequency control signal CS2 is a rectangular wave signal with a frequency f = 13.56 [MHz], and the coupling coefficient of the transmission coil Lt and the power receiving coil Lr is k. = 0.2, 0.4, and the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and 100 [μF]. The self-inductances L _t and L _r are both 0.6 [μH], and the capacitances C _t and C _r are both 230 [pF]. Further, in this simulation, the characteristic impedance Z ₀ = 50 [Ω] of the transmission line TL'is set, and the line length is set to 1/8 (x = 0.125) of the transmission wavelength λ. The self-inductance L of the inductor L11'and the capacitance C of the capacitors C11'and C12'are calculated as l = (0.25-0.125) λ in the above equations (32) and (33), and the self-inductance L is 0.415 [μH] and the capacitance C is 97 [pF]. FIG. 13 (a) shows the waveform of the drain-source voltage V _ds of the switching element Qs, and FIG. 13 (b) shows the waveform of the output current I ₁ of the high frequency power supply device 1. Further, FIG. 13 (c) shows the waveform of the output current I ₃ of the power receiving unit 3, and FIG. 13 (d) shows the waveform of the current I _LOAD flowing through the capacitor of the rectifying smoothing circuit 4. In FIGS. 13 (c) and 13 (d), the one having the larger amplitude is the one when the coupling coefficient is k = 0.2. In FIGS. 13 (a) to 13 (d), there is a slight phase shift between the case where the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and the case where the capacitance is 100 [μF], but the same waveform is obtained. ing.

Where the amplitude of the current I ₁ is about 1.2 [A] as shown in FIG. 13 (b), the amplitude of the current I ₃ is as shown in FIG. 13 (c). When k = 0.4, it is about 2.5 [A], and when k = 0.2, it is about 5 [A]. That is, it was confirmed that the magnitude of the current I ₃ is larger than the magnitude of the current I ₁ , and the smaller the coupling coefficient k, the larger the magnitude of the current I ₂ .

Also in the third embodiment, the same effect as that of the first embodiment can be obtained. Further, in the third embodiment, since the filter circuit may be designed according to the length of the transmission line TL', the transmission line TL'can be set to a desired length.

FIG. 14 is a circuit diagram showing the configuration of the non-contact power transmission system S4 according to the fourth embodiment. Since the configuration of the power receiving device B is the same as that of the power receiving device B according to the first embodiment, only the power transmitting device A4 is described. The non-contact power transmission system S4 differs from the current-voltage conversion circuit 51 of the non-contact power transmission system S1 according to the first embodiment in the configuration of the current-voltage conversion circuit 54.

The current-voltage conversion circuit 54 has the positions of the filter circuit and the transmission line TL'of the current-voltage conversion circuit 53 according to the third embodiment interchanged. That is, in the current-voltage conversion circuit 54, a filter circuit in which the inductor L11'and the two capacitors C11'and C12'are arranged in a π shape is connected to the power transmission unit 2 side, and the line length is x times the transmission wavelength λ ( The transmission line TL'which is xλ) is connected to the high frequency power supply device 1. The self-inductance of the inductor L11'and the capacitance of the two capacitors C11' and C12'are designed according to the length xλ of the transmission line TL', as in the case of the third embodiment. Therefore, the current-voltage conversion circuit 54 is equivalent to a transmission line having a line length of 0.25λ (a quarter of the transmission wavelength λ), and can convert the current output into a voltage output.

Also in the non-contact power transmission system S4 according to the fourth embodiment, the current supplied to the power storage device D is substantially constant regardless of the charging state of the power storage device D. Further, by making the impedance Z _M (= jω ₀ M) of the mutual inductance M due to the magnetic coupling between the power transmitting coil Lt and the power receiving coil Lr smaller than the characteristic impedance Z ₀ , the magnitude of the output current of the power receiving unit 3 can be reduced. It can be made larger than the magnitude of the output current of the high frequency power supply device 1.

Also in the fourth embodiment, the same effect as that of the first embodiment can be obtained. Further, in the fourth embodiment, the filter circuit may be designed according to the length of the transmission line TL', so that the transmission line TL'can be set to a desired length.

In the third and fourth embodiments, the case where the filter circuit of the current-voltage conversion circuit 53 (54) is a filter circuit in which the inductor L11 and the two capacitors C11 and C12 are arranged in a π shape has been described. The circuit configuration of the filter circuit is not limited to that described above. For example, it may be a filter circuit in which two inductors and one capacitor are arranged in a π type (see FIG. 9A), or a filter circuit in which one inductor and two capacitors are arranged in a T type (FIG. 9). (See (b)) may be used, or a filter circuit (see FIG. 9 (c)) in which two inductors and one capacitor are arranged in a T shape may be used.

The design of the filter circuit when the filter circuit is a filter circuit in which two inductors and one capacitor are arranged in a T shape (see FIG. 15) will be described.

The filter circuit is designed to be equivalent to a transmission line with a line length l = (0.25-x) λ. That is, the filter circuit and the transmission line TL'are combined to be 0.25λ (a quarter of the transmission wavelength λ).

When the impedance of the inductor L11 "is Z1, the impedance of the capacitor C11" is Z2, and the impedance of the inductor L12 "is Z3, the F parameter of the filter circuit is given by the following equation (34).

On the other hand, the F parameter of the transmission line having the characteristic impedance Z ₀ and the line length l is given by the following equation (35). Note that β is a phase constant (2π / λ).

The self-inductance of the two inductors L11 "and L12" and the capacitance of the capacitor C11 "are designed so that the F parameters of the above equations (34) and (35) match. The self-inductance of the inductor L11" and the inductor The self-inductance of L12 ”is the same. From the above equations (34) and (35), the following equations (36a) to (36d) can be obtained.

From the above equation (36c), it becomes the following equation (37), and from the above equation (36d) and the following equation (37), it becomes the following equation (38 equation).

Assuming that the self-inductance of the inductors L11 "and L12" is L and the capacitance of the capacitor C11 "is C, Z1 = Z3 = j2πf ₀ L and Z2 = 1 / (j2πf ₀ C), so equations (37) and (38). Therefore, the following equations (39) and (40) are obtained.

By designing the self-inductance L of the inductors L11 "and L12" and the capacitance C of the capacitor C11 "to be the above equations (39) and (40), the current-voltage conversion circuit 53 (54) has a line length. It is equivalent to a transmission line of 0.25λ (a quarter of the transmission wavelength λ), and the current output can be converted into a voltage output.

FIG. 16 shows each waveform when a simulation is performed in the circuit shown in FIG. In FIG. 15, the description of the power storage device D and the power receiving device B is omitted. Each condition of the simulation is the same as the simulation shown in FIG. That is, the DC voltage output by the DC power supply device 11 is 40 [V], the high frequency control signal CS2 is a rectangular wave signal with a frequency f = 13.56 [MHz], and the coupling coefficient of the transmission coil Lt and the power receiving coil Lr is k. = 0.2, 0.4, and the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and 100 [μF]. The self-inductances L _t and L _r are both 0.6 [μH], and the capacitances C _t and C _r are both 230 [pF]. Further, in this simulation, the characteristic impedance Z ₀ = 50 [Ω] of the transmission line TL'is set, and the line length is set to 0.15 times the transmission wavelength λ (x = 0.15). The self-inductance L of the inductors L11 "and L12" and the capacitance C of the capacitor C11 "are calculated as l = (0.25-0.15) λ in the above equations (39) and (40), and the self-inductance. L is 0.299 [μH] and the capacitance C is 190 [pF]. FIG. 16A shows a waveform of the drain-source voltage V _ds of the switching element Qs, and FIG. 16 (b). ) Shows the waveform of the output current I ₁ of the high-frequency power supply device 1. FIG. 16 (c) shows the waveform of the output current I ₃ of the power receiving unit 3, and FIG. 16 (d) shows the waveform of the output current I ₃ . The waveform of the current I _LOAD flowing through the capacitor of the rectifying smoothing circuit 4 is shown. In FIGS. 16C and 16D, the larger amplitude is the one when the coupling coefficient is k = 0.2. In FIGS. 16A to 16D, there is a slight phase shift between the case where the capacitance of the capacitor of the rectifying smoothing circuit 4 is 1 [μF] and the case where the capacitance is 100 [μF], but the same waveform is obtained. It has become.

Where the amplitude of the current I ₁ is about 1.2 [A] as shown in FIG. 16 (b), the amplitude of the current I ₃ is as shown in FIG. 16 (c). When k = 0.4, it is about 2.5 [A], and when k = 0.2, it is about 5 [A]. That is, it was confirmed that the magnitude of the current I ₃ is larger than the magnitude of the current I ₁ , and the smaller the coupling coefficient k, the larger the magnitude of the current I ₂ .

In the first to fourth embodiments, the case where the current-voltage conversion circuit as hardware is provided between the high frequency power supply device 1 and the power transmission unit 2 has been described, but the present invention is not limited to this. For example, the current output from the high frequency power supply device 1 may be converted into a voltage output by control by FPGA (field-programmable gate array) or the like.

In the first to fourth embodiments, the case where the high frequency power supply device 1 uses an amplifier in which the resonance circuit LC2 is added to the so-called class E amplifier has been described, but the present invention is not limited to this. For example, a so-called class E amplifier in which the resonance circuit LC2 is not provided may be used. The high frequency power supply device 1 may be a so-called constant current source.

In the first to fourth embodiments, the case where the power transmission devices A1 to A4 (hereinafter referred to as “power transmission device A”) is embedded in the floor surface (see FIG. 1) has been described, but the present invention is limited to this. I can't. For example, only the power transmission coil may be embedded in the floor surface, or the power transmission coil may be arranged on the floor surface without being embedded in the floor surface. Further, the present invention is not limited to the case where the power transmission coil and the power reception coil are provided so as to be substantially parallel to the floor surface. For example, as shown in FIG. 17A, the power receiving device B is arranged at the rear part of the vehicle body, the power transmitting device A is arranged on the wall surface of the garage, and the power transmitting coil and the power receiving coil are substantially perpendicular to the floor surface. It may be. Further, as shown in FIG. 17B, the power receiving device B is arranged on the side surface of the vehicle body, the power transmitting device A is arranged on the wall surface of the garage, and the power transmitting coil and the power receiving coil are substantially perpendicular to the floor surface. It may be. In short, the power transmission coil and the power reception coil may be arranged in the vehicle body and the garage (parking lot), respectively, so that they can be arranged at positions facing each other in a substantially parallel manner.

In the first to fourth embodiments, the case where the non-contact power transmission system according to the present invention is used for charging the power storage device built in the electric vehicle has been described as an example, but the present invention is not limited to this. For example, it can also be used to charge the power storage device of an automatic guided vehicle used for transportation in a factory. The present invention can also be applied to the case of charging a power storage device of an electric product such as a power tool or a notebook computer. The present invention can also be applied to a case where power is directly supplied to a load such as an electric product connected to a power receiving device instead of charging the power storage device. In this case, a constant current can be supplied to the load. Further, if the high frequency power is supplied to the load as it is, the rectifying smoothing circuit 4 may not be provided. Further, the rectified DC power may be converted into an appropriate AC power by the inverter circuit and used.

The non-contact power transmission system according to the present invention is not limited to the above-described embodiment. The specific configuration of each part of the non-contact power transmission system according to the present invention can be freely redesigned.

A1, A2, A3, A4, A', A Transmission device 1 High frequency power supply device 11 DC power supply device 12 Control device C1, C2, C3, C4, C10 Capacitor L1, L2, L3 Inductor LC1 filter LC2 Resonance circuit (second Resonant circuit)
LC3 resonant circuit (first resonant circuit)
Qs switching element D1 diode 51, 52, 53, 54 Current-voltage conversion circuit (conversion means)
L11, L11', L11 ", L12" Inductor C11, C12, C11', C12', C11 "Capacitor TL, TL' Transmission line 2 Transmission unit Lt Transmission coil Ct Resonance capacitor B, B'Power receiving device 3,3' Unit Lr Power receiving coil Cr Resonance capacitor 4 Rectification smoothing circuit S1, S2, S3, S4, S'Non-contact power transmission system D Power storage device

Claims (5)

A high-frequency power supply that outputs a constant high-frequency current,
A power transmission device that transmits the power output by the high-frequency power supply device, and
A power receiving device that receives power transmitted from the power transmission device in a non-contact manner with the power transmission device,
With
The power transmitting device, current converts the current output from the high frequency power supply voltage output - see including a voltage conversion circuit,
The current-voltage conversion circuit
A filter circuit in which the inductor and capacitor are arranged in a T-type or π-type,
Transmission line and
With
The self-inductance of the inductor and the capacitance of the capacitor are designed so as to be equivalent to the case where the length of the transmission line is approximately one-fourth of the transmission wavelength of the transmission line.
A non-contact power transmission system characterized by that. The power transmission device includes a power transmission unit having a power transmission coil for transmitting electric power and a resonance capacitor connected in series to the power transmission coil to form a resonance circuit.
The power receiving device includes a power receiving unit having a power receiving coil for receiving power and a resonance capacitor connected in series to the power receiving coil to form a resonance circuit.
The non-contact power transmission system according to claim 1 . The impedance of the mutual inductance due to the magnetic coupling between the power transmitting coil and the power receiving coil is smaller than the characteristic impedance of the current-voltage conversion circuit.
The non-contact power transmission system according to claim 2 . In the filter circuit , one inductor and two capacitors are arranged in a π type .
When the length of the transmission line is x times the transmission wavelength λ of the transmission line at the frequency f ₀ output by the high frequency power supply device,
The self-inductance L of the inductor and the capacitance C of the capacitor are designed based on the following equation.
The non-contact power transmission system according to any one of claims 1 to 3 .
The characteristic impedance of the transmission line is Z _0, and β is a phase constant (2π / λ).
In the filter circuit , two inductors and one capacitor are arranged in a T shape .
When the length of the transmission line is x times the transmission wavelength λ of the transmission line at the frequency f ₀ output by the high frequency power supply device,
The self-inductance L of the inductor and the capacitance C of the capacitor are designed based on the following equation.
The non-contact power transmission system according to any one of claims 1 to 3 .
The characteristic impedance of the transmission line is Z _0, and β is a phase constant (2π / λ).