JP5726349B2 - Power module - Google Patents

Description

  The present invention relates to a power module used for motor control of industrial / consumer equipment.

  In power switching semiconductor devices such as IGBTs (Insulated Gate Bipolar Transistors) and power MOSFETs (Metal-Oxide Semiconductor Field-Effect Transistors), a power switching semiconductor device is connected to a main element through which a main current flows for overcurrent detection. Sense current output from the output terminal (sense emitter) of the current sense element is converted into a voltage by a resistor (current detection resistor) and detected. A method is adopted that determines whether the detected voltage is normal or abnormal (overcurrent level) by comparing the detected voltage with a predetermined reference voltage.

  Here, the current sense element has a collector (drain) in common with the main element, but with respect to the area of the emitter (source) of the main element so that the sense current flows with a constant shunt ratio with respect to the main current. Thus, the emitter (source) area is set at a predetermined area ratio.

  For example, when the emitter area ratio of the current sense element to the main element is 1/10000, a current of 1/10000 of the main element flows through the current sense element, and current detection can be performed with a relatively small resistance. it can.

  Here, when a current detection resistor is connected to the current sense element, a difference occurs in the voltage applied to the gate of the main element and the current sense element, and the current shunt ratio varies. Since the fluctuation of the shunt ratio increases when the current detection resistance is large, detection with a small resistance value is required.

  However, if detection is performed with a small resistance value, the threshold voltage (reference voltage) for determining overcurrent becomes small, causing malfunction (false detection).

  In FIG. 1 of Patent Document 1, a sense current is not directly detected by a resistor, but a sense current is received by a current mirror circuit composed of an N-channel MOS transistor, and a mirror current (current I4) obtained by the current mirror circuit is shown. ) Is converted into a voltage by a current detection resistor (resistor R1) connected to the power supply (voltage V3) of the current mirror circuit to be a detection voltage (voltage V1).

  In this configuration, the detection voltage V1 is V1 = V3− (I4 × R1), and the detection voltage V1 depends on the voltage V3 of the power supply. Therefore, the detection voltage V1 varies due to the variation of the voltage V3, and the current detection accuracy decreases. There was a possibility.

  A similar problem is the combination of a current mirror circuit that receives a sense current to generate a mirror current and a current mirror circuit that generates a reference current as the mirror current, and the overcurrent depends on the magnitude relationship between the mirror current of the sense current and the reference current. This also occurs in Patent Document 2 (FIGS. 1 and 2) that determines whether or not there is any. Also in this case, if the power supply voltage of the current mirror circuit fluctuates, the mirror current fluctuates, which may reduce the current detection accuracy.

Japanese Patent Laid-Open No. 10-322185 Japanese Patent Laid-Open No. 1-193909

  As described above, in the conventional configuration for detecting the overcurrent of the power switching semiconductor device, when the sense current is detected by the resistor, the fluctuation increases as the current detection resistor is increased, and the current detection resistor is decreased. Then, there is a problem that erroneous detection is likely to occur, and when the sense current is detected using a current mirror circuit, there is a problem that current detection accuracy decreases due to the influence of power supply fluctuation.

  The present invention has been made to solve the above-described problems. Whether the sense current is detected by a resistor or detected using a current mirror circuit, erroneous detection or current It is an object to provide a power module that does not cause a decrease in detection accuracy.

An aspect of the power module according to the present invention includes a main element through which a main current flows and a current sense element configured to allow a part of the main current to flow, and a sense current is output from an output terminal of the current sense element. An output power switching semiconductor device, a first transistor having a first main electrode connected to the output terminal of the current sensing element, and one end connected to a second main electrode of the first transistor A current detection resistor having the other end connected to a common connection, and a control electrode of the first transistor connected to the potential of the output end of the main element, and the current detection resistor Is detected as a current detection voltage, and is compared with a predetermined threshold voltage, and the power switching semiconductor is determined by the magnitude relationship between the two. An overcurrent determination circuit that determines whether or not an overcurrent flows in the device, and a drive circuit that generates a control signal applied to a control electrode of the power switching semiconductor device, wherein the first transistor is bipolar The first main electrode is an emitter electrode when the first transistor is a bipolar transistor; the first main electrode is a source electrode when the first transistor is a MOSFET; The second main electrode is a collector electrode when the first transistor is the bipolar transistor, and a drain electrode when the first transistor is a MOSFET.

  According to the aspect of the power module according to the present invention, when the on-voltage of the power switching semiconductor device is low, when a current detection resistor is connected to the current sense element, there is a difference in the voltage applied to the control electrode of the main element and the current sense element. And current shunt ratio fluctuates. As a result, an accurate sense current cannot be obtained. However, since the first transistor is connected to the output terminal of the current sense element, voltage fluctuation at the output terminal of the current sense element is caused by the on-resistance of the first transistor. For example, it is suppressed to about 0.7V. As a result, the voltage difference between the voltage applied to the control electrode of the main element and the current sense element is suppressed to about 0.7 V and stabilized, and the detection accuracy of the sense current is improved. Since the voltage difference between the voltage applied to the control electrode of the main element and the current sense element is suppressed to about 0.7 V, there is no need to consider that the current shunt ratio fluctuates between the main element and the current sense element. Since the resistance value of the current detection resistor can be set arbitrarily, erroneous detection can be prevented by increasing the resistance value of the current detection resistor.

It is a circuit diagram which shows the structure of the power module of Embodiment 1 which concerns on this invention. It is the figure which set the simulation conditions of the power module of Embodiment 1 which concerns on this invention. It is a figure which shows the simulation result of the power module of Embodiment 1 which concerns on this invention. It is a circuit diagram which shows the structure of the power module of Embodiment 2 which concerns on this invention. It is the figure which set the simulation conditions of the power module of Embodiment 2 which concerns on this invention. It is a figure which shows the simulation result of the power module of Embodiment 2 which concerns on this invention. It is a figure which shows the structure of the modification 1 of Embodiment 1 which concerns on this invention. It is a figure which shows the modification 1 structure of Embodiment 1 which concerns on this invention. It is a figure which shows the structure of the modification 2 of Embodiment 1 which concerns on this invention. It is a figure which shows the structure of the modification 2 of Embodiment 1 which concerns on this invention. It is a figure which shows the structure of the modification 2 of Embodiment 1 which concerns on this invention. It is a figure which shows the structure of the modification 2 of Embodiment 1 which concerns on this invention. It is sectional drawing which shows the structure of RC-IGBT.

  Hereinafter, an IGBT will be described as an example of a power switching semiconductor device, but the present invention can also be applied to other power switching semiconductor devices such as MOSFE and bipolar transistors. In addition, although the conductive type of the power switching semiconductor device is described as being an N channel type, it goes without saying that it may be a P channel type.

<Embodiment 1>
<Device configuration>
FIG. 1 is a circuit diagram showing a configuration of a power module 100 according to Embodiment 1 of the present invention. As shown in FIG. 1, the power module 100 includes a drive control circuit 10 that drives the IGBT 1 to an on state or an off state by controlling a voltage (gate voltage) supplied between the gate and the emitter of the IGBT 1. .

  The drive control circuit 10 includes a drive circuit C1 that uses the DC power supply V1 as a drive power supply, an overcurrent determination circuit C2, and a current detection circuit C3.

  A free wheel diode 2 is connected in reverse parallel to the IGBT 1 between the collector C and the emitter E. The freewheeling diode 2 receives a return current from the main circuit when the IGBT 1 is in an off state.

  In the power module 100 shown in FIG. 1, the main power supply potential VCC is applied to the collector C of the IGBT 1, the emitter E is grounded, and the ground potential GND (first reference potential) is applied. The output of the driver DR of the drive circuit C1 is given as a control signal to the gate G of the IGBT1.

  The driver DR uses the DC power supply V1 as a drive power supply, and the negative electrode of the DC power supply V1 is connected to the common connection portion BP of the drive control circuit 10 to provide the reference potential of the drive control circuit 10, and the driver DR The two input parts are respectively connected to the positive electrode of the DC power source V1 and the common connection part BP.

  The IGBT 1 has a main element through which a main current flows and a current sense element configured to allow a part of the main current to flow for detection of an overcurrent. A sense current is output from an output terminal (sense emitter) of the current sense element. Is output.

  The current sensing element has a collector (drain) common to the main element, but a predetermined current ratio with respect to the area of the emitter (source) of the main element so that a sense current flows at a constant shunt ratio with respect to the main current. The area of the emitter (source) is set by the area ratio.

  The current detection circuit C3 has an emitter connected to the sense emitter SE of the IGBT 1, a PNP transistor Q5 whose base is grounded, a collector connected to the collector of the PNP transistor Q5, and another end connected to the common connection BP. And a current detection resistor SR. A potential difference based on the common connection portion BP generated by the current detection resistor SR is defined as a current detection voltage Vs.

  The overcurrent determination circuit C2 includes a comparator CP that operates based on the power supply potential Vc with the potential of the common connection BP as a reference potential. One input of the comparator CP is one of the collector of the PNP transistor Q5 and the current detection resistor SR. One input of the comparator CP is connected to the positive electrode of the DC power supply V3 that supplies an arbitrary threshold voltage, and the negative electrode of the DC power supply V3 is connected to the common connection portion BP. Yes.

  Note that the comparator CP compares the current detection voltage Vs with the threshold voltage, determines whether an overcurrent is flowing through the IGBT 1 based on the magnitude relationship between the two, and the result is given to the drive circuit C1 to provide the driver DR Used for control. When the current detection voltage Vs indicates an overcurrent level, control such as turning off the IGBT 1 is performed. However, since it is not related to the present invention, further explanation is omitted.

  Further, the positive electrode of the DC power source V2 included in the drive circuit C1 is grounded, and the negative electrode is connected to the common connection portion BP. The positive electrode of the DC power supply V2 is grounded together with the emitter E of the IGBT1. Note that a P-channel MOSFET transistor can be used instead of the PNP transistor Q5. In that case, transistors other than the IGBT 1 are also formed of MOSFETs.

<Device operation>
Next, the current detection operation of the power module 100 will be described. In the power module 100, the drive control circuit 10 has its own common connection BP, and a negative bias is applied from the DC power source V2 to the drive circuit reference potential (second reference potential). Since the DC power supply V1 drives the driver DR with reference to the drive circuit reference potential, a positive bias and a negative bias are applied as control signals to the gate of the IGBT1. Note that the DC power supply V2 sets a negative potential, and may be referred to as potential setting means.

  FIG. 3 shows a simulation result of the current detection operation in the power module 100 when the positive bias and the negative bias are applied to the gate of the IGBT 1 in this way. FIG. 2 is a diagram in which the simulation conditions are set by specifying the components of the current detection circuit C3 and the drive circuit C1 for performing the simulation. In FIG. 2, the same components as those in FIG.

  In FIG. 2, the IGBT 1 is divided into a main element MT and a current sensing element ST, and the gate-emitter voltage (gate voltage) of the main element MT is represented as Vge and the collector-emitter voltage is represented as Vce. Further, the current flowing through the entire IGBT 1 is represented as a main current Ic, and the current flowing through the current sense element ST is represented as a sense current Is. Further, the current flowing through the PNP transistor Q5 is represented as a current Ie.

  In the drive circuit C1, the driver DR includes an NPN transistor Q1 having a collector connected to the positive electrode of the DC power supply V1, an emitter connected to the gate of the current sensing element ST via the resistor R1, and a collector connected to the common connection BP. A PNP transistor Q2 having an emitter connected to the gate of the current sensing element ST via a resistor R2, and a pulse signal source for applying a pulse signal having a height of 0V to 20V to the bases of the NPN transistor Q1 and the PNP transistor Q2 VP. The pulse signal source VP is connected to the common connection portion BP and uses the drive circuit reference potential as a reference. The resistor R1 is a resistor that sets a switching speed when the IGBT 1 is on, and the resistor R2 is a resistor that sets a switching speed when the IGBT 1 is off.

  The DC power source V1 is a power source that generates 20V as the potential B, and the DC power source V2 is a power source that generates −5V as the potential A. The main power supply potential VCC is set to 200V.

  Further, the inductance L1 of the load between the collector of the IGBT 1 and the positive electrode of the main power supply PW is set to 500 μH, and the resistance value of the current detection resistor SR is set to 12Ω.

  As for the result of the simulation performed under the simulation conditions shown in FIG. 2, the waveform of the gate voltage Vge is shown in FIG. 3A, and the waveform of the collector-emitter voltage Vce and the main current Ic is shown in FIG. FIG. 3C shows the waveform of the sense current Is, FIG. 3D shows the waveform of the current Ie flowing through the PNP transistor Q5, and FIG. 3E shows the waveform of the current detection voltage Vs. Shown in

  As shown in FIG. 3B, the IGBT 1 is turned on and off and the IGBT 1 is turned on in response to the rise and fall of the gate voltage Vge, which is the pulse signal shown in FIG. 3A. Thus, the main current Ic flows, and at the same time, the sense current Is flows as shown in part (c) of FIG. Similar to the sense current Is, the current Ie flows through the PNP transistor Q5 as shown in FIG. 3 (d), and the current detection voltage Vs is accordingly changed as shown in FIG. 3 (e). can get.

  Here, as shown in FIG. 3A, the gate voltage Vge has a pulse waveform to which not only a positive bias of 0V to 15V but also a negative bias of 0V to −5V is applied. As described above, by using the pulse signal formed with the positive bias and the negative bias as the gate voltage, the IGBT 1 can be turned off more reliably.

  That is, even if the pulse signal is formed only with a positive bias, the IGBT can be turned off if the IGBT gate-emitter voltage is equal to or lower than the threshold voltage of the IGBT, but is formed including a negative bias. If it is a pulse signal, the IGBT can be turned off more reliably.

  Further, when a pulse signal formed with a positive bias and a negative bias is used as a gate voltage, even when an on-voltage of a power switching device such as an IGBT is lower than when a pulse signal formed only with a positive bias is used. There is also an advantage that malfunction becomes difficult.

  In addition, when the on-voltage of the power switching device is low, if a current detection resistor is connected to the current sense element, a voltage difference (ΔVge) between the voltage applied to the main element and the gate of the current sense element by the voltage drop across the current detection resistor Occurs. Since the voltage drop at the current detection resistor increases as the current flows, the voltage drop increases particularly when overcurrent is detected. ΔVge also increases and the current shunt ratio varies. As a result, an accurate sense current cannot be obtained. However, since the PNP transistor Q5 is connected to the sense emitter SE of the IGBT 1, voltage fluctuation at the sense emitter SE is caused to be on resistance of the PNP transistor Q5, for example, about 0.7V. Can be suppressed. As a result, ΔVge is suppressed to about 0.7 V and stabilized, and the detection accuracy of the sense current is improved.

  Further, since ΔVge can be suppressed to about 0.7 V regardless of the resistance value of the current detection resistor SR, it is not necessary to consider that the current shunt ratio fluctuates between the main element MT and the current sense element ST. Since the resistance value of SR can be arbitrarily set, erroneous detection can be prevented by increasing the resistance value of current detection resistor SR.

  In the overcurrent determination circuit C2, the overcurrent state is determined by comparing the current detection voltage Vs with an arbitrary threshold voltage by the comparator CP. However, the threshold voltage is a common connection having the lowest potential (negative potential). Since the drive circuit reference potential which is the potential of the part BP is used as a reference, the drive circuit reference potential does not fluctuate even if the DC power supply V1 fluctuates, and highly accurate current detection is possible. When the voltage of the DC power supply V2 fluctuates, the drive circuit reference potential also fluctuates. However, when the drive circuit reference potential fluctuates, not only the DC power supply V3 but also the reference potentials of all the circuits fluctuate in the same way. The fluctuation becomes zero, and high-precision current detection can be maintained.

<Embodiment 2>
<Device configuration>
FIG. 4 is a circuit diagram showing a configuration of a power module 200 according to the second embodiment of the present invention. As shown in FIG. 4, the power module 200 includes a drive control circuit 20 that drives the IGBT 1 to an on state or an off state by controlling a voltage (gate voltage) supplied between the gate and the emitter of the IGBT 1. . In addition, the same code | symbol is attached | subjected about the structure same as the power module 100 shown in FIG. 1, and the overlapping description is abbreviate | omitted.

  The drive control circuit 20 includes a drive circuit C1 that uses the DC power supply V1 as a drive power supply, an overcurrent determination circuit C2, and a current detection circuit C4. The current detection circuit C4 is different from the drive control circuit 10 shown in FIG.

  The current detection circuit C4 includes PNP transistors Q3 and Q4 whose emitters are connected to the sense emitter SE of the IGBT1, and a current detection resistor whose one end is connected to the collector of the PNP transistor Q4 and whose other end is connected to the common connection portion BP. SR. The bases of the PNP transistors Q3 and Q4 are commonly connected to the collector of the PNP transistor Q3, and the PNP transistors Q3 and Q4 constitute a current mirror circuit.

  The collector of the PNP transistor Q3 is grounded, the negative electrode of the DC power supply V2 is connected to the common connection BP, and the positive electrode of the DC power supply V2 is grounded together with the emitter E of the IGBT1.

  A connection node ND between the collector of the PNP transistor Q4 and one end of the current detection resistor SR is connected to one input of the comparator CP.

<Device operation>
Next, the current detection operation of the power module 200 will be described. In the power module 200, the drive control circuit 20 has its own common connection BP, and a negative bias is applied from the DC power source V2 to the drive circuit reference potential. Since the DC power supply V1 drives the driver DR with reference to the drive circuit reference potential, a positive bias and a negative bias are applied as control signals to the gate of the IGBT1.

  FIG. 6 shows a simulation result of the current detection operation in the power module 200 when the positive bias and the negative bias are applied to the gate of the IGBT 1 in this way.

  FIG. 5 is a diagram in which the simulation conditions are set by specifying the components of the current detection circuit C4 and the drive circuit C1 for performing the simulation. In FIG. 5, the same components as those in FIG. 1 are denoted by the same reference numerals, the simulation conditions are also the same, and a duplicate description is omitted.

  In FIG. 5, the current flowing through the entire IGBT 1 is represented as a main current Ic, the current flowing through the current sensing element ST is represented as a sense current Is, and the currents flowing through the PNP transistors Q3 and Q4 are represented as a current Ie. Note that the transistor characteristics of the PNP transistors Q3 and Q4 are the same, and the current Ie is a half of the sense current Is.

  As a result of the simulation performed under the simulation conditions shown in FIG. 5, the waveform of the gate voltage Vge is shown in FIG. 6 (a), and the waveform of the collector-emitter voltage Vce and the main current Ic is shown in FIG. 6 (b). FIG. 6C shows the waveform of the sense current Is, FIG. 6D shows the waveform of the current Ie flowing through the PNP transistor Q4, and FIG. 6E shows the waveform of the current detection voltage Vs. Shown in

  As shown in FIG. 6B, the IGBT 1 is turned on and off and the IGBT 1 is turned on in response to the rise and fall of the gate voltage Vge which is the pulse signal shown in FIG. 6A. Thus, the main current Ic flows, and at the same time, the sense current Is flows as shown in part (c) of FIG. Then, as shown in FIG. 6 (d), about half of the sense current Is, current Ie flows to the PNP transistor Q4, and accordingly, the current detection voltage Vs is shown in FIG. 6 (e). Is obtained.

  As described above, in the drive control circuit 20, since the current mirror circuit receives the output of the sense emitter SE of the IGBT 1 by the current mirror circuit, the current Ie, which is about half of the sense current Is, flows through the current detection resistor SR. The power consumption at the resistor SR can be reduced.

  For example, if it is determined that an overcurrent occurs when the main current Ic is 100 A, the shunt ratio of the current sense element to the main element is 1/10000, and the current detection voltage Vs is 0.5 V, the first embodiment will be described. In the drive control circuit 10, the power consumption in the current detection resistor SR is Vs × Is = 0.5 × (100/10000) = 5 mW. On the other hand, in the drive control circuit 20, the power consumption in the current detection resistor SR is Vs × (1/2) Is = 0.5 × (50/10000) = 2.5 mW.

  In this way, by adopting a configuration in which the output of the sense emitter SE of the IGBT 1 is received by the current mirror circuit, it is possible to change the transistor size (size ratio) of the current mirror circuit or to provide a plurality of transistors that generate mirror currents. It is possible to arbitrarily change the current flowing through the detection resistor SR.

  For example, if the size ratio of the PNP transistor Q4 to the PNP transistor Q3 is 10 to 1, a current Ie that is about 1/10 of the sense current Is flows through the PNP transistor Q4.

<Modification 1>
In the first and second embodiments described above, the drive circuit reference potential is set by applying a negative bias from the DC power supply V2 to the common connection portion BP. However, instead of the DC power supply V2, the potential B of the DC power supply V1 is a resistance. A configuration may be adopted in which a negative bias is obtained by dividing, and a negative bias is obtained by a Zener diode.

  FIG. 7 shows a configuration for obtaining a negative bias by resistance division, and FIG. 8 shows a configuration for obtaining a negative bias by a resistor and a Zener diode. 7 and 8, the same components as those shown in FIGS. 2 and 5 are denoted by the same reference numerals, and redundant description is omitted.

  In the power module 100A shown in FIG. 7, resistors R4 and R5 that divide the potential B of the DC power supply V1 are inserted in series in the order of resistors R4 and R5 between the positive electrode of the DC power supply V1 and the common connection BP. The connection node between the resistors R4 and R5 is connected to the base of the PNP transistor Q5 and grounded together with the emitter E of the IGBT1.

  In such a configuration, there is an advantage that a negative bias determined by the resistance division ratio, for example, −5 V can be applied to the common connection portion BP with the potential A as a reference, and the DC power supply V2 is not necessary. Since the negative potentials can be set by the resistors R4 and R5, these are referred to as potential setting means PS.

  In the power module 100B shown in FIG. 8, a resistor R4 and a Zener diode Z1 are inserted in series in this order between the positive electrode of the DC power supply V1 and the common connection portion BP. The anode of the Zener diode Z1 is connected to the common connection BP, the cathode of the Zener diode Z1 is connected to the resistor R4, and the connection node is connected to the base of the PNP transistor Q5 and grounded together with the emitter E of the IGBT1. Yes.

  In such a configuration, there is an advantage that a negative bias determined by the Zener voltage of the Zener diode Z1, for example, −5V, can be applied to the common connection portion BP with the potential A as a reference, and the DC power supply V2 becomes unnecessary.

  Since the negative bias is defined by the Zener voltage of the Zener diode Z1, the negative bias can be easily set by using a Zener diode having a desired Zener voltage. Since the negative potential can be set by the resistor R4 and the Zener diode Z1, these are called potential setting means PS.

  7 and 8 are shown as modified examples of the power module 100, but may be applied to the power module 200.

<Modification 2>
The power module 100 of the first embodiment shows a configuration in which the positive electrode of the DC power supply V2 is connected to the base of the PNP transistor Q5. In the power module 200 of the second embodiment, the positive electrode of the DC power supply V2 is connected to the PNP transistor. Although the configuration connected to the collector of Q3 is shown, as shown in FIGS. 2 and 5, the gate voltage (gate-emitter voltage) of the current sense element ST is the base-emitter voltage of the PNP transistor Q5 and the PNP transistor Q4. Therefore, the voltage is about 0.7 V lower than the gate voltage (gate-emitter voltage) of the main element MT. For this reason, the current shunt ratio of the current sense element ST with respect to the main element MT may fluctuate, and the current detection accuracy may decrease.

  In order to avoid this, the configuration shown in FIGS. 9 and 10 may be adopted. That is, FIG. 9 shows a configuration of a power module 100C that applies a potential D obtained by lowering the potential A of the DC power supply V2 to the base of the PNP transistor Q5, and FIG. 10 shows the potential A of the DC power supply V2 as a predetermined. The configuration of a power module 200A that applies a potential D that has been lowered to the base of a PNP transistor Q4 is shown. 9 and 10, the same components as those illustrated in FIGS. 2 and 5 are denoted by the same reference numerals, and redundant description is omitted.

  In the power module 100C shown in FIG. 9, a diode D2 and a resistor R6 are inserted in series in this order between the positive electrode of the DC power source V2 and the common connection BP, and the connection between the diode D2 and the resistor R6 is connected. The node is connected to the base of the PNP transistor Q5.

  The diode D2 is connected to the DC power supply V2 in the forward direction, and can generate a potential D obtained by dropping the potential A by a diode built-in voltage (voltage between pn), that is, about 0.7V. By applying this to the base of the PNP transistor Q5, the gate voltage drop of the current sense element ST is canceled, and the gate voltage (gate-emitter voltage) can be matched between the main element MT and the current sense element ST. Therefore, it is possible to suppress the fluctuation of the current shunt ratio of the current sense element ST with respect to the main element MT, and it is possible to detect the current with higher accuracy.

  In the power module 200A shown in FIG. 10, a diode D2 and a resistor R6 are inserted in series in this order between the positive electrode of the DC power supply V2 and the common connection BP, and the connection between the diode D2 and the resistor R6 is connected. The node is connected to the base of the PNP transistor Q4.

  The diode D2 is connected to the DC power supply V2 in the forward direction, and can generate a potential D obtained by dropping the potential A by a diode built-in voltage (voltage between pn), that is, about 0.7V. By applying this to the base of the PNP transistor Q4, the gate voltage drop of the current sense element ST is canceled, and the gate voltage (gate-emitter voltage) can be matched between the main element MT and the current sense element ST. Therefore, it is possible to suppress the fluctuation of the current shunt ratio of the current sense element ST with respect to the main element MT, and it is possible to detect the current with higher accuracy.

  9 and 10 show a configuration in which a diode D2 and a resistor R6 are interposed in series between the positive electrode of the DC power supply V2 and the common connection BP, but instead of the diode D2, FIG. As shown in FIG. 12, a diode-connected transistor may be used.

  In the power module 100D shown in FIG. 11, a PNP transistor Q6 and a resistor R6 are inserted in series in this order between the positive electrode of the DC power source V2 and the common connection BP, and the emitter of the PNP transistor Q6 is the base. The PNP transistor Q6 functions as a diode. A connection node between the PNP transistor Q6 and the resistor R6 is connected to the base of the PNP transistor Q5.

  In the power module 200B shown in FIG. 12, a PNP transistor Q6 and a resistor R6 are inserted in series in this order between the positive electrode of the DC power supply V2 and the common connection BP, and the emitter of the PNP transistor Q6. Are connected to the base, and the PNP transistor Q6 functions as a diode. A connection node between the PNP transistor Q6 and the resistor R6 is connected to the base of the PNP transistor Q4.

  By adopting such a configuration, a potential D obtained by dropping the potential A by the built-in voltage can be created, as in the case of using a diode. In addition, by configuring the PNP transistor Q6 with the same transistor as the PNP transistor Q5 and PNP transistor Q4 (preferably a transistor in the same manufacturing lot), individual differences due to temperature characteristics and process variations between the transistors are reduced. Thus, the voltage drop at the PNP transistor Q6 can be made the same as the voltage drop at the PNP transistor Q5 and the PNP transistor Q4, and more accurate current detection is possible.

<Intelligent power module>
In each of power modules 100 and 200 described in the first and second embodiments, drive control circuits 10 and 20 are configured by a configuration excluding IGBT 1, freewheeling diode 2, a power supply that provides main power supply potential VCC, and DC power supply V1. However, the whole or a part of the drive control circuits 10 and 20 may be built in the control IC.

  Such a control IC, IGBT 1 and free wheel diode 2 integrated in one package is called an intelligent power module (IPM).

  By making the drive control circuits 10 and 20 into an IC, the circuit scale can be reduced and the entire power modules 100 and 200 can be downsized.

  Moreover, since the power module is configured by the IGBT 1, the free wheel diode 2, and the drive control circuit 10 or 20 by making the entire drive control circuits 10 and 20 into an IC, the number of parts is reduced, and individual differences of parts are also increased. The percentage of defects is reduced.

  In addition, by reducing the number of parts, assembly errors are reduced, the probability of failure during assembly is reduced, and the failure rate is reduced.

  Further, if the number of parts is reduced, parts management and assembly are facilitated, and the manufacturing cost can be reduced.

  In addition, since the number of parts is reduced, the accuracy of current detection can be improved if the individual difference of parts is reduced.

  Here, as an example in which a part of the drive control circuits 10 and 20 is made into an IC, when the NPN transistor Q1 and the PNP transistor Q2 constituting the driver DR are made into an IC, in addition to the NPN transistor Q1 and the PNP transistor Q2, the resistor R1 And R2 may be integrated into an IC.

  Further, when the NPN transistor Q1 and the PNP transistor Q2 and the DC power supply V1 are integrated, it is conceivable that the resistors R1 and R2 are also integrated in addition to the NPN transistor Q1 and the PNP transistor Q2 and the DC power supply V1. The DC power supply V1 is built in the IC as a regulator.

  In addition, the configuration of the drive control circuits 10 and 20 other than the current detection resistor SR may be formed as an IC. Since the resistance value of the current detection resistor SR has to be set strictly in order to perform highly accurate detection and cannot be changed if it is made into an IC, it is desirable to have a separated configuration.

  This also applies to the resistors R1 and R2 that set the switching speed of the IGBT 1. In some cases, the resistors R1 and R2 are separated in preparation for changing the switching speed for each product.

<Use of semiconductor with wide band gap>
In the power modules 100 and 200 described in the first and second embodiments, the materials of the IGBT 1 and the free wheel diode 2 are not mentioned, but the IGBT 1 and the free wheel diode 2 are formed on a silicon (Si) substrate. The IGBT 1 may be a silicon semiconductor device, and the free wheel diode 2 may be a silicon carbide semiconductor device formed on a silicon carbide (SiC) substrate or a gallium nitride (GaN) -based material. A gallium nitride semiconductor device formed over a substrate to be formed may be used.

  SiC and GaN are wide band gap semiconductors, and semiconductor devices composed of wide band gap semiconductors have high voltage resistance and high allowable current density, and thus can be made smaller than silicon semiconductor devices. By using these miniaturized semiconductor devices, it is possible to reduce the size of a power module incorporating these.

  In addition, since the heat resistance is high, it is possible to reduce the size of the heat dissipating fins of the heat sink and to cool by air cooling instead of water cooling, thereby further reducing the size of the power module.

  Further, since the size is smaller than that of the silicon semiconductor device, the drive control circuits 10 and 20 can be reduced in size with the same rating.

  Conversely, the free wheel diode 2 may be configured as a silicon semiconductor device, and a switching device (including a bipolar transistor or a MOSFET) such as an IGBT 1 may be a wide band gap semiconductor device such as a silicon carbide semiconductor device or a gallium nitride semiconductor device. . In this case, the same effect as described above can be obtained.

  In addition, when the switching device is a silicon semiconductor device, since the on-voltage is low, the current shunt ratio is likely to fluctuate due to the voltage difference (ΔVge) applied to the gate of the main element and the current sensing element, but the wide band gap semiconductor device is switched. In the case of a device, the on-voltage is increased, the fluctuation of the current shunt ratio due to ΔVge is suppressed, and improvement in current detection accuracy can be expected.

  Of course, it goes without saying that both the IGBT 1 and the free wheel diode 2 may be formed of wide band gap semiconductor devices.

<Use of RC-IGBT>
In the power modules 100 and 200 described in the first and second embodiments, the configuration in which the free wheel diode 2 is connected in reverse parallel to the IGBT 1 is shown. An RC-IGBT (Reverse Conducting Insulated Gate Bipolar Transistor) integrally having a diode connected in reverse parallel may be used.

Here, the configuration of the RC-IGBT will be described with reference to FIG. Figure 13 shows a cross-sectional view of a semiconductor chip 31 which incorporates the IGBT and the diode, the semiconductor chip 31, n ^- are formed by using a substrate 32. An n-type impurity layer 33 containing n-type impurities is provided on the n ⁻ substrate 32, and a p base layer 34 containing p-type impurities is selectively provided thereon.

On the p base layer 34, an emitter region 35 containing a high-concentration n-type impurity is selectively formed. A trench 36 that penetrates from the emitter region 35 to the p base layer 34 and the n-type impurity layer 33 and reaches the n ⁻ substrate 32 is formed. A gate insulating film 37 is formed on the inner wall of the trench 36, and a polysilicon gate electrode 38 is formed on the inner side thereof.

An interlayer insulating film 39 is provided on the emitter region 35. An emitter electrode 40 is provided so as to contact a part of the emitter region 35 and the p base layer 34. An n ⁺ cathode layer 41 and a p ⁺ collector layer 42 are provided on the back surface of the n ⁻ substrate 32, and a collector electrode 43 is provided on the back surface of these layers. In this structure, a diode is formed in a region where the n ⁺ cathode layer 41 exists, and an IGBT is formed in a region where the p ⁺ collector layer 42 exists. In this way, the IGBT and the diode connected in reverse parallel to the IGBT are formed in the same chip, and the RC-IGBT is configured.

  The diode of the semiconductor chip 31 shown in FIG. 13 is turned on when the voltage between the p base layer 34 and the n-type impurity layer 33 exceeds the built-in potential of the pn junction. When the gate of the IGBT is turned on, the n-type impurity layer 33 and the emitter region 35 become conductive and have the same potential. However, since the emitter region 35 has a common contact with the p base layer 34, a voltage is not easily applied to the pn junction formed by the p base layer 34 and the n-type impurity layer 33 by turning on the gate. Become. For this reason, hole injection at the pn junction hardly occurs and the forward voltage drop (Vf) increases.

  As described above, by using the RC-IGBT in which the IGBT and the diode are formed in the same chip, the number of parts is further reduced as compared with the case where the individual IGBT and the diode are used, and the assemblability of the power module is improved.

  Note that the RC-IGBT may be configured as a silicon semiconductor device, but may be configured as a silicon carbide semiconductor device or a gallium nitride semiconductor device.

  C1 drive circuit, C2 overcurrent determination circuit, C3, C4 current detection circuit, SR current detection resistor, 10,20 drive control circuit.

Claims (2)

A power switching semiconductor device including a main element through which a main current flows and a current sense element configured to allow a part of the main current to flow; and a sense current is output from an output terminal of the current sense element;
A first transistor having a first main electrode connected to the output terminal of the current sense element; a first end connected to the second main electrode of the first transistor; and the other end connected to a common connection. A current detection circuit, wherein the control electrode of the first transistor is connected to the potential of the output terminal of the main element ;
A potential difference based on the common connection portion generated by the current detection resistor is detected as a current detection voltage, compared with a predetermined threshold voltage, and an overcurrent flows in the power switching semiconductor device due to the magnitude relationship between the two. An overcurrent determination circuit for determining whether or not
A drive circuit for generating a control signal applied to a control electrode of the power switching semiconductor device,
The first transistor is a bipolar transistor or MOSFET;
The first main electrode is an emitter electrode when the first transistor is the bipolar transistor, and a source electrode when the first transistor is a MOSFET.
The power module, wherein the second main electrode is a collector electrode when the first transistor is the bipolar transistor, and a drain electrode when the first transistor is a MOSFET. The first transistor includes:
It corresponds to a transistor through which a mirror current of a current mirror circuit that receives the sense current from the current sense element flows,
The current mirror circuit is:
A first transistor having a first main electrode connected to the output terminal of the current sensing element and a second main electrode connected to a potential of an output terminal of the main element ;
2. The power module according to claim 1, wherein the control electrodes of the first and second transistors are commonly connected to the potential of the output terminal of the main element .

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