JP5415211B2 - Control device for voltage source inverter - Google Patents

Description

The present invention relates to a current control device for a voltage source inverter.

FIG. 6 shows an example of the control configuration of the speed control device for the induction motor. In FIG. 6, 1 is a DC power supply for supplying driving power to the motor, 2 is a smoothing capacitor, 3 is an inverter for converting DC power to three-phase AC power, 4 is an induction motor, 5U, 5V, and 5W are for detecting motor current. Hole CT, 6 is a rotation sensor for detecting the position of the rotor of the electric motor, 7 is a rotation detection circuit for obtaining and outputting an electrical angle signal θr and speed ωr corresponding to the position of the rotor from the output signal of the rotation sensor. , 8 is a subtracter that takes the difference between the speed reference ωr * and the speed detection ωr, and 9 is a speed deviation output from the subtracter 8 to adjust the torque command Trq * so that the speed ωr follows the speed reference ωr *. A speed control circuit 10 for calculating and outputting a torque component current reference Iq *, a flux component current reference Id *, and a slip angle θS * from the magnetic flux reference Φ * and the torque reference Trq *, and 11 a rotation Child An adder that adds the signal θr and the slip angle θS * to output the magnetic flux position signal θ0, 12 adjusts the output signal of the Hall CT (5U, 5V, 5W) to the scaling in the control circuit, and detects the current detection signals Iu, IV, The current detection circuit 13 outputs as IW, and 13 converts the current detection signals Iu, IV, and IW into the magnetic flux current detection Id and the torque current detection Iq on the dq axis coordinate synchronized with the magnetic flux of the motor using the magnetic flux position signal θ0. Coordinate conversion circuit 14d, 14q is a subtractor that takes the difference between the current reference Id *, Iq * and current detection Id, Iq for each of the d-axis and q-axis,
15d and 15q are steady-state deviation correction control circuits that amplify the current deviation output from the subtractors 14d and 14q, 16 is a limiter that limits the correction signal Iqc * output from the q-axis side correction control circuit 15q, and 17d is a correction control circuit. An adder that adds the correction signal Idc * output by 15d and the current reference Id * to output the current reference PWM current reference Id **, and 17q adds the limiter 16 output and Iq * to obtain the current tracking type. An adder that outputs the PWM current reference Iq **, 18 is a subtractor that takes the difference between the q-axis side correction controller 15q output Iqc * and the weakening control start level IqcLIM *, and 19 amplifies the output of the subtractor 18 A magnetic flux weakening control circuit 20 is a limiter that sets the lower limit of the output of the magnetic flux weakening control circuit to zero.

A subtractor 21 subtracts the magnetic flux weakening control circuit output from the strong magnetic flux command value Φ ** to output a magnetic flux command Φ * corresponding to the state of the motor.

Current reference Id ** and Iq ** for current follow-up type PWM are converted into three-phase current references Iu *, IV * and IW * of the stator stationary coordinates by the coordinate conversion circuit 22, and are subtracted by the subtractors 23U, 23V and 23W. Differences from each of the three-phase current detections Iu, IV, and IW are taken and applied to the current tracking type PWM control circuit 24.

The current follow-up type PWM control circuit 24 generates a PWM signal in which the current detection Iu, Iv, IW follows the current reference Iu *, Iv *, IW *, and this PWM signal controls on / off of the constituent switching elements of the inverter 3. To do.

In this method, a carrier wave is not used, and a PWM signal in which the current follows the command value is directly generated, so that the current response is extremely fast. Since the PWM waveform is automatically and continuously switched according to the operating frequency, there is no need for intentional PWM control switching. There is a feature such as a transition to one-pulse operation continuously without falling out of control in the high-speed region.

  However, there is a steady error in principle in the current tracking type PWM. The current tracking type PWM compares the current reference on the stator stationary coordinates with the current detection, and generates a PWM signal based only on the magnitude relationship, so the proportional gain is infinite. If operated as it is, the switching frequency becomes too high. For this reason, a dead zone due to hysteresis and a delay time due to a timer are provided to limit the switching frequency. A steady-state error occurs in this dead zone and delay time.

The lower the switching frequency, the larger the steady-state error and the greater the influence on the control performance of the motor.

The current tracking type PWM has a great merit that a high-speed response can be obtained regardless of the switching frequency. Applying current tracking PWM to large industrial drives that use high-current switching elements, train main machine drives, etc. can speed up the current response without increasing the switching frequency, which can dramatically improve performance. . It is conceivable to actively lower the switching frequency to improve both performance and efficiency. However, in order to apply the current tracking type PWM to these applications, it is necessary to realize current control without a steady deviation. The steady deviation correction control circuits 15d and 15q are for this purpose.

  If the current detections Id and Iq are smaller than the current references Id * and Iq *, the correction control circuits 15d and 15q increase the outputs Idc * and Iqc *. As a result, the current follow-up type PWM control circuit correction current reference Id **, Iq ** increases, so that the motor current Id, Iq increases due to the current follow-up type PWM, and the current reference Id *, Iq * The difference becomes smaller. If the correction control circuits 15d and 15q have an integral element, even if the deviation output from the subtractors 14d and 14q is very small, it is integrated to correct the correction current references Id ** and Iq **. Therefore, the steady deviation can be made zero for both the d-axis and the q-axis.

  Since the correction signal Iqc * output from the correction control circuit 15q is very small in the middle / low speed region of the motor, the output signal of the weakening control circuit 19 becomes negative and is limited to the lower limit value 0 by the limiter 20. Therefore, the magnetic flux command Φ * given to the vector calculation circuit 10 is equal to the strong magnetic flux command Φ **.

When the rotation speed of the motor increases, the induced voltage increases and no current can be supplied to the motor. For this reason, the output signal of the steady deviation correction circuit 15q on the q-axis side increases rapidly. When the 15q output signal exceeds the weakening start level Iq * CLIM, the output of the subtractor 18 turns positive, and the output of the weakening control circuit 19 begins to increase. As a result, the magnetic flux command Φ * input to the vector arithmetic circuit 10 becomes a value obtained by subtracting the output of the control circuit 19 from the strong magnetic flux command Φ ** and starts to weaken the magnetic flux. The Id * output from the vector arithmetic circuit 10 becomes smaller and the motor magnetic flux Φ becomes smaller. As a result, the increase in the induced voltage is limited, and the value of the output signal of the correction control circuit 15q is controlled to the limit value Iq * CLIM.

  The q-axis side correction control circuit 15q output Iqc * is added to the current reference Iq * via the limiter 16, but without this limiter 16, the flux weakening may not be in time and control may be impossible. For example, when a large-amplitude power running torque command is given, Iqc * becomes a large signal transiently, so that the magnetic flux weakening operates, the d-axis side current reference is narrowed, and becomes smaller than the current detection Id. As a result, Idc * swings to the negative side. On the other hand, Iqc * swings to the positive side. Because of the step change, the current deviation on the q-axis side is large and the increasing speed of Iqc * is fast. In one pulse region, q-axis current cannot flow unless the magnetic flux is weakened. For this reason, Iqc * increases in the positive direction. On the other hand, when the magnetic flux current reference Idc * is reduced, Idc * also increases to the negative side.

If the current tracking type PWM control tries to control the current on both the d-axis side and the q-axis side without steady deviation based on this corrected current reference, both the d-axis side and the q-axis side are in accordance with the standard. The current can no longer flow.

The limiter 16 controls the q-axis side while allowing a deviation transiently. On the other hand, since the d-axis side is not limited, the magnetic flux is continuously weakened while Iqc * is weakened and exceeds the control start level. Eventually, the magnetic flux is sufficiently weakened so that Iqc * is controlled to be constant at the weakening control start level. Since the steady deviation correction control circuit has an integral element, the fact that Iqc * is controlled to a constant value means that Iq is controlled to be equal to Iq *. On the other hand, since the current on the d-axis side is controlled so that there is no deviation, both the currents Id and Iq can be controlled according to the reference value even in one pulse region.

This makes it possible to control the current with high accuracy while using the current tracking type PWM having excellent current response, and to realize high-performance vector control excellent in both accuracy and response.

Also, in the conventional combination method of PI control type dq-axis current control and triangular wave comparison PWM control, the q-axis current control output (voltage reference) is weakened early so as not to exceed the q-axis voltage that can actually be output. In this method, the magnetic flux is weakened only after detecting that the voltage for current control is insufficient. This makes it possible to control the flux weakening with a complete one-pulse voltage, and increase the output capacity and improve the efficiency in the weakened region.

Japanese Patent No. 3267524 JP 2003-235270 A

Ogasawara et al., “Current-Controlled PWM Inverter Enabling Harmonic Suppression and High-Speed Current Response”, IEEJ Transactions B, Vol. 106, No. 2

When the switching frequency is limited by hysteresis in the current tracking control type PWM, the switching frequency varies depending on various factors such as the inductance of the motor, the inverter DC voltage, the motor rotation speed, and the current magnitude. When the switching frequency of the large-capacity switching element fluctuates, the cooling capacity is required and the cooling fin is increased in size. When the inverter loss is reduced at a switching frequency lower than that of the prior art by utilizing the high-speed tracking capability of the present system, fluctuations in the switching frequency are not preferable.

Non-patent document 1 introduces a method of controlling the switching frequency by current tracking control type PWM using hysteresis.

The switching frequency is detected, matched with the command value, the deviation is integrated, and the reciprocal is used as hysteresis. If the switching frequency is higher than the command value, the deviation becomes negative and the integrator output becomes small, so the hysteresis which is the reciprocal thereof becomes large. If the hysteresis region is widened, the switching frequency is decreased, and if the hysteresis region is narrowed, the switching frequency is increased. Thus, the switching frequency can be controlled.

  If a switching frequency of about several kilohertz is allowed and full range PWM control is performed, there is no problem with this method. However, an inverter using a large switching element that only allows a switching frequency of several hundred Hz causes a problem. At low frequencies such as several hundred Hz, sampling for switching frequency detection is not sufficiently fast, so that control cannot be performed quickly. For this reason, there is a risk that the response is slow, the control is not in time when the current command step changes, sudden acceleration / deceleration, etc., and the switching frequency rises transiently and causes thermal damage to the element.

Further, in the vicinity of one pulse region, the switching frequency is controlled to decrease so as to increase the switching frequency until it becomes equal to the command value by reducing the hysteresis, but this is an unnecessary operation. When the switching frequency is lowered in a region where the operating frequency is low, the current ripple becomes large, so it is necessary to control the switching frequency so as not to fall. However, in the vicinity of one pulse region, the switching frequency decreases even with a certain hysteresis, and the problem that the current ripple increases does not occur. Rather, it is preferable to reduce the switching loss closer to one pulse.

The first object of the present invention is to keep the switching frequency substantially constant by controlling the hysteresis rapidly at the time of step change of the current command and sudden acceleration / deceleration even if the response of the switching frequency control is slow in the PWM region of the motor.

The second object of the present invention is to control the hysteresis so that the switching frequency naturally falls to the 1-pulse mode when approaching the 1-pulse region.

The problem is that the currents Iu, IV, IW are changed to the PWM current references Iu *, IV *, IW * based on the comparison result between the currents Iu, IV, IW and the PWM current references Iu *, IV *, IW *. In a voltage source inverter control device using a current tracking type PWM control circuit that directly generates a tracking PWM signal, a switching frequency detection circuit for detecting a switching frequency of the inverter, and the switching frequency detection follows a switching frequency command value Feedback control circuit that controls the magnitude of the hysteresis, and a hysteresis table that shows the relationship between the hysteresis for each current reference and the rotation speed of the motor , and the magnitude of the hysteresis so that the desired switching frequency is obtained. A feedforward control circuit for controlling Based the Dobakku control circuit to the feed-forward control circuit, the calculated hysteresis governing the switching frequency of the current-following PWM control circuit may be achieved by providing the current-following PWM circuit.

The above problem is that the currents Iu, IV, IW are changed to the PWM current references Iu *, IV *, IW * based on the comparison result between the currents Iu, IV, IW and the PWM current references Iu *, IV *, IW *. In a voltage source inverter control device using a current tracking type PWM control circuit that directly generates a tracking PWM signal, a switching frequency detection circuit for detecting a switching frequency of the inverter, and the switching frequency detection follows a switching frequency command value The integral type control circuit for controlling the magnitude of the hysteresis, the limit circuit for limiting the integral type control circuit output so as not to become negative, the rotational speed of the motor and the current reference, and the rotational speed and the current reference Each current reference so that sometimes the sum with the limit circuit output is a hysteresis to obtain the desired switching frequency. A look-up table circuit that outputs a hysteresis somewhat smaller than the optimum hysteresis using a hysteresis table indicating the relationship between the hysteresis of the motor and the rotation speed of the motor , and adding the limit circuit output and the look-up table output This value can be achieved by applying a value to the current tracking PWM circuit as a hysteresis governing the switching frequency of the current tracking PWM control circuit.

According to the present invention, even when the response of the switching frequency control is slow in the PWM region of the motor, the hysteresis can be rapidly controlled at the time of step change of the current command, sudden acceleration / deceleration, and the switching frequency can be kept substantially constant.

Further, the hysteresis can be controlled so that the switching frequency is naturally lowered to the one-pulse mode when approaching one pulse region.

Control configuration of an embodiment of an inverter control device of the present invention Detailed configuration of switching frequency control circuit as main part of the present invention Illustration of hysteresis curve used in current tracking PWM control circuit Explanatory diagram of the operation of the present invention Another embodiment of switching frequency control circuit Control configuration of motor control device using conventional current tracking type PWM control circuit

(First embodiment)
(Constitution)
A motor control device according to a first embodiment of the present invention will be described in detail with reference to the drawings. FIG. 1 is a control configuration diagram of an electric motor control apparatus according to a first embodiment based on the present invention. FIG. 2 is a detailed configuration diagram of a switching frequency control circuit which is a main part of the present invention. In addition, about the thing of the same structure as FIG. 6, the same code | symbol is attached | subjected and description is abbreviate | omitted.

One feature of the motor control device according to the first embodiment of the present invention is that 25 switching frequency control circuits are newly provided in the configuration of the conventional motor control device shown in FIG.

The switching frequency control circuit 25 receives the gate signals Gu, GV, GW, Gx, Gy, Gz, the speed ωr, and the torque current command Iq *, and the hysteresis Hys that determines the width of the allowable error region of the current tracking type PWM control circuit. Is output. In a battery drive device or a train where the inverter DC voltage Vdc varies over a wide range, it is necessary to adjust the hysteresis Hys also by the DC voltage Vdc.

FIG. 2 shows details of the switching frequency control circuit 25. In FIG. 2, 30 is a switching frequency detection circuit, 31 is a subtractor, 32 is an integrator, 33 is an inverse circuit, 34 is a limiter, 35 is a look-up table circuit, and 36 is an adder. The feedback control circuit includes a switching frequency detection circuit 30, a subtractor 31, an integrator 32, an inverse circuit 33, and a limiter 34, and the feedforward control circuit includes a lookup table circuit 35.

The switching frequency detection circuit 30 measures the number of rises and falls during a predetermined period for each of the Gu, GV, GW, Gx, Gy, and Gz gate signals, and outputs the sum of them as the switching frequency detection FsW. Although it is possible to detect from a three-phase PWM signal instead of the 6-arm gate signal, it is preferable to detect from a gate signal. Since the gate signal is a signal after a dead time for preventing the short circuit between the upper and lower arms, the count of pulses that are included in the PWM signal but are eliminated in the dead time is eliminated in the gate signal. In addition, since the pulse with a very narrow width included in the PWM signal damages the element, the minimum width of the pulse is secured in the gate signal. The narrow pulse may be expanded to the minimum width, but may be removed due to the dead time, and the number of pulses differs between the PWM signal and the gate signal. For this reason, it is more accurate to count using the gate signal actually applied to the element. In the case of a gate signal, there is an advantage that an accurate detection circuit can be easily made because there is no counting error if counting is performed with clock pulses having a period equal to or less than the minimum width.

The subtractor 31 calculates the difference (FsW * −FsW) between the switching frequency detection FsW and the switching frequency command value FsW *, and the deviation (FsW * −FsW) is integrated by the integrator 32. In order to lower the switching frequency, the hysteresis must be widened, so that the integral value is reciprocal by the reciprocal circuit 33. A limiter 34 limits the lower limit value of the reciprocal circuit output signal to 0 to obtain a switching frequency control signal HysI by feedback control.

The look-up table circuit 35 has a built-in hysteresis table that indicates the relationship between the hysteresis having a predetermined switching frequency for each of several current values of Iq * and the operating frequency ωr. A switching frequency control signal HysFF corresponding to Iq * is output.

The two switching frequency control signals HysI and HysFF are added by an adder 36, and are given as hysteresis Hys to the current tracking type PWM control circuit 24 of FIG.

Here, hysteresis will be described. In the case of an induction machine load, the hysteresis for keeping the switching frequency at a certain value is a curve as schematically shown in FIG. When the inverter DC voltage is a rated value, the largest hysteresis is required in the vicinity of 1/2 to 2/3 of the motor rated speed. Since the induced voltage of the motor is small near the rotation speed of zero, the current decay due to the induced voltage is delayed. On the other hand, when the rated rotational speed is approached, the difference between the inverter output voltage and the induced voltage becomes small, so that the increase in current becomes moderate. Therefore, the curve is as shown in FIG.

  In the control device for the voltage type inverter of the present embodiment, the hysteresis output from the lookup table is as shown in FIG. That is, the first lower limit value is provided for the hysteresis in the low rotation range below the first predetermined operation frequency, and the second lower limit value is provided for the hysteresis in the high rotation range above the second predetermined operation frequency.

Further, with respect to the finally required value of FIG. 3A, the value output from the lookup table circuit 35 is, for example, about 0.7 to 0.9 with respect to the value of FIG. The value multiplied. This may be a value obtained by multiplying the hysteresis value recorded in the table by about 0.7 to 0.9 from the beginning, and a hysteresis value of 1 times is written in the table and is output to the output stage of the lookup table circuit. A multiplier for multiplying by 0.7 to 0.9 may be provided.

(Function)
The only difference between the voltage type inverter control device of the first embodiment shown in FIG. 1 and the conventional inverter control device of FIG. 2 is the presence or absence of switching frequency control. This will be described below.

4A shows the relationship between the hysteresis output from the switching frequency control circuit 25 and the operating frequency, and FIG. 4B shows the switching frequency at that time. (A) also shows a breakdown of the hysteresis HysFF output from the feedforward side of the switching frequency control circuit 25 and the hysteresis HysI output from the feedback side.

  In section (A), feedforward HysFF and feedback HysI are added to the section marked with T2 and supplied to the current follow-up type PWM control circuit. Therefore, the switching frequency in the corresponding section in (B) is the switching frequency command FsW for feedback control. Hysteresis is controlled so that the value of * is obtained.

  HysFF larger than the value for setting the switching frequency to FsW * is output from the feedforward side in the low-speed T1 period. The feedback control side outputs a negative signal in an attempt to control the switching frequency equal to the command value, but the limiter 34 limits the negative signal to zero. For this reason, HysFF is output as it is as the final hysteresis. As a result, as shown in (B), the switching frequency at the boundary point between T1 and T2 is equal to FsW *, but the switching frequency decreases as the rotational speed decreases.

In the section T3 having a frequency higher than T2, HysFF larger than the value for setting the switching frequency to FsW * is output from the feedforward side in the same manner as the section T1. HysI on the feedback side is zero. For this reason, since a constant HysFF is supplied to the current follow-up type PWM control circuit, as shown in (B), as the rotational speed increases, the switching frequency gradually decreases to one pulse.
All the intervals of T4 are one pulse.

  The constant HysFF value used in the T3 and T4 intervals determines the width of the T3 interval. If it is made smaller, the section of T3 becomes narrower, the slope of the switching frequency decrease from FsW * of the switching frequency shown in (B) to one pulse becomes steeper, and if it is made larger, the slope becomes gentler.

  In the T1 section, the switching frequency is not suddenly set to FsW * at the time of start-up, but gradually approaches FsW * with an inclination.

  When switching from the T1 section to the T2 section or from the T3 section to the T2 section, the feedback control starts to operate from a state in which the deviation of the switching frequency is zero. The amount can be reduced. Further, as can be seen from FIGS. 3A and 4A, both T1 and T3 are sections in which hysteresis is smaller than T2, and are areas where the hysteresis changes greatly depending on the power running / regenerative state even at the same rotation speed. . For this reason, there is a possibility that the switching frequency becomes too high because the control is not in time with the feedback control alone. In the present invention, since these sections are only feedforward, a stable current waveform can be obtained.

(effect)
The switching frequency is directly linked to the physique and weight of the element cooling device. For this reason, it is extremely important in a drive device using a large semiconductor element having a large switching loss.

  Unlike the conventional combination method of dq axis current control and triangular wave comparison PWM control, current tracking type PWM control has a feature that the speed of current response is not affected by the switching frequency. Therefore, if current tracking type PWM control is used, the cooling frequency can be reduced by reducing the switching frequency and suppressing the heat generation of the element without lowering the control performance.

  However, in order to design a specific cooling device, it is not only possible to reduce the switching frequency as compared with the prior art, but also a clear guarantee that the switching frequency does not rise above the predetermined frequency is necessary. Management becomes necessary.

The present invention has been made to achieve this.

If hysteresis that affects the switching frequency of current tracking PWM control is performed only by integral control, overshooting of the switching frequency is unavoidable when the current reference step changes, the frequency suddenly changes, or the like. In particular, when the switching frequency must be controlled to be low, the control sampling becomes long, so that the response is slow and the period during which the switching frequency is equal to or higher than a predetermined value is also long. If only integral control is used, the switching frequency is controlled to the command value up to the vicinity of one pulse region. Since the operation frequency and the switching frequency are equal in one pulse region, the switching frequency control becomes unstable near the switching point.

On the other hand, the hysteresis necessary for maintaining the switching frequency at a predetermined value fluctuates due to various factors. Therefore, feedforward control that can cope with all factors is extremely difficult.

In the present invention, feedforward control and feedback control are combined, an approximate value is given by feedforward, and is precisely matched by feedback control, thereby satisfying both the transient response of the switching frequency and the control accuracy.

In the present invention, the feedback control functions only to increase the hysteresis output by the feedforward control, and the hysteresis width output by the feedforward control is kept constant around zero rotation and near one pulse. I made it. As a result, as shown in FIG. 4A, in that region, the feedback control output naturally approaches zero and shifts to an operation of only feedforward, so that the switching frequency is stably controlled to a frequency equal to or less than the command value.

(Other embodiments)
The voltage of the overhead line voltage that supplies power to the train, the battery that serves as the power source for the electric vehicle, etc., vary greatly. In the case of inverters using these as power sources, the switching frequency control circuit of FIG. 2 cannot keep the switching frequency constant. The switching frequency control circuit of FIG. 5 corresponds to this.

In FIG. 5, the inverter DC voltage detection Vdc is newly added. The constant VdcTBL appearing in the elements 37 and 40 is a DC voltage assumed when the hysteresis table is created.

The motor rotation speed ωr is multiplied by a constant VdcTBL by a multiplier 37, divided by a DC voltage detection Vdc by a divider 38, and then input to a lookup table circuit 35. Thus, instead of ωr, ω'r = (VdcTBL / Vdc) · ωr
The table is drawn using. When the DC voltage is 1500V, the motor that shifts from PWM to one pulse at 60Hz shifts to one pulse at 30Hz when the DC voltage becomes 750V. For this reason, (1500/750) · ωr is used when a table created with a DC voltage of 1500 V is read out at 750 V. As a result, hysteresis at 750 V-30 Hz and 1500 V-60 Hz can be read.

However, since the current change rate is (voltage / inductance), if the voltage is halved, the hysteresis may be halved. Therefore, the look-up table circuit output HysFF is multiplied by the DC voltage detection Vdc by the multiplier 39 and multiplied by (1 / VdcTBL) by the multiplier 40. So instead of HysFF
HysFF ′ = (Vdc / VdcTBL) · HysFF
Is used. HysFF '= (750/1500) · HysFF at 750V, half the hysteresis at 1500V. This HysFF ′ is added to the hysteresis HysI by feedback control by the adder 36 and is given to the current follow-up type PWM 24 as the hysteresis Hys.

  As a result, the switching frequency can be kept almost constant even in applications where the DC voltage fluctuation is large, and the current tracking type PWM is applied to the inverter using a large-capacity switching element, and the switching frequency is reduced. While reducing loss, it is possible to speed up the current response and improve performance.

  Further, the hysteresis curve stored in the look-up table circuit is not used as it is, but is used in the sum with the feedback control, so that accuracy is not required. This makes it easy to create a table.

  Note that the present invention can also be applied to an inverter using a high-speed switching element having a small and medium capacity. According to the present invention, the control gain of the switching frequency control can be lowered so as not to cause interference with other control systems such as current control and speed control. Even small and medium-capacity inverters can be designed with a smaller cooling frequency if the switching frequency is small, and a smaller and cheaper inverter can be provided.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying constituent elements without departing from the scope of the invention in the implementation stage. Further, various inventions can be formed by appropriately combining a plurality of constituent elements disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, you may combine the component covering different embodiment suitably.

1: DC power supply, 2: capacitor, 3: inverter, 4: motor
5U, 5V, 5W: Hall CT, 6: Rotation sensor, 7: Rotation detection circuit 8: Subtractor, 9: Speed control circuit, 10: Magnetic flux weakening function generator,
10: vector operation circuit, 11: adder, 12: current detector,
13: coordinate conversion circuit, 14d, 14q: subtractor,
15d, 15q: steady deviation correction control circuit, 16: limiter,
17d, 17q: Adder, 18: Subtractor, 19: Magnetic flux weakening control circuit, 20: Limiter 21: Subtractor, 22: Coordinate conversion circuit, 23U, 23V, 23W: Subtractor 24: Current tracking type PWM control circuit,
25: Switching frequency control circuit,
30: switching frequency detection circuit, 31: subtractor, 32: integrator, 33: reciprocal circuit,
34: Limiter, 35: Look-up table circuit, 36: Adder 37: Multiplier, 38: Divider, 39: Multiplier, 40: Multiplier

Claims (5)

The currents Iu, IV, IW follow the PWM current standards Iu *, IV *, IW * based on the comparison result between the currents Iu, IV, IW and the PWM current standards Iu *, IV *, IW *. In a voltage source inverter control device using a current follow-up type PWM control circuit that directly generates a PWM signal,
A switching frequency detection circuit for detecting the switching frequency of the inverter;
A feedback control circuit for controlling the magnitude of hysteresis so that the switching frequency detection follows a switching frequency command value;
It has a hysteresis table showing the relationship between the hysteresis for each current reference and the rotation speed of the motor, and has a feedforward control circuit that controls the magnitude of the hysteresis so as to obtain a desired switching frequency,
A control device for a voltage source inverter, wherein a hysteresis governing a switching frequency of the current tracking type PWM control circuit is calculated based on the feedback control circuit and the feedforward control circuit, and applied to the current tracking type PWM circuit. . The currents Iu, IV, IW follow the PWM current standards Iu *, IV *, IW * based on the comparison result between the currents Iu, IV, IW and the PWM current standards Iu *, IV *, IW *. In a voltage source inverter control device using a current follow-up type PWM control circuit that directly generates a PWM signal,
A switching frequency detection circuit for detecting the switching frequency of the inverter, an integral type control circuit for controlling the magnitude of hysteresis so that the switching frequency detection follows a switching frequency command value,
Limit circuit for limiting the output of the integral type control circuit so as not to be negative, and hysteresis for obtaining the desired switching frequency when the motor speed and current reference are input and the sum of the speed and the limit circuit output when the current is used Using a hysteresis table that shows the relationship between the hysteresis for each current reference and the rotation speed of the motor, and a look-up table circuit that outputs a hysteresis somewhat smaller than the optimum hysteresis,
A control apparatus for a voltage source inverter, wherein a value obtained by adding the limit circuit output and the look-up table output is given to the current tracking type PWM circuit as a hysteresis governing the switching frequency of the current tracking type PWM control circuit. . 3. A control apparatus for a voltage source inverter according to claim 1, wherein most of the PWM region of the motor is somewhat smaller than the optimum hysteresis so that the sum of the PWM region and the limit circuit output is a hysteresis for obtaining a desired switching frequency. A control apparatus for a voltage source inverter, which uses a look-up table circuit that outputs a hysteresis but outputs a hysteresis having a constant value regardless of a current value at a rotational speed higher than the vicinity of one pulse region. 3. A control apparatus for a voltage source inverter according to claim 1, wherein most of the PWM region of the motor is somewhat smaller than the optimum hysteresis so that the sum of the PWM region and the limit circuit output is a hysteresis for obtaining a desired switching frequency. Hysteresis is output, but in the low speed region near zero rotation, the current value,
A voltage source inverter control device using a look-up table circuit that outputs a hysteresis having a constant value regardless of the number of revolutions. In addition to the rotational speed of the motor and the current reference, an inverter DC voltage detection signal is input to the voltage source inverter control device according to claim 1 or 2, and the DC voltage used when creating the hysteresis curve is converted into a DC voltage detection signal. A value obtained by multiplying the rotation number by the rotation speed is input to the lookup table circuit as a correction rotation speed, and a value obtained by dividing the DC voltage detection signal by the DC voltage used when creating the hysteresis curve is output from the lookup table circuit. A control apparatus for a voltage source inverter, wherein the hysteresis is multiplied by a correction hysteresis, and the correction hysteresis is given to the current follow-up type PWM control circuit.

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