JP2021163548A - High frequency power supply system - Google Patents

Abstract

To propose a technology for suppressing periodic fluctuation of reflection wave power caused from output of a bias power supply.SOLUTION: A high frequency power supply system 1 is provided which supplies high frequency power to a load 2 connected thereto. The high frequency power supply system 1 comprises: a bias power supply for outputting bias power at a first frequency; a source power supply for outputting source power at a second frequency higher than the first frequency; and an impedance matching unit 104 for matching source power supply side impedance with load side impedance. The source power supply performs increase/decrease control on an output frequency to make a reactance value of the load side impedance be closer to a predetermined target value.SELECTED DRAWING: Figure 1

Description

The present disclosure relates to a high frequency power supply system.

In the field of semiconductor manufacturing, high-density mounting is required as electronic devices become smaller and more sophisticated, and the connection of elements to the mounting board is miniaturized, and more reliable mounting is required. ing.

One of the methods for ensuring the reliability of mounting is a surface modification method using plasma. For example, when the substrate to be treated is subjected to plasma treatment, contamination by organic substances adhering to the surface of the substrate can be removed, the bonding strength of wire bonding can be improved, the wettability is improved, and the adhesion between the substrate and the sealing resin can be improved. Can be improved. In order to perform such plasma processing, it is necessary to connect a power supply device to the plasma processing device.

As a high-frequency power supply system that supplies power to the plasma processing device, for example, a configuration including two high-frequency power supply devices can be used. That is, the configuration includes a source power supply (operating at the second frequency: for example, 40.68 MHz) and a bias power supply (operating at the first frequency lower than the first frequency: for example, 400 kHz), but the outputs of the two high-frequency power supply devices. If there is a large difference in frequency, the first frequency of the bias power supply causes a periodic reflected wave on the source power supply side, and the high frequency power output from the source power supply is transmitted to the plasma processing device. The problem of difficulty arises.

In order to solve this problem, for example, Patent Document 1 uses a source power supply (operating at a second frequency: for example, 40.68 MHz) and a bias power supply (operating at a first frequency lower than the second frequency: for example, 400 kHz). In the high frequency power supply system provided, the source power supply detects a trigger event which is an impedance fluctuation of the plasma, and discloses that a frequency offset is added to the RF output signal of the source power supply in response to the trigger event. By doing so, it becomes possible to suppress the generation of the reflected wave caused by the first frequency.

Special Table 2018-536295

In Patent Document 1, the reflected wave power is controlled to approach 0 (zero) by adding a frequency offset to the RF output signal of the source power source. That is, it is necessary to control the resistance value of the impedance on the load side to approach the target value (usually 50Ω) and the reactance value to approach 0 (zero), so it is appropriate to add a frequency offset. It is difficult to understand. Therefore, control will be complicated. As a solution, the appropriate value of the frequency offset is stored in the lookup table, but it is necessary to obtain the appropriate value in advance.
In view of such a situation, the present disclosure proposes a technique that makes it possible to reduce the reflected wave caused by the first frequency by simply controlling the output frequency.

In order to solve the above problems, the present disclosure is a high-frequency power supply system that provides high-frequency power to a connected load, and is a bias power supply that outputs bias power at the first frequency, and a bias power supply that outputs bias power at the first frequency, and more than the first frequency. It is equipped with a source power supply that outputs source power at a high second frequency and an impedance matching box that matches the impedance on the source power supply side with the impedance on the load side. The source power supply is controlled by increasing or decreasing the output frequency. Provided is a high frequency power supply system that brings the reactance value of the load side impedance close to a predetermined target value.

Further features relating to this disclosure will become apparent from the description herein and the accompanying drawings. In addition, the aspects of the present disclosure are achieved and realized by the combination of elements and various elements, as well as the following detailed description and the aspects of the appended claims.
It should be understood that the description herein is merely a typical example and does not limit the scope or application of the claims in any way.

According to the technique of the present disclosure, in a power supply system that provides a dual frequency power supply, it is possible to suppress fluctuations in the reflected wave power generated due to the traveling wave power supplied from the bias power supply to the load. This control is easy because it only controls the reactance value of the load-side impedance so as to approach a predetermined target value.

It is a figure which shows the state which connected the plasma load 2 to the power supply system (also called a high frequency power supply system) 1 by 1st Embodiment. It is a figure which shows the internal structure example of a phase difference detector 105. It is a figure for demonstrating the state of zero phase difference. It is a figure which shows the state which connected the plasma load 2 to the power supply system (high frequency power supply system) 100 by 2nd Embodiment. It is a figure which shows the internal structure example of the frequency controller 106 by the modification. It is a figure which shows the internal structure example of the impedance matching box 104 by the modification.

Hereinafter, embodiments of the present disclosure will be described with reference to the accompanying drawings. In the attached drawings, functionally the same elements may be displayed with the same number. The accompanying drawings show specific embodiments and implementation examples in accordance with the principles of the present disclosure, but these are for the purpose of understanding the present disclosure and in order to interpret the present disclosure in a limited manner. Not used.

In this embodiment, the description is given in sufficient detail for those skilled in the art to implement the present disclosure, but other implementations and embodiments are also possible and do not deviate from the scope and spirit of the technical idea of the present disclosure. It is necessary to understand that it is possible to change the structure / structure and replace various elements. Therefore, the following description should not be construed as limited to this.

Further, the embodiment of the present disclosure may be implemented by software running on a general-purpose computer, or may be implemented by dedicated hardware or a combination of software and hardware.

(1) First Embodiment In the first embodiment, the phase difference between the traveling wave voltage Vf and the reflected wave voltage Vr is detected, the frequency shift amount in the voltage control oscillator is calculated based on the phase difference, and the voltage is controlled. We propose a power supply system that controls the frequency of the high-frequency signal output from the oscillator.

<Configuration example of power supply system 1>
FIG. 1 is a diagram showing a state in which a plasma load 2 is connected to a power supply system (also referred to as a high frequency power supply system) 1 according to the present embodiment. The power supply system 1 includes a voltage control oscillator (VCO: first oscillator) 101, a first amplifier 102, a directional coupler 103, an impedance matcher (first matcher) 104, a phase difference detector 105, and a frequency controller 106. , Source power supply (supplying high frequency output) including reflection level measuring instrument 107, frequency detector 108, motor control processing unit 109, and motor group 110, bias oscillator (second oscillator) 201, second amplifier 202, and A bias power supply (supplying a low frequency output) including a second matching unit 203, and a traveling wave power (source power) output from the source power supply 10 and a traveling wave power (bias power) output from the bias power supply 20 are provided. It is a system that superimposes (example) and supplies the plasma load 2.

In the source power supply, the voltage controlled oscillator 101 generates a high frequency signal at the oscillation frequency (initial value f1) and outputs it to the first amplifier 102. This oscillation frequency can be changed within a predetermined range.
In the present embodiment, the frequency of the high frequency signal output from the voltage controlled oscillator 101 has 40.68 MHz as the center frequency. The center frequency of the high-frequency signal output from the voltage-controlled oscillator 101 is not limited to 40.68 MHz, but is, for example, a frequency in the industrial RF band (Radio Frequency) such as 13.56 MHz and 27.12 MHz. May be good.
Further, in the present embodiment, the frequency of the signal output from the bias oscillator is a frequency of 400 kHz, but the frequency is not limited to this frequency.

The first amplifier 102 amplifies the high frequency signal supplied from the voltage controlled oscillator 101 (high frequency output) and outputs it to the directional coupler 103.
The high frequency output output from the first amplifier 102 is the so-called traveling wave power Pf, and its voltage component is the traveling wave voltage Vf. Further, the voltage component of the reflected wave power Pr reflected from the plasma load 2 side is the reflected wave voltage Vr. Further, the output frequency, which is the frequency of the traveling wave power Pf (progressing wave voltage Vf) output from the first amplifier 102, is determined by the oscillation frequency of the voltage controlled oscillator 101. Therefore, the output frequency can also be changed within a predetermined range. Then, apart from the impedance matching in the impedance matching box (first matching box) 104, which will be described later, the impedance on the plasma load 2 side is adjusted by changing the output frequency of the first amplifier 102, and the reflected wave power Pr is reduced. It becomes possible to do.

A narrow band pass filter (not shown) may be provided between the voltage controlled oscillator 101 and the first amplifier 102. When following the frequency of the voltage controlled oscillator 101, a frequency shift from the center frequency of the narrow band pass filter occurs, and the effect of the filter is reduced. Therefore, in order to eliminate the inconvenience in this case, the narrow band pass filter changes the center frequency (within the band of the first amplifier 102) according to the tracking frequency, and signals from the voltage controlled oscillator 101 (within the band of the first amplifier 102). It may be configured so that a high frequency signal) can be acquired. As the narrow band pass filter, for example, a voltage-controlled tunable filter or a filter bank can be used (inserted).

The directional coupler 103 detects the traveling wave voltage Vf, which is a voltage component of the traveling wave power Pf in the first amplifier 102 (the output end of the first amplifier 102 may be the input end of the impedance matching device 104). , The reflected wave voltage Vr which is a voltage component of the reflected wave power Pr at the output end of the first amplifier 102 is detected, and the detected traveling wave voltage Vf and the reflected wave voltage Vr are used as the phase difference detector 105 and the reflection level measuring device 107. Output to. Further, the detected traveling wave voltage Vf is also supplied to the frequency detector 108.
The traveling wave voltage Vf and the reflected wave voltage Vr output from the directional coupler 103 are actually a detection signal of the traveling wave voltage Vf and a detection signal of the reflected wave voltage Vr, but the description is simplified. Therefore, in the following, it will be described simply as a traveling wave voltage Vf and a reflected wave voltage Vr.

The phase difference detector 105 detects the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr supplied from the directional coupler 103, and outputs the information of the phase difference φ to the frequency controller 106.
FIG. 2 is a diagram showing an example of the internal configuration of the phase difference detector 105. As shown in FIG. 2, the phase difference detector 105 includes a first limiter circuit 1051 that limits the amplitude of the traveling wave voltage Vf within a certain range, and a second limiter circuit 1051 that limits the amplitude of the reflected wave voltage Vr within a certain range. From the limiter circuit 1052, the multiplier 1053 that multiplies the traveling wave voltage Vf and the reflected wave voltage Vr whose amplitudes are limited by the first and second limiter circuits 1051 and 1052, and the multiplication result by the multiplier 1053, the cutoff frequency is higher than the cutoff frequency. It includes a low-pass filter LPF1054 that cuts off high frequency components, and a calculation unit 1055 that calculates the phase difference between the traveling wave voltage Vf and the reflected wave voltage Vr.
Here, the amplitude of the traveling wave voltage Vf is A, the angular frequency is ω, and the time is t, the amplitude of the reflected wave voltage Vr is B, the angular frequency is ω, and the time is t, and the traveling wave voltage Vf and the reflected wave voltage Vr. Assuming that the phase difference with and is φ, it can be expressed as Vf = A · cos (ω · t) and Vr = B · cos (ω · t + φ). At this time, since the output s of the low-pass filter LPF1054 is s = cosφ, the phase difference φ is calculated by φ = arccos (s) in the calculation unit 1055.
Further, assuming that the phase angle of the traveling wave voltage Vf is φf and the phase angle of the reflected wave voltage Vr is φr, the calculation can be performed with the phase difference φ = φr−φf.

The frequency controller 106 calculates a frequency shift (error signal) corresponding to the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr detected by the phase difference detector 105 (for example, the detected phase difference φ). The frequency shift value can be calculated based on the differential and ± (difference) with respect to the target value of the phase difference φ (for example, zero phase difference)), and the frequency is shifted to correct the frequency shift. A voltage (command signal) is applied to the voltage controlled oscillator 101. As a result, the voltage controlled oscillator 101 outputs a high frequency signal at a frequency f1 + Δf shifted by an amount corresponding to the difference from the previous oscillation frequency (initial value f1) with respect to the target value of the phase difference φ. Note that Δf indicates a frequency correction amount. Further, the frequency controller 106 recognizes the frequency of the oscillation signal of the voltage controlled oscillator 101. Therefore, when a frequency value is required for the impedance matching operation in the impedance matching device 104, the frequency of the oscillation signal recognized by the frequency controller 106 can be used.

FIG. 3 is a diagram for explaining zero phase difference. FIG. 3 is a diagram in which a resistance axis (horizontal axis) and a reactance axis (vertical axis) are added to a simple representation of the so-called Smith chart, but the scales of the reactance circle, the reactance circle, etc. are omitted. doing. Further, the region surrounded on the right side of the reactance axis shows an example in which the load side impedance (impedance seen from the first amplifier 102 (strictly speaking, the directional coupler 103) on the plasma load 2 side) can be taken. ..
A description for making the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr zero will be described with reference to FIG.
In the example shown in FIG. 3, it is assumed that the phase difference φ indicated by the arrow is the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr at the present time. The phase difference φ has a correspondence relationship with the phase angle when the reflection coefficient Γ is expressed by the absolute value and the phase angle.
In the example shown in FIG. 3, since the phase difference φ indicated by the arrow is above the resistance axis, the load-side impedance shows inductiveness. In such a case, since the output frequency of the source power supply is in a region higher than the resonance frequency, it is possible to approach the resonance frequency by lowering the output frequency of the source power supply. That is, by correcting the output frequency of the source power supply by the frequency corresponding to the phase difference φ, the output frequency of the source power supply can be brought close to the resonance frequency so that the phase difference φ = 0. In the example shown in FIG. 3, since the phase difference φ is positive (plus), the frequency correction amount Δf is a negative (minus) value.

Contrary to the above, when the phase difference φ is lower than the resistance axis, the load side impedance shows capacitance. Therefore, in order to bring the output frequency of the source power supply closer to the resonance frequency, the frequency correction amount Δf should be positive. Therefore, the output frequency of the source power supply may be increased. As a result, the output frequency of the source power supply can be brought close to the resonance frequency so that the phase difference φ = 0.
That is, the frequency controller 106 executes the increase / decrease control of the oscillation frequency based on the sign (±) of the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr.
The frequency correction amount Δf can be, for example, a frequency corresponding to the phase difference φ. In this case, the corresponding frequency may be obtained in advance by experiment or simulation. Of course, the frequency correction amount Δf can be set to a frequency smaller than the frequency corresponding to the phase difference φ.

Here, a problem in supplying traveling wave power having different output frequencies from the source power supply and the bias power supply shown in FIG. 1 will be supplemented.
When plasma is generated in the plasma load 2, the thickness of the plasma sheath is affected by the traveling wave power supplied from the bias power supply to the plasma load 2 and the thickness of the plasma sheath corresponds to the output frequency of the bias power supply (for example, the bias power supply). It fluctuates periodically at the same frequency as the output frequency of 400 kHz).

Since the plasma sheath is considered as an electrical insulator, the plasma sheath can be considered as a capacitor (capacitance). That is, when the plasma load 2 is shown by an equivalent circuit, it can be considered that the equivalent circuit includes a capacitor (capacitance). Therefore, the fact that the thickness of the plasma sheath fluctuates periodically means that the capacitance component of the plasma load 2 fluctuates periodically, so that the reactance value fluctuates periodically. That is, since the impedance of the plasma load 2 fluctuates, the reflected wave power Pr is generated. Since the magnitude of this reflected wave power Pr fluctuates in a very fast cycle (400 kHz in the above example), it cannot be reduced by the matching operation in the impedance matching box 104 using a motor, and therefore a countermeasure has been desired.

The fact that the reactance value fluctuates periodically and the reflected wave power Pr is generated means that the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr fluctuates periodically, so that the output frequency of the source power supply If the reactance frequency can be brought close to the resonance frequency so that the phase difference φ = 0, the periodic fluctuation of the reflected wave power Pr due to the traveling wave power supplied from the bias power supply to the plasma load 2 can be reduced. ..

Therefore, the phase difference φ may be constantly monitored, and the output frequency of the source power supply may be controlled so that the phase difference φ approaches 0 (zero).
For that purpose, it is necessary to monitor the phase difference φ and control the output frequency of the source power supply in a cycle faster than the above-mentioned fluctuation cycle. For example, when the output frequency of the bias power supply is 400 kHz, the output frequency of the source power supply may be controlled at a cycle corresponding to a frequency (4 MHz or more) of at least several times, preferably 10 times or more.
As described above, since the output frequency of the source power supply can be changed at high speed, it is possible to reduce the fluctuation of the reflected wave power caused by the traveling wave power supplied from the bias power supply to the plasma load 2.

Further, as can be seen from the above explanation, since the reflected wave power Pr fluctuates periodically due to the periodic fluctuation of the phase difference φ, the source so that the phase difference φ is not necessarily 0 (zero). It is not necessary to control the output frequency of the power supply, and it may be controlled so as to have another target value. However, as described above, by controlling the output frequency of the source power supply so that the phase difference φ becomes 0 (zero), the output frequency of the source power supply can be brought close to the resonance frequency, which is preferable. However, if the load-side impedance is capacitive, the problem that the current waveform is distorted with the switching operation tends to occur. Therefore, the reference phase difference φ may be made larger than 0 (zero). Further, if the phase difference φ is within the permissible value set for the target value (for example, zero), the output frequency of the source power supply may not be changed.

The reflection level measuring instrument 107 calculates the reflection level amount from the reflection coefficient defined by the ratio (= Vr / Vf) of the traveling wave voltage Vf and the reflected wave voltage Vr, and outputs it to the motor control processing unit 109 as the control amount. .. After calculating the magnitude of the traveling wave power Pf based on the traveling wave voltage Vf and calculating the magnitude of the reflected wave power Pr based on the reflected wave voltage Vr, the reflected wave power Pr with respect to the traveling wave voltage Pf The amount of reflection level defined by the ratio (= Pr / Pf) may be calculated.
Further, the frequency detector 108 detects the frequency value (= output frequency of the source power supply) of the traveling wave voltage Vf supplied to the plasma load 2 and outputs it to the motor control processing unit 109.

The motor control processing unit 109 is composed of, for example, a processor such as a CPU or an MPU, and has an impedance value based on the amount of reflection level from the reflection level measuring instrument 107 and the traveling wave power frequency value from the frequency detector 108. The rotation control amount of each motor (for example, motors M1 to M3) included in the motor group 110 is determined so as to have an appropriate value (for example, 50Ω), and each motor is controlled. Specifically, the motor control processing unit 109 compares the reflection level amount with a predetermined threshold value and executes impedance matching (by rotating the motors M1 to M3, the variable capacitor value and the variable inductance value are changed. (Is it necessary?) Judge. Then, when it is determined that impedance matching is to be performed, the amount of rotation of the motors M1 to M3 is adjusted in order to adjust the variable capacitor value and the variable inductance value so that the load side impedance and the output impedance of the source power supply approach the matching state. To control. Further details of determining the amount of rotation of the motors M1 to M3 will be described later.
The frequency value detected by the frequency detector 108 is not always necessary, and may be used when the frequency value is required for the impedance matching operation of the impedance matching device 104. Therefore, when the frequency value is unnecessary, the frequency detector 108 is not always necessary.

The impedance matching box 104 includes, for example, a plurality of (two in FIG. 1) variable capacitors and a variable inductor, and the impedance matching box 104 is input at the frequency of the direct wave (traveling wave power Pf) from the directional coupler 103. Calculate the output impedance. Then, in the impedance matching box 104, the values of the variable capacitor and the variable inductor are adjusted so as to operate at the resonance frequency (optimal frequency: initial value f1) of the plasma load 2 from this calculation point. The adjustment of the values of the variable capacitor and the variable inductor is performed by the motor group (M1 to M3) 110 controlled by the motor control processing unit 109 as described above.

<Rotation amount control of motors M1 to M3>
As described above, the motor control processing unit 109 has an impedance value of an appropriate value (for example, 50Ω) based on the amount of reflection level from the reflection level measuring device 107 and the traveling wave power frequency value from the frequency detector 108. The rotation control amount of each motor (for example, motors M1 to M3) included in the motor group 110 is determined so as to be, and each motor is controlled. For the motor control, for example, the following method can be considered.

(I) Control method 1
A plurality of motors (motors M1 to M3) are moved (rotated) by a predetermined amount at the same time, and the reflected wave power level when each of the motors is moved is monitored by the reflection level measuring device 107, and the reflected wave power level is set. The amount of rotation of the motor when it reaches the minimum value is adopted. After that, each motor is moved in order by a predetermined amount (for example, the motor M1, the motor M2, and the motor M3 are moved in this order), and the rotation amount that minimizes the reflected wave power level is adopted in each of them.

(Ii) Control method 2
A plurality of motors (motors M1 to M3) are moved (rotated) in a predetermined order, for example, by a predetermined predetermined amount, and at each motor position (rotation amount) at which the reflected wave power level is minimized. stop. After that, the reflected wave power level is measured by the reflection level measuring device 107, and it is determined whether or not the value is equal to or less than a predetermined threshold value (threshold value for checking whether the reflected wave power level is within the allowable range). If it is determined that it is larger than the threshold value, the motor rotation amount control is executed from the beginning (the operation of moving the motors M1 to M3 in order to check the reflected wave power level is executed again).

(Iii) Control method 3
Stop at the position (rotation amount) of each combination of motors (motors M1 to M3) that minimizes the reflected wave power level in any combination (for example, M1 and M2, M2 and M3, M3 and M1). .. Then, it is determined whether or not the total reflected wave power is equal to or less than a predetermined threshold value, and if it is determined to be larger than the threshold value, the motor rotation amount control is executed from the beginning (reflection is performed by any combination of the motors M1 to M3). The operation to check the wave power level is executed again).

The motor control processing unit 109 can perform impedance matching at the resonance frequency or in the vicinity of the resonance frequency by determining the rotation control amount of the motor by any of the methods (i) to (iii). As described above, impedance matching is usually performed in a state where the phase difference φ = 0 or the phase difference φ is equal to or less than the allowable value set for the target value. The resistance value of the load side impedance is controlled so as to approach the target value (for example, 50Ω). Therefore, the amount of control is smaller than bringing both the resistance value and reactance value of the load-side impedance closer to the target value. Therefore, it is possible to shorten the time for impedance matching.

<Technical effect of the first embodiment>
According to the first embodiment, the phase difference detector 105 is provided, the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr is detected, and the phase difference φ is brought close to the target value (preferably zero). The output frequency of the source power supply is controlled to increase or decrease. By doing so, the reactance value of the load-side impedance can be brought close to a predetermined target value, so that the reflected wave power Pr generated due to the traveling wave power supplied from the bias power supply to the plasma load 2 can be obtained. Fluctuations can be suppressed. This control is easy because it only controls the phase difference φ so as to approach the target value.
Further, in the control of the variable capacitor value and the variable inductance value of the impedance matching device 104 by the motor control processing unit 109, it is possible to shorten the time for impedance matching.

(2) Second Embodiment In the first embodiment, the power supply system 1 including the phase difference detector 105 for detecting the phase difference using the traveling wave voltage Vf and the reflected wave voltage Vr has been described. In the second embodiment, a configuration example including a current detector (current waveform signal detector) and a voltage detector (voltage waveform signal detector) for detecting the phase difference between voltage and current will be described.

<Configuration example of power supply system>
FIG. 4 is a diagram showing a state in which the plasma load 2 is connected to the power supply system (high frequency power supply system) 100 according to the second embodiment. The power supply system 100 replaces the directional coupler 103, the phase difference detector 105, the frequency controller 106, the reflection level measuring instrument 107, and the frequency detector 108 of the power supply system 1 according to the first embodiment. 1 The current waveform signal detector 1001 for detecting the current waveform signal I1 of the current at the output end of the amplifier 102 (the output end of the first amplifier 102 may be the input end of the impedance matching device 104), and the first amplifier 102. The voltage (voltage waveform signal V1) and current (current waveform signal I1) and current (current waveform signal I1) are used by the voltage waveform signal detector 1002 that detects the voltage waveform signal V1 of the voltage at the output end, and the detected voltage waveform signal V1 and current waveform signal I1. ), And a control unit 1003 that detects the phase difference θ and calculates the reflection coefficient.

Further, the control unit 1003 calculates (for example, is detected) a frequency shift (error signal) corresponding to the phase difference θ based on the information of the phase difference θ between the voltage (voltage waveform signal V1) and the current (current waveform signal I1). The frequency shift value can be calculated based on the differentiation of the phase difference θ and ± (difference) with respect to the target value of the phase difference θ (for example, zero phase difference)), and the frequency is shifted to obtain the above frequency shift. A voltage (command signal) that corrects the above is applied to the voltage controlled oscillator 101. As a result, the voltage controlled oscillator 101 outputs a high frequency signal at a frequency f1 + Δf shifted by an amount corresponding to the difference from the previous oscillation frequency (initial value f1) with respect to the target value of the phase difference θ. Note that Δf indicates a frequency correction amount.
Further, the control unit 1003 recognizes the frequency of the oscillation signal of the voltage controlled oscillator 101. Therefore, when a frequency value is required for the impedance matching operation in the impedance matching device 104, the frequency of the oscillation signal recognized by the control unit 1003 can be used.
Further, the control unit 1003 outputs the reflection coefficient information to the motor control processing unit 109. Specifically, the control unit 1003 calculates the impedance Z1 on the impedance matching device 104 side from the current waveform signal I1 and the voltage waveform signal V1 by Z1 = V1 / I1. Here, it can be defined as Z1 = R1 + jX1 (j is an imaginary number). Then, the control unit 1003 obtains the phase difference θ between the voltage waveform signal V1 and the current waveform signal I1 by θ = tan ^-1 (X1 / R1). Further, the control unit 1003 obtains the reflection coefficient Γ1 by Γ1 = (Z1-Z0) / (Z1 + Z0). Here, Z0 indicates the characteristic impedance between the impedance matching box 104 and the plasma load 2, and is usually 50Ω.
Further, assuming that the phase angle of the voltage waveform signal V1 is θv and the phase angle of the current waveform signal I1 is θi, the calculation can be performed with the phase difference θ = θi−θv.

The voltage controlled oscillator 101 controls the output frequency of the source power supply based on the command signal output from the control unit 1003. Specifically, if the phase difference θ between the voltage waveform signal V1 and the current waveform signal I1 is 0 (zero), the voltage and current are in phase, so the reactance value of the load-side impedance is 0 (zero). .. Therefore, if the output frequency of the source power supply is controlled so that the phase difference θ between the voltage waveform signal V1 and the current waveform signal I1 becomes 0 (zero), the same effect as described in the first embodiment can be obtained. Can be done.
Of course, as in the first embodiment, the phase difference θ between the voltage waveform signal V1 and the current waveform signal I1 does not necessarily have to be controlled to 0 (zero), and the output frequency of the source power supply does not necessarily have to be controlled. It may be controlled so as to reach the target value.
However, by controlling the output frequency of the source power supply so that the phase difference θ becomes 0 (zero), the output frequency of the source power supply can be brought close to the resonance frequency, which is preferable.
However, if the load-side impedance is capacitive, the problem that the current waveform is distorted with the switching operation tends to occur. Therefore, the reference phase difference θ may be made larger than 0 (zero) so that the current lags behind the voltage.
Further, if the phase difference θ is within the permissible value set for the target value (for example, zero), the output frequency of the source power supply may not be changed.
Further, when the value of the reflection coefficient from the control unit 1003 is equal to or higher than a predetermined threshold value, the motor control processing unit 109 can be used as a motor group 110 for changing the values of the variable capacitor and the variable inductor included in the impedance matching box 104. The number of revolutions of the included motors M1 to M3 is determined. Since the operation of the motor control processing unit 109 is the same as that described above, the description thereof will be omitted.

<Technical effect of the second embodiment>
According to the second embodiment, the voltage and the current are based on the current and the current at the output end of the first amplifier 102 (the output end of the first amplifier 102 may be the input end of the impedance matching box 104). The phase difference θ is detected. Then, the phase difference θ is controlled so as to be a target value (preferably zero (reactance zero)).
By doing so, it is possible to stably supply high frequency power to the plasma load 2 by suppressing fluctuations in the reflected wave power Pr generated due to the traveling wave power supplied from the bias power supply to the plasma load 2. become able to. This control is easy because it only controls the phase difference θ so as to approach the target value.
Further, in the control of the variable capacitor value and the variable inductance value of the impedance matching device 104 by the motor control processing unit 109, it is possible to shorten the time for impedance matching.

(3) Modification Example (i) Internal Configuration Example of Voltage Controlled Oscillator 101 FIG. 5 is a diagram showing an internal configuration example of the frequency controller 106 according to the modification.

The frequency controller 106 according to the modified example includes a filter 1011, a multiplier 1012, a filter 1013, a frequency divider 1014, a multiplier 1016, a fixed oscillator 1015, a gain adjuster 1017, a filter 1018, and voltage control. It includes an oscillator 101.
In this modification, as in the first embodiment, the case where the oscillation frequency of the voltage controlled oscillator 101 is changed according to the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr will be described as an example. ..

The voltage-controlled oscillator 101 can change the frequency of the oscillating signal to be output according to the output of the gain regulator 1017 output via the low-pass filter 1018, and has the same configuration as the voltage-controlled oscillator 101 of FIG. be. In the present embodiment, the frequency of the high frequency signal output from the voltage controlled oscillator 101 has 40.68 MHz as the center frequency.

The filter 1011 cuts off or sufficiently attenuates unnecessary frequency components included in the information of the phase difference φ output from the phase difference detector 105 and outputs the filter 1011. As the filter 1011 for example, a low-pass filter or a band-pass filter can be used.

As described above, the phase difference φ fluctuates due to the traveling wave power supplied from the bias power supply to the plasma load 2. Therefore, the information of the phase difference φ is a signal whose magnitude fluctuates depending on the output frequency of the bias power supply (400 kHz in this embodiment), but since it contains unnecessary frequency components other than the output frequency of the bias power supply, The filter 1011 is used to extract the signal of the output frequency component of the bias power supply.

The multiplier 1012 is a signal of a difference frequency component among the sum frequency component and the difference frequency component generated when the oscillation signal output from the voltage controlled oscillator 101 and the output signal of the filter 1011 (information on the phase difference φ) are mixed. Is output. Therefore, when the frequency of the high frequency signal output from the voltage controlled oscillator 101 is 40.68 MHz and the frequency of the output signal of the filter 1011 is 400 kHz, it is 40.28 MHz (40.68 MHz-0.4 MHz = 40.28 MHz). Outputs a signal with a frequency component.

The filter 1013 is a filter designed to pass the above-mentioned difference frequency component, and for example, a low-pass filter or a band-pass filter can be used.
The frequency divider 1014 is for dividing the signal output from the filter 1013, and is divided into, for example, 1/4. The frequency division ratio is not limited to 1/4, and may be another frequency division ratio. Further, it may have a function of changing the division ratio according to a set signal from the outside.

The fixed oscillator 1015 is an oscillating unit that generates an oscillating signal having a fixed frequency, and outputs an oscillating signal having a frequency of, for example, 10.17 MHz.
The frequency of the oscillation signal of the fixed oscillator 1015 is determined by the frequency of the oscillation signal of the voltage controlled oscillator 101 and the frequency division ratio of the frequency divider 1014. In the above example, the central frequency of the high frequency signal output from the voltage controlled oscillator 101 is 40.68 MHz, and the frequency division ratio of the frequency divider 1014 is 1/4, so 10.17 MHz (40.68 MHz / 4 = 10. It is set to 17 MHz). In consideration of changing the frequency division ratio of the frequency divider 1014, it may have a function of changing the frequency of the oscillation signal of the fixed oscillator 1015.

The multiplier 1016 outputs a signal of a difference frequency component among the sum frequency component and the difference frequency component generated when the output signal of the fixed oscillator 1015 and the output signal of the frequency divider 1014 are mixed and mixed.

The gain adjuster 1017 controls the increase / decrease of the oscillation frequency with respect to the voltage controlled oscillator 101 based on the output signal of the multiplier 1016. At this time, if necessary, the voltage controlled oscillator 101 is controlled to increase or decrease the oscillation frequency via the filter 1018 (for example, a low-pass filter). Hereinafter, this increase / decrease control will be described.

The amplitude of the output signal of the frequency divider 1014 changes according to the phase difference φ. On the other hand, since the amplitude of the output signal of the fixed oscillator 1015 is constant, the amplitude of the output signal of the multiplier 1016 changes according to the phase difference φ. Further, since the amplitude (target amplitude) of the output signal of the multiplier 1016 when the phase difference φ is the target value (for example, zero) is known in advance, the amplitude of the output signal of the multiplier 1016 becomes the target amplitude. The voltage (command signal) that corrects the frequency shift by shifting the frequency of the output signal of the voltage control oscillator 101 by the amount corresponding to the difference between the target amplitude and the amplitude of the output signal of the multiplier 1016 is applied to the voltage control oscillator 101. By applying to the voltage control oscillator 101, the increase / decrease control of the oscillation frequency is performed.

In this way, even if the reflected wave power Pr generated periodically fluctuates due to the output of the bias power supply, the frequency of the output signal of the voltage controlled oscillator 101 is canceled so as to cancel the reflected wave power Pr. Is shifted, so that the phase difference φ changes at the target value or near the target value. As a result, the periodic fluctuation of the reflected wave power Pr generated due to the output of the bias power supply is suppressed.
In the above, an example of performing control based on the amplitude of the output signal of the multiplier 1016 is shown, but the amplitude represents the magnitude of the amplitude such as the average value, the effective value, the peak value, and the peak peak value. All you need is.

Further, if the multiplier 1016 is a digital processing method, the amplitude difference information between the output signal of the fixed oscillator 1015 and the output signal of the frequency divider 1014, and the output signal of the fixed oscillator 1015 and the output signal of the frequency divider 1014 The phase difference information of may be output.
In this case, since the amplitude difference information when the phase difference φ is the target value is known in advance, the gain adjuster 1017 is based on the amplitude difference information output from the multiplier 1016, and the frequency shift is the same as described above. By applying a voltage (command signal) that corrects the above to the voltage controlled oscillator 101, it is possible to control the increase / decrease of the oscillation frequency with respect to the voltage controlled oscillator 101.

Further, since the above phase difference information fluctuates due to the frequency (400 kHz) of the output signal of the filter 1011, if the phase difference information is set to a constant value, it is supplied from the bias power supply to the plasma load 2. Fluctuations in the reflected wave power Pr generated due to the traveling wave power can be suppressed. Therefore, since the phase difference information when the phase difference φ is the target value is known in advance, a voltage that corrects the frequency deviation (similarly to the above) based on the phase difference information output from the multiplier 1016. By applying the command signal) to the voltage controlled oscillator 101, it is possible to control the increase / decrease of the oscillation frequency with respect to the voltage controlled oscillator 101.

As described above, since the phase difference φ is controlled so as to reach (approach) the target value (preferably zero), the phase difference φ changes at the target value or near the target value. That is, according to the present embodiment, when the phase difference φ changes to the target value or the vicinity of the target value, the output signal of the filter 1011 becomes a substantially constant value. Of course, when the phase difference φ is controlled to be zero (reactance zero) (close to it), the output signal of the filter 1011 becomes substantially zero.

Further, in the description of the modified example of FIG. 5 above, the oscillation frequency of the voltage controlled oscillator 101 is changed according to the phase difference φ between the traveling wave voltage Vf and the reflected wave voltage Vr, as in the first embodiment. However, as in the first embodiment, it is also applied to the case where the oscillation frequency of the voltage controlled oscillator 101 is changed according to the phase difference θ between the voltage waveform signal V1 and the current waveform signal I1. Can be done.

(Ii) Example of Internal Configuration of Impedance Matching Box 104 FIG. 6 is a diagram showing an example of internal configuration of the impedance matching box 104 according to a modified example.
The impedance matching box 104 according to the modified example includes a fixed capacitance first capacitor 401, a fixed capacitance second capacitor 402, a fixed reactance coil 403, and electronic variable capacitance capacitors 404 to 406. The impedance matching box 104 according to the modified example is not provided with a variable capacitor or a variable reactor for adjusting the capacitance value by the motors M1 to M3 as in the first embodiment, but is an electronic type that changes the capacitance value by voltage application control. A variable capacitance capacitor is used. When an electronic variable capacitor is used, it is desirable that the frequency detected by the frequency detector 108 is close to the optimum frequency of the plasma load 2.
As described above, the impedance matching box 104 can be controlled at high speed by using the electronic variable capacitor.

Various inventions can be formed by an appropriate combination of the plurality of components disclosed in the present embodiment. For example, some components may be deleted from all the components shown in the present embodiment. In addition, components across different embodiments may be combined as appropriate.

Further, in the above-described embodiment, the control lines and information lines are shown as necessary for explanation, and the product does not necessarily show all the control lines and information lines. All configurations may be interconnected.

For example, in the above embodiment, the impedance matching box 104 exemplifies a type in which the variable capacitor value is adjusted by a motor, but a type in which the variable capacitor value is adjusted by a semiconductor switch (for example, a PIN diode) may be used. ..
Further, in the above embodiment, the voltage controlled oscillator 101 is exemplified as the oscillator, but the present invention is not limited to this. For example, a direct digital synthesizer (DDS) may be used as the oscillator.
Further, in the above embodiment, three examples of the variable element (variable capacitor, variable inductor) provided in the impedance matching box 104 are shown, but the present invention is not limited to this. For example, there may be two variable capacitors.

1,100 power supply system (high frequency power supply system)
2 Plasma load 101 Voltage controlled oscillator 102 First amplifier 103 Directional coupler 104 Impedance matching box (first matching box)
105 Phase difference detector 106 Frequency controller 107 Reflection level measuring instrument 108 Frequency detector 109 Motor control processing unit 110 Motor group 201 Bias oscillator 202 Second amplifier 203 Second matching device 1001 Current waveform signal detector 1002 Voltage waveform signal detection Instrument 1003 Control unit

Claims (6)

A high frequency power supply system that provides high frequency power to the connected load.
A bias power supply that outputs bias power at the first frequency,
A source power source that outputs source power at a second frequency higher than the first frequency, and
An impedance matching device that matches the impedance on the source power supply side with the impedance on the load side is provided.
The source power supply is a high-frequency power supply system that brings the reactance value of the load-side impedance closer to a predetermined target value by controlling the output frequency in an increase or decrease. In claim 1,
The source power supply detects the phase difference between the detection signal corresponding to the traveling wave power supplied to the load and the detection signal corresponding to the reflected wave power returned from the load, and sets the phase difference to a predetermined target value. A high-frequency power supply system that brings the reactance value of the load-side impedance closer to a predetermined target value by increasing or decreasing the output frequency so that it approaches. In claim 1,
The source power supply detects the phase difference between the detection signal corresponding to the current of the traveling wave power supplied to the load and the detection signal corresponding to the voltage, and outputs the frequency so that the phase difference approaches a predetermined target value. A high-frequency power supply system that brings the reactance value of the load-side impedance closer to a predetermined target value by controlling the increase or decrease of. In claim 2 or 3,
The detected phase difference signal is converted into a low frequency signal using a frequency divider and a multiplier, and the output frequency is increased / decreased and controlled so that the phase difference approaches a predetermined target value based on the converted signal. High frequency power supply system. In any one of claims 1 to 4,
A high frequency power supply system in which the target value of the reactance is zero. In any one of claims 1 to 5,
A high frequency power supply system that changes the values of a plurality of variable elements included in the impedance matching box so that the level of reflected wave power returned from the load is reduced.

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