JP2001102820A - High frequency circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a high frequency circuit capable of wire bonding at the frequencies above several tens GHz. SOLUTION: A filter is formed at a line terminal by utilizing the inductance of a bonding wire. A coplanar line 20 is formed on a ceramic substrate 10, a coplanar line 20B continuously thinning a central conductor 21a and a coplanar line 20C widening the central conductor are formed. When the bonding wire is connected to this coplanar line, an LCL T filter is provided. At such a time, while using a transmission matrix, the line of a thinned part 21b and the line parameters (impedance Z and electric length θL) of a widened part 21c are designed so as to match the impedance with a line connected to that filter at input and output terminals thereof. Thus, reflection loss is reduced and the wire bonding at the frequencies above several tens GHz is enabled.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a high-frequency circuit formed on a substrate. In particular, the present invention relates to a high-frequency circuit in which a pattern having a different line impedance is formed at an end portion thereof, a filter is formed in combination with an inductive component of a bonding wire, and impedance matching is performed with a connected line. The present invention can be applied to, for example, connection between coplanar / coplanar lines having different or equal line impedances, connection between microstrip / coplanar lines, connection between microstrip / microstrip lines, and the like.

[0002]

2. Description of the Related Art Conventionally, there has been a high frequency circuit composed of a microstrip line / coplanar line. They are generally used by being connected to a high-frequency element or another line by connection means such as wire bonding. At this time, the bonding wire is an inductive component for the high-frequency signal,
The magnitude of the parasitic reactance defines the upper limit frequency of the applicable high-frequency signal. Therefore, various proposals have been made. For example, “Microwave / millimeter wave packaging technology” (IEICE General Conference, TC-1
1, 1999) have proposed wire connection shortening, wire ribbonization, flip chip bonding using through holes, and the like as mounting techniques.

Further, "DC-10 GHz GaAs mixer IC for coherent optical heterodyne receiver" (IEICE Spring Conference, C-630, 1991) is 240
It is stated that an nF microcapacitor chip capacitor is provided between the bonding wires to form a T-type filter, thereby improving characteristics.

[0004]

However, the former shortening of the wire length is physically limited by the bonding apparatus. In addition, the inductive component is reduced only by forming the wire into a ribbon and using flip-chip bonding using bumps, but the inductive component is not eliminated, and is not a fundamental solution. In addition, the latter mounting of the small-capacity chip capacitor increases the number of mounting steps and complicates the configuration. Therefore, this configuration increases the manufacturing cost. In both cases, the applied frequency is about 40 GHz, and 70 GHz.
It was not applicable to the above high frequency signals.

An object of the present invention is to change the line impedance at the end of a line by patterning, form a filter including an inductive component of a bonding wire, and achieve impedance matching with a line to be connected. Is to reduce the reflection loss. And 70GHz
A high-frequency circuit that can be applied to the above high-frequency signals is provided.

[0006]

According to a first aspect of the present invention, there is provided a high-frequency circuit comprising a line formed of a center conductor and a ground conductor formed on a substrate, and another line connected to an end of the line. In a high-frequency circuit to which a line is connected, a filter is formed at an end by changing a line impedance by partial shape deformation of at least one of a center conductor and a ground conductor, and an impedance is formed at an input end and an output end of the filter. It is characterized by being matched.

The high-frequency circuit according to claim 2 is characterized in that an inductive component and a capacitive component are generated by a change in line impedance due to shape deformation to form a filter.

Further, in the high-frequency circuit according to the third aspect, the change in line impedance is caused by a thin line portion in which a part of the center conductor is thinned and a widened portion in which the line width of the center conductor is widened. The line parameters (line impedance and electrical length) of the thin line portion and the widened portion are determined so that the line impedance is impedance-matched at the input end and the output end of the filter.

Further, the high-frequency circuit according to claim 4 is characterized in that the end is connected to another line by a predetermined connecting means, and the filter includes the predetermined connecting means. That is, the filter includes a portion where the line impedance changes, for example, a thin line portion, a widened portion, and a predetermined connection means. Here, the predetermined connection means means all connection means having an inductive component such as a bonding wire and a ribbon formed by ultrasonic bonding and via holes, through holes, and bumps formed by solder bonding.

According to the high-frequency circuit of the present invention, the line is a microstrip line having a ground conductor formed on the back surface of the substrate and a center conductor formed on the upper surface of the substrate.

According to the high frequency circuit of the present invention, the line is formed of a first ground conductor on the back surface of the substrate, a coplanar conductor on the upper surface of the substrate, and second ground conductors formed on both sides of the center conductor. It is a track.

Further, according to the high frequency circuit of the present invention, the line parameters (line impedance and electric length) of the lines constituting the filter having different line impedances.
Means that the line impedances of both connected lines are Z ₁ , Z
_2. The elements of the transmission matrix W of the formed filter are represented by W ₁₁ =
When A, W ₁₂ = B, W ₂₁ = C, and W ₂₂ = D, the impedance matching condition Z ₁ = (AZ
₂ + B) / (CZ ₂ + D) and the matching condition Z ₂ =
(DZ ₁ + B) / (CZ ₁ + A) is determined.

Further, according to the high frequency circuit of the present invention, the line parameters (line impedance and electrical length) of the lines constituting the filter having different line impedances.
Represents the line impedance of both connected lines as Z ₀ , and the elements of the transmission matrix W of the filter as W ₁₁ = A, W ₁₂ = B, W ₂₁
= C and W ₂₂ = D, the voltage transmission coefficient S ₁₂ = 2 / (A + B / Z ₀ + Z ₀ · C + D) in the filter is determined to be approximately 1.

[0014]

According to the high frequency circuit of the first aspect, a filter is formed at the end by changing the line impedance by partially deforming at least one of the center conductor and the ground conductor. The impedance is matched between the input terminal and the output terminal of the filter. As a result, the line impedance looking upstream from the input end of the filter matches the impedance looking downstream. Therefore, reflection of the high-frequency signal at the input end is reduced. In addition, the impedance seen from the output end of the filter and the impedance seen from the downstream thereof match each other.
Therefore, reflection at the output end is also reduced. Therefore, a high-frequency circuit that can connect lines having different line impedances without reflection is provided. The partial shape deformation means changing the width of the center conductor, changing the distance between the center conductor and the ground conductor, and forming a branch from the ground conductor to the center conductor. Generally, the inductive component is determined by the relative width and length of the center conductor, and the capacitance component is determined by the relative spacing between the ground conductor and the center conductor. In short, it includes everything that forms an inductive component and a capacitive component based on the line width, line length, bending, division, branch line addition, and positional relationship. Also, an open stag shorter than λ / 4 provided in parallel from the ground conductor to the center conductor also generates an inductive component and changes the line impedance.

According to the high frequency circuit of the present invention, the filter is formed by generating an inductive component and a capacitive component by a change in the line impedance due to the shape deformation.
If the inductive component becomes excessive with respect to the characteristic impedance,
It is equivalent to the inductive component inserted into the line, and if the capacitive component is excessive, it is equivalent to the capacitive component inserted between the center conductor and the ground conductor. Thus, an L-type filter composed of a series induction component and a parallel capacitance component is equivalently configured. This L-type filter can have an impedance conversion function. The magnitudes of the inductive component and the capacitive component are determined so that impedance matching is performed at the input terminal and the output terminal of the filter.

According to the high-frequency circuit of the third aspect, the change in the line impedance is caused by the thin line portion in which a part of the center conductor is thinned and the widened portion wider than the thin line portion. The thin line portion and the widened portion form a line having a different line impedance together with the ground conductor. That is, the filter is composed of a plurality of lines having different line impedances. The line parameters (line impedance Z and electrical length θ _L ) of the thin line portion and the line parameters of the widening portion can be determined from their transmission matrices and the impedance matching conditions of the line to which the filter is connected. That is, it is determined that the line impedance seen from the input terminal of the filter and the impedance seen from the downstream end match each other. Further, the impedance is determined so that the impedance viewed from the output end of the filter and the impedance viewed from the downstream thereof match each other. By setting the above parameters, the impedance can be matched at the input terminal and the output terminal of the filter, so that the reflection loss of the high-frequency signal can be minimized.

According to the high frequency circuit of the present invention, the filter formed at the end further includes a connection means. That is, the filter is constituted by predetermined connection means for connecting the filter and another line by partial deformation of at least one of the center conductor and the ground conductor. At this time, the predetermined connection means is regarded as a transmission line in terms of high frequency.

For example, it is assumed that the predetermined connecting means is a bonding wire and, for example, another line is connected via a filter composed of the above-mentioned thin line portion and the above-mentioned widened portion, which is an example of the case where the line impedance is different. At this time, the impedance of the bonding wire is an inductive component.
Therefore, equivalently, a T-type filter including a series induction component, a parallel capacitance component, and a series induction component by wire bonding is formed.

This T-type filter can be an impedance conversion filter. That is, if the line parameters (impedance Z and electrical length θ _L ) of the thin line portion and the line parameters of the widening portion are selected, the line impedance viewed from the input end of the T-type filter and the impedance viewed downstream thereof are matched. can do. Similarly, it is possible to match the impedance looking upstream from the output end (wire end on the other line side) of the T-type filter with the impedance looking downstream. That is, impedance matching can be achieved between the input end and the output end of the T-type filter. Even if the impedance seen from the right side of the wire deviates from the characteristic impedance, impedance matching can be achieved at the output end. That is, the insertion loss due to any wire bonding can be reduced. This effect is the same even if the wire is replaced with a connecting means of another conductive material. Thereby, for example, the reflection by the wire is reduced, so that the wire bonding connection is possible even for a high-frequency circuit pattern of, for example, 70 GHz or more. Therefore, with existing wire bonding technology,
This is a high-frequency circuit that enables connection between lines in the millimeter wave band.

According to the high frequency circuit of the present invention, the line is a microstrip line having a ground conductor formed on the back surface of the substrate and a center conductor formed on the upper surface of the substrate. The microstrip line has a ground conductor on the back surface of the substrate. Therefore, for example, if the line width of the center conductor is made thinner to produce a thin line portion, the inductive component thereof can be easily increased to change the line impedance.
Further, if the line width of the thinned center conductor is further widened to form the widened portion, the capacitance component can be further increased and the line impedance can be changed. Therefore, a high-frequency circuit for reducing the reflection loss due to the bonding wire can be easily formed.

According to the high frequency circuit of the present invention, the line is composed of a first ground conductor on the back surface of the substrate, a center conductor on the top surface of the substrate, and second coplanar conductors formed on both sides of the center conductor. Tracks. The line impedance of the coplanar line can be adjusted not only by the line width of the center conductor but also by the separation distance from the second ground conductor. For example, by adjusting the above-mentioned separation distance with respect to the center conductor having the same line width, the induction component and the capacitance component can be changed over a wider range. Therefore, impedance matching between the input terminal and the output terminal can be more flexibly achieved for various lines. Therefore, a more convenient high-frequency circuit is provided.

Further, according to the high frequency circuit of the present invention, the line parameters (line impedance and electrical length) of the lines constituting the filter having different line impedances.
Means that the line impedances of both connected lines are Z ₁ , Z
_2. The elements of the transmission matrix W of the formed filter are represented by W ₁₁ =
When A, W ₁₂ = B, W ₂₁ = C, and W ₂₂ = D, the impedance matching condition Z ₁ = (AZ
₂ + B) / (CZ ₂ + D) and the matching condition Z ₂ =
It is determined so that (DZ ₁ + B) / (CZ ₁ + A) holds.

The above matrix elements A, B, C, and D are
The change in impedance is composed of, for example, a thin line portion and a widened portion.
In this case, the parameters (impedance Z
_a, Electrical length θ_a) And the parameters of the widening section (impedance
Z_b, Electrical length θ_b). In addition,
For elements A, B, C, and D, a line connected to the filter
Impedance matching conditions with the path, for example, the filter input end
Impedance Z looking upstream at ₁And downstream
If the dances are equal, Z₁= (AZ_Two+ B) / (C
Z_Two+ D). Also, downstream at the output end of the filter
Seen impedance Z_TwoAnd the impedance looking upstream
If equal, Z_Two= (DZ₁+ B) / (CZ₁+
A) holds. Further, regarding the transmission matrix, AD-B
C = 1 holds.

Here, the line impedance Z₁, Z_TwoIs
Since the matrix elements A, B, C, and D are known,
The engagement is established. Therefore, the parameters of the thin line part and the widened part
Substituting A, B, C, D represented by the parameters of
The above four parameters (Z_a, Θ _a, Z_b, Θ_b) 3 in between
Is established. Therefore, the above four parameters
If one of them is determined arbitrarily, the other three parameters are unique
Is determined. For example, the electrical length θ of the widened portion_bArbitrarily decided
If set, the impedance Z of the widened part_b, And fine line
Impedance Z_aAnd electrical length θ_aIs determined. That is,
The shapes of the widened portion and the thin line portion are easily determined. Note that
In order to make it easier to understand,
As described above, using the thin line portion and the widened portion,
The change in the impedance is not limited to this thin line portion and widened portion.
No. Also, the part where the line impedance changes is 3 or more
There may be. In this way, the line impedance
If the line parameters of different lines are determined,
Impedance at both input and output
Since the matching is performed, it is not possible to reflect high-frequency signals.
No. Therefore, the line parameters that minimize the reflection loss are acceptable.
A high-frequency circuit that can be easily created.

The connected lines include the case where the line impedances Z ₁ and Z ₂ are equal. In this case, the impedance matching conditions at the input terminal and the output terminal of the filter are such that the voltage reflection coefficient S ₁₁ = S ₂₂ = 0 and the voltage transmission coefficient S ₁₂ =
It has the same meaning as S ₂₁ = 1. When the line impedances to be connected are equal, the matching condition includes such a deformation condition. In particular, when the line impedance of both lines is equal to Z ₀ , the element of the transmission matrix W of the filter is represented by W
_{When 11} = A, W ₁₂ = B, W ₂₁ = C, and W ₂₂ = D, the voltage transmission coefficient S ₁₂ = 2 / (A + B / Z ₀ +
Z ₀ · C + D) is determined to be approximately 1.

[0026]

Embodiments of the present invention will be described below with reference to the drawings. Note that the present invention is not limited to the following examples. (First Embodiment) FIG. 1 shows an example of a high-frequency circuit according to the present invention. The figure is a top view. The high-frequency circuit of the present invention is a coplanar line 20 formed continuously on a ceramic substrate 10 of alumina or the like.
And coplanar lines 20B, 2
0C. These are formed at the end of the line for use in input and output.

The normal coplanar line 20A has a line width 66
μm center conductor 21a and line widths 33 formed on both sides thereof
It is composed of a second ground conductor 22 of 0 μm, and its length is arbitrary. The coplanar line 20 </ b> B includes a thin line portion 21 b having a line width of 30 μm and a length of about 140 μm in which the center conductor is thinned, and the second ground conductor 22. The subsequent coplanar line 20C includes a widened portion 21c of 100 μm × 115 μm in which the line width of the center conductor is widened in reverse, and the second ground conductor 22. The ceramic substrate 10 has a thickness of about 200 μm and a dielectric constant of 9.6, and has a first ground conductor formed on the back surface. That is, the coplanar lines 20A, 20B, 20C are grounded coplanar lines (GCPW). The above elements are manufactured by a semiconductor planar technology such as a lithography technology, a CVD film forming technology, and an etching technology.

FIG. 2 shows a mounting example of the high-frequency circuit of this embodiment. FIG. 2A is an overhead view showing the mounting order, and FIG. 2B is a cross-sectional view after mounting. The ceramic substrate 10 is first joined to a ground metal plate 110 such as brass. Next, the semiconductor chip 50 is inserted into the recess 111 formed on the ground metal plate 110 through the opening 12 provided in the ceramic substrate 10 (FIG. 2A). Finally, a predetermined portion of the semiconductor chip 50 is connected to an end of the high-frequency circuit by a bonding wire 40 (FIG. 2B).

FIG. 3 is an enlarged view of the connection portion. The figure shows that the coplanar line 20 of the present embodiment and the coplanar line 30 formed on the semiconductor chip 50 are bonded to a bonding wire 40.
FIG. FIG. 3A is a top view thereof, and FIG.
(B) shows a sectional view thereof. The widened portion 21c of the coplanar line 20 and the center conductor 31 of the coplanar line 30, and the second ground conductor 22 and the ground conductor 32 are connected by bonding wires 40, respectively. The semiconductor chip 5
0 is, for example, GaAs having a thickness of 450 μm and a dielectric constant of 11.9.
This is a switch element formed on the s substrate. Although not shown, for example, an element such as a PIN diode is formed between the center conductor 31 and the ground conductor 32 of the coplanar line 30. The center conductor width of the coplanar line 30 is about 56 μm, the ground conductor width is about 330 μm,
The separation distance is about 42 μm. The ceramic substrate 10 is grounded by the first ground conductor 11 on the back surface (FIG. 3B).

The bonding wire 40 connecting the coplanar line 20 and the coplanar line 30 is an inductive component. This is the main cause of the insertion loss of the high-frequency element. FIG. 4 shows the impedance of the bonding wire 40 and the resulting insertion loss and reflection loss. FIG. 4A shows the insertion loss, and FIG. 4B shows the reflection loss. This means that a 50 ohm line has an impedance Z _C = 100Ω,
This is a characteristic when three types of bonding wires 40 of 200Ω and 300Ω are connected. It can be seen that the insertion loss and the reflection loss increase as the bonding wire length increases. Next, a method of designing a high-frequency circuit for canceling the inductive component of the bonding wire 40 will be described.

FIG. 5 is a simplified diagram of the center conductor 21a, the thin wire portion 21b, the widened portion 21c, the bonding wire 40, and the line connected thereto. This is the coplanar line 20B, 2
0C, the bonding wire 40 is represented by line parameters, and the impedance matching conditions at the input terminal and the output terminal of the filter composed of the three lines are to be obtained. In other words, the coplanar lines 20B, 20C
Is manufactured, and an impedance conversion filter is formed by using the inductive component of the bonding wire 40 to reduce the reflection loss.

[0032] In detail, the transmission line as shown in FIG. 5, line impedance of the coplanar line having a fine line portion 21b Z _a, the electrical length θ first transmission line H _a of _a, a coplanar line having a widened portion 21c Is the line impedance Z
_b , the second transmission line H _{b having} the electrical length θ _b , and the bonding wire 40 as the line impedance Z _c and the transmission line H _C having the electrical length θ _c will be described.

The transmission matrix F _a of the first transmission line H _a is expressed by the following equation.

[Number 1] _{[cosθ a j Z a sin θ} a] [jsinθ a / Z a cosθ a] ... (1) It should be noted that [] represents the components of each row, it expresses the whole matrix. _Fa is a 2 × 2 matrix. Θ _a is an electrical length, and if the physical length of the line is L _a , θ
_a = 2πL _a / λ. Here, λ is the wavelength of the high frequency signal.

[0034] Similarly, the transmission matrix F _b of the second transmission line H _b
Is represented by the following equation.

[Cos θ _b j Z _b sin θ _b ] [jsin θ _b / Z _b cos θ _b ] (2) where θ _b = 2πL _b / λ

[0035] Then, the transmission matrix F _c of the bonding wire 40 is obtained by the following equation.

[1 jωL] [0 1] (3) where L is the inductance of the bonding wire 40,
ω is the angular frequency of the high frequency signal.

Therefore, the transmission matrix W as a filter is F
It is expressed by _a F _b F _c. Let the elements of the transmission matrix W be W ₁₁ =
When A, W ₁₂ = B, W ₂₁ = C, and W ₂₂ = D, each component can be expressed by the following equation.

[0037]

Equation 4] _{A = - Z a sin θ a} sin θ b / Z b + cosθ a cos θ b ... (4)

B = jωL (cos θ _a cos θ _b -Z _a sin θ _a sin θ _b / Z _b ) + j (Z _b cos θ _a sin θ _b + Z _a sin θ _a cos θ _b ) (5) )

C = jcos θ _a sin θ _b / Z _b + jsin θ _a cos θ _b / Z _a (6)

D = −ωL (sin θ _a cos θ _b / Z _a + cos θ _a sin θ _b / Z _b ) + (− Z _b sin θ _a sin θ _b / Z _a + cos θ _a cos θ _b ) (7)

Here, in the four-terminal network, the condition under which the high-frequency signal propagates without loss is that the lines connected to the filter are matched. That is, put an impedance viewed impedance Z ₁ and the downstream viewed upstream in the filter input are equal, the following equation is established.

[Expression 8] Z ₁ = (AZ ₂ + B) / (CZ ₂ + D) (8) Also, if it is assumed that the impedance Z ₁ looking downstream at the output end is equal to the impedance looking upstream, then the following equation holds. I do.

Z ₂ = (DZ ₁ + B) / (CZ ₁ + A) (9)

The matrix elements A, B, C, and D have
The following equation holds from | w | = 1.

AD-BC = 1 (10) Since the line impedances Z ₁ and Z ₂ are known, such as 50Ω, the matrix elements A, B, C, and D are connected by three relational expressions.

Here, if the matrix elements A, B, C, and D are substituted into the above equations (8), (9), and (10), the four parameters Z _b , θ _b , Z _c , and θ _c become Also, three relational expressions hold. Therefore, if arbitrarily determine an impedance Z _b of the line H _b including, for example, widened portion 21c of the 4 parameters,
The other three parameters are uniquely determined. That is, the shape of the widened portion and the shape of the thin line portion are easily determined. This provides a high-frequency circuit that can easily create line parameters that minimize reflection loss. The above-mentioned arbitrary determination is selected in consideration of a manufacturing limit such as a line width. Then, a plurality of solutions can be obtained within the range. Further, in detail, the whole electromagnetic field simulation is performed, and the electric length is finely adjusted. The high-frequency circuit of the present embodiment is designed and manufactured in this manner.

Next, the above design method will be described qualitatively. FIG. 6A shows the positions A, B, C, and D in the connected coplanar lines 20 and 30, and FIG. 6B shows the reflection coefficient at that position. FIG. 6B is a Smith chart. The Smith chart shows the reflection coefficient 伝 送
Where the horizontal axis is a real number and the vertical axis is an imaginary number.
The reflection coefficient Γ is given by the following equation.

[0042]

１１ (x) = (Z−Z ₀ ) / (Z + Z ₀ ) · exp (−2jβx) = u + jv (11) where: x: position on the line Z: impedance at the position x Z ₀ : line Impedance β: wave number u: real part v: imaginary part

In FIG. 6A, the observation position is A → B →
It is assumed that the movement moves from C to D. First, at point A, Z =
Since it is Z ₀ , Γ (x) = 0 from the above equation (11).
Therefore, in FIG. 6B, the position A ′ representing the reflection coefficient at point A is the origin. Next, at point B, since the impedance of the bonding wire 40 passes through the impedance Z = jωL, from the above equation (11), it moves rightward on the circle with the normalized resistance = 1, and the reflection coefficient Ｂ at point B becomes the first. Move to point B 'in the quadrant. Next, at point C, the capacitance component Z =
Since the signal passes through −j / ωC, it moves rightward on the equivalent normalized conductance circle and moves to the point C ′ in the fourth quadrant. And finally, the impedance Z = jω by the thin wire portion 21b
By passing through L ′, it moves rightward on the circle with the normalized resistance = 1, and point D ′ (point representing the reflection coefficient at point D)
Is moved to the origin A ′. This means that the reflection coefficient is set to 0 again at point D. In other words, it means that impedance matching is performed and a high-frequency signal is propagated. A series of operations based on the transmission matrix means such a transition of the reflection coefficient on the Smith chart. This confirms that a high-frequency circuit that reduces the reflection loss can be easily designed.

The effects of the high-frequency circuit of this embodiment designed and manufactured by the above method will be described. In order to demonstrate the effect, a comparison between the characteristics of a single high-frequency switch element (MMIC) and the characteristics when a high-frequency switch element is mounted is shown. The characteristics are an insertion loss characteristic and an isolation characteristic. FIG. 7A shows the insertion loss characteristics and isolation characteristics of the high-frequency switch element alone, and FIG. 7B shows the same when the same high-frequency switch element is mounted on the high-frequency circuit of the present embodiment. The horizontal axis represents frequency (GHz), and the vertical axis represents isolation characteristics and insertion loss characteristics. In both cases, a curve a shown by a solid line shows the insertion loss, and a curve b shown by a broken line shows the isolation.

As can be seen from the figure, the insertion loss at 76.5 GHz is about -1.5 dB by itself and about -1.9 dB when mounted on a high-frequency circuit. This means that the loss at the line length (about 3 mm) other than the connection part (loss at the coplanar line 20A in FIG. 1) is expected to be about 0.4 dB, so that there is almost no loss at the connection part. are doing. or,
Excellent isolation characteristics are also shown. As described above, if a filter is formed at the end and the filter is designed so that the impedance is matched using the transmission matrix, a high-frequency circuit having excellent insertion loss and isolation can be obtained.

(Second Embodiment) The first embodiment is a high-frequency circuit for connecting two coplanar lines. The present embodiment is a high-frequency circuit for connecting two microstrip lines. FIG. 8 shows a high-frequency circuit according to the second embodiment. The figure is a top view of the circuit pattern. The high-frequency circuit according to this embodiment includes a central conductor 121a, a thin line portion 121b having a reduced width, and a widened portion 121c having a wider conductor width. Although not shown, this high-frequency circuit is formed on a ceramic substrate having a ground conductor on the back surface as in the first embodiment, and a microstrip line is formed with the ground conductor. Then, it is connected to another microstrip line 60 by a bonding wire 40.

Even with this configuration, the transmission matrix of the microstrip line having the thin line portion 121b is Fa, the transmission matrix of the widened portion 121c is Fb, and the bonding wire 40
Is the transmission matrix of Fc, and the line impedance on both sides of the filter consisting of the transmission matrix FaFbFc is Z
₁ and Z ₂ , the same discussion as in the first embodiment holds. Therefore, also in a line using a microstrip line, if the thin line portion 121b and the widened portion 121c are formed at the ends, a high-frequency circuit with reduced insertion loss can be obtained as in the first embodiment.

(Third Embodiment) The first and second embodiments are examples of a high-frequency circuit when two lines are connected by a bonding wire or the like. This embodiment is an example in which two types of different lines are formed on the same substrate and the bonding wire length is set to 0. That is, this is a high-frequency circuit in which different types of lines are continuously connected by a film forming technique such as vapor deposition. The present invention can be applied to such a line conversion.

FIG. 9 shows a top view of the circuit pattern.
The high-frequency circuit according to the present embodiment includes a coplanar line 80, a constricted portion 75 formed continuously from the coplanar line 80, and having a reduced line width of the second ground conductor 73, and a line converter 70 having a length of about λ / 4 or more. Consists of In the line converter 70,
As in the first and second embodiments, a thin line portion 71 in which the line width of the center conductor 81 is thinned and a widened portion 72 in which the width is increased.
Are formed. Further, the ground conductor 73 of the line conversion unit 70 is separated from the center conductor 81 by a predetermined distance (about 50
.mu.m), the width thereof is gradually increased in the direction of the coplanar line 80, and the constricted portion 75 is formed.
Has a length between the ends thereof set to λ / 4. The microstrip line 60 is continuously formed (connected) on the widened portion 72 by a film forming technique such as sputtering.
It is a configuration that is performed. The ground conductor 73 in the figure is the center conductor 8
1, but only one is described.

In this case, the line converter 70 is a coplanar line 70 divided into three parts by the line width of the central conductor 81.
A, 70B, 70C can be seen as a cascade connection. The transmission matrices Fa, Fb, and Fc are applied to these, respectively, and the impedances at the input and output terminals are Z ₁ and Z, respectively.
_{If the number 2} is matched, the same discussion as in the first embodiment is established. Thereby, the thin line portion 71 and the widened portion 72
If the line parameters such as the impedance Z and the electrical length θ _L are determined, the line converter 70 does not reflect the high-frequency signal.

The configuration in which the line width of the ground conductor 73 is gradually increased in the direction of the coplanar line 80 and has the constricted portion 75 with a length of λ / 4 is a coplanar line type electric field mode which is a microstrip line. It is a configuration that gradually and completely converts to the electric field mode of the mold or vice versa. In the ground conductor 73, the constricted portion 75 formed at a length of λ / 4 from the open end causes the potential of the second ground conductor formed on both sides of the center conductor to be a virtual short. This is a configuration in which the electric field of the signal is converted from the center conductor to this virtual short. For example, the propagation mode of the signal that has propagated through the microstrip line 60 is now completely converted to that of the coplanar line and propagated to the coplanar line 80. Normally, when lines having different propagation modes are connected, reflection and leakage occur due to discontinuity of the mode. The above arrangement gradually and completely converts the two different electric field modes. This reduces reflection and leakage of the propagating signal. Therefore, reflection loss and leakage loss due to discontinuity of the propagation mode can be avoided. Therefore, a high-frequency circuit can be provided which can connect different types of lines formed on the same substrate without loss.

(Modifications) Although one embodiment of the present invention has been described above, various other modifications are conceivable. For example, in the first embodiment, the inductive component is formed by thinning the center conductor, but as shown in FIG. 10, an open stub (open branch) shorter than λ / 4 is provided in the second ground conductor formed on both sides. It may be formed. One example of various induction component forming methods is shown in FIG. FIG. 10A shows a stub 90 between the center conductor and the second ground conductor, and FIG.
Is formed. If the stub length is set to λ / 4, the stub branching portion becomes short-circuited, and the high-frequency signal is reflected near the stub. Conversely, if the stub length is λ / 4 or less, an inductive component is formed. FIG.
If a concave portion is formed at a predetermined position of the second ground conductor as shown in (c) and the distance between the center conductor and the second ground conductor is set to be larger than a predetermined value, the line impedance increases at that position. . In this way, an induction component can be produced. Further, as shown in FIG. 10D, a stub having a length of λ / 4 or less may be formed on the center conductor, or an inductive component may be produced.

In the above embodiment, an alumina-based ceramic substrate is used, but another ceramic substrate such as a zirconia-based ceramic substrate may be used. Regardless of the material, the dielectric substrate is not particularly limited as long as it has a dielectric constant equivalent to that of the first embodiment.

In the first to third embodiments, the material of the line to be connected is not mentioned, but the material may be the same or different. One may be an aluminum alloy wiring and the other may be a gold wiring or the like. Further, the line impedances have been described as being different, but it is needless to say that the line impedances may be the same.

[Brief description of the drawings]

FIG. 1 is a top view of a high-frequency circuit according to a first embodiment of the present invention.

FIG. 2 is an explanatory view of mounting a high-frequency circuit according to the first embodiment of the present invention.

FIG. 3 is a connection top view of the high-frequency circuit according to the first embodiment and other lines.

FIG. 4 is a diagram showing the relationship between bonding wires and loss characteristics.

FIG. 5 is an explanatory diagram in which each line is represented by a transmission matrix.

FIG. 6 is a Smith chart showing a reflection coefficient position with respect to each position of a connected line.

FIG. 7 is a comparative diagram showing characteristics of the high-frequency circuit according to the first embodiment.

FIG. 8 is a connection top view of the high-frequency circuit according to the second embodiment and other lines.

FIG. 9 is a connection top view of the high-frequency circuit according to the third embodiment and other lines.

FIG. 10 is a coplanar line pattern forming an inductive component according to a modification.

DESCRIPTION OF SYMBOLS 10 Ceramic substrate 11 First ground conductor 20, 20A Coplanar line 20B, 20C Coplanar line 21a, 121a Center conductor 21b, 121b Thin line portion 21c, 121c Widening portion 22 Second ground conductor 30 Coplanar line 31 Center conductor 32 Ground conductor 40 Bonding wire 50 Semiconductor chip 60 Microstrip line 70 Line conversion unit 80 Coplanar line 90 Stub 110 Ground metal plate

Claims (8)

[Claims] 1. A high-frequency circuit comprising a center conductor and a ground conductor formed on a substrate, wherein another end of the line is connected to another end of the line. A high-frequency circuit wherein a filter is formed by changing a line impedance by partial deformation of at least one of conductors, and impedance matching is performed at an input terminal and an output terminal of the filter. 2. The high-frequency circuit according to claim 1, wherein the filter is formed by generating an inductive component and a capacitive component by a change in line impedance due to the shape deformation. 3. The change in the line impedance is caused by a thin line portion in which a part of the center conductor is thinned and a widened portion in which the line width of the center conductor is widened. 2. The line parameters (line impedance and electrical length) of the filter are determined so that impedance is matched between an input terminal and an output terminal of the filter.
Or the high-frequency circuit according to claim 2. 4. The filter according to claim 1, wherein the end is connected to the other line by a predetermined connection unit, and the filter includes the predetermined connection unit. 2. The high-frequency circuit according to claim 1. 5. The line according to claim 1, wherein the line is a microstrip line having a ground conductor formed on the back surface of the substrate and a center conductor formed on the upper surface of the substrate.
The high-frequency circuit according to the item. 6. The line according to claim 1, wherein the line is a coplanar line including a first ground conductor on the back surface of the substrate, and the center conductor on the upper surface of the substrate and second ground conductors formed on both sides of the center conductor. The high-frequency circuit according to any one of claims 1 to 5. 7. The line parameters (line impedance and electrical length) of the lines constituting the filter having different line impedances include the line impedances Z ₁ and Z _{2 of the} two lines to be connected, and the element of the transmission matrix W of the filter. To W ₁₁
= A, W ₁₂ = B, W ₂₁ = C, W ₂₂ = D, the impedance matching condition Z ₁ at the input end of the filter
= (AZ ₂ + B) / (CZ ₂ + D) and the matching condition Z ₂ = (DZ ₁ + B) / (CZ ₁ + A) at the output end are determined. Claim 6
The high-frequency circuit according to any one of the above. 8. The line parameters (line impedance and electrical length) of the lines constituting the filter having different line impedances are Z ₀ , the line impedance of both connected lines, and W _11, the element of the transmission matrix W of the filter. = A,
When W ₁₂ = B, W ₂₁ = C, and W ₂₂ = D, the voltage transmission coefficient S ₁₂ = 2 / (A + B) in the filter.
/ Z ₀ + Z ₀ · C + D) is determined to be approximately 1. The high-frequency circuit according to any one of claims 1 to 6, wherein: