WO2021059350A1 - Electric motor driving device and refrigeration cycle application apparatus - Google Patents

Abstract

The present invention comprises an inverter (30) for supplying an AC voltage with a variable frequency and voltage value to an electric motor (7) that drives a load element the load torque of which fluctuates periodically, and a control device (100) for controlling the inverter (30), wherein the control device (100) includes: an excitation current command value generation unit (113) which generates a γ-axis current command value in a rotating coordinate system; a speed control unit (610) which generates a first δ-axis current command value in the rotating coordinate system; a limiting unit (630) which limits the first δ-axis current command value by using a limit value, and generates a second δ-axis current command value; a compensation value calculation unit (504) which generates a δ-axis current compensation value such that the output torque of the electric motor (7) follows the periodic fluctuation of the load torque; and a vibration suppression control unit (640) which generates a third δ-axis current command value by using the second δ-axis current command value, the limit value, and the δ-axis current compensation value, and controls the inverter (30) by using the γ-axis current command value and the third δ-axis current command value.

Description

Electric motor drive and refrigeration cycle applicable equipment

The present invention relates to an electric motor drive device and a refrigeration cycle applicable device including the electric motor drive device.

Permanent magnet synchronous motors (hereinafter referred to as PM (Permanent Magnet) motors) have higher efficiency characteristics than induction motors, so their application range is expanding to the fields of home appliances, industrial equipment, electric vehicles, etc. .. Equipment equipped with a PM motor is required to have high efficiency in the low speed rotation range, that is, with a light load, in accordance with the movement to prevent global warming and save energy, and in order to improve the usability of the equipment, the high speed rotation range, That is, it is also required to expand the drive range under high load. As a method of improving efficiency in the low speed rotation range of the motor, there is a low speed design by increasing the amount of magnets and windings of the motor. However, when the motor is designed at a low speed, the induced voltage generated in the high speed rotation range increases. To solve such a problem, a method of expanding the high-speed rotation range of the motor by using the weakening magnetic flux control is known.

The PM motor is used, for example, in a compressor provided in a refrigerating and air-conditioning device for operating a compression mechanism for compressing a refrigerant. Refrigerating and air-conditioning equipment has a characteristic that the load torque fluctuates periodically according to the operation process in which the compressor compresses the refrigerant. In general, refrigeration and air conditioning equipment suppresses the speed fluctuation of the PM motor by matching the output torque of the PM motor with the load torque of the compressor that fluctuates periodically, and performs vibration suppression control that reduces the vibration of the compressor. Is going. According to Patent Document 1, a motor drive device mounted on a refrigerating and air-conditioning device suppresses the occurrence of step-out in the weakened magnetic flux control region, and secures necessary torque even when vibration suppression control is performed in the weakened magnetic flux control region. The technology to control it so that it can be performed is disclosed.

Japanese Unexamined Patent Publication No. 2018-14854

However, according to the above-mentioned conventional technique, if the vibration suppression control in the weakened magnetic flux control region is not properly performed, overcurrent interruption occurs in the refrigerating and air-conditioning equipment, and the operation is inefficient. There was a problem.

The present invention has been made in view of the above, and it is desired to obtain an electric motor drive device capable of performing highly efficient operation while suppressing the occurrence of overcurrent and step-out in the entire operating range of the electric motor. The purpose.

In order to solve the above-mentioned problems and achieve the object, the electric motor drive device according to the present invention supplies an AC voltage having a variable frequency and voltage value to an electric motor that drives a load element in which the load torque fluctuates periodically. The inverter is provided with a control device for controlling the inverter. The control device includes a γ-axis current command value generator that generates a γ-axis current command value in a rotational coordinate system having a γ-axis and a δ-axis, and a speed control unit that generates a first δ-axis current command value in the rotational coordinate system. The limit value is used to limit the first δ-axis current command value, and the limit unit that generates the second δ-axis current command value, and the output torque of the motor follow the periodic fluctuation of the load torque. A compensation value calculation unit that generates a δ-axis current compensation value, and a vibration suppression control unit that generates a third δ-axis current command value using the second δ-axis current command value, limit value, and δ-axis current compensation value. And, the inverter is controlled by using the γ-axis current command value and the third δ-axis current command value.

The electric motor drive device according to the present invention has an effect that highly efficient operation can be performed while suppressing the occurrence of overcurrent and step-out in the entire operating area of the electric motor.

The figure which shows the structural example of the electric motor drive device which concerns on Embodiment 1. The figure which shows the configuration example of the inverter included in the electric motor drive device which concerns on Embodiment 1. The figure which shows the operation state when the vibration suppression control is not performed in the electric motor drive device which concerns on Embodiment 1. FIG. The figure which shows the operation state at the time of having the vibration suppression control in the electric motor drive device which concerns on Embodiment 1. A block diagram showing a configuration example of a control device included in the electric motor drive device according to the first embodiment. A block diagram showing a configuration example of a voltage command value calculation unit included in the control device according to the first embodiment. A block diagram showing a configuration example of a compensation value calculation unit included in the voltage command value calculation unit according to the first embodiment. The figure which shows the voltage vector which shows the state of the voltage applied to the electric motor when the electric motor is rotating in a high speed region in the electric motor drive device which concerns on Embodiment 1. The figure which shows the voltage vector which shows the state of the voltage applied to the electric motor when the electric motor is rotating in a high speed region in the electric motor drive device which concerns on Embodiment 1, and vibration suppression control is performed about the limiter value of a δ-axis current command. The figure which shows the difference of the γ-axis current by the magnitude of the limiter value of the δ-axis current command in the electric motor drive device which concerns on Embodiment 1. A first block diagram showing a configuration example of a torque current command value generation unit included in the voltage command value calculation unit according to the first embodiment. The first flowchart which shows the operation of the torque current command value generation part which concerns on Embodiment 1. A flowchart showing an operation in which the limiting unit according to the first embodiment selects a limiter value. The first figure which shows the operating state of the electric motor drive device which concerns on Embodiment 1. A second block diagram showing a configuration example of a torque current command value generation unit included in the voltage command value calculation unit according to the first embodiment. A second flowchart showing the operation of the torque current command value generation unit according to the first embodiment. FIG. 2 is a second diagram showing an operating state of the electric motor drive device according to the first embodiment. The figure which shows an example of the hardware configuration which realizes the control device provided in the electric motor drive device which concerns on Embodiment 1. The figure which shows the structural example of the refrigerating cycle application apparatus which concerns on Embodiment 2.

Hereinafter, the electric motor drive device and the refrigeration cycle applicable device according to the embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to this embodiment.

Embodiment 1.
FIG. 1 is a diagram showing a configuration example of an electric motor drive device 2 according to a first embodiment of the present invention. FIG. 2 is a diagram showing a configuration example of an inverter 30 included in the electric motor drive device 2 according to the first embodiment. The electric motor drive device 2 is connected to the AC power supply 1 and the electric motor 7. The electric motor drive device 2 rectifies the AC voltage supplied from the AC power supply 1, converts it into an AC voltage again, supplies the AC voltage to the electric motor 7, and drives the electric motor 7. The electric motor drive device 2 includes a reactor 4, a rectifier circuit 10, a smoothing capacitor 20, an inverter 30, a bus voltage detection unit 82, a bus current detection unit 84, and a control device 100.

The rectifier circuit 10 includes four diodes D1, D2, D3, and D4. The four diodes D1 to D4 are bridge-connected to form a diode bridge circuit. The rectifier circuit 10 rectifies the AC voltage supplied from the AC power supply 1 by a diode bridge circuit composed of four diodes D1 to D4. In the rectifier circuit 10, one end of the input terminal is connected to the AC power supply 1 via the reactor 4, and the other end of the input terminal is connected to the AC power supply 1. Further, in the rectifier circuit 10, the output terminal is connected to the smoothing capacitor 20.

The smoothing capacitor 20 smoothes the output voltage of the rectifier circuit 10. One electrode of the smoothing capacitor 20 is connected to the first output terminal of the rectifier circuit 10 and the DC bus 22a on the high potential side, that is, the positive side. The other electrode of the smoothing capacitor 20 is connected to the second output terminal of the rectifier circuit 10 and the DC bus 22b on the low potential side, that is, the negative side. The voltage smoothed by the smoothing capacitor 20 is referred _{to as a bus voltage V dc.}

The inverter 30 receives the voltage across the smoothing capacitor 20, that is, the bus voltage V _dc , generates a three-phase AC voltage having a variable frequency and voltage value, and supplies the voltage to the electric motor 7 via the output lines 331 to 333. As shown in FIG. 2, the inverter 30 includes an inverter main circuit 310 and a drive circuit 350. The input terminal of the inverter main circuit 310 is connected to the DC bus 22a and 22b. The inverter main circuit 310 includes switching elements 311 to 316. Rectifier elements 321 to 326 for reflux are connected in antiparallel to each of the switching elements 311 to 316.

The drive circuit 350 generates drive signals Sr1 to Sr6 based on the PWM (Pulse Width Modulation) signals Sm1 to Sm6 output from the control device 100. The drive circuit 350 controls the on / off of the switching elements 311 to 316 by the drive signals Sr1 to Sr6. As a result, the inverter 30 can supply the frequency-variable and voltage-variable three-phase AC voltage to the electric motor 7 via the output lines 331 to 333.

The PWM signals Sm1 to Sm6 are signals having a signal level of a logic circuit, that is, a magnitude of 0V to 5V. The PWM signals Sm1 to Sm6 are signals using the ground potential of the control device 100 as a reference potential. On the other hand, the drive signals Sr1 to Sr6 are signals having a voltage level required for controlling the switching elements 311 to 316, for example, a magnitude of -15V to + 15V. The drive signals Sr1 to Sr6 are signals whose reference potential is the potential of the negative terminal of the corresponding switching element, that is, the emitter terminal.

The electric motor 7 is, for example, a three-phase permanent magnet synchronous motor. In the present embodiment, it is assumed that the electric motor 7 drives a load element in which _{the load torque T l fluctuates periodically.} In the following description, the electric motor may be referred to as a motor.

The bus voltage detection unit 82 detects the voltage between the DC bus 22a and 22b as the bus voltage _Vdc . The bus voltage detection unit 82 includes, for example, a voltage dividing circuit that divides the voltage by resistors connected in series. The bus voltage detection unit 82 _{converts the detected bus voltage V dc} into a voltage suitable for processing by the control device 100 using a voltage dividing circuit, for example, a voltage of 5 V or less, and uses it as a voltage detection signal which is an analog signal. Output to the control device 100. The voltage detection signal output from the bus voltage detection unit 82 to the control device 100 is converted from an analog signal to a digital signal by an AD (Analog to Digital) conversion unit (not shown) in the control device 100, and is internally processed by the control device 100. Used for.

The bus current detection unit 84 includes a shunt resistor inserted in the DC bus 22b. The bus current detection unit 84 uses a shunt resistor to detect the _{current input to the inverter 30 as a direct current I dc.} The bus current detection unit 84 outputs the detected direct current I _dc to the control device 100 as a current detection signal which is an analog signal. The current detection signal output from the bus current detection unit 84 to the control device 100 is converted from an analog signal to a digital signal by an AD conversion unit (not shown) in the control device 100, and is used for internal processing in the control device 100.

The control device 100 generates PWM signals Sm1 to Sm6 in order to control the inverter 30. The control device 100 outputs PWM signals Sm1 to Sm6 to the inverter 30 to control the inverter 30. Specifically, the control device 100 controls the inverter 30 to change the angular frequency ω and the voltage value of the output voltage of the inverter 30.

The angular frequency ω of the output voltage of the inverter 30 is represented by the same code ω as the angular frequency of the output voltage, and determines the rotational angular velocity at the electric angle of the electric motor 7. _{The rotational angular velocity ω m} at the mechanical angle of the electric motor 7 is equal to the rotational angular velocity ω at the electric angle of the electric motor 7 divided by the _{pole log number P m. Therefore, there is a relationship represented by the following equation (1) between the rotational angular velocity ω m} at the mechanical angle of the electric motor 7 and the angular frequency ω of the output voltage of the inverter 30. In the following description, the rotational angular velocity may be simply referred to as a rotational velocity, and the angular frequency may be simply referred to as a frequency.

Controller 100, the phase current _i u flowing through the electric motor _7, i v, based on the _{i w} generates an excitation current command value i _gamma ^*, exciting current command value i _gamma ^* gamma-axis voltage command value V _gamma based on ^{* Is} generated. Further, the control unit 100, the frequency estimate omega _est of the electric motor 7 calculates a torque current command value i _[delta] ^* to match the frequency command value omega _e ^*, [delta] axes based on the torque current command value i _[delta] ^* Generates the voltage command value V _δ ^*. The control device 100 controls the inverter 30 based on the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^*. As described above, in the present embodiment, the control device 100 controls in the rotating coordinate system having the γ axis and the δ axis.

When the electric motor 7 drives a load element in which _{the load torque T l} fluctuates periodically, the control device 100 _{causes the output torque T m} of the electric motor 7 to follow the periodic fluctuation of the load torque T _{l, that is, pulsation.} It is desirable to control the inverter 30. The control device 100 may generate a torque current compensation value in order to make _{the output torque T m} follow the periodic fluctuation of the load torque T _{l, that is, the pulsation.} In the control device 100, the generated torque current compensation value is used to correct the _{torque current command value i δ} ^*.

Here, the necessity and operation of vibration suppression control in the electric motor drive device 2 will be described with reference to FIGS. 3 and 4. FIG. 3 is a diagram showing a state of operation of the electric motor drive device 2 according to the first embodiment when vibration suppression control is not performed. FIG. 4 is a diagram showing a state of operation of the electric motor drive device 2 according to the first embodiment when vibration suppression control is provided. _{3 and 4 show the load torque T l} of the single rotary compressor assumed as the load element driven by the electric motor 7 in one rotation of the mechanical angle of the electric _{motor 7, the output torque T m} of the electric motor 7, and the electric motor in the single rotary compressor. It is a figure which shows the relationship between the rotation speed of 7 and a torque current compensation value. FIG. 3 shows a state in which _{the control device 100 constantly controls the output torque T m} of the electric motor 7. FIG. 4 shows a state in which the control device 100 controls the _{torque current compensation value so that the output torque T m} of the electric motor 7 matches the load torque T _l , and controls the rotation speed to be constant.

As can be seen from FIG. 3, when the control device 100 _{constantly controls the output torque T m} of the electric motor 7, the rotation speed fluctuates due to the difference between _{the output torque T m} of the electric motor 7 and the load torque T _l. When the rotation speed fluctuates, not only vibration and noise are generated in the single rotary compressor, but also when the rotation speed fluctuates greatly, the electric motor 7 may step out and stop.

Therefore, in the present embodiment, the control device 100 controls the output torque T _m of the electric motor 7 so as to match the load torque T _{l in the vibration suppression control shown in FIG.} The control device 100 can control the rotation speed to be constant by eliminating excess or deficiency of torque between _{the output torque T m} and the load torque T _{l of the electric motor 7.} However, the control device 100, in order to realize the vibration suppression control, as shown in FIG. 4, it is necessary to be changed in accordance a torque current compensation value i _{Deruta_trq} the load torque T _l.

The configuration of the control device 100 will be described. FIG. 5 is a block diagram showing a configuration example of a control device 100 included in the electric motor drive device 2 according to the first embodiment. The control device 100 includes an operation control unit 102 and an inverter control unit 110.

The operation control unit 102 receives the command information Q _e from the outside and generates the frequency command value ω _e ^* based on the command information Q _e. As shown in the following equation (2), the frequency command value ω _e ^* can be obtained by multiplying the _{rotation angular velocity command value ω m} ^* , which is the command value of the rotation speed of the electric motor 7, by the pole logarithm P _m. ..

Controller 100, when controlling the air conditioner as a refrigeration cycle application device, controls the operation of each unit of the air conditioner based on the command information Q _e. The command information Q _e includes, for example, a temperature detected by a temperature sensor (not shown), information indicating a set temperature indicated by a remote controller which is an operation unit (not shown), operation mode selection information, operation start and operation end instruction information, and the like. Is. The operation mode is, for example, heating, cooling, dehumidification, and the like. The operation control unit 102 may be outside the control device 100. That is, the control device 100 may be configured to acquire the _{frequency command value ω e} ^{* from the outside.}

The inverter control unit 110 includes a current restoration unit 111, a three-phase two-phase conversion unit 112, an exciting current command value generation unit 113, a voltage command value calculation unit 115, an electric phase calculation unit 116, and a two-phase three-phase conversion. A unit 117 and a PWM signal generation unit 118 are provided.

_{The current restoration unit 111 restores the phase currents i u} , _iv , i _w flowing through the motor 7 based on the _{direct current I dc} detected by the bus current detection unit 84. _{The current restoration unit 111 samples the direct current I dc} detected by the bus current detection unit 84 at a timing determined based on the PWM signals Sm1 to Sm6 generated by the PWM signal generation unit 118, whereby the phase current i _u , _iv , i _w can be restored.

The three-phase two-phase conversion unit 112 uses _{the electric phase θ e} generated by the electric phase calculation unit 116, which will be described later, to _{convert the phase currents i u} , _iv , i _w restored by the current restoration unit 111 to the γ-axis. It is converted into an exciting current i _γ , which is a current, and a torque current i _δ , which is a δ-axis current, that is, a current value on the γ-δ axis.

The exciting current command value generation unit 113 generates the exciting current command value i _γ ^* in the above-mentioned rotating coordinate system. Specifically, the exciting current command value generation unit 113 obtains the _{optimum exciting current command value i γ} ^* that is most efficient for driving the electric motor 7 based on the _{torque current i δ.} Based on the _{torque current i δ} , the exciting current command value generator 113 sets the _{output torque T m} to be greater than or equal to or maximum than the specified value, that is, the current phase β _{m to be equal to or less than or minimum to the specified value.} The exciting current command value i _γ ^* is output. Here, the exciting current command value generation unit 113 obtains the exciting current command value i _γ ^* _{based on the torque current i δ} , but this is an example and is not limited thereto. The same effect can be obtained even if the exciting current command value generation unit 113 obtains the exciting current command value i _γ ^* _{based on the exciting current i γ, the} frequency command value ω _e ^{*, or the like.} Further, the exciting current command value generation unit 113 may determine the _{exciting current command value i γ} ^{* by the weakening magnetic flux control as described later.} In the following description, the exciting current command value may be referred to as a γ-axis current command value, and the exciting current command value generating unit may be referred to as a γ-axis current command value generating unit.

The voltage command value calculation unit 115 includes a frequency command value ω _e ^* _{acquired from the operation control unit 102, an exciting current i γ} and a torque current i _δ acquired from the three-phase two-phase conversion unit 112, and an exciting current command value generation unit. Based on the exciting current command value i _γ ^* obtained from 113, the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^* are generated. Further, the voltage command value calculation unit 115 sets the frequency estimation value ω _{est based on the γ-axis voltage command value V γ} ^* , the δ-axis voltage command value V _δ ^* , the exciting current i _γ, and the torque current i _δ. To estimate.

The electric phase calculation unit 116 calculates the electric phase θ _e by _{integrating the frequency estimation value ω est} acquired from the voltage command value calculation unit 115.

_{The two-phase three-phase conversion unit 117 performs electrical phase calculation on the γ-axis voltage command value V γ} ^* and the δ-axis voltage command value V _δ ^* acquired from the voltage command value calculation unit 115, that is, the voltage command value of the two-phase coordinate system. _{Using the electric phase θ e} acquired from the unit 116, the voltage is converted into the _{three-phase voltage command values V u} ^* , V _v ^* , and V _w ^* , which are the output voltage command values of the three-phase coordinate system.

The PWM signal generation unit 118 combines the three-phase voltage command values V _u ^* , V _v ^* , V _w ^* _{acquired from the two-phase three-phase conversion unit 117 and the bus voltage V dc} detected by the bus voltage detection unit 82. By comparing, PWM signals Sm1 to Sm6 are generated. The PWM signal generation unit 118 can also stop the electric motor 7 by not outputting the PWM signals Sm1 to Sm6.

The configuration of the voltage command value calculation unit 115 will be described. FIG. 6 is a block diagram showing a configuration example of the voltage command value calculation unit 115 included in the control device 100 according to the first embodiment. The voltage command value calculation unit 115 includes a frequency estimation unit 501, a subtraction unit 502, a torque current command value generation unit 503, a compensation value calculation unit 504, a subtraction unit 509, 510, an excitation current control unit 511, and a torque. A current control unit 512 is provided.

The frequency estimation unit 501 is the frequency of the voltage supplied to the electric motor 7 based on _{the exciting current i γ} , the torque current i _δ , the γ-axis voltage command value V _γ ^*, and the δ-axis voltage command value V _δ ^*. _Is estimated and output as the frequency estimated value ω est.

The subtraction unit 502 calculates the difference (ω _e ^* −ω _est ) of _{the frequency estimation value ω est} estimated by the frequency estimation unit 501 with respect to the frequency command value ω _e ^*.

The compensation value calculation unit 504 generates a _{torque current compensation value i δ_trq so that the output torque T m} of the electric motor 7 follows the periodic fluctuation of the load torque T _l. Specifically, the compensation value calculation unit 504 generates the torque current compensation value i _{δ_trq based on the frequency estimation value ω est acquired from the frequency estimation unit 501.} The torque current compensation value i _{δ_trq} is for suppressing the pulsating component of the frequency estimation value ω _est , particularly the pulsating component having a frequency of ω _mn. Here, the " _{pulsating component of the frequency estimated value ω est} , particularly the pulsating component having a frequency of ω _mn " is a pulsating component of a DC amount which is a value representing the _{frequency estimated value ω est , particularly the pulsating frequency is ω mn} . It means a pulsating component. Note that m is a parameter related to the amount of DC, and n is a parameter indicating the load driven by the electric motor 7. Regarding n, for example, the load driven by the electric motor 7 is 1 in the case of a single rotary compressor and 2 in the case of a twin rotary compressor. Further, n may be 3 or more. In the following description, the torque current compensation value i _{δ_trq} may be referred to as a δ-axis current compensation value.

The torque current command value generation unit 503 generates the torque current command value i _δ ^*** in the above-mentioned rotating coordinate system. Specifically, the torque current command value generation unit 503 performs proportional integration calculation, that is, PI (Proportional Integral) control on _{the difference (ω e} ^* -ω _{est) calculated by the subtraction unit 502, and the difference.} Find the torque current command value i _δ ^* that brings (ω _e ^* _{-ω est) close to zero. By generating the torque current command value i δ} ^* in this way, the torque current command value generation unit 503 controls to match the frequency estimation value ω _est with the frequency command value ω _e ^*. Further, the torque current command value generation unit 503 is generated by the pulsation of the load torque T _{l by correcting the torque current command value i δ} ^* _{using the torque current compensation value i δ_trq} acquired from the compensation value calculation unit 504. It is possible to suppress the speed pulsation. The torque current command value generation unit 503 generates and outputs the torque current command value i _δ ^*** _{corrected by using the torque current compensation value i δ_trq.} In the following description, the torque current command value may be referred to as a δ-axis current command value, and the torque current command value generation unit may be referred to as a δ-axis current command value generation unit.

The subtraction unit 509 calculates the difference (i _γ ^* −i _γ ) of the exciting current i _γ with respect to the exciting current command value i _γ ^*. The subtraction unit 510 calculates the difference (i _δ ^*** −i _δ ) of the torque current i _δ with respect to the torque current command value i _δ ^***.

The exciting current control unit 511 performs a proportional integration operation on _{the difference (i γ} ^* -i _{γ ) calculated by the subtraction unit 509 to bring the difference (i γ} ^* -i _γ ) close to zero. Generates the value V _γ ^*. _{By generating the γ-axis voltage command value V γ} ^* in this way, the exciting current control unit 511 controls to match the exciting current i _γ with the exciting current command value i _γ ^*.

The torque current control unit 512 performs a _{proportional integration calculation on the difference (i δ} ^*** -i _δ ) calculated by the subtraction unit 510 to bring the difference (i _δ ^*** -i _δ ) close to zero. Generates the δ-axis voltage command value V _δ ^*. _{By generating the δ-axis voltage command value V δ} ^* in this way, the torque current control unit 512 controls the torque current i _δ to match the torque current command value i _δ ^***.

The configuration of the compensation value calculation unit 504 will be described. FIG. 7 is a block diagram showing a configuration example of the compensation value calculation unit 504 included in the voltage command value calculation unit 115 according to the first embodiment. The compensation value calculation unit 504 includes a calculation unit 550, a cosine calculation unit 551, a sine calculation unit 552, a multiplication unit 535, 554, a low- pass filter 555, 556, a subtraction unit 557, 558, and a frequency control unit 559, It includes a 560, a multiplication unit 561, 562, and an addition unit 563.

The calculation unit 550 calculates the _{mechanical angle phase θ mn} indicating the rotation position of the electric motor 7 by integrating _{the frequency estimation value ω est} and dividing by the pole logarithm. Cosine calculation unit 551, based on the mechanical angle phase theta _mn, calculates the cosine cos [theta] _mn. Sine calculator 552, based on the mechanical angle phase theta _mn, calculates a sine sin [theta _mn.

Multiplying unit 553 multiplies the cosine cos [theta] _mn in frequency estimate omega _est, calculates the cosine component ω _est · cosθ _mn frequency estimate omega _est. Multiplying unit 554 multiplies the sine sin [theta _mn in frequency estimate omega _est, it calculates a sine component ω _est · sinθ _mn frequency estimate omega _est. The cosine component ω _est · cosθ _mn and sine component ω _est · sinθ _mn is calculated by multiplying unit 553 and 554, other pulsating component frequency is omega _mn, pulsating component having a frequency higher than the frequency omega _mn, i.e. Harmonic components are included.

The low- pass filters 555 and 556 are first-order lag filters whose _{transfer function is represented by 1 / (1 + s · T f).} Here, s is a Laplace operator. T _f is a time constant and is defined to remove pulsating components at frequencies higher than _{the frequency ω mn.} In addition, "removal" includes a case where a part of the pulsating component is attenuated, that is, reduced. The time constant T f is set by the operation control unit 102 based on the speed command value, and the operation control unit 102 may notify the low- pass filters 555 and 556, or the low- pass filters 555 and 556 hold the _{time constant T f.} Good. Regarding the low- pass filters 555 and 556, the first-order lag filter is an example, and may be a moving average filter or the like, and the type of filter is not limited as long as the pulsating component on the high frequency side can be removed.

The low-pass filter 555 performs low-pass filtering on the cosine component ω _est · cos θ _mn , removes the pulsating component having a frequency higher than the frequency ω _mn , and outputs the _{low frequency component ω est_cos.} The low frequency component ω _{est_cos} is a DC amount representing a cosine component having a _{frequency of ω mn} among the pulsating components of the frequency estimation value ω _est.

The low-pass filter 556 performs low-pass filtering on the sine component ω _est · sin θ _mn , removes a pulsating component having a frequency higher than the frequency ω _mn , and outputs a _{low frequency component ω est_sin.} The low frequency component ω _{est_sin} is a DC amount representing a sine component having a _{frequency of ω mn} among the pulsating components of the frequency estimation value ω _est.

The subtraction unit 557 calculates the difference (ω _{est_cos} −0) _{between the low frequency component ω est_cos} and 0 output from the low-pass filter 555. The subtraction unit 558 calculates the difference (ω _{est_sin −} 0) _{between the low frequency component ω est_sin} and 0 output from the low-pass filter 556.

_{The frequency control unit 559 performs a proportional integration calculation on the difference (ω est_cos-} 0) calculated by the subtraction unit 557, and _{sets the cosine component i δ_trq_cos} of the current command value that brings the difference (ω _{est_cos-} 0) close to zero. calculate. The frequency control unit 559 controls the low frequency component ω _{est_cos} to match 0 _{by generating the cosine component i δ_trq_cos in this way.}

The frequency control unit 560 performs a proportional integration operation on _{the difference (ω est_sin-} 0) calculated by the subtraction unit 558, and _{sets the sine component i δ_trq_sin} of the _{current command value that brings the difference (ω est_sin-} 0) close to zero. calculate. The frequency control unit 560 controls the low frequency component ω _{est_sin} to match 0 _{by generating the sine component i δ_trq_sin in this way.}

Multiplication unit 561 generates a _i δ_trq_cos · cosθ _mn by multiplying the cosine cos [theta] _mn to the cosine component i _{Deruta_trq_cos} output from the frequency control unit 559. i _{δ_trq_cos} · cos θ _mn is an AC component having a frequency n · ω _est.

Multiplying unit 562 multiplies the sine sin [theta _mn to produce a _i δ_trq_sin · sinθ _mn sine component i _{Deruta_trq_sin} output from the frequency control unit 560. i _{δ_trq_sin} · sinθ _mn is an AC component having a frequency n · ω _est.

The addition unit 563 obtains the sum of _{i δ_trq_cos} · cos θ _mn output from the multiplication unit 561 and _{i δ_trq_sin} · sin θ _mn output from the multiplication unit 562. The compensation value calculation unit 504 outputs what is obtained by the addition unit 563 as the torque current compensation value i _{δ_trq.}

_{The torque current command value generation unit 503 adds the} torque current compensation value i δ_trq obtained in the compensation value calculation unit 504 as described above to the torque current command value in the middle of calculation, and adds the addition result to the corrected torque current. By using it as the command value i _δ ^*** , the pulsating component can be suppressed.

The operation of the torque current command value generation unit 503 will be described in detail. Generally, in an electric motor drive device that controls a refrigeration cycle application device, a limiter value is set for a δ-axis current command for the purpose of vibration suppression control or the like. In the present embodiment, the torque current command value generation unit 503 uses the limiter values iδ_lim1, iδ_lim2, iδ_trq_lim as the limiter value for the δ-axis current command. The limiter values iδ_lim1, iδ_lim2, and iδ_trq_lim are represented by the following equations (3) to (5), respectively.

The limiter value iδ_lim1 is assumed to be limited based on the current value of the electric motor 7 when the rotation speed of the electric motor 7 is in the low speed region. In the formula (3), _Ie is an effective value of the overcurrent cutoff value of the phase current determined by the demagnetization limit of the motor 7. The torque current i _[delta], because to be a priority constituting the exciting current i _gamma, has a configuration obtained by subtracting the excitation current i _gamma from overcurrent break value of the phase current. That is, the limiter value iδ_lim1 is defined by the current limit value with respect to the phase current of the electric motor 7 and the exciting current i _γ.

The limiter value iδ_lim2 is assumed to be limited based on the voltage value of the electric motor 7 when the rotation speed of the electric motor 7 is in the medium-high speed region. In the formula (4), L _γ is the γ-axis inductance of the above-mentioned rotating coordinate system, and L _δ is the δ-axis inductance of the above-mentioned rotating coordinate system. Generally, there is a limit to the maximum AC voltage that the inverter 30 can output to the motor 7. Therefore, when the limit value of the γδ-axis voltage is set to _Vol , the relationship between the exciting current i _γ and the torque current i _{δ is} , Is expressed as in equation (6). The limit value _Vom may be a value obtained by subtracting, for example, the winding resistance of the electric motor 7 and the voltage drop of the switching elements 311 to 316 of the inverter 30. Strictly speaking, the output limit range of the inverter 30 is a hexagonal shape, but here it is approximated by a circle. In the present embodiment, the discussion is made on the premise that the approximation is made by a circle, but it goes without saying that the discussion may be made by strictly considering a hexagon.

Equation (4) can be derived by solving equation (6) with _{respect to the torque current i δ.} Since the delta-axis current command of the equation (4) can take into consideration the voltage limit and the effectiveness of the weakening magnetic flux control, a more optimum delta-axis current is compared with, for example, the number 6 described in Patent Document 1. It can be said that it is the limiter value of the command. In other words, the limiter value iδ_lim2 the voltage limit value the inverter 30 is the limiting value V _om based on the output enable voltage to the motor 7, the rotational speed of the motor 7, the ?? axes flux linkage [Phi _a motor _7, the rotation of the above It is defined by the γ-axis inductance of the coordinate system and the δ-axis inductance of the rotating coordinate system described above.

In the present embodiment, the circle whose radius centered on the origin is the limit value _Vom is referred to as the voltage limit circle 21. Incidentally, the limit value _{V om,} when the inverter 30 is a PWM inverter, it is known to vary with the value of the bus voltage _{V dc.}

Since the velocity electromotive force ω _e Φ _a becomes very large in the high-speed region, in order to increase the _{torque current i δ , the exciting current i γ} is passed in the negative direction, and ^{the amplitude of the voltage command vector v *} is set in the voltage limiting circle 21. Must be within range. As described above, the control method of generating the γ-axis stator magnetic flux L _γ i _γ in the direction opposite to the γ δ-axis magnetic flux chain crossover number Φ _a to reduce the voltage amplitude is generally called weakening magnetic flux control.

FIG. 8 is a diagram showing a voltage vector representing a state of voltage applied to the electric motor 7 when the electric motor 7 is rotating in a high speed region in the electric motor drive device 2 according to the first embodiment. In FIG. 8, the arc-shaped dotted line is the above-mentioned limit value _Vom , that is, the voltage limit circle 21. FIG. 8 shows the difference between the present embodiment and Patent Document 1 regarding the limiter value of the δ-axis current command. When iδ_lim2 is used as the limiter value of the δ-axis current command as in the present embodiment, the δ-axis current command according to the effect of the weakening magnetic flux control by _{ω e} L _γ i _γ with respect to _{ω e} Φ _a. The limiter value can be uniquely determined. On the other hand, when iδ_lim4 is used as the limiter value of the δ-axis current command according to Patent Document 1, the limiter value of the δ-axis current command cannot be uniquely determined, resulting in excess or deficiency.

FIG. 9 shows the state of the voltage applied to the electric motor 7 when the electric motor 7 is rotating in the high-speed region in the electric motor drive device 2 according to the first embodiment and vibration suppression control is performed for the limiter value of the δ-axis current command. It is a figure which shows the representative voltage vector. The voltage command fluctuates in the γ-axis direction depending on the torque / current compensation value i _{δ_trq.} Further, when the voltage command vector v ^* is as large as the voltage limit circle 21 as shown in FIG. 9, and the torque current compensation value i _{δ_trq} increases in the positive direction and the γ-axis voltage increases in the negative direction, the electric motor The drive device 2 controls the weakening magnetic flux so as to be within the voltage limiting circle 21.

In FIG. 9, for example, when iδ_lim4 is excessively large, the voltage command vector becomes v ^** , which greatly deviates from the voltage limiting circle 21. In this case, the possibility of step-out in the electric motor 7 increases, and the control of the electric motor drive device 2 with respect to the electric motor 7 tends to become unstable. When the voltage command vector is v ^** , the electric motor drive device 2 needs to perform weak magnetic flux control in order to stably drive the electric motor 7. _{However, if the exciting current i γ} is passed through the electric motor driving device 2 more than necessary, the operation becomes inefficient when controlling the electric motor 7. As a result of the weakening magnetic flux control, the voltage command becomes v_lim4 ^* . 9, the voltage command vector v _{ω e} L γ i γ by flux-weakening control for ^**, it is understood that greater than _{ω e} L γ i γ by flux-weakening control for the voltage command vector v ^*. Further, in FIG. 9, the amplitude of the γ-axis direction with respect to the voltage command vector v ^** It can be seen that is larger than the amplitude of the γ-axis direction with respect to the voltage command vector v ^*.

FIG. 10 is a diagram showing a difference in γ-axis current depending on the magnitude of the limiter value of the δ-axis current command in the motor drive device 2 according to the first embodiment. FIG. 10 (a) shows a state in which the weakening magnetic flux control was performed when the limiter value iδ_lim4 of the δ-axis current command was performed, and FIG. Indicates the state. Comparing FIG. 10A and FIG. 10B, when the limiter value iδ_lim4 of the δ-axis current command is excessively large, the γ-axis current flows more than when the limiter value iδ_lim2 of the δ-axis current command is set. It can be seen that the efficiency is getting worse. It can be seen in FIG. 10 (a) that the current peak is −10 A, whereas in FIG. 10 (b), the current peak is reduced to −5 A, that is, the increase in copper loss is suppressed. Further, if the limiter value iδ_lim4 of the δ-axis current command is excessively large, it is possible that the motor drive device 2 may be protected against overcurrent.

Weak magnetic flux control will be described. The simplest method for weakening magnetic flux control is to determine the γ-axis current command based on the voltage equation. Eq. (7) is obtained by solving Eq. (6) with _{respect to the exciting current i γ.}

_{However, the weakened magnetic flux control that can obtain the exciting current i γ} represented by the equation (7) has a drawback that it is vulnerable to changes and variations in the motor constant, and is not widely used in the industrial world.

An integral type weakening magnetic flux control is used instead of the weakening magnetic flux control in which the _{exciting current i γ} represented by the equation (7) can be obtained. For example, a method of determining the exciting current command value i _γ ^{* by integrating and controlling the difference between the amplitude | v *} | of the voltage command vector and the limit value _{vom is known.} In this method, when the voltage command vector amplitude | v ^* | is _{larger than the limit value Vol, the} exciting current command value i _γ ^* is increased in the negative direction, and conversely, the voltage command vector amplitude | v ^* | is limited. If it is smaller than the value _Vol , reduce the _{exciting current command value i γ} ^*. As a general rule, a limiter is appropriately applied to the exciting current command value i _γ ^*. This is to prevent the motor 7 from being demagnetized due to an excessive excitation current command value i _γ ^*. _{Further, in order to prevent a positive exciting current i γ} from flowing in a region where the rotation speed of the electric motor 7 is low to medium speed, a limiter in the positive direction may be applied. The limiter value in the positive direction is generally zero or "current command value for maximum torque / current control".

The specific configuration and operation of the torque current command value generation unit 503 will be described. FIG. 11 is a first block diagram showing a configuration example of a torque current command value generation unit 503 included in the voltage command value calculation unit 115 according to the first embodiment. Note that FIG. 11 also includes the subtraction unit 502 in the previous stage. The torque current command value generation unit 503 includes a speed control unit 610, a vibration suppression control unit 620, and a limiting unit 630.

The speed control unit 610 generates the _{torque current command value i δ} ^* in the above-mentioned rotating coordinate system. Specifically, the speed control unit 610 includes a proportional control unit 611, an integration control unit 612, and an addition unit 613. The proportional control unit 611 performs proportional control on the difference (ω _e ^* -ω _{est ) between the frequency command value ω e} ^* and the frequency estimated value ω _{est acquired from the subtraction unit 502, and sets} the proportional term i ^{δ_p *} . Output. The integration control unit 612 performs integration control on the difference (ω _e ^* -ω _{est ) between the frequency command value ω e} ^* and the frequency estimation value ω _{est acquired from the subtraction unit 502, and sets} the integration term i ^{δ_i *} . Output. The addition unit 613 adds the proportional term i _{δ_p} ^* _{acquired from the proportional control unit 611 and the integral term i δ_i} ^* acquired from the integral control unit 612 to generate the torque current command value i _δ ^*. In the following description, the torque current command value i _δ ^* may be referred to as the first δ-axis current command value.

The vibration suppression control unit 620 includes an addition unit 621. The addition unit 621 adds the torque current command value i _δ ^* _{generated by the speed control unit 610 and the torque current compensation value i δ_trq} acquired from the compensation value calculation unit 504, and adds the torque current command value i _δ ^**. To generate.

The limiting unit 630 includes a storage unit 631, a selection unit 632, and a limiter 633. The storage unit 631 stores the limiter values iδ_lim1 and iδ_lim2. That is, the limiting unit 630 has limiter values iδ_lim1 and iδ_lim2. The selection unit 632 selects one of the limiter values iδ_lim1 and iδ_lim2 stored in the storage unit 631 and sets the limiter value iδ_lim. The limiter 633 outputs the torque current command value i _δ ^** generated by the vibration suppression control unit 620, which is limited by the limiter value i δ_lim, as the torque current command value i _δ ^*** . The limiting unit 630 may store the limiter values iδ_lim1 and iδ_lim2 calculated by itself in the storage unit 631 or may be acquired from the outside, for example, the operation control unit 102 and stored in the storage unit 631. You may memorize it. In the following description, the limiter value iδ_lim may be referred to as a limit value, the limiter value iδ_lim1 may be referred to as a first limit value, and the limiter value iδ_lim2 may be referred to as a second limit value.

FIG. 12 is a first flowchart showing the operation of the torque current command value generation unit 503 according to the first embodiment. In the torque current command value generation unit 503, the speed control unit 610 generates the torque current command value i _δ ^* from the difference (ω _e ^* -ω _{est ) between the frequency command value ω e} ^* and the frequency estimation value ω _{est (ω e * -ω est).} Step S1). The vibration suppression control unit 620 _{adds the torque current command value i δ} ^* and the torque current compensation value i _{δ_trq} to generate the torque current command value i _δ ^** (step S2). When the limiter value i δ_lim is _{smaller than the torque current command value i δ} ^** (step S3: No), the _{limiting unit 630 reduces the integral term i δ_i} ^* of the speed control unit 610 (step S4). Specifically, the limiting unit 630 instructs the speed control unit 610 to set _{"i δ_i} ^* = i _{δ_lim} ^{-i δ_p *".} Limiting section 630, if the limiter value iδ_lim is more ^** torque current command value i _[delta] (Step S3: Yes), the torque current command value i _[delta] ^***, outputs a torque current command value i _[delta] ^** ( Step S5).

The operation in which the limiting unit 630 selects either the limiter value iδ_lim1 or the limiter value iδ_lim2 as the limiter value iδ_lim will be described. FIG. 13 is a flowchart showing an operation in which the limiting unit 630 according to the first embodiment selects a limiter value. In the limiting unit 630, the selecting unit 632 selects the limiter value based on, for example, the modulation factor of the electric motor 7. The modulation factor is a value obtained by dividing the line voltage of each phase of the motor 7 by _{the peak voltage of the bus voltage V dc.} When the modulation factor exceeds 1 (step S11: Yes), the selection unit 632 selects the limiter value iδ_lim2 (step S12). When the modulation factor is 1 or less (step S11: No), the selection unit 632 selects the limiter value iδ_lim1 (step S13). The operation shown in FIG. 13 is an example, and the selection unit 632 may select the limiter value by another method. The selection unit 632 may select the limiter value iδ_lim2, for example, when the rotation speed of the electric motor 7, the load, etc. are large and the magnetic flux control is required. Further, the limiting unit 630 may compare the limiter value iδ_lim1 and the limiter value iδ_lim2 and select the smaller one.

Here, in the case of step S3: No in the flowchart shown in FIG. 12, the torque current command value generation unit 503 _operates a control called anti-windup control that reduces ^{the integration term i δ_i * of the speed control unit 610.} The operating state of the electric motor drive device 2 in this case is shown in FIG. FIG. 14 is a first diagram showing an operating state of the electric motor drive device 2 according to the first embodiment. FIG. 14 shows an operating state when the configuration of the torque current command value generation unit 503 of the electric motor drive device 2 is FIG. 11. From FIG. 14, it can be seen that the actual speed cannot follow the speed command value because _{the integral term i δ_i} ^* of the speed control unit 610 becomes small. In the configuration of the torque current command value generation unit 503 shown in FIG. 11, since the limiting unit 630 is located after the vibration suppression control unit 620, the vibration suppression control is prioritized and the speed control is not properly performed. If the speed control cannot be performed properly, the electric motor drive device 2 cannot produce the desired capacity, and there is a high possibility that the control will fail.

The specific configuration and operation of the torque current command value generation unit 503 when speed control is prioritized will be described. FIG. 15 is a second block diagram showing a configuration example of the torque current command value generation unit 503 included in the voltage command value calculation unit 115 according to the first embodiment. Note that FIG. 15 also includes the subtraction unit 502 in the previous stage. The torque current command value generation unit 503 includes a speed control unit 610, a limiting unit 630, and a vibration suppression control unit 640.

The limiting unit 630 outputs the torque current command value i _δ ^* generated by the speed control unit 610, which is limited by the limiter value ^{i δ_lim, as the torque current command value i δ_lim *} . That is, the limiting section 630, using the limiter value Aideruta_lim limits the torque current command value i? ^*, For generating a torque current command value iδ_lim ^*. In the following description, the torque current command value iδ_lim ^* may be referred to as the second δ-axis current command value. In the example of FIG. 15, the restriction unit 630 has a different target of restriction as compared with the example of FIG. 11, but the content of the operation is the same as the content of the operation in the case of the example of FIG.

The vibration suppression control unit 640 generates a _{torque current command value i δ} ^*** using the torque current command value i ^{δ_lim *} , the limiter value i δ_lim, and the torque current compensation value i _{δ_ trq.} Specifically, the vibration suppression control unit 640 includes a subtraction unit 641, a limiter 642, and an addition unit 643. The subtraction unit 641 calculates the difference between the limiter value iδ_lim acquired from the limit unit 630 and the torque current command value iδ_lim ^*, and calculates the limiter value iδ_trq_lim * with respect to the torque current compensation value. The limiter 642 outputs what is limited by the limiter value iδ_trq_lim with respect to the torque current compensation value i _{δ_trq as} ^{the torque current compensation value iδ_trq_lim *} after the limiter. The addition unit 643 adds the torque current command value iδ_lim ^* and the torque current compensation value iδ_trq_lim ^* after the limiter to generate the torque current command value I _δ ^***. In the following description, the torque current command value i _δ ^*** may be referred to as a third δ-axis current command value.

In the example shown in FIG. 15, the torque current command value generation unit 503 is provided with a limit unit 630 after the speed control unit 610, and the limiter value iδ_trq_lim with respect to the torque current compensation value is set to iδ_lim-iδ_lim ^* . As a result, the torque current command value generation unit 503 can secure the delta-axis current command for the amount that can follow the speed command, and use the surplus for the delta-axis current command for the vibration suppression control.

FIG. 16 is a second flowchart showing the operation of the torque current command value generation unit 503 according to the first embodiment. In the torque current command value generation unit 503, the speed control unit 610 generates the torque current command value i _δ ^* from the difference (ω _e ^* -ω _{est ) between the frequency command value ω e} ^* and the frequency estimation value ω _{est (ω e * -ω est).} Step S21). When the limiter value i δ_lim is _{smaller than the torque current command value i δ} ^* (step S22: No), the _{limiting unit 630 reduces the integration term i δ_i} ^* of the speed control unit 610 (step S23). Specifically, the limiting unit 630 instructs the speed control unit 610 to set _{"i δ_i} ^* = i _{δ_lim} ^{-i δ_p *".} Limiting section 630, if the limiter value Aideruta_lim is more ^* torque current command value i _[delta] (step S22: Yes), the torque current command value after limiter Aideruta_lim ^*, outputs a torque current command value i _[delta] ^* (step S24 ).

The vibration suppression control unit 640 calculates the value obtained ^{by subtracting the torque current command value iδ_lim *} from the limiter value iδ_lim as the limiter value iδ_trq_lim with respect to the torque current compensation value i _{δ_trq (step S25).} When the limiter value iδ_trq_lim is _{equal to} or greater than the torque current compensation value i δ_trq (step S26: Yes), the ^{vibration suppression control unit 640 sets the torque current compensation value iδ_trq_lim *} after the limiter as the torque current compensation value i _{δ_trq} (step S27). When the limiter value iδ_trq_lim is _{less than the torque current compensation value i δ_trq} (step S26: No), the ^{vibration suppression control unit 640 sets the torque current compensation value iδ_trq_lim *} after the limiter as the limiter value iδ_trq_lim (step S28). The vibration suppression control unit 640 adds the torque current command value iδ_lim ^* and the torque current compensation value iδ_trq_lim ^* after the limiter to generate the torque current command value i _δ ^*** (step S29).

The operation of the selection unit 632 in the limiting unit 630 to select either the limiter value iδ_lim1 or the limiter value iδ_lim2 as the limiter value iδ_lim is the same as described above.

FIG. 17 is a second diagram showing an operating state of the electric motor drive device 2 according to the first embodiment. FIG. 17 shows an operating state when the configuration of the torque current command value generation unit 503 of the electric motor drive device 2 is FIG. 15. Unlike the case of FIG. 14, it can be seen from FIG. 17 that the actual speed can follow the speed command value.

In the motor drive device 2, the control device 100 generates the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^* _{using the exciting current command value i γ} ^* and the torque current command value i _δ ^***. Further, the inverter 30 is controlled by converting the three-phase voltage command values V _u ^* , V _v ^* , and V _w ^* and then generating PWM signals Sm1 to Sm6. In this way, the electric motor drive device 2 follows the speed command value and suppresses step-out by setting the torque current command value generation unit 503 to the configuration shown in FIG. 15 and providing a limiter value for the δ-axis current. At the same time, it is possible to perform efficient vibration suppression control.

In the present embodiment, the electric motor drive device 2 is configured to restore the phase currents i _u , _iv , i _{w from the direct current I dc} on the input side of the inverter 30, but is not limited to this. The electric motor drive device 2 may detect the phase current by providing a current detector on the output lines 331, 332, 333 of the inverter 30. In this case, the electric motor drive device 2 may use the current value detected by the current detector instead of the current restored by the current restoration unit 111.

In the electric motor drive device 2, the switching elements 311 to 316 of the inverter main circuit 310 are assumed to be IGBTs (Insulated Gate Bipolar Transistors), MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors), etc., but switching is performed. Any element that can be used may be used. In the motor drive device 2, when the switching elements 311 to 316 are MOSFETs, the MOSFET has a parasitic diode due to its structure, so that the same effect can be obtained without connecting the rectifying elements 321 to 326 for circulation in antiparallel. Can be done.

The materials constituting the switching elements 311 to 316 are made of not only silicon (Si) but also silicon carbide (SiC), gallium nitride (GaN), diamond, etc., which are wide bandgap semiconductors. It is possible to reduce the loss.

Next, the hardware configuration of the control device 100 included in the electric motor drive device 2 will be described. FIG. 18 is a diagram showing an example of a hardware configuration that realizes the control device 100 included in the electric motor drive device 2 according to the first embodiment. The control device 100 is realized by the processor 201 and the memory 202.

The processor 201 is a CPU (Central Processing Unit, central processing unit, processing unit, arithmetic unit, microprocessor, microcomputer, processor, DSP (Digital Signal Processor)), or system LSI (Large Scale Integration). The memory 202 is non-volatile or volatile such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), and EEPROM (registered trademark) (Electrically Erasable Programmable Read-Only Memory). The semiconductor memory of the above can be illustrated. The memory 202 is not limited to these, and may be a magnetic disk, an optical disk, a compact disk, a mini disk, or a DVD (Digital Versatile Disc).

As described above, according to the present embodiment, in the electric motor drive device 2, the control device 100 determines the output torque of the electric motor 7 when the electric motor 7 drives a load element in which _{the load torque T l fluctuates periodically.} The _{inverter 30 is controlled so that T m} follows a periodic fluctuation of the load torque T _l , that is, a pulsation. The control device 100 appropriately sets the limiter value and _{generates the torque current command value i δ} ^*** according to the effectiveness of the weakening magnetic flux control, thereby realizing low vibration in the entire operating range of the motor 7. However, highly efficient operation can be performed while suppressing the occurrence of overcurrent and step-out.

Embodiment 2.
FIG. 19 is a diagram showing a configuration example of the refrigeration cycle application device 900 according to the second embodiment. The refrigeration cycle application device 900 according to the second embodiment includes the electric motor drive device 2 described in the first embodiment. The refrigeration cycle application device 900 according to the second embodiment can be applied to products including a refrigeration cycle such as an air conditioner, a refrigerator, a freezer, and a heat pump water heater. In FIG. 19, the components having the same functions as those of the first embodiment are designated by the same reference numerals as those of the first embodiment.

In the refrigeration cycle application device 900, the compressor 901 incorporating the electric motor 7 in the first embodiment, the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, and the outdoor heat exchanger 910 form a refrigerant pipe 912. It is attached via.

Inside the compressor 901, a compression mechanism 904 for compressing the refrigerant and an electric motor 7 for operating the compression mechanism 904 are provided.

The refrigeration cycle applicable device 900 can perform a heating operation or a cooling operation by switching the four-way valve 902. The compression mechanism 904 is driven by an electric motor 7 that is controlled at a variable speed.

During the heating operation, as shown by the solid arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, and passes through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910 and the four-way valve 902. Return to the compression mechanism 904.

During cooling operation, as shown by the broken arrow arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, and passes through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906, and the four-way valve 902. Return to the compression mechanism 904.

During the heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During the cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 depressurizes the refrigerant and expands it.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is one of the configurations without departing from the gist of the present invention. It is also possible to omit or change the part.

1 AC power supply, 2 electric motor drive, 4 reactor, 7 electric motor, 10 rectifier circuit, 20 smoothing condenser, 22a, 22b DC bus, 30 inverter, 82 bus voltage detector, 84 bus current detector, 100 controller, 102 operation Control unit, 110 inverter control unit, 111 current restoration unit, 112 3-phase 2-phase conversion unit, 113 excitation current command value generation unit, 115 voltage command value calculation unit, 116 electrical phase calculation unit, 117 2-phase 3-phase conversion unit, 118 PWM signal generator, 310 inverter main circuit, 311-316 switching element, 321-326 rectifier element, 331-333 output line, 350 drive circuit, 501 frequency estimation unit, 502,509,510,557,558,641 subtraction Unit, 503 torque current command value generation unit, 504 compensation value calculation unit, 511 exciting current control unit, 512 torque current control unit, 550 calculation unit, 551 cosine calculation unit, 552 sine calculation unit, 533, 554, 561, 562 multiplication Unit, 555,556 low pass filter, 559,560 frequency control unit, 563,613,621,643 addition unit, 610 speed control unit, 611 proportional control unit, 612 integration control unit, 620,640 vibration suppression control unit, 630 limit Unit, 631 storage unit, 632 selection unit, 633,642 limiter, 900 refrigeration cycle applicable equipment, 901 compressor, 902 four-way valve, 904 compression mechanism, 906 indoor heat exchanger, 908 expansion valve, 910 outdoor heat exchanger, 912 Rectifier piping, D1 to D4 diodes.

Claims (3)

1. An inverter that supplies an AC voltage with variable frequency and voltage value to an electric motor that drives a load element whose load torque fluctuates periodically.
   A control device that controls the inverter and
   With
   The control device is
   A γ-axis current command value generator that generates a γ-axis current command value in a rotating coordinate system having a γ-axis and a δ-axis,
   A speed control unit that generates a first delta-axis current command value in the rotating coordinate system,
   A limiting unit that limits the first δ-axis current command value using the limit value and generates a second δ-axis current command value,
   A compensation value calculation unit that generates a delta-axis current compensation value so that the output torque of the electric motor follows the periodic fluctuation of the load torque.
   A vibration suppression control unit that generates a third δ-axis current command value using the second δ-axis current command value, the limit value, and the δ-axis current compensation value.
   An electric motor drive device that controls the inverter by using the γ-axis current command value and the third δ-axis current command value.
2. The restriction part is
   The current limit value for the phase current of the motor and the first limit value defined by the γ-axis current,
   Defined from the voltage limit value based on the voltage that the inverter can output to the electric motor, the rotation speed of the electric motor, the magnetic flux chain intersection of the electric motor, the γ-axis inductance of the rotating coordinate system, and the δ-axis inductance of the rotating coordinate system. The second limit to be done and
   Have,
   When the value obtained by dividing the line voltage of the motor by the peak voltage of the bus voltage is used as the modulation factor, the first limit value is selected as the limit value when the modulation factor is smaller than 1, and the modulation factor is selected. The electric motor drive device according to claim 1, wherein when is 1 or more, the second limit value is selected.
3. A refrigeration cycle applicable device including the electric motor drive device according to claim 1 or 2.

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