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WO2019001217A1 - 一种有源钳位反激式开关电源电路 - Google Patents

一种有源钳位反激式开关电源电路 Download PDF

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Publication number
WO2019001217A1
WO2019001217A1 PCT/CN2018/089577 CN2018089577W WO2019001217A1 WO 2019001217 A1 WO2019001217 A1 WO 2019001217A1 CN 2018089577 W CN2018089577 W CN 2018089577W WO 2019001217 A1 WO2019001217 A1 WO 2019001217A1
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WO
WIPO (PCT)
Prior art keywords
primary winding
power supply
capacitor
channel fet
clamp
Prior art date
Application number
PCT/CN2018/089577
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English (en)
French (fr)
Inventor
王保均
Original Assignee
广州金升阳科技有限公司
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Publication of WO2019001217A1 publication Critical patent/WO2019001217A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/01—Resonant DC/DC converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the field of switching power supplies, and more particularly to flyback switching power supplies that use active clamping.
  • the rectifier bridge 101, the filter circuit 200, and the basic flyback topology unit circuit 300 are also referred to as a main power stage.
  • the practical circuit is further provided with a varistor, an NTC thermistor, and an EMI (Electromagnetic Interference) in front of the rectifier bridge. ) Protect the circuit to ensure that the electromagnetic compatibility of the flyback power supply meets the requirements for use. Under normal circumstances, the flyback switching power supply requires that the leakage inductance between the primary and secondary windings is as small as possible, so that the conversion efficiency is high, and the withstand voltage of the primary side main power switch V is also reduced, for using the RCD network as a go Magnetic flyback converters, the loss of the RCD network is also reduced.
  • RCD absorption refers to the absorption circuit composed of resistors, capacitors and diodes.
  • the literature in China is the same as the international one.
  • the letter R is used to give the resistance number and represents the resistance.
  • the letter C is used to number the capacitor and represent the capacitor.
  • the letter D is used to give the diode. Numbered and represents the diode, the resistor and capacitor are connected in parallel, and then connected in series with the diode to form an RCD network.
  • the rectifier bridge 101 is generally composed of four diodes. When there is no rectifier bridge 101, 200, 300 can constitute a DC/DC switching power supply or converter. Because it is DC power supply, there is no power factor requirement, and the power can be more than 75W. . In fact, the use of flyback topology in low-voltage DC/DC switching power supplies is not the mainstream. This is because the input current of the flyback power supply is discontinuous and the ripple is large at low voltage, which requires higher requirements for the power supply equipment of the former stage.
  • Low-voltage DC/DC switching power supplies generally refer to input voltages below 48V. Some low-voltage DC/DC switching power supplies can operate up to 160V DC, such as railway power supplies.
  • the inductance of the primary winding is also low. It is often found that the calculated number of turns cannot be tiled to the left to the right of the slot of the full frame.
  • the sandwich series winding method can be used at a low operating voltage.
  • it is forced to adopt the sandwich parallel winding method. Since the two primary windings are not in the same layer, there is a leakage inductance between the two primary windings, and the leakage inductance will cause loss, thereby making the switching power supply less efficient.
  • Loss caused by leakage inductance between two parallel primary windings this will exist during excitation and demagnetization; if the third winding is used for demagnetization, it is not good to choose the third winding and the two parallel Whoever winding in the side winding can only use two third windings, which are respectively wound with two parallel primary windings, and then connected in parallel to form a "third winding". The process is complicated, and the two windings are connected in parallel. The presence of windings also induces unequal voltages, causing losses and large electromagnetic interference.
  • the advantage is non-destructive demagnetization, the efficiency is higher, but the choice of the wire diameter of the third winding is also a problem: the selection is relatively thin, and the winding of the primary winding is more troublesome, easy to put The thin wire is broken; if the same wire diameter is selected as the primary winding, the cost is high.
  • the third winding demagnetization flyback converter is also referred to as a "three-winding absorption flyback converter".
  • the application numbers are: 201710142832.0, 201710142797.2, the two names are "a flyback switching power supply", respectively, showing the technical solutions of Figure 1-2, Figure 1-3, solving the above problem, namely
  • the primary winding can be used without two separate parallel connections, which can allow the leakage between the primary and secondary windings to be large, without using the third winding for demagnetization, while the conversion efficiency is not reduced, and the loss during excitation and demagnetization. reduce.
  • the demagnetization mode of Figure 1-2 requires strict leakage inductance. Otherwise, the energy of the excitation may be directly returned from the DC power source U DC by D1 instead of the secondary winding N S.
  • the demagnetization circuit itself is a more classic topology, the duty cycle can be greater than 0.5, and the leakage inductance energy is not recycled.
  • the inventors defined the topology used in the flyback switching power supply with the Chinese application numbers 201710142832.0 and 201710142797.2, including the forward topology, and the basic topology excluding the demagnetization mode is defined as: LCL converter, source Its two primary side magnetizing inductances and a capacitor in series with them.
  • the present invention is to solve the above-mentioned shortcomings of the existing low-voltage flyback switching power supply, and to provide an active clamped flyback switching power supply circuit, the duty ratio can be greater than 0.5, and the demagnetization circuit is realized at the same time. Energy recovery, further, realizes zero voltage switching of the main power switch tube, further reducing losses and improving conversion efficiency.
  • an active clamped flyback switching power supply circuit including a transformer, a first N-channel FET, a first capacitor, a second capacitor, a first diode, and a clamp a bit network
  • the transformer comprises a first primary winding, a second primary winding and a secondary winding
  • the clamping network comprises at least an anode and a cathode
  • the secondary winding is connected to the first diode anode, the first diode
  • the cathode is connected to one end of the second capacitor and forms an output positive, the same-side end of the secondary winding is connected with the other end of the second capacitor, and the output is negative
  • the positive terminal of the input DC power supply is simultaneously the same name as the first primary winding
  • the clamp network The cathode of the first primary winding is connected to the drain of the first N-channel field effect transistor; the anode of the clamp network is connected to the opposite end of the second primary winding, and the source of the first N-channel FET
  • One end of the third capacitor is the cathode of the clamp network, the other end of the third capacitor is connected to the drain of the second N-channel FET, and the source of the second N-channel FET is the anode of the clamp network, a gate connection clamp control signal of the two N-channel FET;
  • one end of the third capacitor is the anode of the clamp network, the other end of the third capacitor is connected to the source of the second N-channel FET, and the drain of the second N-channel FET is the cathode of the clamp network, The gate of the two N-channel FET is connected to the clamp control signal.
  • the active clamp flyback switching power supply circuit works by: the first N-channel FET is turned on every cycle, and the second N-channel FET is turned on when the active clamp flyback switching power supply is near full load. Each cycle is open;
  • the first N-channel FET is turned on every cycle, and the second N-channel FET is turned on every m cycles when the active clamp flyback switching power supply is lightly loaded, m is a positive integer, and the active clamp The lighter the load of the flyback switching power supply, the larger the value of m.
  • a flyback switching power supply circuit includes a transformer, a first N-channel FET, a first capacitor, and a second a capacitor, a first diode, a clamp network, the transformer includes a first primary winding, a second primary winding, and a secondary winding, the clamp network includes at least an anode and a cathode, and the secondary winding has a different name and a first
  • the anode of the pole tube is connected, the cathode of the first diode is connected to one end of the second capacitor, and the output is positive, the end of the secondary winding is connected with the other end of the second capacitor, and the output is negative; the positive terminal of the input DC power is simultaneously
  • the drain of the N-channel FET and the second-side winding are connected at different ends, and the source of the first N-channel FET is connected to the same-name end of the first primary winding; the second-side winding
  • One end of the third capacitor is the cathode of the clamp network, the other end of the third capacitor is connected to the drain of the second N-channel FET, and the source of the second N-channel FET is the anode of the clamp network, a gate connection clamp control signal of the two N-channel FET;
  • one end of the third capacitor is the anode of the clamp network, the other end of the third capacitor is connected to the source of the second N-channel FET, and the drain of the second N-channel FET is the cathode of the clamp network, The gate of the two N-channel FET is connected to the clamp control signal.
  • the active clamp flyback switching power supply circuit works by: the first N-channel FET is turned on every cycle, and the second N-channel FET is turned on when the active clamp flyback switching power supply is near full load. Each cycle is open;
  • the first N-channel FET is turned on every cycle, and the second N-channel FET is turned on every m cycles when the active clamp flyback switching power supply is lightly loaded, m is a positive integer, and the active clamp The lighter the load of the flyback switching power supply, the larger the value of m.
  • the second N-channel field effect transistor can be at an integer m and a value larger than the integer when the active clamp flyback switching power supply is lightly loaded (m) +1) change back and forth.
  • a low-voltage drop and fast recovery in the same direction as the body diode of the second N-channel FET are connected in parallel between the drain and the source of the second N-channel FET. Diode.
  • the first primary winding and the second primary winding have the same wire diameter.
  • the physical path of the excitation current of the first primary winding and the second primary winding is reversed in the PCB layout.
  • the invention has the beneficial effects that the duty ratio can be greater than 0.5, and at the same time, the energy recovery of the demagnetization circuit is realized, and further, the zero voltage turn-on of the main power switch tube is realized, further reducing the loss and improving the conversion efficiency, especially at light load. When the conversion efficiency is improved.
  • Figure 1-1 is a schematic diagram of a conventional flyback switching power supply for AC to DC;
  • Figure 1-2 is a schematic diagram of the disclosed technical solution of the Chinese application number 201710142832.0;
  • Figure 1-3 is a schematic diagram of the disclosed technical solution of the Chinese application number 201710142797.2;
  • 2-1 is a schematic diagram of a first embodiment of the present invention, and the clamp network adopts (1) mode;
  • 2-2 is a second schematic diagram of the first embodiment of the present invention, and the clamp network adopts (2) mode;
  • 2-3 is a schematic diagram of generating two excitation currents 41, 42 when Q1 is saturated in the first embodiment
  • 2-4 is a schematic diagram of the Q1 cutoff in the first embodiment, generating a freewheeling current 43 and a demagnetizing current 44;
  • 3-1 is a schematic diagram of a second embodiment of the present invention, and the clamp network adopts (1) mode;
  • 3-2 is a second schematic diagram of a second embodiment of the present invention, and the clamp network adopts the (2) mode.
  • 2-1 and 2-2 illustrate a schematic diagram of an active clamp flyback switching power supply circuit according to a first embodiment of the present invention, including a transformer B, a first N-channel FET Q1, and a first capacitor.
  • C1, a second capacitor C2, a first diode D2, a clamp network 400, and a transformer B includes a first primary winding N P1 , a second primary winding N P2 and a secondary winding N S
  • the clamp network 400 includes at least An anode and a cathode, the secondary winding N S different end is connected to the first diode D2 anode, the first diode D2 cathode is connected to one end of the second capacitor C2, and forms an output positive, which is the + end of Vout in the figure,
  • the secondary winding N S has the same name end connected to the other end of the second capacitor C2, and forms an output negative, which is the end of Vout in the figure;
  • the input DC power supply U DC hereinafter also referred to as DC power
  • One end of the third capacitor C3 is the cathode of the clamp network 400, the other end of the third capacitor C3 is connected to the drain d of the second N-channel FET Q2, and the source s of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in FIG. 2-1;
  • One end of the third capacitor C3 is the anode of the clamp network 400, the other end of the third capacitor C3 is connected to the source s of the second N-channel FET Q2, and the drain d of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in Figure 2-2.
  • the anode of the clamp network 400, the cathode, and the body diode of the second N-channel FET Q2 therein are corresponding.
  • the anode of the body diode of Q2 is the anode of 400.
  • the cathode of Q2's body diode passes through C3 and is the cathode of 400.
  • the cathode of Q2's body diode is 400 cathode
  • the anode of Q2's body diode passes through C3 and is 400 anode.
  • Heterogeneous end one end of the winding in the figure where there is no black mark
  • Driving control signal including various pulse waves such as PWM pulse width modulation signal and PFM pulse frequency modulation;
  • Clamp control signal includes various square waves such as PWM pulse width modulation signal and PFM pulse frequency modulation, but appears differently from the drive control signal;
  • Transformer B the first primary winding N P1 and the second primary winding N P2 are in the figure, the cores are connected by a broken line, indicating that they are wound around a transformer and share the same core, not a separate transformer, just for The graphics are clear and the connection relationship is simple, and the drawing method in the figure is used.
  • the source of the N-channel FET Q1 is connected to the same end of the second primary winding N P2 , and the connection point is simultaneously connected to the negative terminal of the input DC power supply U DC - that is, the FET
  • the source of Q1 is connected to the negative terminal of the input DC power supply U DC - which does not exist directly in practical applications. This is because in the field of switching power supply, the analysis of the working principle of the basic topology will omit unnecessary factors.
  • the source of the FET is connected to a current sense resistor or a current transformer to detect the average current or peak current to implement various control strategies.
  • the current sense resistor or current transformer is connected to the source.
  • the current transformer can appear anywhere in the excitation circuit, such as the drain of a FET, such as the same or different end of the first primary winding, and the current transformer has a conventional primary side. It is also a Hall sensor that is a "wire" and a magnetic core transformer whose secondary side is a multi-turn coil.
  • the second N-channel FET Q2 does not work.
  • Q1 does not work because it does not receive the drive control signal. It is equivalent to an open circuit, then the power supply U DC passes through the first original.
  • the side winding N P1 charges C1, and the current is simultaneously returned to the negative end of the power source U DC through the second primary winding N P2 , and the charging current of the first primary winding N P1 is: flowing from the same name end to the different name end;
  • the charging current of the primary winding N P2 is: flowing from the different name end to the same name end;
  • N P1 and N P2 are two lines and winding, the two currents are equal in magnitude, and the generated magnetic flux is opposite, completely canceled, that is, at the time of power-on
  • the power supply U DC charges C1 through the two windings of the transformer B.
  • C1 is equivalent to the DC internal resistance through N P1 and N P2 in parallel with the power supply U DC , and C1 still starts. To the power supply filtering, decoupling; over time, the terminal voltage of C1 is equal to the voltage of U DC , left and right negative.
  • Q1 When Q1 receives the control signal normally, taking one cycle as an example, when the gate of Q1 is high, Q1 is saturated and its internal resistance is equal to the on-state internal resistance R ds(ON) . For the convenience of analysis, this is the case. As a straight-through, it is a wire. As shown in Figure 2-3, Q2 is in the off state and does not participate in the work. In the figure, 400 is drawn as an open state; at this time, two excitation currents are generated, 41 in Figure 2-3. And 42;
  • the excitation currents of 41 and 42 are in parallel. Since the inductances of N P1 and N P2 are the same, the excitation voltages are the same, and they are equal to U DC , 41 and 42 are completely equal.
  • the secondary winding N S is pressed.
  • the induced voltage is the same.
  • the induced voltage is: a positive voltage is induced at the same name, and a negative voltage is induced at the opposite end.
  • the magnitude is equal to U DC multiplied by the turns ratio n, that is, N S induces a positive and negative voltage.
  • N S induces a positive and negative voltage.
  • the currents of 41 and 42 increase linearly upward; the current direction flows from the same name end to the different name end in the inductance;
  • the gate of Q1 changes from high level to low level, Q1 also turns from saturation conduction to off. Since the current in the inductor cannot be abrupt, even though Q1 is off at this time, the currents of 41 and 42 still flow from the same name end. At the opposite end, since the current loop of the primary side has been cut, the energy in the core flows from the same name to the opposite end on the secondary side. Referring to Figure 2-4, the secondary winding N S appears to flow from the same name end to the different name end.
  • the initial magnitude of the current (the sum of 41 and 42 at the instant of Q1 turn-off) / ⁇ ratio n, which causes D2 to conduct a forward conduction, and through the D2 of the forward conduction, Capacitor C2 is charged and Vout establishes voltage or continuously outputs energy. This process is also the process of demagnetization.
  • the output of the flyback switching power supply is named after the primary winding is disconnected from the power supply.
  • the transformer B is not the function of the voltage conversion, but is the isolated version of the Buck-Boost converter. Therefore, transformer B is often referred to as a flyback transformer;
  • the circuit for demagnetizing the leakage inductance of the present invention is composed of a clamp network 400 composed of Q2 and C3 and a second primary winding N P2 , and the working principle is:
  • the first primary winding N P1 and the second primary winding N P2 are double-wired and wound, and the leakage inductance between the two windings is zero.
  • the electric energy of the leakage inductance in the second primary winding N P2 is in the same direction as the direction of the excitation, flowing from the same end to the opposite end, that is, in FIG. 2-4, flowing from bottom to top, opening the body of Q2 a diode, a current flowing from the source s of Q2 to the drain d, and this electrical energy is charged to C3 to form a leakage inductance demagnetization current as indicated by 44;
  • the leakage energy of the first primary winding N P1 is coupled to the second primary winding N P2 without leakage inductance, and is demagnetized by the body diode of Q2, and the leakage inductance demagnetization current shown by 44 is also formed. ;
  • the output voltage Vout is divided by the turns ratio n, which is the "reflected voltage” formed on the primary side when the secondary winding N S is turned on at D2. Since there is C3 blocking, the reflected voltage is greater than the value of the DC power source U DC , the circuit It can also work normally.
  • C2 is equivalent to a voltage source, which is "excited” to the secondary winding N S and the "reflected voltage” formed by the primary side.
  • the primary winding is equivalent to a voltage source having a voltage equal to the reflected voltage. In series with the leakage inductance, it is felt that the current in D2 drops to zero, and the primary winding is restored to the series connection of the magnetizing inductance and the leakage inductance.
  • the working method of the flyback switching power supply of the present invention is defined as: the first N-channel FET Q1 is turned on every period, and the second N-channel FET Q2 is in a cycle when the flyback switching power supply is nearly full.
  • Turn-on that is, Q2 and body diodes are synchronously turned on or lag-on.
  • the primary windings N P2 and N P1 are in a state in which the voltage source and the leakage inductance are connected in series. At this time, the leakage inductance and C3 resonate.
  • the terminal voltage of C3 will be close to zero or equal to zero at a certain time.
  • the primary windings N P2 and N P1 are double wound and wound, the leakage between the two is felt.
  • zero that is, the terminal voltages of the primary windings N P2 and N P1 remain equal at any time.
  • the terminal voltage of C3 is close to or equal to zero, Q2 is in a saturated conduction state, and its terminal voltage is also zero, then the clamp
  • the terminal voltage of the bit network 400 is close to or equal to zero, that is, the terminal voltage of the primary winding N P2 is equal to the voltage of the DC power source U DC , and is up and down negative, then the primary winding N P1 is lower positive and negative, the end of Q2 voltage at this point is twice the voltage U DC; and when the terminal voltage of the resonance C3
  • the voltage of the source electrode i.e., Q2 is s -U DC, since the terminal voltage of C1 is negative to the left of the right has been positive, and equal to U DC, at the moment, C1 left
  • ZVS Zero Voltage Switch
  • Quasi-ZVS also known as soft-switching technology, realizes the recycling of primary leakage energy.
  • the energy on the output capacitor of Q1 is also transferred due to resonance, realizing the recycling of primary leakage energy.
  • the capacity of the C3 is relatively large in order to achieve a long resonance time. Because C3 is large and the terminal voltage can be increased, the voltage of the terminal voltage and the voltage of the DC power supply U DC are in series. Using the volt-second balance law, the duty ratio can be greater than 0.5, and all can work normally.
  • the terminal voltage of C3 will be close to or equal to twice the U at a certain time.
  • the DC voltage is up and down, since the terminal voltage of C1 is always left positive and negative and equal to U DC , at this moment, the left terminal voltage of C1 is zero volt, that is, the terminal voltage of Q1 is also zero volt. If Q1 is saturated and turned on at this time, then zero voltage turn-on of Q1 is realized, and this mode must be the current interrupt mode, and the time of Q1 re-conduction is extremely easy to be detected.
  • the principle is: when the switching power supply is lightly loaded, the excitation current of the primary side controlled by the loop is small, and the leakage inductance is stored. The energy in the middle is relatively small, and the capacity of C3 is relatively large. Thus, when Q1 is turned off, Q2 is not turned on, and the leakage inductance energy is small, and the leakage inductance energy is: 0.5 ⁇ leakage inductance amount ⁇ square of excitation current; After the leakage inductance energy is charged to C3 through the body diode of Q2, if the energy absorbed by the clamped capacitor C3 reaches the threshold value, the next cycle Q2 is to be turned on, which can be preset by the load condition, that is, The size of the duty ratio is set in advance.
  • C3 does not reach the threshold, the circuit repeats the last cycle of action. Since the leakage inductance energy stored in C3 in the previous cycle is not released, the C3 energy increases again on the basis of the previous cycle, and the voltage across C3 rises again to a stable value. After several cycles, the voltage across C3 will increase in a step-like manner until the threshold is reached, that is, Q2 will be turned on in the next cycle to achieve resonance and zero voltage turn-on. Thus, the drive loss is further reduced. reduce.
  • m is a positive integer
  • this mode of operation is also allowed: the second N-channel FET Q2 can be back and forth between an integer m and a value larger than the integer (m+1) when the flyback switching power supply is lightly loaded. Change to better balance the design.
  • the notebook power adapter its working frequency is mostly 65KHz
  • the well-known theory believes that the loop response of its switching power supply can generally achieve one tenth of it, namely 6.5KHz
  • the principle of this part can refer to Dr. Zhang Xingzhu's paper "Switching Power Supply”
  • loop response wants to be high, affected by optocoupler delay in switching power supply, etc., it is difficult to improve, that is, in the switching power supply, the duty cycle of the main power conversion circuit is in ten working cycles.
  • the first cycle and the tenth cycle will produce significant changes. In other words, the duty cycle of the main power conversion circuit cannot be abruptly changed in adjacent cycles.
  • the wire diameters of the first primary winding and the second primary winding are the same, so that the winding is convenient, the wire diameters described herein are the same, and they are all of the same size Litz wire, the color can be Different, that is, multi-strand stranding, for the convenience of identification, the same specification wire including the Litz wire can have different colors. As the operating frequency increases, the high frequency current tends to flow on the surface of the enameled wire. In this case, the Litz wire can solve this problem.
  • the present invention has many differences, mainly: the duty ratio can be greater than 0.5, and at the same time, the energy recovery of the demagnetization circuit is realized, and further, the zero voltage turn-on of the main power switch tube is realized. Further reducing the loss and improving the conversion efficiency, especially at light loads, the conversion efficiency is improved.
  • an active clamp flyback switching power supply circuit includes a transformer B and a first N-channel.
  • the secondary winding N S has the same name end connected to the other end of the second capacitor C2, and forms an output negative, which is the end of Vout in the figure; the positive terminal of the input DC power supply U DC + simultaneously with the N groove
  • the drain of the MOSFET Q1 and the second primary winding N P2 are connected to each other.
  • the source of the N-channel FET Q1 is connected to the same end of the first primary winding N P1 ; the second primary winding N P2 is of the same name.
  • the clamp network 400 is connected to the cathode, the anode connected to a first primary winding N P1 dotted end of the clamp network 400, while the connection point An input connected to the negative terminal of the DC power source U DC; connection control signal gate N-channel MOSFET Q1; N Pl a first primary winding and a second primary winding of bifilar N P2, the end of the first capacitor C1 Connected to the same end of the first primary winding N P1 , the other end of the first capacitor C1 is connected to the same end of the second primary winding N P2 , and the clamp network 400 includes at least a third capacitor C3 and a second N-channel FET Q2 The third capacitor C3 and the second N-channel FET Q2 are connected in series, and the series connection is one of the following two ways:
  • One end of the third capacitor C3 is the cathode of the clamp network 400, the other end of the third capacitor C3 is connected to the drain d of the second N-channel FET Q2, and the source s of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in FIG. 3-1;
  • One end of the third capacitor C3 is the anode of the clamp network 400, the other end of the third capacitor C3 is connected to the source s of the second N-channel FET Q2, and the drain d of the second N-channel FET Q2
  • the gate g of the second N-channel FET Q2 is connected to the clamp control signal, as shown in Figure 3-2.
  • the working method of the flyback switching power supply is as follows: the first N-channel FET Q1 is turned on every period, and the second N-channel FET Q2 is turned on every cycle when the flyback switching power supply is nearly full load;
  • the first N-channel FET Q1 is turned on every cycle, and the second N-channel FET Q2 is turned on every m cycles when the flyback switching power supply is lightly loaded, m is a positive integer, and the flyback switching power supply The lighter the load, the larger the value of m.
  • the second embodiment is a modification of the first embodiment: on the basis of FIG. 2-1 of the first embodiment, the series devices of the excitation circuit are interchanged, that is, the positions of N P1 and Q1 are interchanged, and the clamps are simultaneously
  • the bit network 400 and N P2 are interchanged, and C1 is still connected between the connection points of the two series devices, and the circuit of FIG. 3-1 of the second embodiment is obtained. Since the source voltage of Q1 is fluctuating, this The circuit is floating drive, but the direct drive of the second N-channel FET Q2 for clamping is obtained, and the driving cost is low.
  • the C3 in the clamp network 400 is Q2 swaps the position to get the circuit of Figure 3-2.
  • the terminal voltage of C1 is equal to the voltage of U DC , right and left negative;
  • the first way is: the positive end of the power supply U DC enters through the drain of Q1, the source of Q1 is out, and then enters through the same name of the first primary winding N P1 , and the different name of N P1 is output, returning to the power supply U DC Negative end
  • the second way is: the right positive end of the capacitor C1 passes through the same name of the second primary winding N P2 , the different name of N P2 is output, the drain of Q1 enters, the source of Q1 is out, and the left negative end of capacitor C1 is returned. ;
  • the first and second excitation currents are in parallel relationship. Since the inductances of N P1 and N P2 are the same and the excitation voltages are the same, they are equal to U DC , and the two paths are completely equal.
  • the secondary winding N S The induced voltage is also generated according to the ⁇ ratio, the positive voltage is induced by the same name end, and the negative voltage is induced by the different name.
  • the size is equal to U DC multiplied by the ⁇ ratio n, that is, N S induces a positive and negative voltage, this voltage and C2
  • the terminal voltage is connected in series, and is applied to both ends of D2, and D2 is reverse biased and not turned on. At this time, the secondary side is equivalent to no load, and no output;
  • the first and second excitation currents increase linearly upward; the current direction flows from the same name end to the different name end in the inductance;
  • the circuit for demagnetizing the leakage inductance is composed of the clamp network 400 and the second primary winding N P2 , and the operation principle is the same as that of the first embodiment.
  • the second embodiment is a modification of the first embodiment, and the working principle is equivalent, and the object of the invention is also achieved.
  • an N-channel FET it can also be realized by a P-channel field effect transistor.
  • the P-channel FET has a low cost at a low operating voltage, and at this time, the basis of the first embodiment described above.
  • the polarity of the power source, the diode, and the end of the same name are reversed, and the output rectification portion is not reversed.
  • the third and fourth embodiments are obtained. Since it is a common change in the technical field, it should be regarded as the protection scope of the claims.

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Abstract

一种有源钳位反激式开关电源电路，在LCL反激变换器基础上，保持变压器B中的N P1同名端接电源，第二原边绕组N P2同名端接地，N P1和N P2为双线并绕，电容C1的一端与N P1异名端相连，另一端与N P2异名端相连，N P2异名端通过场效应管Q2与C3串联的钳位网络400接电源，这样实现了：当Q1饱和导通时，N P1和N P2都激磁，当Q1关断时，副边N S输出能量，原边呈电压源与漏感串联，Q2导通，C3与漏感谐振，让Q1实现零电压开通；轻载时，Q2隔几个周期才导通，期间C3的电压会呈现一个阶梯状的增长，Q2导通时，C3与漏感谐振，让Q1实现零电压开通，同时进一步降低轻载时驱动Q2的功耗，实现占空比可以大于0.5、去磁电路的能量回收，特别是在轻载时，变换效率得到提高。

Description

一种有源钳位反激式开关电源电路 技术领域
本发明涉及开关电源领域，特别涉及使用有源钳位的反激式开关电源。
背景技术
开关电源应用很广，对于输入功率在75W以下，对功率因数(PF，Power Factor，也称功率因素)不作要求的场合，反激式(Fly-back)开关电源具有迷人的优势：电路拓扑简单，输入电压范围宽。由于元件少，可靠性相对就高，所以应用很广。为了方便，很多文献也称为反激开关电源、反激电源、反激变换器，日本和台湾地区又称返驰式变换器、返驰式开关电源、返驰电源。用于AC/DC变换器的常见拓扑如图1-1，该图原型来自张兴柱博士所著的书号为ISBN978-7-5083-9015-4的《开关电源功率变换器拓扑与设计》第60页。由整流桥101、滤波电路200、以及基本反激拓扑单元电路300组成，300也称为主功率级，实用的电路在整流桥前还加有压敏电阻、NTC热敏电阻、EMI(Electromagnetic Interference)等保护电路，以确保反激电源的电磁兼容性达到使用要求。一般情况下，反激式开关电源要求原副边绕组之间的漏感越小越好，这样变换效率高，而且原边主功率开关管V承受的耐压也降低，对于使用RCD网络作为去磁的反激变换器，RCD网络的损耗也降低。注：RCD吸收是指电阻、电容、二极管组成的吸收电路，我国的文献同国际上一样，一般用字母R给电阻编号并代表电阻，用字母C给电容编号并代表电容，用字母D给二极管编号并代表二极管，电阻和电容并联，再与二极管串联后形成RCD网络。
整流桥101一般由四个二极管组成，当不存在整流桥101时，200、300可以构成DC/DC开关电源或变换器，因为是直流供电，不存在功率因数的要求，功率可以做到75W以上。事实上，低压DC/DC开关电源中采用反激拓扑的并非主流，这是因为在低压时，反激电源的输入电流不连续，纹波较大，对前级的供电设备的要求较高；输出电流也不连续，纹波很大，对后面的滤波电容的容量要求高；特别是当输入电压较低时，由于激磁电流变大，原边绕组得采用多股线并绕；通常采用两个并联的原边绕组应用于低压DC/DC，低压DC/DC开关电源一般指输入电压在48V以下，部分用途的低压DC/DC开关电源可工作到直流160V，如铁路电源。
原边绕组的电感量也较低，经常出现计算出来的匝数不能平铺绕满骨架的线槽的左边到右边，当工作电压较高时可以采用三明治串联绕法的方案，在低工作电压下而被迫采用三明治并联绕法的方案，由于两个原边绕组不在同一层，这两个原边绕组之间就有漏感，这个漏感会产生损耗，从而让开关电源的效率变低，两个并联的原边绕组之间的漏感引发 的损耗问题：这在激磁和去磁时都会存在；若使用第三绕组去磁的话，不好选择第三绕组是和两个并联的原边绕组中的谁并绕，只能采用两个第三绕组，分别与两个并联的原边绕组并绕，然后再并联成“第三绕组”，工艺复杂，由两个绕组并联的第三绕组也存在会感应出不相等的电压，从而引起损耗和较大的电磁干扰。
其实，对于常见的第三绕组去磁，优点为无损去磁，效率较高，但是第三绕组的线径选择也是一个问题：选得比较细，与原边绕组的并绕比较麻烦，容易把细线拉断；若选得和原边绕组相同线径，成本高。第三绕组去磁反激变换器，又作“三绕组吸收反激变换器”。
在中国申请号分别为：201710142832.0、201710142797.2的二份名称均为《一种反激式开关电源》中，分别示出了图1-2、图1-3的技术方案，解决了上述问题，即：原边绕组可以不采用两个分开的并联，即可以允许原、副边绕组之间的漏感较大，不使用第三绕组去磁，同时变换效率不降低，激磁和去磁时的损耗降低。但这两个方案中，图1-2这种去磁方式，对漏感要求很严格，否则，激磁的能量可能都由D1直接返回直流电源U DC，而不出现在副边绕组N S中，造成副边D2中没有电流，从而使得输出电压低或无输出；且要求D2导通时产生的反射电压不能大于直流电源U DC，再如占空比无法大于0.5，导致功率密度不能进一步提高。对于图1-3，去磁电路本身是较为经典的拓扑，占空比可大于0.5，漏感的能量并没有被回收利用。
为了方便，发明人对中国申请号分别为201710142832.0、201710142797.2的反激式开关电源所使用的拓扑进行了定义，包括正激拓扑，不包括去磁方式的基本拓扑都定义为：LCL变换器，源于其两个原边激磁电感和一个与它们串联的电容。
发明内容
有鉴于此，本发明要解决现有的低压反激式开关电源存在的上述不足，提供一种使用有源钳位反激式开关电源电路，占空比可大于0.5，同时实现去磁电路的能量回收，进一步地，实现主功率开关管的零电压开关，进一步地降低损耗，提高变换效率。
本发明的目的是这样实现的，一种有源钳位反激式开关电源电路，包括一变压器，第一N沟道场效应管，第一电容、第二电容，第一二极管，一钳位网络，变压器包括第一原边绕组、第二原边绕组和副边绕组，钳位网络至少包括阳极和阴极，副边绕组异名端与第一二极管阳极连接，第一二极管阴极与第二电容一端连接，并形成输出正，副边绕组同名端与第二电容另一端连接，并形成输出负；输入直流电源的正端同时与第一原边绕组同名端、钳位网络的阴极相连，第一原边绕组异名端与第一N沟道场效应管的漏极相连；钳位网络的阳极与第二原边绕组异名端相连，第一N沟道场效应管的源极连接第二原边绕组同名端，连接点同时连接输入直流电源的负端；第一N沟道场效应管的栅极连接驱动控制信 号；第一原边绕组和第二原边绕组为双线并绕，第一电容的一端与第一原边绕组异名端相连，第一电容的另一端与第二原边绕组异名端相连，其特征在于：钳位网络至少包括第三电容和第二N沟道场效应管，第三电容和第二N沟道场效应管串联，串联方式为以下两种方式之一：
(1)第三电容的一端为钳位网络的阴极，第三电容的另一端连接第二N沟道场效应管的漏极，第二N沟道场效应管的源极为钳位网络的阳极，第二N沟道场效应管的栅极连接钳位控制信号；
(2)第三电容的一端为钳位网络的阳极，第三电容的另一端连接第二N沟道场效应管的源极，第二N沟道场效应管的漏极为钳位网络的阴极，第二N沟道场效应管的栅极连接钳位控制信号。
有源钳位反激式开关电源电路的工作方法为：第一N沟道场效应管每个周期都开通，第二N沟道场效应管在有源钳位反激式开关电源接近满载时，每个周期都开通；
第一N沟道场效应管每个周期都开通，第二N沟道场效应管在有源钳位反激式开关电源轻载时，隔m个周期才开通，m为正整数，且有源钳位反激式开关电源负载越轻，m的值越大。
本发明还提供上述方案一的等同方案，方案二：本发明目的还可以这样实现的，一种反激式开关电源电路，包括一变压器，第一N沟道场效应管，第一电容、第二电容，第一二极管，一钳位网络，变压器包括第一原边绕组、第二原边绕组和副边绕组，钳位网络至少包括阳极和阴极，副边绕组异名端与第一二极管阳极连接，第一二极管阴极与第二电容一端连接，并形成输出正，副边绕组同名端与第二电容另一端连接，并形成输出负；输入直流电源的正端同时与第一N沟道场效应管的漏极、第二原边绕组异名端相连，第一N沟道场效应管的源极与第一原边绕组同名端相连；第二原边绕组同名端与钳位网络的阴极相连，第一原边绕组异名端与钳位网络的阳极相连，连接点同时连接输入直流电源的负端；第一N沟道场效应管的栅极连接驱动控制信号；第一原边绕组和第二原边绕组为双线并绕，第一电容的一端与第一原边绕组同名端相连，第一电容的另一端与第二原边绕组同名端相连，其特征在于：钳位网络至少包括第三电容和第二N沟道场效应管，第三电容和第二N沟道场效应管串联，串联方式为以下两种方式之一：
(1)第三电容的一端为钳位网络的阴极，第三电容的另一端连接第二N沟道场效应管的漏极，第二N沟道场效应管的源极为钳位网络的阳极，第二N沟道场效应管的栅极连接钳位控制信号；
(2)第三电容的一端为钳位网络的阳极，第三电容的另一端连接第二N沟道场效应管的源极，第二N沟道场效应管的漏极为钳位网络的阴极，第二N沟道场效应管的栅极连接钳位控制信号。
有源钳位反激式开关电源电路的工作方法为：第一N沟道场效应管每个周期都开通，第二N沟道场效应管在有源钳位反激式开关电源接近满载时，每个周期都开通；
第一N沟道场效应管每个周期都开通，第二N沟道场效应管在有源钳位反激式开关电源轻载时，隔m个周期才开通，m为正整数，且有源钳位反激式开关电源负载越轻，m的值越大。
作为上述二种方案的改进，其特征在于：第二N沟道场效应管在有源钳位反激式开关电源轻载时，m可以在一个整数m和一个比该整数大1的数值(m+1)之间来回变动。
作为上述二种方案的改进，其特征在于：在第二N沟道场效应管的漏极、源极之间并联一只和第二N沟道场效应管的体二极管方向相同的低压降、快恢复的二极管。
作为上述二种方案的改进，其特征在于：第一原边绕组和第二原边绕组的线径相同。
优选地，PCB布线时第一原边绕组和第二原边绕组的激磁电流的物理路径的方向相反。
工作原理将结合实施例，进行详细地阐述。本发明的有益效果为：占空比可以大于0.5，同时实现去磁电路的能量回收，进一步地，实现主功率开关管的零电压开通，进一步地降低损耗，提高变换效率,特别是在轻载时，变换效率得到提高。
附图说明
图1-1为现有的反激式开关电源用于交流变直流的原理图；
图1-2为中国申请号201710142832.0的公开的技术方案原理图；
图1-3为中国申请号201710142797.2的公开的技术方案原理图；
图2-1本发明第一实施例原理图之一，钳位网络采用(1)方式；
图2-2本发明第一实施例原理图之二，钳位网络采用(2)方式；
图2-3为第一实施例中Q1饱和导通时，产生两路激磁电流41、42的示意图；
图2-4为第一实施例中Q1截止，产生续流电流43、去磁电流44的示意图；
图3-1为本发明第二实施例原理图之一，钳位网络采用(1)方式；
图3-2为本发明第二实施例原理图之二，钳位网络采用(2)方式。
具体实施方式
第一实施例
图2-1和图2-2示出了本发明第一实施例的有源钳位反激式开关电源电路的原理图，包括一变压器B，第一N沟道场效应管Q1，第一电容C1、第二电容C2，第一二极管D2，钳 位网络400，变压器B包括第一原边绕组N P1、第二原边绕组N P2和副边绕组N S，钳位网络400至少包括阳极和阴极，副边绕组N S异名端与第一二极管D2阳极连接，第一二极管D2阴极与第二电容C2一端连接，并形成输出正，为图中Vout的+端，副边绕组N S同名端与第二电容C2另一端连接，并形成输出负，为图中Vout的-端；输入直流电源U DC(下文也称作直流电源U DC、电源U DC，或U DC)的正端+同时与第一原边绕组N P1同名端、钳位网络400的阴极相连，第一原边绕组N P1异名端与N沟道场效应管Q1的漏极d相连；钳位网络400的阳极与第二原边绕组N P2异名端相连，N沟道场效应管Q1的源极s连接第二原边绕组N P2同名端，连接点同时连接输入直流电源U DC的负端-；N沟道场效应管Q1的栅极g连接驱动控制信号；第一原边绕组N P1和第二原边绕组N P2为双线并绕，还包括第一电容C1，第一电容C1的一端与第一原边绕组N P1异名端相连，第一电容C1的另一端与第二原边绕组N P2异名端相连，钳位网络400至少包括第三电容C3和第二N沟道场效应管Q2，第三电容C3和第二N沟道场效应管Q2串联，串联方式为以下两种方式之一：
(1)第三电容C3的一端为钳位网络400的阴极，第三电容C3的另一端连接第二N沟道场效应管Q2的漏极d，第二N沟道场效应管Q2的源极s为钳位网络400的阳极，第二N沟道场效应管Q2的栅极g连接钳位控制信号，如图2-1所示；
(2)第三电容C3的一端为钳位网络400的阳极，第三电容C3的另一端连接第二N沟道场效应管Q2的源极s，第二N沟道场效应管Q2的漏极d为钳位网络400的阴极，第二N沟道场效应管Q2的栅极g连接钳位控制信号，如图2-2所示。
可以看到，钳位网络400的阳极、阴极，和其内部的第二N沟道场效应管Q2的体二极管是对应的，在图2-1中，Q2的体二极管的阳极就是400的阳极，Q2的体二极管的阴极通过C3后就是400的阴极，在图2-2中，Q2的体二极管的阴极就是400的阴极，Q2的体二极管的阳极通过C3后就是400的阳极，当Q2更换为P沟道场效应管时，要保证P沟道场效应管内部的体二极管与图2-1或图2-1中的体二极管方向一致，即可正常工作。
同名端：图中绕组中以黑点标记的一端；
异名端：图中绕组中没有黑点标记的一端；
驱动控制信号：包括PWM脉冲宽度调制信号、PFM脉冲频率调制等各种方波；
钳位控制信号：包括PWM脉冲宽度调制信号、PFM脉冲频率调制等各种方波，但与驱动控制信号不同时出现；
变压器B：第一原边绕组N P1和第二原边绕组N P2在图中，其磁心用虚线相连，表示其为绕在一只变压器上，共用同一只磁心，并非独立的变压器，只是为了图形清晰、连接关系简单，才使用了图中的画法。
在图2-1、图2-2中，N沟道场效应管Q1的源极连接第二原边绕组N P2同名端，连接点同时连接输入直流电源U DC的负端-，即场效应管Q1的源极连接输入直流电源U DC的负端-，这在实际应用中并不直接存在，这是因为在开关电源领域中，基本拓扑的工作原理分析都会略去不必要的因素。在实际应用中，场效应管的源极都会接入电流检测电阻或电流互感器来检测平均电流或峰值电流来实现各种控制策略，这种通过电流检测电阻或电流互感器与源极相连，等同与源极相连，这是本技术领域的公知技术，本申请遵循业界默认的规则。若使用电流互感器，电流互感器可以出现在激磁回路的任何一个地方，如场效应管的漏极，如第一原边绕组的同名端或异名端，而且电流互感器除了传统的原边为一匝的“导线”、副边为多匝线圈的磁心式互感器，还可以是霍尔传感器。
工作原理：参见图2-1、图2-2中，当钳位网络400用一只和Q2(为了分析方便，按教科书的标准，以下简称为效应管Q2或Q2，其它器件同)体二极管方向相同的二极管替代时，就是图1-2的现有技术电路，但是本发明加了钳位网络400后，电路的工作原理与现有技术比，完全不同；
图2-1、图2-2电路在上电时，第二N沟道场效应管Q2不工作，Q1因没有收到驱动控制信号也不工作，相当于开路，那么电源U DC通过第一原边绕组N P1向C1充电，该电流同时通过第二原边绕组N P2回到电源U DC的负端，第一原边绕组N P1的充电电流为：从同名端流向异名端；第二原边绕组N P2的充电电流为：从异名端流向同名端；N P1和N P2为双线并绕，这两个电流大小相等，产生的磁通相反，完全抵消，即在上电时，电源U DC通过变压器B两个绕组向C1充电，这两个绕组因为互感作用而抵消，不起作用，C1相当于通过N P1和N P2的直流内阻与电源U DC并联，C1仍起到电源滤波、退耦的作用；随着时间的推移，C1的端电压等于U DC的电压，左正而右负。
当Q1正常收到控制信号时，以一个周期为例，Q1的栅极为高电平时，Q1饱和导通，其内阻等于通态内阻R ds(ON)，为了分析方便，把这种情况看作是直通，是一条导线，如图2-3所示，Q2处于截止状态，不参与工作，图中把400画为开路状态；这时产生两路激磁电流，图2-3中的41和42所示；
可见，41和42两路激磁电流是并联关系，由于N P1和N P2感量相同，激磁电压相同，都等于U DC，41和42完全相等，在激磁过程中，副边绕组N S按匝比同样产生感应电压，这个感应电压是：同名端感应出正电压，异名端感应出负电压，大小等于U DC乘以匝比n，即N S感应出下正上负的电压，这个电压与C2的端电压串联，加在D2的两端，D2反偏而不导通，这时副边相当于空载，无输出；
在激磁过程中，41和42电流呈线性向上增加；电流方向在电感中是从同名端流向异名端；
Q1的栅极由高电平变为低电平，Q1也由饱和导通变为截止，由于电感中的电流不能突变，尽管这时Q1已截止，但是41和42电流仍要从同名端流向异名端，由于原边的电流回路已被切断，磁心里的能量在副边从同名端流向异名端，参见图2-4，副边绕组N S出现从同名端流向异名端的电流，如图2-4中43所示，该电流的初始大小＝(41和42在Q1关断瞬间之和)/匝比n，该电流促使D2正向导通，并通过正向导通的D2，向电容C2充电，Vout建立电压或持续输出能量。这个过程也是去磁的过程。
反激式开关电源的输出端在原边绕组断开电源时获得能量故而得名，变压器B并不是变换电压的作用，而是隔着磁心续流的作用，是Buck-Boost变换器的隔离版本；所以变压器B通常又称为反激式变压器；
由于原边绕组与副边绕组，在一般情况下不可能是双线并绕，一定存在漏感。原边绕组激磁电感上储存的能量，在Q1关断后通过变压器B被传输到副边绕组N S、输出端，但是漏感上的能量没有传递，造成Q1管两端过压并损坏Q1管。本发明对漏感进行去磁的电路由Q2和C3组成钳位网络400和第二原边绕组N P2组成，工作原理为：
第一原边绕组N P1和第二原边绕组N P2为双线并绕，这两个绕组之间的漏感为零，在Q1关断瞬间及以后，漏感上的能量没有传递到副边，第二原边绕组N P2中漏感的电能量，其电流方向同激磁时的方向，从同名端流向异名端，即在图2-4中，由下向上流动，开通Q2的体二极管，电流从Q2的源极s流向漏极d，且这个电能量向C3充电，形成44所示的漏感去磁电流；
第一原边绕组N P1中漏感的电能量，通过无漏感地耦合到第二原边绕组N P2中，通过Q2的体二极管实现去磁，同样形成44所示的漏感去磁电流；
在图2-4中，Q2不起作用的部分画成浅色，起作用的体二极管用深色表示。
显而易见，输出电压Vout除以匝比n，这就是在副边绕组N S在D2导通时在原边形成的“反射电压”，由于存在C3隔直，反射电压大于直流电源U DC的值，电路也是可以正常工作的。当D2续流时，C2相当于电压源，该电压源向副边绕组N S“激磁”，原边形成的“反射电压”，这时，原边绕组相当于一个电压等于反射电压的电压源和漏感串联，感到D2中的电流下降为零，原边绕组才恢复为激磁电感和漏感串联。
那么，在D2开通续流期间，就会出现多种工作模式，即C3吸收了漏感的能量后，电路的工作模式有很多种，这里陈述一下工作原理：
本发明的反激式开关电源的工作方法限定为：第一N沟道场效应管Q1每个周期都开通，第二N沟道场效应管Q2在反激式开关电源接近满载时，每个周期都开通,即Q2和体二极管同步导通或滞后导通，在D2导通期间，原边绕组N P2和N P1呈电压源和漏感串联的状态，这时，漏感和C3出现谐振，在设计时，计算好C3的值，谐振的过程中，C3的端电压在特定的时间会接近零或等于零，由于原边绕组N P2和N P1是双绕并绕，两者之间的漏感为零，即原边绕组N P2和N P1的端电压在任何时刻保持相等，那么，C3的端电压接近或等于零的时刻，Q2是处于饱和导通状态，其端电压也为零，那么钳位网络400的端电压接近或等于零，即原边绕组N P2的端电压等于直流电源U DC的电压，且为上正下负，那么，原边绕组N P1是下正上负，Q2的端电压在此刻为两倍的U DC电压；而当C3的端电压谐振到为两倍的U DC电压时，且为上正下负时,即Q2的源极s电压为-U DC，由于C1的端电压一直为左正右负，且等于U DC，此刻，C1的左端子电压为零伏，即Q1的端电压也为零伏，若Q1在这个时刻饱和导通，那么，就实现了Q1的零电压开通(Zero Voltage Switch缩写为ZVS，这里只实现零电压开通，为准ZVS)，又称软开关技术，实现原边漏感能量的回收利用，Q1的输出电容上的能量，也因为谐振而被转移，实现原边漏感能量的回收利用。
由于漏感较小，为了实现较长的谐振时间，C3的容量是比较大的。正因为C3较大，且端电压可以升高，端电压和直流电源U DC的电压为串联关系，利用伏秒平衡定律,占空比可以大于0.5，都可以正常工作。
这种方式，显然是电流连续模式，Q1再导通的时间极难把握，若等D2关断后，C3和原边电感谐振，C3的端电压在特定的时间会接近或等于两倍的U DC电压时，且为上正下负时,由于C1的端电压一直为左正右负，且等于U DC，此刻，C1的左端子电压为零伏，即Q1的端电压也为零伏，若Q1在这个时刻饱和导通，那么，就实现了Q1的零电压开通，而这种方式一定是电流断续模式，Q1再导通的时间极容易被检测而实现。
而在本发明的反激式开关电源轻载时，若Q2每个周期都开通，那么Q2驱动损耗就不能忽视，Q2驱动损耗P Q21为：P Q21＝0.5×C iss×(U Q2) 2×f，C iss为Q2的输入电容，U Q2为Q2的栅极驱动电压，f为工作频率。可见，Q1每个周期都开通，Q1也可以采用优化降频的方式来降低损耗，但这种方式仍符合每个周期都开通的定义，只不过周期被拉长了；Q2在反激式开关电源轻载时，隔几个周期才开通，这样可以降低在电源轻载时的损耗，其原理为：开关电源轻载时，受环路控制的原边的激磁电流较小，储存在漏感中的能量较小，C3的容量是比较大的，这样，当Q1关断时，Q2并不开通，漏感能量较小，漏感能量为：0.5×漏感感量×激磁电流的平方；漏感能量通过Q2的体二极管对C3充电后，若钳位用的电容C3吸收的能量达到了门限值，下个周期Q2要导通，这可以通过负载的情况而预先设定，即通过 占空比的大小而预先设定。若C3没有达到门限值，则电路重复上个周期的动作。由于上一个周期存储在C3中的漏感能量没有释放，所以C3能量在上一个周期的基础上再次增加，C3两端的电压也再次上升到一个稳定值。经过几个周期的时间，C3两端的电压会呈现一个阶梯状的增长，直到达到门限值，即下个周期Q2要导通,实现谐振,实现零电压开通，这样，驱动损耗得到了进一步的降低。
为了进一步提高效率，在场效应管Q2的漏极、源极之间并联一只和Q2体二极管方向相同的低压降、快恢复的二极管，这种改进为公知技术，应视为和体二极管等效，本发明不再以实施例保护；
显然，m为正整数，且反激式开关电源负载越轻，激磁电流就越小,那么m的值越大，C3可以吸收更多周期的漏感能量，这样驱动损耗得到了进一步的降低，那么这种工作方式也是允许的：第二N沟道场效应管Q2在反激式开关电源轻载时，m可以在一个整数m和一个比该整数大1的数值(m+1)之间来回变动，以更好地兼顾设计。
在轻载工作时，或许本技术领域的人会有一个疑问，如：原来电路以Q1工作5个周期，Q2才参与导通一次，那么，在Q1工作的5个周期中，其中有一个周期占空比突然加大，激磁电流急聚上升，漏感能量过大，可能会引起C3两端的电压超过门限值而引起电路损坏。事实上，这种情况是不存在的，在中国申请号为201410186940.4的专利申请文件第0050段中，论述过：主功率变换电路的占空比在相邻的周期中不能突变，如目前较为流行的笔记本电源适配器，其工作频率多采用65KHz，公知理论认为，其开关电源的环路响应一般可以做到其十分之一，即6.5KHz，这部分的原理可以参考张兴柱博士的论文《开关电源的动态小信号分析与设计》，环路响应想做高，受开关电源中光耦延时等影响，很难提高，即开关电源中，主功率变换电路的占空比在十个工作周期中，第一个周期和第十个周期才会产生明显的变化，换一种说法，就是主功率变换电路的占空比在相邻的周期中不能突变。
由于41和42的电流相同，第一原边绕组和第二原边绕组的线径相同，这样绕制方便，这里所述的线径相同，还包括它们本身都是相同规格利兹线，颜色可以不同，即多股线绞合，为了方便识别，包括利兹线的同规格线材其颜色可以不同。随着工作频率的提升，高频电流更趋于在漆包线的表面流动，这种情况下，利兹线可以解决这一问题。当然，使用两种不同颜色的漆包线先做成利兹线，直接绕制，再按颜色分出第一原边绕组和第二原边绕组，或这两个绕组的线径和股数都不相同，都同样实现发明目的。
可见，与现有的LCL变换器相比，本发明有很多不同，主要为：占空比可以大于0.5，同时实现去磁电路的能量回收，进一步地，实现主功率开关管的零电压开通，进一步地降低损耗，提高变换效率,特别是在轻载时，变换效率得到提高。
第二实施例
本发明还提供上述第一实施例的等同方案，对应方案二，参见图3-1、图3-2，一种有源钳位反激式开关电源电路，包括一变压器B，第一N沟道场效应管Q1，第一电容C1、第二电容C2，第一二极管D2，钳位网络400，变压器B包括第一原边绕组N P1、第二原边绕组N P2和副边绕组N S，钳位网络400至少包括阳极和阴极，副边绕组N S异名端与第一二极管D2阳极连接，第一二极管D2阴极与第二电容C2一端连接，并形成输出正，为图中Vout的+端，副边绕组N S同名端与第二电容C2另一端连接，并形成输出负，为图中Vout的-端；输入直流电源U DC的正端+同时与N沟道场效应管Q1的漏极、第二原边绕组N P2异名端相连，N沟道场效应管Q1的源极与第一原边绕组N P1同名端相连；第二原边绕组N P2同名端与钳位网络400的阴极相连，第一原边绕组N P1异名端与钳位网络400的阳极相连，连接点同时连接输入直流电源U DC的负端；N沟道场效应管Q1的栅极连接控制信号；第一原边绕组N P1和第二原边绕组N P2为双线并绕，第一电容C1的一端与第一原边绕组N P1同名端相连，第一电容C1的另一端与第二原边绕组N P2同名端相连，钳位网络400至少包括第三电容C3和第二N沟道场效应管Q2，第三电容C3和第二N沟道场效应管Q2串联，串联方式为以下两种方式之一：
(1)第三电容C3的一端为钳位网络400的阴极，第三电容C3的另一端连接第二N沟道场效应管Q2的漏极d，第二N沟道场效应管Q2的源极s为钳位网络400的阳极，第二N沟道场效应管Q2的栅极g连接钳位控制信号，如图3-1所示；
(2)第三电容C3的一端为钳位网络400的阳极，第三电容C3的另一端连接第二N沟道场效应管Q2的源极s，第二N沟道场效应管Q2的漏极d为钳位网络400的阴极，第二N沟道场效应管Q2的栅极g连接钳位控制信号，如图3-2所示。
反激式开关电源的工作方法为：第一N沟道场效应管Q1每个周期都开通，第二N沟道场效应管Q2在反激式开关电源接近满载时，每个周期都开通；
第一N沟道场效应管Q1每个周期都开通，第二N沟道场效应管Q2在反激式开关电源轻载时，隔m个周期才开通，m为正整数，且反激式开关电源负载越轻，m的值越大。
事实上，第二实施例是第一实施例的变形：在第一实施例的图2-1基础上，把激磁回路的串联器件互换一下，即N P1和Q1互换位置，同时把钳位网络400和N P2互换位置，C1仍接在两个串联器件的连接点中间，就得到了第二实施例图3-1的电路，由于Q1的源极电压是变动的，所以，这个电路是浮地驱动，但却获得了钳位用的第二N沟道场效应管Q2的直接驱动，驱动的成本较低,在图3-1的基础上，把钳位网络400中的C3和Q2互换位置，即可得到图3-2的电路。
其工作原理简述：
参见图3-1和图3-2，电路在上电时，Q2不工作，Q1也不工作，相当于开路，那么电源U DC通过第一原边绕组N P1向C1充电，那么电源U DC通过N P2向C1充电，该电流同时通过N P1回到电源U DC的负端，同样在上电时，电源U DC通过变压器B两个绕组向C1充电，这两个绕组因为互感作用而抵消，不起作用，C1相当于通过N P2和N P1的直流内阻与电源U DC并联，C1仍起到电源滤波、退耦的作用；
随着时间的推移，C1的端电压等于U DC的电压，右正而左负；
当Q1饱和导通，其内阻等于通态内阻R ds(ON)，同前文看作是一条导线，这时产生两路激磁电流；
第一路为：电源U DC正端通过Q1的漏极进，Q1的源极出，再通过第一原边绕组N P1的同名端进，N P1的异名端出，回到电源U DC负端；
第二路为：电容C1右正端通过第二原边绕组N P2的同名端进，N P2的异名端出，Q1的漏极进，Q1的源极出，回到电容C1左负端；
可见，第一路和第二路激磁电流是并联关系，由于N P1和N P2感量相同，激磁电压相同，都等于U DC，这两路完全相等，在激磁过程中，副边绕组N S按匝比同样产生感应电压，同名端感应出正电压，异名端感应出负电压，大小等于U DC乘以匝比n，即N S感应出下正上负的电压，这个电压与C2的端电压串联，加在D2的两端，D2反偏而不导通，这时副边相当于空载，无输出；
在激磁过程中，第一路和第二路激磁电流呈线性向上增加；电流方向在电感中是从同名端流向异名端；
Q1截止时，电感中的电流不能突变，磁心里的能量在副边从同名端流向异名端，副边绕组N S出现从同名端流向异名端的电流，该电流通过正向导通的D2，向电容C2充电，Vout建立电压或持续输出能量。这个过程也是去磁的过程。
第二实施例中，对漏感进行去磁的电路由钳位网络400和第二原边绕组N P2组成，工作原理同第一实施例。
第二实施例为第一实施例的变形，工作原理等效，同样实现发明目的。作为用N沟道场效应管的技术方案，还可以用P沟道场效应管来实现，P沟道场效应管在低工作电压下，成本也是比较低的，这时，在上述第一实施例的基础上，电源、二极管、同名端的极性要反过来，输出整流部分不用反过来，那么得到第三、第四实施例，由于是本技术领域的常见变换，应该视为权利要求的保护范围。
以上仅是本发明的优选实施方式，应当指出的是，上述优选实施方式不应视为对本发 明的限制。对于本技术领域的普通技术人员来说，在不脱离本发明的精神和范围内，还可以做出若干改进和润饰，如加入控制环路实现输出的稳压，这是通过现有技术显而易见得到的，如采用其它符号的开关管Q1等，副边输出加入多路输出，滤波使用π型滤波；这些改进和润饰也应视为本发明的保护范围，这里不再用实施例赘述，本发明的保护范围应当以权利要求所限定的范围为准。

Claims (6)

  1. 一种有源钳位反激式开关电源电路，包括一变压器，第一N沟道场效应管，第一电容、第二电容，第一二极管，一钳位网络，变压器包括第一原边绕组、第二原边绕组和副边绕组，钳位网络至少包括阳极和阴极，副边绕组异名端与第一二极管阳极连接，第一二极管阴极与第二电容一端连接，并形成输出正，副边绕组同名端与第二电容另一端连接，并形成输出负；输入直流电源的正端同时与第一原边绕组同名端、钳位网络的阴极相连，第一原边绕组异名端与第一N沟道场效应管的漏极相连；钳位网络的阳极与第二原边绕组异名端相连，第一N沟道场效应管的源极连接第二原边绕组同名端，连接点同时连接输入直流电源的负端；第一N沟道场效应管的栅极连接驱动控制信号；第一原边绕组和第二原边绕组为双线并绕，第一电容的一端与第一原边绕组异名端相连，第一电容的另一端与第二原边绕组异名端相连，其特征在于：所述的钳位网络至少包括第三电容和第二N沟道场效应管，所述的第三电容和所述的第二N沟道场效应管串联，串联方式为以下两种方式之一：
    (1)所述的第三电容的一端为所述的钳位网络的阴极，所述的第三电容的另一端连接所述的第二N沟道场效应管的漏极，所述的第二N沟道场效应管的源极为所述的钳位网络的阳极，所述的第二N沟道场效应管的栅极连接钳位控制信号；
    (2)所述的第三电容的一端为所述的钳位网络的阳极，所述的第三电容的另一端连接所述的第二N沟道场效应管的源极，所述的第二N沟道场效应管的漏极为所述的钳位网络的阴极，所述的第二N沟道场效应管的栅极连接钳位控制信号；
    所述的第一N沟道场效应管每个周期都开通，所述的第二N沟道场效应管在有源钳位反激式开关电源接近满载时，每个周期都开通；所述的第二N沟道场效应管在有源钳位反激式开关电源轻载时，隔m个周期才开通，m为正整数，且有源钳位反激式开关电源负载越轻，m的值越大。
  2. 根据权利要求1所述的一种有源钳位反激式开关电源电路，其特征在于：输入直流电源的正端同时与所述的第一N沟道场效应管的漏极、所述的第二原边绕组异名端相连，所述的第一N沟道场效应管的源极与所述的第一原边绕组同名端相连；所述的第二原边绕组同名端与所述的钳位网络的阴极相连，所述的第一原边绕组异名端与所述的钳位网络的阳极相连，连接点同时连接输入直流电源的负端；所述的第一N沟道场效应管的栅极连接驱动控制信号；所述的第一原边绕组和所述的第二原边绕组为双线并绕，所述的第一电容 的一端与所述的第一原边绕组同名端相连，所述的第一电容的另一端与所述的第二原边绕组同名端相连。
  3. 根据权利要求1或2所述的一种有源钳位反激式开关电源电路，其特征在于：所述的第二N沟道场效应管在有源钳位反激式开关电源轻载时，m可以在一个整数m和一个比该整数大1的数值(m+1)之间来回变动。
  4. 根据权利要求3所述的一种有源钳位反激式开关电源电路，其特征在于：在第二N沟道场效应管的漏极与源极之间并联一只和第二N沟道场效应管的体二极管方向相同的二极管。
  5. 根据权利要求4所述的一种有源钳位反激式开关电源电路，其特征在于：所述的第一原边绕组和所述的第二原边绕组的线径相同。
  6. 根据权利要求5所述的一种有源钳位反激式开关电源电路，其特征在于：所述的第一原边绕组和所述的第二原边绕组的激磁电流的物理路径的方向相反。
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