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WO2012093423A1 - Non-contact charging system power supply device - Google Patents

Non-contact charging system power supply device Download PDF

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Publication number
WO2012093423A1
WO2012093423A1 PCT/JP2011/001901 JP2011001901W WO2012093423A1 WO 2012093423 A1 WO2012093423 A1 WO 2012093423A1 JP 2011001901 W JP2011001901 W JP 2011001901W WO 2012093423 A1 WO2012093423 A1 WO 2012093423A1
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WO
WIPO (PCT)
Prior art keywords
power
switching elements
power supply
output voltage
voltage
Prior art date
Application number
PCT/JP2011/001901
Other languages
French (fr)
Japanese (ja)
Inventor
秀樹 定方
藤田　篤志
Original Assignee
パナソニック株式会社
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Filing date
Publication date
Application filed by パナソニック株式会社 filed Critical パナソニック株式会社
Publication of WO2012093423A1 publication Critical patent/WO2012093423A1/en

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J7/007—Regulation of charging or discharging current or voltage
    • H02J7/00712—Regulation of charging or discharging current or voltage the cycle being controlled or terminated in response to electric parameters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/80—Circuit arrangements or systems for wireless supply or distribution of electric power involving the exchange of data, concerning supply or distribution of electric power, between transmitting devices and receiving devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to a power feeding device in a contactless charging system that charges a secondary battery mounted in, for example, an electric propulsion vehicle (electric vehicle or hybrid vehicle) in a contactless manner.
  • a contactless charging system that charges a secondary battery mounted in, for example, an electric propulsion vehicle (electric vehicle or hybrid vehicle) in a contactless manner.
  • each of the power feeding device and the power receiving device includes a resonance unit that resonates an AC signal, thereby reducing the restriction on the positional relationship between the power feeding device and the power receiving device (for example, see Patent Document 1).
  • the present invention has been made in view of such problems of the prior art, and not only can reduce the current / voltage ripple of the commercial frequency component in the output of the power supply apparatus, but also reduce the number of parts of the power supply apparatus. Accordingly, it is an object of the present invention to provide a power supply device for a non-contact charging system that can reduce the size or cost of the power supply device and can reduce power supply loss as much as possible.
  • the present invention provides a power supply device that supplies power to a power receiving device in a non-contact manner, a power factor improvement circuit that outputs a DC voltage by improving the power factor, and an output of the power factor improvement circuit
  • a DC voltage When a DC voltage is applied, it has an electrolytic capacitor that accumulates electric charge and a plurality of switching elements, operates using the electrolytic capacitor as a power source, generates an AC signal, and receives an AC signal from the inverter circuit Then, a resonance capacitor and an inductor that perform power supply to the power receiving device in a non-contact manner, and a power supply side control circuit that changes at least one of the energization rate and the drive frequency of the plurality of switching elements are provided.
  • the power supply apparatus can be reduced in size or cost by reducing the number of parts of the power supply apparatus, and power supply loss can be minimized. Can be reduced.
  • FIG. 1 is a circuit diagram of a contactless charging system according to an embodiment of the present invention.
  • FIG. 2 is a circuit diagram of a feeding power detection unit provided in the non-contact charging system of FIG.
  • FIG. 3 is a diagram showing waveforms of respective parts of a conventional non-contact charging system for comparison with waveforms of respective parts in the non-contact charging system of FIG.
  • FIG. 4 is a diagram illustrating waveforms of respective parts in the non-contact charging system when the energization rates of a plurality of switching elements provided in the non-contact charging system of FIG. 1 are changed.
  • FIG. 5 is a diagram showing waveforms of respective parts in the non-contact charging system when the driving frequency of a plurality of switching elements is changed.
  • FIG. 6 is a diagram showing waveforms of respective parts in the non-contact charging system when both the energization rate and the driving frequency of the plurality of switching elements are changed.
  • FIG. 7 is a diagram showing operation waveforms of the main part of the power feeding device when the input power to the inverter circuit 4 of FIG. 1 is high.
  • FIG. 8 is a diagram illustrating operation waveforms of main parts of the power feeding device when the input power to the inverter circuit 4 in FIG. 1 is low.
  • FIG. 9 is a circuit diagram of another contactless charging system according to the present invention.
  • the present invention is a power supply device that supplies power to a power receiving device in a non-contact manner.
  • the power factor improvement circuit that performs power factor improvement and outputs a DC voltage, and the output DC voltage of the power factor improvement circuit is given a charge.
  • an inverter circuit that operates using the electrolytic capacitor as a power source to generate an AC signal, and when an AC signal from the inverter circuit is input, the power receiving device is A resonance capacitor and an inductor that perform power supply by contact, and a power supply-side control circuit that changes at least one of a conduction rate and a drive frequency of the plurality of switching elements are provided.
  • the power supply device further includes a power supply power detection unit that detects a current flowing through the inductor or a voltage of the inductor, and the power supply side control circuit performs a plurality of switching operations based on the current or voltage detected by the power supply power detection unit. You may change at least one of the electricity supply rate and drive frequency of an element.
  • the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic energization rate of the plurality of switching elements so as to become the received power command value, and When the output voltage is high, the energization rate of the plurality of switching elements may be set lower than the basic energization rate, and when the output voltage is low, the energization rate of the plurality of switching elements may be set higher than the basic energization rate.
  • the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic drive frequency of the plurality of switching elements so as to become the received power command value, and detects the power supply power When the output voltage of the unit is high, the driving frequency of the plurality of switching elements can be set higher than the basic driving frequency, and when the output voltage is low, the driving frequency of the plurality of switching elements can be set lower than the basic driving frequency.
  • the power supply side control circuit has an output voltage detection unit that detects the output voltage of the power factor correction circuit, and based on the output voltage of the power factor correction circuit detected by the output voltage detection unit, the energization of a plurality of switching elements. At least one of the rate and the driving frequency may be changed.
  • the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic energization rate of the plurality of switching elements so as to become the received power command value, and outputs the power command value.
  • the energization rate of the plurality of switching elements is set lower than the basic energization rate, and when low, the energization rate of the plurality of switching elements is set higher than the basic energization rate.
  • the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic drive frequency of the plurality of switching elements so as to become the received power command value, and detects the output voltage.
  • the driving frequency of the plurality of switching elements can be set higher than the basic driving frequency
  • the driving frequency of the plurality of switching elements can be set lower than the basic driving frequency
  • FIG. 1 is a circuit diagram of the non-contact charging system in the present embodiment.
  • the contactless charging system includes, for example, a power feeding device installed in a parking space and a power receiving device mounted in, for example, an electric propulsion vehicle.
  • the non-contact charging system includes a commercial power source 1, a first rectifier circuit 2, a power factor correction circuit 3, an inverter circuit 4, a feed power detection unit 5, a first resonance capacitor 6, and a first power supply device.
  • control circuit 13 The inductor 7 and the control circuit 13 on the power feeding device side (hereinafter simply referred to as “control circuit 13”), and the configuration on the power receiving device side include a second inductor 8, a second resonance capacitor 9, a second rectifier circuit 11, A load (battery) 12, a control circuit 14 on the power receiving device side (hereinafter simply referred to as “control circuit 14”), and a received power detection unit 10 are provided.
  • the power factor improving circuit 3 is a circuit for improving the power factor of the commercial power source 1, and includes a bypass capacitor 29, a choke coil 15 serving as a first choke coil, and a first switching element 16 (in the present embodiment, a MOSFET). ), A diode 17 that is a first diode, and a smoothing capacitor (electrolytic capacitor) 18.
  • the commercial power source 1 is a 200 V commercial power source that is a low-frequency AC power source, and is connected to the input terminal of the first rectifier circuit 2 including a bridge diode and an input filter.
  • the high potential terminal (positive electrode side) output terminal of the first rectifier circuit 2 is connected to the high potential terminal of the bypass capacitor 29 and the input terminal of the choke coil 15. Further, the high potential side terminal (drain) of the switching element 16 is connected to a connection line between the output side terminal of the choke coil 15 and the anode side terminal of the diode 17.
  • a low potential side terminal of the bypass capacitor 29, a low potential side terminal (source) of the switching element 16, and a low potential side terminal of the smoothing capacitor 18 are connected to the low potential side (negative electrode side) output terminal of the first rectifier circuit 2.
  • the high potential side terminal of the smoothing capacitor 18 is connected to the cathode side terminal of the diode 17.
  • the power factor correction circuit 3 is supplied with the output voltage of the first rectifier circuit 2 as a DC power supply, and the voltage fluctuation of the input output voltage of the first rectifier circuit 2 is suppressed by the bypass capacitor 29.
  • 15 and the switching element 16 are turned on and off, and a DC voltage having a peak value larger than the peak value and boosted to an arbitrary voltage is supplied across the smoothing capacitor 18 and smoothed.
  • a MOSFET having a high switching speed is used as a typical example of the switching element 16 in order to increase the power factor improvement effect by operating the power factor improvement circuit 3 at a high frequency.
  • a diode is attached to the MOSFET in the opposite direction, it is not shown in the figure because it does not affect the basic operation of the present embodiment even without this diode.
  • the output voltage of the smoothing capacitor 18 is supplied between the input terminals of the inverter circuit 4.
  • the input terminal of the inverter circuit 4 is connected to the output terminal of the power factor correction circuit 3, that is, to both ends of the smoothing capacitor 18.
  • a series connection body of switching elements (second and third switching elements) 19 and 20 and a series connection body of switching elements (fourth and fifth switching elements) 24 and 26 are parallel. Connected to.
  • Diodes (second and third diodes) 21 and 22 are connected in antiparallel to the switching elements 19 and 20, respectively (the high potential side terminal (collector) of the switching element and the cathode side terminal of the diode are connected). ). Further, a snubber capacitor 23 is connected in parallel to the switching element 20 (which may be the switching element 19).
  • diodes (fourth and fifth diodes) 25 and 27 are connected in antiparallel to the switching elements 24 and 26 (the high potential side terminal (collector) of the switching element and the cathode side terminal of the diode are connected to each other). Connected). A snubber capacitor 28 is connected in parallel to the switching element 26 (which may be the switching element 24).
  • a series connection body of the first resonance capacitor 6, the first inductor 7, and the feeding power detection unit 5 is connected to the connection line of the switching element 19 and the switching element 20 and the connection line of the switching element 24 and the switching element 26. Is done.
  • the second inductor 8 is disposed so as to face the first inductor 7 as the electric propulsion vehicle moves, for example.
  • a second resonant capacitor 9 is connected to the high potential side of the second inductor 8, and the low potential side of the second inductor 8 and the second resonant capacitor 9 include a second rectifier circuit 11 including a smoothing filter.
  • the received power detection unit 10 is connected to the high potential side of the second rectifier circuit 11, and the load (battery) 12 is connected to the low potential side of the received power detection unit 10 and the second rectifier circuit 11. .
  • the supply power detection unit 5 in the present embodiment includes a current detection unit 30, a voltage detection unit 31, and a power calculation unit 32.
  • the current detection unit 30 and the voltage detection unit 31 may be used when the power supply power can be estimated using either the current or the voltage.
  • the feed power detection unit 5 is connected in series to the series resonance circuit of the first inductor 7 and the first resonance capacitor 6, only one of the current and voltage is correlated. The power supply can be estimated by detection.
  • the received power detection unit 10 may have the same configuration as the power supply power detection unit 5.
  • the control circuit 13 receives a power command value from the control circuit 14 by wireless communication.
  • the control circuit 13 compares the supplied power detected by the supplied power detection unit 5 with the received power command value, and the switching elements 19 and 20 and the switching elements 24 and 26 of the inverter circuit 4 are obtained so that the power command value is obtained.
  • the switching element 16 of the power factor correction circuit 3 is driven.
  • a dedicated control IC may be used for controlling the switching element 16 of the power factor correction circuit 3.
  • the control circuit 14 determines a power command value according to the remaining voltage of the battery 12 detected by the received power detection unit 10 and transmits it to the control circuit 13 by wireless communication. Further, the received power is detected by the received power detection unit 10 during operation of the power supply apparatus, and the control circuit 14 changes the power command value to the control circuit 13 so that the load (battery) 12 is not overcurrent or overvoltage.
  • a battery for an electric propulsion vehicle is used as the load 12 of the first embodiment.
  • Battery charging is performed by supplying a voltage equal to or higher than the remaining voltage of the battery, but when the power supply voltage exceeds the remaining battery voltage, a charging current suddenly flows. This means that the load impedance as viewed from the power supply device varies greatly depending on the remaining battery voltage and the power supply voltage.
  • FIG. 3A is a schematic diagram showing an AC voltage waveform of the commercial power source 1
  • FIG. 3B is a schematic diagram showing an output voltage waveform of the DC power source, that is, an output voltage waveform of the first rectifier circuit 2. It is. This voltage is input to the power factor correction circuit 3, boosted, and then output to the smoothing capacitor 18.
  • FIG. 3C is a schematic diagram showing a waveform applied to the smoothing capacitor 18, that is, an output voltage waveform of the power factor correction circuit 3 and an input voltage waveform of the inverter circuit 4.
  • FIG. 3D is a schematic diagram showing a high-frequency current waveform generated in the first inductor 7
  • FIG. 3E is a schematic diagram showing a power waveform fed from the power feeding device to the power receiving device.
  • 3F is a schematic diagram illustrating an output current waveform of the second rectifier circuit 11, that is, an input current waveform of the load 2.
  • 3 (g) and 3 (h) are schematic diagrams showing an energization rate (duty ratio) and an operating frequency, respectively.
  • FIG. 4 shows voltage waveforms, current waveforms, etc. of each part of the non-contact charging system according to the present invention.
  • FIGS. 4 (a) to 4 (h) are respectively shown in FIGS. 3 (a) to 3 (h). It corresponds.
  • the commercial power source 1 shown in FIG. 4A is full-wave rectified by the first rectifier circuit 2 to form a DC power source as shown in the voltage waveform of FIG.
  • This DC power supply is supplied between the input terminals of the power factor correction circuit 3.
  • the diode 17 included in the power factor correction circuit 3 and the bridge diode of the first rectifier circuit 2 are turned on when the instantaneous value of the DC power supply voltage is smaller than the voltage of the smoothing capacitor 18. If not, the input current waveform is distorted and the power factor is significantly reduced. At that time, the control circuit 13 improves the power factor by turning the switching element 16 on and off.
  • the power factor correction circuit 3 has not only a power factor correction function but also a boosting function at the same time. For this reason, as shown in FIG.
  • the voltage of the smoothing capacitor 18 is the peak value of the input voltage of the power factor correction circuit 3 whose peak value is the peak value of the commercial power source 1, that is, the peak value of the DC power source. The voltage becomes higher and is supplied to the inverter circuit 4 through the smoothing capacitor 18.
  • FIGS. 3 (a) to 3 (c) and FIGS. 4 (a) to 4 (c) the commercial power supply in the power transmission system described in Patent Document 1 and the non-contact charging system according to the present invention. There is no significant difference between the AC voltage waveform of 1, the output voltage waveform of the first rectifier circuit 2, and the output voltage waveform of the power factor correction circuit 3.
  • the smoothed DC voltage output to both ends of the smoothing capacitor 18 connected between the output ends of the power factor correction circuit 3 shown in FIG. 4C is supplied to the inverter circuit 4.
  • the inverter circuit 4 is shown in FIG. 4D by the first resonant capacitor 6 and the first inductor 7 depending on whether the switching elements 19 and 20 are turned on / off and the switching elements 24 and 26 are turned on / off. Thus, a high-frequency current having a predetermined frequency is generated.
  • the on / off control of the switching elements 19 and 20 and the on / off control of the switching elements 24 and 26 are performed by the control circuit 13 applying an on signal to the gates of the switching elements 19, 20, 24 and 26. .
  • FIG. 7 and 8 show enlarged operation waveforms of the inverter circuit 4 at high input power and low input power, respectively.
  • (A), (c), and (d) are the switching elements 19 and 26 and the diode 21, respectively. , 27, the voltage of the switching elements 19, 26, and the gate voltage of the switching elements 19, 26, respectively, (b) and (e) flow in the switching elements 20, 24 and the diodes 22, 25. The current and the gate voltage of the switching elements 20 and 24 are shown.
  • (f) shows the current flowing through the first inductor 7, and the current flowing through the switching elements 19 and 26 and the diodes 21 and 27 flows during the Ton period in the figure, and the remainder of one cycle (in the figure) During the period of (T ⁇ Td ⁇ Ton), the current flowing through the switching elements 20 and 24 and the diodes 22 and 25 flows. During the dead time Td described later, the resonance current of the first inductor 7, the first resonance capacitor 6, and the snubber capacitors 23 and 28 flows.
  • the two switching elements 19 and 20 connected in series are exclusively energized, and the two switching elements 24 and 24 connected in series to the two switching elements 19 and 20 are connected in parallel. 26 is energized exclusively by shifting the drive signal phase of the switching elements 19 and 20.
  • the switching element 19 and the switching element 26 are repeatedly turned on and off in synchronization.
  • the switching element 20 and the switching element 24 are turned off, and the switching element 19 and the switching element 26 are turned on.
  • the switching element 20 and the switching element 24 are turned on, the switching element 20 and the switching element 24 are repeatedly turned on and off in synchronization.
  • the ON period of the switching elements 19 and 24 and the switching element 20 are set so that the switching element 19 and the switching element 20 are not turned on simultaneously, and so that the switching element 24 and the switching element 26 are not turned on simultaneously.
  • the dead time Td is set so as not to overlap.
  • the switching elements 19 and 26 are turned off from the on state, the snubber capacitor 23 is discharged with a gentle inclination due to resonance of the first inductor 7, the first resonance capacitor 6, and the snubber capacitor 23.
  • the switching elements 19, 26 realize a zero volt switching (ZVS) turn-off operation.
  • ZVS zero volt switching
  • the snubber capacitor 28 is discharged with a gentle slope due to resonance of the first inductor 7, the first resonant capacitor 6, and the snubber capacitor 28.
  • the elements 20 and 24 realize a ZVS turn-off operation.
  • the snubber capacitor 23 is charged and the snubber capacitor 28 is completely discharged, the diodes 21 and 27 are turned on, and an on signal is sent to the gates of the switching elements 19 and 26 during the period in which the diodes 21 and 27 are on.
  • the switching elements 19 and 26 and the switching elements 20 and 24 are alternately turned on / off by providing a dead time Td (for example, about 2 ⁇ s) so as not to short-circuit the smoothing capacitor 18.
  • Td for example, about 2 ⁇ s
  • the drive (operation) frequency of the switching elements 19, 20, 24, and 26 is made constant, and as shown in FIG. 4 (g), the energization rate (duty ratio).
  • the high frequency power is controlled by controlling.
  • the “energization rate” here means the switching element 19 with respect to the time required for one cycle of ON / OFF of the switching elements 19 and 26 (or the switching elements 20 and 24), as shown in FIGS. , 26 (or switching elements 20, 24).
  • the inverter circuit 4 has a voltage ripple of 120 Hz as shown in FIG.
  • a current ripple is generated in the current of the first inductor 7. Therefore, the feed power fluctuates as shown in FIG. 3E, and a 120-Hz current ripple occurs in the input current of the load 12 as shown in FIG.
  • the inverter circuit 4 when the input voltage including the voltage ripple shown in FIG. 4C is applied to the inverter circuit 4, the current (or the first current) of the first inductor 7 detected by the feed power detector 5.
  • the energization rates (duty ratios) of the switching elements 19, 20, 24, and 26 are controlled so that the magnetic field generated by the inductor 7 becomes substantially constant.
  • the electric current which flows through the 1st inductor 7, and the electric power feeding electric power by the side of an electric power feeder become substantially constant, as FIG.4 (d) and (e) show.
  • the control circuit 13 compares the power supply power detected by the power supply power detection unit 5 with the received power command value to obtain a power command value.
  • the switching elements 19 and 20 of the inverter circuit 4 the switching elements 24 and 26, and the switching element 16 of the power factor correction circuit 3 are driven.
  • the energization rate is set so that the power supply power increases when the energization rates of the switching elements 19 and 20 and the switching elements 24 and 26 are increased.
  • the basic energization rate is determined so that the power command value is obtained.
  • the control circuit 13 lowers the energization rate below the basic energization rate when the output voltage of the power factor correction circuit 3 is high, and sets the energization rate at the low voltage. Control is performed so that it is set higher than the rate (see FIG. 4G). As a result, the current flowing through the first inductor 7 and the feed power can be made substantially constant.
  • the basic energization rate is selected so that the supply power matches the power command value when the energization rate control is performed so that the supply power is substantially constant.
  • the basic energization rate is 50% and the output is maximum, and the basic energization rate is variable in the range of 0 to 50%. Therefore, when the feed power that becomes the power command value cannot be obtained by the energization rate control, the control circuit 13 changes the drive frequency. When the feed power is low, the drive frequency is set lower, and the control circuit 13 resets the basic energization rate. On the contrary, when the feed power is high, the drive frequency is set higher and the basic energization rate is also reset.
  • the energization rate is set to be higher at high input power than at low input power (for example, 50%, 40%, etc.
  • the control circuit 13 controls the switching elements 19, 20, 24, and 26, so that the current flowing through the first inductor 7 and the feeding power are made substantially constant. can do.
  • the control circuit 14 determines command values such as charging current, voltage, and power according to the remaining voltage of the battery detected by the received power detection unit 10 at the start of charging, and transmits the command value to the control circuit 13 by wireless communication. Even during charging, information such as charging current, voltage, and power is transmitted to the control circuit 13 by wireless communication, and the control circuit 13 performs control based on the received information such as charging current, voltage, and power.
  • the power transmission efficiency between the first inductor 7 and the second inductor 8 can be increased by causing the second inductor 8 and the second resonant capacitor 9 to resonate. This is because, when the impedance component due to the leakage inductance that cannot be magnetically coupled to the first inductor 7 among the second inductors 8 is canceled by the second resonance capacitor 9, the impedance on the secondary side is lowered and it becomes easy to transmit power. Can also explain. Even if the second resonant capacitor 9 is not provided, the present invention is not affected.
  • control circuit 13 drives and controls the power factor correction circuit 3 and the inverter circuit 4 so that the power command value and the detection result of the feed power detection unit 5 are matched by the above-described operation.
  • the switching elements 19, 20, 24, and 26 are controlled based on the current value detected by the power supply power detection unit 5, but the power supply voltage detection unit 5 detects the power supply voltage. By controlling the switching elements 19, 20, 24, and 26 so that the detected voltage value becomes substantially constant, the voltage applied to the first inductor 7 and the feeding power can be made substantially constant.
  • the driving frequency of the switching elements 19, 20, 24, and 26 is made constant, and the energization rate is controlled to control the high frequency power, but in the present embodiment, FIG. As shown in FIG. 4, the high-frequency power is controlled by controlling the drive frequency of the switching elements 19, 20, 24, and 26 while keeping the energization rate constant.
  • the driving frequency is set so that the feeding power increases when the driving frequency is lowered (the operation is performed at a frequency higher than the frequency at which the feeding power becomes maximum), and is detected by the feeding power detection unit 5.
  • the control circuit 13 is set so that the drive frequency is set high when the output voltage of the power factor correction circuit 3 is high and the drive frequency is set low when the output voltage is low (see FIG. 5H).
  • the control circuit 13 compares the power supply power detected by the power supply power detection unit 5 with the received power command value to obtain a power command value.
  • the switching elements 19 and 20 of the inverter circuit 4 the switching elements 24 and 26, and the switching element 16 of the power factor correction circuit 3 are driven.
  • the driving frequency is set so that the feeding power increases when the driving frequency of the switching elements 19 and 20 and the switching elements 24 and 26 is lowered (maximum feeding power).
  • the control circuit 13 determines the basic drive frequency so that the power command value is obtained.
  • the control circuit 13 sets the drive frequency higher than the basic frequency when the output voltage of the power factor correction circuit 3 is high, and sets the drive frequency higher than the basic frequency when the output voltage is low. Control is performed so that it is set low (see FIG. 5H). As a result, the current flowing through the first inductor 7 and the feed power can be made substantially constant.
  • the basic drive frequency is selected so that the feed power matches the power command value when such drive frequency control is performed to make the feed power substantially constant.
  • the energization rate is basically 50%. However, when the energization rate is set higher, the power supply becomes larger, while when the energization rate is set lower, the power supply is reduced. Yes. By changing this energization rate, the supplied power and the power command value can be made more consistent.
  • the current ripple in the load is generated when the output voltage of the power factor correction circuit 3, that is, the voltage of the electrolytic capacitor, changes as shown in FIG. Since the magnitude of this voltage increases as the feed power increases, in a charging system in which the power command value sent by wireless communication from the control circuit 14 varies as in this embodiment, the current ripple in the load Will also change depending on the power command value. For example, when the power command value increases, the voltage change width of the electrolytic capacitor increases and the load current ripple also increases.
  • the total power command value is The load current ripple is improved and the operation is performed so that the power supply becomes substantially constant.
  • the drive frequency is high when the output voltage of the power factor correction circuit 3 is high.
  • the control circuit 13 controls the drive frequency of the switching elements 19, 20, 24, and 26 so that the drive frequency is low when the voltage is low, the voltage ripple is reduced and the power supply voltage can be made substantially constant.
  • the high frequency power is controlled by controlling both the energization rate and the drive frequency of the switching elements 19, 20, 24, and 26.
  • control circuit 13 controls the switching elements 19, 20, 24, and 26 so as to reduce the current ripple, thereby reducing the current ripple and making the current flowing through the first inductor 7 substantially constant.
  • FIG. 9 is a circuit diagram of the non-contact charging system in the present embodiment. 9 is that the control circuit 13 is provided with an output voltage detector 33 that detects the output voltage of the smoothing capacitor 18 of the power factor correction circuit 3 (input voltage of the inverter circuit 4). It is different from the circuit diagram.
  • control circuit 13 controls the switching elements 19, 20, 24, and 26 based on the output voltage of the smoothing capacitor 18 detected by the output voltage detector 33 (energization rate control and / or / (Or frequency control), the feeding current or feeding voltage is controlled to a desired value.
  • the switching elements 19, 20, 24, and 26 are controlled as follows, as in the above-described first to third embodiments.
  • (I) Energization rate control (drive frequency is constant) In this control, the control circuit 13 switches the switching elements 19, 20, 24, and 26 so that the energization rate is small when the output voltage of the smoothing capacitor 18 detected by the output voltage detection unit 33 is high and the energization rate is large when the output voltage is low. By controlling the above, the current or voltage of the first inductor 7 and the feeding power are made substantially constant.
  • Energization rate control and drive frequency control In this control, the energization rate is small and the drive frequency is high when the output voltage of the smoothing capacitor 18 detected by the output voltage detector 33 is high, and the energization rate is large when the output voltage is low.
  • the control circuit 13 controls the switching elements 19, 20, 24, and 26 so that the drive frequency is lowered, so that the current or voltage of the first inductor 7 and the feeding power are made substantially constant.
  • the power feeding device of the non-contact charging system can not only reduce the current / voltage ripple of the power feeding device, but also reduce the number of parts of the power feeding device to reduce the size or cost of the power feeding device. Since the power loss can be reduced as much as possible, it is useful for power feeding to a power receiving device of an electric propulsion vehicle, for example.

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Abstract

Provided is a non-contact power supply device comprising: a power factor improvement circuit (3) which performs power factor improvement and outputs DC voltage; an electrolytic capacitor (18) which is connected with the output terminal of the power factor improvement circuit (3); an inverter circuit (4) which has a plurality of switching elements (19, 20, 24, 26) and generates an AC signal, using the voltage of the electrolytic capacitor (18) as a power source; a first resonance capacitor (6) and a first inductor (7) which are connected with the output terminal of the inverter circuit (4); and a power supply device control circuit (13). Furthermore, when power is supplied to a power-receiving device, the first inductor (7) is made to be opposite a second inductor (8) of the power-receiving device, and the power supply device control circuit (13) changes the current passage time and/or the drive frequency of the plurality of switching elements (19, 20, 24, 26) in accordance with the output of the inverter circuit (4).

Description

非接触充電システムの給電装置Power supply device for contactless charging system
　本発明は、例えば電気推進車両（電気自動車やハイブリッド車）などに搭載される二次電池に非接触で充電する非接触充電システムにおける給電装置に関する。 The present invention relates to a power feeding device in a contactless charging system that charges a secondary battery mounted in, for example, an electric propulsion vehicle (electric vehicle or hybrid vehicle) in a contactless manner.
　非接触で電力伝送するための技術として、磁界、電界、電波などを用いる技術が開発されており、このような非接触電力伝送技術によって、給電装置と受電装置とを接続する電線が不要となるため、ユーザにとっては、接続の手間が省けたり、雨天時などの漏電や感電の心配がなくなったりする。 As a technique for transmitting power without contact, a technique using a magnetic field, an electric field, a radio wave, and the like has been developed, and such a contactless power transmission technique eliminates the need for an electric wire that connects the power feeding device and the power receiving device. Therefore, it is possible for the user to save the trouble of connection or to worry about electric leakage or electric shock during rainy weather.
　ところで、非接触電力伝送では、例えば高効率化のため、給電装置と受電装置との位置関係が重要となる。この問題に対処するため、従来、給電装置及び受電装置のそれぞれに交流信号を共振させる共振部を備えることで、給電装置と受電装置との位置関係の制約を低減する技術が提案されている（例えば、特許文献１参照）。 By the way, in non-contact power transmission, for example, the positional relationship between the power feeding device and the power receiving device is important for high efficiency. In order to cope with this problem, conventionally, a technique has been proposed in which each of the power feeding device and the power receiving device includes a resonance unit that resonates an AC signal, thereby reducing the restriction on the positional relationship between the power feeding device and the power receiving device ( For example, see Patent Document 1).
　しかしながら、特許文献１に記載の技術の場合、給電装置から出力される給電電力には商用周波数成分が重畳することから、給電装置の出力には商用周波数成分の電流リプルあるいは電圧リプルが発生し、電解コンデンサの寿命が短くなるなどの課題がある。 However, in the case of the technique described in Patent Document 1, since the commercial frequency component is superimposed on the feed power output from the power feeding device, a current ripple or voltage ripple of the commercial frequency component is generated at the output of the power feeding device. There are problems such as shortening the life of electrolytic capacitors.
　そこで、直列接続された４段の回路からなる三つの列回路（コンバータ）を複数の平滑コンデンサを共有して並列接続し、各列回路毎に位相を２π／３（rad）ずらして各列回路を駆動し、複数の平滑コンデンサへの充放電電流を列回路間で融通し合うことにより、電圧リプルを低減するようにしたものも提案されている（例えば、特許文献２参照）。 Therefore, three column circuits (converters) consisting of four stages connected in series are connected in parallel by sharing a plurality of smoothing capacitors, and each column circuit is shifted in phase by 2π / 3 (rad) for each column circuit. Has been proposed in which voltage ripple is reduced by interfacing charge / discharge currents to a plurality of smoothing capacitors between column circuits (see, for example, Patent Document 2).
特開２００９－２９６８５７号公報JP 2009-296857 A 特開２００８－２６３７１５号公報JP 2008-263715 A
　特許文献２に記載の構成の場合、電圧リプルを低減することはできるものの、列回路（コンバータ）を複数必要とすることから、給電装置の部品点数が多く、給電装置の大型化やコストアップを招き、給電損失が増加するという問題がある。 In the case of the configuration described in Patent Document 2, although the voltage ripple can be reduced, since a plurality of column circuits (converters) are required, the number of parts of the power supply device is large, which increases the size and cost of the power supply device. There is a problem that power supply loss increases.
　本発明は、従来技術の有するこのような問題点に鑑みてなされたものであり、給電装置の出力における商用周波数成分の電流・電圧リプルを低減できるばかりでなく、給電装置の部品点数を少なくして給電装置の小型化あるいはコストダウンを図ることができ、給電損失を極力低減することができる非接触充電システムの給電装置を提供することを目的としている。 The present invention has been made in view of such problems of the prior art, and not only can reduce the current / voltage ripple of the commercial frequency component in the output of the power supply apparatus, but also reduce the number of parts of the power supply apparatus. Accordingly, it is an object of the present invention to provide a power supply device for a non-contact charging system that can reduce the size or cost of the power supply device and can reduce power supply loss as much as possible.
　上記目的を達成するために、本発明は、受電装置に非接触で給電を行う給電装置であって、力率改善を行って直流電圧を出力する力率改善回路と、力率改善回路の出力直流電圧が与えられると、電荷を蓄積する電解コンデンサと、複数のスイッチング素子を有しており、電解コンデンサを電源として動作し、交流信号を発生するインバータ回路と、インバータ回路からの交流信号が入力されると、受電装置に非接触で給電を行う共振コンデンサ及びインダクタと、複数のスイッチング素子の通電率及び駆動周波数の少なくとも一方を変化させる給電側制御回路とを備えている。 In order to achieve the above object, the present invention provides a power supply device that supplies power to a power receiving device in a non-contact manner, a power factor improvement circuit that outputs a DC voltage by improving the power factor, and an output of the power factor improvement circuit When a DC voltage is applied, it has an electrolytic capacitor that accumulates electric charge and a plurality of switching elements, operates using the electrolytic capacitor as a power source, generates an AC signal, and receives an AC signal from the inverter circuit Then, a resonance capacitor and an inductor that perform power supply to the power receiving device in a non-contact manner, and a power supply side control circuit that changes at least one of the energization rate and the drive frequency of the plurality of switching elements are provided.
　本発明によれば、上記構成により給電装置の電流・電圧リプルを低減できるばかりでなく、給電装置の部品点数を少なくして給電装置の小型化あるいはコストダウンを図ることができ、給電損失を極力低減することができる。 According to the present invention, not only can the current / voltage ripple of the power supply apparatus be reduced by the above-described configuration, but also the power supply apparatus can be reduced in size or cost by reducing the number of parts of the power supply apparatus, and power supply loss can be minimized. Can be reduced.
図１は本発明の一実施の形態に係る非接触充電システムの回路図FIG. 1 is a circuit diagram of a contactless charging system according to an embodiment of the present invention. 図２は図１の非接触充電システムに設けられた給電電力検知部の回路図FIG. 2 is a circuit diagram of a feeding power detection unit provided in the non-contact charging system of FIG. 図３は図１の非接触充電システムにおける各部の波形と比較するための従来の非接触充電システムの各部の波形を示す図FIG. 3 is a diagram showing waveforms of respective parts of a conventional non-contact charging system for comparison with waveforms of respective parts in the non-contact charging system of FIG. 図４は図１の非接触充電システムに設けられた複数のスイッチング素子の通電率を変えた場合の非接触充電システムにおける各部の波形を示す図FIG. 4 is a diagram illustrating waveforms of respective parts in the non-contact charging system when the energization rates of a plurality of switching elements provided in the non-contact charging system of FIG. 1 are changed. 図５は複数のスイッチング素子の駆動周波数を変えた場合の非接触充電システムにおける各部の波形を示す図FIG. 5 is a diagram showing waveforms of respective parts in the non-contact charging system when the driving frequency of a plurality of switching elements is changed. 図６は複数のスイッチング素子の通電率及び駆動周波数の両方を変えた場合の非接触充電システムにおける各部の波形を示す図FIG. 6 is a diagram showing waveforms of respective parts in the non-contact charging system when both the energization rate and the driving frequency of the plurality of switching elements are changed. 図７は図１のインバータ回路４への入力電力が高い場合における、給電装置の要部の動作波形を示す図FIG. 7 is a diagram showing operation waveforms of the main part of the power feeding device when the input power to the inverter circuit 4 of FIG. 1 is high. 図８は図１のインバータ回路４への入力電力が低い場合における、給電装置の要部の動作波形を示す図FIG. 8 is a diagram illustrating operation waveforms of main parts of the power feeding device when the input power to the inverter circuit 4 in FIG. 1 is low. 図９は本発明に係る別の非接触充電システムの回路図FIG. 9 is a circuit diagram of another contactless charging system according to the present invention.
　本発明は、受電装置に非接触で給電を行う給電装置であって、力率改善を行って直流電圧を出力する力率改善回路と、力率改善回路の出力直流電圧が与えられると、電荷を蓄積する電解コンデンサと、複数のスイッチング素子を有しており、電解コンデンサを電源として動作し、交流信号を発生するインバータ回路と、インバータ回路からの交流信号が入力されると、受電装置に非接触で給電を行う共振コンデンサ及びインダクタと、複数のスイッチング素子の通電率及び駆動周波数の少なくとも一方を変化させる給電側制御回路とを備えている。 The present invention is a power supply device that supplies power to a power receiving device in a non-contact manner. The power factor improvement circuit that performs power factor improvement and outputs a DC voltage, and the output DC voltage of the power factor improvement circuit is given a charge. And an inverter circuit that operates using the electrolytic capacitor as a power source to generate an AC signal, and when an AC signal from the inverter circuit is input, the power receiving device is A resonance capacitor and an inductor that perform power supply by contact, and a power supply-side control circuit that changes at least one of a conduction rate and a drive frequency of the plurality of switching elements are provided.
　この構成により、給電装置の電流・電圧リプルを低減できるばかりでなく、給電装置の部品点数を少なくして給電装置の小型化あるいはコストダウンを図ることができ、給電損失を極力低減することができる。 With this configuration, not only can the current / voltage ripple of the power supply apparatus be reduced, but also the number of parts of the power supply apparatus can be reduced to reduce the size or cost of the power supply apparatus, thereby reducing power supply loss as much as possible. .
　また、給電装置は、インダクタに流れる電流、又は該インダクタの電圧を検知する給電電力検知部をさらに備え、給電側制御回路は、給電電力検知部が検知した電流あるいは電圧に基づいて、複数のスイッチング素子の通電率及び駆動周波数の少なくとも一方を変化させてもよい。 The power supply device further includes a power supply power detection unit that detects a current flowing through the inductor or a voltage of the inductor, and the power supply side control circuit performs a plurality of switching operations based on the current or voltage detected by the power supply power detection unit. You may change at least one of the electricity supply rate and drive frequency of an element.
　また、給電側制御回路は、受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、複数のスイッチング素子の基本通電率を決定し、給電電力検知部の出力電圧が高い時には、複数のスイッチング素子の通電率を基本通電率より低く設定し、低い時には複数のスイッチング素子の通電率を基本通電率より高く設定してもかまわない。 Further, the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic energization rate of the plurality of switching elements so as to become the received power command value, and When the output voltage is high, the energization rate of the plurality of switching elements may be set lower than the basic energization rate, and when the output voltage is low, the energization rate of the plurality of switching elements may be set higher than the basic energization rate.
　他にも、給電側制御回路は、受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、複数のスイッチング素子の基本駆動周波数を決定し、給電電力検知部の出力電圧が高い時には、複数のスイッチング素子の駆動周波数を基本駆動周波数より高く設定し、低い時には複数のスイッチング素子の駆動周波数を基本駆動周波数より低く設定することもできる。 In addition, the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic drive frequency of the plurality of switching elements so as to become the received power command value, and detects the power supply power When the output voltage of the unit is high, the driving frequency of the plurality of switching elements can be set higher than the basic driving frequency, and when the output voltage is low, the driving frequency of the plurality of switching elements can be set lower than the basic driving frequency.
　さらに、給電側制御回路は、力率改善回路の出力電圧を検知する出力電圧検知部を有し、出力電圧検知部が検知した力率改善回路の出力電圧に基づいて、複数のスイッチング素子の通電率及び駆動周波数の少なくとも一つを変化させてもかまわない。 Furthermore, the power supply side control circuit has an output voltage detection unit that detects the output voltage of the power factor correction circuit, and based on the output voltage of the power factor correction circuit detected by the output voltage detection unit, the energization of a plurality of switching elements. At least one of the rate and the driving frequency may be changed.
　給電側制御回路は、より具体的には、受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、複数のスイッチング素子の基本通電率を決定し、出力電圧検知部の出力電圧が高い時には、複数のスイッチング素子の通電率を基本通電率より低く設定し、低い時には複数のスイッチング素子の通電率を基本通電率より高く設定する。 More specifically, the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic energization rate of the plurality of switching elements so as to become the received power command value, and outputs the power command value. When the output voltage of the voltage detector is high, the energization rate of the plurality of switching elements is set lower than the basic energization rate, and when low, the energization rate of the plurality of switching elements is set higher than the basic energization rate.
　他にも、給電側制御回路は、受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、複数のスイッチング素子の基本駆動周波数を決定し、出力電圧検知部の出力電圧が高い時には、複数のスイッチング素子の駆動周波数を基本駆動周波数より高く設定し、低い時には複数のスイッチング素子の駆動周波数を基本駆動周波数より低く設定することもできる。 In addition, the power supply side control circuit receives the power command value by wireless communication with the power receiving device side, determines the basic drive frequency of the plurality of switching elements so as to become the received power command value, and detects the output voltage. When the output voltage of the unit is high, the driving frequency of the plurality of switching elements can be set higher than the basic driving frequency, and when the output voltage is low, the driving frequency of the plurality of switching elements can be set lower than the basic driving frequency.
　以下、本発明の実施の形態について、図面を参照しながら説明する。なお、この実施の形態によって本発明が限定されるものではない。 Hereinafter, embodiments of the present invention will be described with reference to the drawings. Note that the present invention is not limited to the embodiments.
（実施の形態１）
　図１は、本実施の形態における非接触充電システムの回路図である。
　図１に示されるように、非接触充電システムは、例えば駐車スペースに設置される給電装置と、例えば電気推進車両に搭載される受電装置とを備えている。非接触充電システムは、給電装置側の構成として、商用電源１、第１の整流回路２、力率改善回路３、インバータ回路４、給電電力検知部５、第１の共振コンデンサ６、第１のインダクタ７及び給電装置側の制御回路１３（以下、単に「制御回路１３」という）と、受電装置側の構成として、第２のインダクタ８、第２の共振コンデンサ９、第２の整流回路１１、負荷（バッテリー）１２、受電装置側の制御回路１４（以下、単に「制御回路１４」という）、及び受電電力検知部１０とを備えている。 (Embodiment 1)
FIG. 1 is a circuit diagram of the non-contact charging system in the present embodiment.
As shown in FIG. 1, the contactless charging system includes, for example, a power feeding device installed in a parking space and a power receiving device mounted in, for example, an electric propulsion vehicle. The non-contact charging system includes a commercial power source 1, a first rectifier circuit 2, a power factor correction circuit 3, an inverter circuit 4, a feed power detection unit 5, a first resonance capacitor 6, and a first power supply device. The inductor 7 and the control circuit 13 on the power feeding device side (hereinafter simply referred to as “control circuit 13”), and the configuration on the power receiving device side include a second inductor 8, a second resonance capacitor 9, a second rectifier circuit 11, A load (battery) 12, a control circuit 14 on the power receiving device side (hereinafter simply referred to as “control circuit 14”), and a received power detection unit 10 are provided.
　以下これらの回路ブロックの構成について説明する。 Hereinafter, the configuration of these circuit blocks will be described.
　まず、力率改善回路３の構成について説明する。力率改善回路３は、商用電源１の力率を改善する回路であって、バイパスコンデンサ２９、第１のチョークコイルであるチョークコイル１５、第１のスイッチング素子１６（本実施の形態においてはＭＯＳＦＥＴが例示）、第１のダイオードであるダイオード１７、及び、平滑コンデンサ（電解コンデンサ）１８を含んでいる。 First, the configuration of the power factor correction circuit 3 will be described. The power factor improving circuit 3 is a circuit for improving the power factor of the commercial power source 1, and includes a bypass capacitor 29, a choke coil 15 serving as a first choke coil, and a first switching element 16 (in the present embodiment, a MOSFET). ), A diode 17 that is a first diode, and a smoothing capacitor (electrolytic capacitor) 18.
　商用電源１は、低周波交流電源である２００Ｖ商用電源であり、ブリッジダイオードと入力フィルタを含む第１の整流回路２の入力端に接続される。第１の整流回路２の高電位側（正極側）出力端子に、バイパスコンデンサ２９の高電位端子とチョークコイル１５の入力側端子が接続される。さらに、チョークコイル１５の出力側端子とダイオード１７のアノード側端子との接続ラインにスイッチング素子１６の高電位側端子（ドレイン）が接続される。第１の整流回路２の低電位側（負極側）出力端子に、バイパスコンデンサ２９の低電位側端子とスイッチング素子１６の低電位側端子（ソース）と平滑コンデンサ１８の低電位側端子が接続される。また、平滑コンデンサ１８の高電位側端子は、ダイオード１７のカソード側端子に接続される。力率改善回路３には、第１の整流回路２の出力電圧が直流電源として入力され、入力された第１の整流回路２の出力電圧はバイパスコンデンサ２９により電圧変動が抑制され、さらにチョークコイル１５とスイッチング素子１６のオン・オフ動作によりそのピーク値より大きいピーク値を有する直流電圧であって任意の電圧に昇圧した電圧が、平滑コンデンサ１８の両端に供給され、平滑される。 The commercial power source 1 is a 200 V commercial power source that is a low-frequency AC power source, and is connected to the input terminal of the first rectifier circuit 2 including a bridge diode and an input filter. The high potential terminal (positive electrode side) output terminal of the first rectifier circuit 2 is connected to the high potential terminal of the bypass capacitor 29 and the input terminal of the choke coil 15. Further, the high potential side terminal (drain) of the switching element 16 is connected to a connection line between the output side terminal of the choke coil 15 and the anode side terminal of the diode 17. A low potential side terminal of the bypass capacitor 29, a low potential side terminal (source) of the switching element 16, and a low potential side terminal of the smoothing capacitor 18 are connected to the low potential side (negative electrode side) output terminal of the first rectifier circuit 2. The The high potential side terminal of the smoothing capacitor 18 is connected to the cathode side terminal of the diode 17. The power factor correction circuit 3 is supplied with the output voltage of the first rectifier circuit 2 as a DC power supply, and the voltage fluctuation of the input output voltage of the first rectifier circuit 2 is suppressed by the bypass capacitor 29. 15 and the switching element 16 are turned on and off, and a DC voltage having a peak value larger than the peak value and boosted to an arbitrary voltage is supplied across the smoothing capacitor 18 and smoothed.
　本実施の形態においては、力率改善回路３を高周波動作させ力率改善効果を高めるために、スイッチング速度の速いＭＯＳＦＥＴをスイッチング素子１６の典型例として使用している。ＭＯＳＦＥＴに逆向きにダイオードが付帯するが、このダイオードが無くても本実施の形態の基本動作に何ら影響を与えないため、図には記載していない。平滑コンデンサ１８の出力電圧はインバータ回路４の入力端子間に供給される。 In the present embodiment, a MOSFET having a high switching speed is used as a typical example of the switching element 16 in order to increase the power factor improvement effect by operating the power factor improvement circuit 3 at a high frequency. Although a diode is attached to the MOSFET in the opposite direction, it is not shown in the figure because it does not affect the basic operation of the present embodiment even without this diode. The output voltage of the smoothing capacitor 18 is supplied between the input terminals of the inverter circuit 4.
　インバータ回路４の入力端子は力率改善回路３の出力端子、つまり平滑コンデンサ１８の両端に接続される。平滑コンデンサ１８の両端には、スイッチング素子（第２及び第３のスイッチング素子）１９，２０の直列接続体とスイッチング素子（第４及び第５のスイッチング素子）２４，２６の直列接続体が、並列に接続される。 The input terminal of the inverter circuit 4 is connected to the output terminal of the power factor correction circuit 3, that is, to both ends of the smoothing capacitor 18. At both ends of the smoothing capacitor 18, a series connection body of switching elements (second and third switching elements) 19 and 20 and a series connection body of switching elements (fourth and fifth switching elements) 24 and 26 are parallel. Connected to.
　スイッチング素子１９，２０には、それぞれダイオード（第２及び第３のダイオード）２１、２２が逆並列に接続される（スイッチング素子の高電位側端子（コレクタ）とダイオードのカソード側端子が接続される）。また、スイッチング素子２０（スイッチング素子１９であってもよい）に並列にスナバコンデンサ２３が接続される。 Diodes (second and third diodes) 21 and 22 are connected in antiparallel to the switching elements 19 and 20, respectively (the high potential side terminal (collector) of the switching element and the cathode side terminal of the diode are connected). ). Further, a snubber capacitor 23 is connected in parallel to the switching element 20 (which may be the switching element 19).
　同様に、スイッチング素子２４，２６には、それぞれダイオード（第４及び第５のダイオード）２５，２７が逆並列に接続される（スイッチング素子の高電位側端子（コレクタ）とダイオードのカソード側端子が接続される）。また、スイッチング素子２６（スイッチング素子２４であってもよい）に並列にスナバコンデンサ２８が接続される。 Similarly, diodes (fourth and fifth diodes) 25 and 27 are connected in antiparallel to the switching elements 24 and 26 (the high potential side terminal (collector) of the switching element and the cathode side terminal of the diode are connected to each other). Connected). A snubber capacitor 28 is connected in parallel to the switching element 26 (which may be the switching element 24).
　さらに、スイッチング素子１９とスイッチング素子２０の接続ラインと、スイッチング素子２４とスイッチング素子２６の接続ラインに、第１の共振コンデンサ６と第１のインダクタ７と給電電力検知部５の直列接続体が接続される。 Further, a series connection body of the first resonance capacitor 6, the first inductor 7, and the feeding power detection unit 5 is connected to the connection line of the switching element 19 and the switching element 20 and the connection line of the switching element 24 and the switching element 26. Is done.
　第２のインダクタ８は、例えば電気推進車両の移動に伴い、第１のインダクタ７と対向するように配置される。また、第２のインダクタ８の高電位側に第２の共振コンデンサ９が接続され、第２のインダクタ８の低電位側と第２の共振コンデンサ９は平滑フィルタを内包する第２の整流回路１１に接続され、第２の整流回路１１の高電位側に受電電力検知部１０が接続され、受電電力検知部１０と第２の整流回路１１の低電位側に負荷（バッテリー）１２が接続される。 The second inductor 8 is disposed so as to face the first inductor 7 as the electric propulsion vehicle moves, for example. A second resonant capacitor 9 is connected to the high potential side of the second inductor 8, and the low potential side of the second inductor 8 and the second resonant capacitor 9 include a second rectifier circuit 11 including a smoothing filter. The received power detection unit 10 is connected to the high potential side of the second rectifier circuit 11, and the load (battery) 12 is connected to the low potential side of the received power detection unit 10 and the second rectifier circuit 11. .
　本実施の形態における給電電力検知部５は、図２に示されるように、電流検知部３０、電圧検知部３１、電力演算部３２で構成されている。ただし、電流または電圧のいずれか一方で給電電力を推定できる場合は、電流検知部３０と電圧検知部３１のいずれか一方だけでもよい。本実施の形態のように、給電電力検知部５を第１のインダクタ７と第１の共振コンデンサ６の直列共振回路に直列接続する場合、電流と電圧には相関があるためいずれか一方だけの検知で給電電力を推定できる。 As shown in FIG. 2, the supply power detection unit 5 in the present embodiment includes a current detection unit 30, a voltage detection unit 31, and a power calculation unit 32. However, only one of the current detection unit 30 and the voltage detection unit 31 may be used when the power supply power can be estimated using either the current or the voltage. As in the present embodiment, when the feed power detection unit 5 is connected in series to the series resonance circuit of the first inductor 7 and the first resonance capacitor 6, only one of the current and voltage is correlated. The power supply can be estimated by detection.
　なお、受電電力検知部１０については詳述しないが、受電電力検知部１０は、給電電力検知部５の構成と同じであってもよい。 Although the received power detection unit 10 is not described in detail, the received power detection unit 10 may have the same configuration as the power supply power detection unit 5.
　次に、制御回路１３の構成について説明する。制御回路１３は、無線通信により、制御回路１４より電力指令値を受信する。制御回路１３は、給電電力検知部５によって検知する給電電力と、受信した電力指令値とを比較し、電力指令値が得られるようにインバータ回路４のスイッチング素子１９，２０とスイッチング素子２４，２６、及び、力率改善回路３のスイッチング素子１６を駆動する。なお、力率改善回路３のスイッチング素子１６の制御には専用の制御ＩＣを用いてもよい。 Next, the configuration of the control circuit 13 will be described. The control circuit 13 receives a power command value from the control circuit 14 by wireless communication. The control circuit 13 compares the supplied power detected by the supplied power detection unit 5 with the received power command value, and the switching elements 19 and 20 and the switching elements 24 and 26 of the inverter circuit 4 are obtained so that the power command value is obtained. And the switching element 16 of the power factor correction circuit 3 is driven. A dedicated control IC may be used for controlling the switching element 16 of the power factor correction circuit 3.
　制御回路１４は、受電電力検知部１０によって検知するバッテリー１２の残電圧に応じて電力指令値を決定し、無線通信により制御回路１３に送信する。また、給電装置動作中に受電電力検知部１０によって受電電力を検知し、負荷（バッテリー）１２に過電流や過電圧がかからないように、制御回路１４は制御回路１３への電力指令値を変更する。 The control circuit 14 determines a power command value according to the remaining voltage of the battery 12 detected by the received power detection unit 10 and transmits it to the control circuit 13 by wireless communication. Further, the received power is detected by the received power detection unit 10 during operation of the power supply apparatus, and the control circuit 14 changes the power command value to the control circuit 13 so that the load (battery) 12 is not overcurrent or overvoltage.
　また、本実施の形態１の負荷１２には、電気推進車両用のバッテリーを用いている。バッテリー充電は、バッテリーの残電圧以上の電圧を供給して充電するが、給電電圧がバッテリー残電圧を超えると、急激に充電電流が流れる。このことは給電装置からみた負荷インピーダンスがバッテリー残電圧や給電電圧によって大きく変動することを意味している。 Moreover, a battery for an electric propulsion vehicle is used as the load 12 of the first embodiment. Battery charging is performed by supplying a voltage equal to or higher than the remaining voltage of the battery, but when the power supply voltage exceeds the remaining battery voltage, a charging current suddenly flows. This means that the load impedance as viewed from the power supply device varies greatly depending on the remaining battery voltage and the power supply voltage.
　以上のように構成された非接触充電システムの動作を以下説明するが、上述した特許文献１に記載の電力伝送システムにおける各部の電圧波形、電流波形等を図３を参照しながらまず説明する。ただし、本発明に係る非接触充電システムと、特許文献１に記載の電力伝送システムは、当然のことながら回路構成は異なっているが、図３において、「第１の整流回路２の出力電圧」、「力率改善回路３の出力電圧」等と記載したのは、特許文献１に記載の電力伝送システムにおける対応部位の出力電圧等を示すためである。 The operation of the non-contact charging system configured as described above will be described below. First, the voltage waveform and current waveform of each part in the power transmission system described in Patent Document 1 will be described with reference to FIG. However, although the circuit configuration of the contactless charging system according to the present invention and the power transmission system described in Patent Document 1 is naturally different, in FIG. 3, “output voltage of the first rectifier circuit 2”. “The output voltage of the power factor correction circuit 3” is described to indicate the output voltage of the corresponding part in the power transmission system described in Patent Document 1.
　図３（ａ）は、商用電源１の交流電圧波形を示す模式図であり、図３（ｂ）は、直流電源の出力電圧波形、すなわち第１の整流回路２の出力電圧波形を示す模式図である。この電圧は、力率改善回路３に入力され、昇圧された後に平滑コンデンサ１８に出力される。図３（ｃ）は、平滑コンデンサ１８に印加される波形、すなわち力率改善回路３の出力電圧波形であり、かつインバータ回路４の入力電圧波形を示す模式図である。図３（ｄ）は、第１のインダクタ７に発生する高周波電流波形を示す模式図であり、図３（ｅ）は、給電装置から受電装置に給電される電力波形を示す模式図である。図３（ｆ）は、第２の整流回路１１の出力電流波形、すなわち負荷２の入力電流波形を示す模式図である。また、図３（ｇ）及び（ｈ）は、それぞれ通電率（デューティ比）及び動作周波数を示す模式図である。 FIG. 3A is a schematic diagram showing an AC voltage waveform of the commercial power source 1, and FIG. 3B is a schematic diagram showing an output voltage waveform of the DC power source, that is, an output voltage waveform of the first rectifier circuit 2. It is. This voltage is input to the power factor correction circuit 3, boosted, and then output to the smoothing capacitor 18. FIG. 3C is a schematic diagram showing a waveform applied to the smoothing capacitor 18, that is, an output voltage waveform of the power factor correction circuit 3 and an input voltage waveform of the inverter circuit 4. FIG. 3D is a schematic diagram showing a high-frequency current waveform generated in the first inductor 7, and FIG. 3E is a schematic diagram showing a power waveform fed from the power feeding device to the power receiving device. FIG. 3F is a schematic diagram illustrating an output current waveform of the second rectifier circuit 11, that is, an input current waveform of the load 2. 3 (g) and 3 (h) are schematic diagrams showing an energization rate (duty ratio) and an operating frequency, respectively.
　一方、図４は、本発明に係る非接触充電システムの各部の電圧波形、電流波形等を示しており、図４（ａ）～（ｈ）は、図３（ａ）～（ｈ）にそれぞれ対応している。 On the other hand, FIG. 4 shows voltage waveforms, current waveforms, etc. of each part of the non-contact charging system according to the present invention. FIGS. 4 (a) to 4 (h) are respectively shown in FIGS. 3 (a) to 3 (h). It corresponds.
　まず、力率改善回路３の動作について説明する。
　図４（ａ）に示される商用電源１は第１の整流回路２により全波整流され、図４（ｂ）の電圧波形に示されるような直流電源が形成される。この直流電源は、力率改善回路３の入力端子間に供給される。力率改善回路３は、この直流電源電圧の瞬時値の大きさが平滑コンデンサ１８の電圧よりも小さい場合に力率改善回路３に含まれるダイオード１７及び第１の整流回路２のブリッジダイオードがターンオンできずに入力電流波形が歪み、力率が著しく低くなる。その際に、制御回路１３は、スイッチング素子１６をターンオン・オフさせることにより力率を改善する。 First, the operation of the power factor correction circuit 3 will be described.
The commercial power source 1 shown in FIG. 4A is full-wave rectified by the first rectifier circuit 2 to form a DC power source as shown in the voltage waveform of FIG. This DC power supply is supplied between the input terminals of the power factor correction circuit 3. In the power factor correction circuit 3, the diode 17 included in the power factor correction circuit 3 and the bridge diode of the first rectifier circuit 2 are turned on when the instantaneous value of the DC power supply voltage is smaller than the voltage of the smoothing capacitor 18. If not, the input current waveform is distorted and the power factor is significantly reduced. At that time, the control circuit 13 improves the power factor by turning the switching element 16 on and off.
　第１のスイッチング素子１６がターンオンしている状態では、商用電源１からチョークコイル１５にエネルギーが蓄えられており、その後、スイッチング素子１６がターンオフし、チョークコイル１５に蓄えられたエネルギーがダイオード１７を介して、平滑コンデンサ１８に供給される。これにより、商用電源１からチョークコイル１５を介して入力電流が流れるようになり、商用電源１側から歪んだ入力電流を流さないようにする。また、本実施の形態では、力率改善回路３は、力率改善機能だけでなく、昇圧機能を同時に有する。このため、図４（ｃ）に示されるように、平滑コンデンサ１８の電圧は、そのピーク値が商用電源１のピーク値すなわち直流電源のピーク値である力率改善回路３の入力電圧のピーク値より高い電圧となり、平滑コンデンサ１８を介してインバータ回路４に供給される。 In a state where the first switching element 16 is turned on, energy is stored in the choke coil 15 from the commercial power source 1, and then the switching element 16 is turned off, and the energy stored in the choke coil 15 causes the diode 17 to be turned on. To the smoothing capacitor 18. As a result, an input current flows from the commercial power source 1 through the choke coil 15 and a distorted input current is prevented from flowing from the commercial power source 1 side. In the present embodiment, the power factor correction circuit 3 has not only a power factor correction function but also a boosting function at the same time. For this reason, as shown in FIG. 4C, the voltage of the smoothing capacitor 18 is the peak value of the input voltage of the power factor correction circuit 3 whose peak value is the peak value of the commercial power source 1, that is, the peak value of the DC power source. The voltage becomes higher and is supplied to the inverter circuit 4 through the smoothing capacitor 18.
　なお、図３（ａ）～（ｃ）及び図４（ａ）～（ｃ）を比較すれば分かるように、特許文献１に記載の電力伝送システムと本発明に係る非接触充電システムにおける商用電源１の交流電圧波形、第１の整流回路２の出力電圧波形、及び、力率改善回路３の出力電圧波形に大きな違いはない。 As can be seen by comparing FIGS. 3 (a) to 3 (c) and FIGS. 4 (a) to 4 (c), the commercial power supply in the power transmission system described in Patent Document 1 and the non-contact charging system according to the present invention. There is no significant difference between the AC voltage waveform of 1, the output voltage waveform of the first rectifier circuit 2, and the output voltage waveform of the power factor correction circuit 3.
　次に、インバータ回路４の動作について説明する。
　図４（ｃ）に示される力率改善回路３の出力端間に接続された平滑コンデンサ１８の両端に出力され平滑された直流電圧はインバータ回路４に供給される。インバータ回路４は、スイッチング素子１９，２０のオン・オフ、及び、スイッチング素子２４，２６のオン・オフによって、第１の共振コンデンサ６と第１のインダクタ７に、図４（ｄ）に示されるように所定の周波数の高周波電流を発生させる。 Next, the operation of the inverter circuit 4 will be described.
The smoothed DC voltage output to both ends of the smoothing capacitor 18 connected between the output ends of the power factor correction circuit 3 shown in FIG. 4C is supplied to the inverter circuit 4. The inverter circuit 4 is shown in FIG. 4D by the first resonant capacitor 6 and the first inductor 7 depending on whether the switching elements 19 and 20 are turned on / off and the switching elements 24 and 26 are turned on / off. Thus, a high-frequency current having a predetermined frequency is generated.
　スイッチング素子１９，２０のオン・オフ制御、及び、スイッチング素子２４，２６のオン・オフ制御は、制御回路１３が各スイッチング素子１９，２０，２４，２６のゲートにオン信号を加えることで行われる。 The on / off control of the switching elements 19 and 20 and the on / off control of the switching elements 24 and 26 are performed by the control circuit 13 applying an on signal to the gates of the switching elements 19, 20, 24 and 26. .
　図７及び図８は、高入力電力時と低入力電力時における拡大したインバータ回路４の動作波形をそれぞれ示しており、（ａ）（ｃ）（ｄ）は、スイッチング素子１９，２６及びダイオード２１，２７に流れる電流、スイッチング素子１９，２６の電圧、及び、スイッチング素子１９，２６のゲート電圧をそれぞれ示しており、（ｂ）（ｅ）は、スイッチング素子２０，２４及びダイオード２２，２５に流れる電流、及び、スイッチング素子２０，２４のゲート電圧をそれぞれ示している。また、（ｆ）は第１のインダクタ７に流れる電流を示しており、図中のＴｏｎ期間中はスイッチング素子１９，２６及びダイオード２１，２７に流れる電流が流れ、１周期の残り（図中のＴ－Ｔｄ－Ｔｏｎ）期間中はスイッチング素子２０，２４及びダイオード２２，２５に流れる電流が流れる。後述するデッドタイムＴｄ期間中は第１のインダクタ７と第１の共振コンデンサ６とスナバコンデンサ２３，２８の共振電流が流れる。 7 and 8 show enlarged operation waveforms of the inverter circuit 4 at high input power and low input power, respectively. (A), (c), and (d) are the switching elements 19 and 26 and the diode 21, respectively. , 27, the voltage of the switching elements 19, 26, and the gate voltage of the switching elements 19, 26, respectively, (b) and (e) flow in the switching elements 20, 24 and the diodes 22, 25. The current and the gate voltage of the switching elements 20 and 24 are shown. Further, (f) shows the current flowing through the first inductor 7, and the current flowing through the switching elements 19 and 26 and the diodes 21 and 27 flows during the Ton period in the figure, and the remainder of one cycle (in the figure) During the period of (T−Td−Ton), the current flowing through the switching elements 20 and 24 and the diodes 22 and 25 flows. During the dead time Td described later, the resonance current of the first inductor 7, the first resonance capacitor 6, and the snubber capacitors 23 and 28 flows.
　図７及び図８に示されるように、直列接続された二つのスイッチング素子１９，２０を排他的に通電し、これら二つのスイッチング素子１９，２０に並列に直列接続された二つのスイッチング素子２４，２６を、スイッチング素子１９，２０の駆動信号位相をずらして排他的に通電している。 As shown in FIGS. 7 and 8, the two switching elements 19 and 20 connected in series are exclusively energized, and the two switching elements 24 and 24 connected in series to the two switching elements 19 and 20 are connected in parallel. 26 is energized exclusively by shifting the drive signal phase of the switching elements 19 and 20.
　すなわち、スイッチング素子１９とスイッチング素子２６は同期してオン・オフを繰り返し、スイッチング素子１９とスイッチング素子２６がオンのときに、スイッチング素子２０とスイッチング素子２４がオフし、スイッチング素子１９とスイッチング素子２６がオフのときに、スイッチング素子２０とスイッチング素子２４がオンすることで、スイッチング素子２０とスイッチング素子２４は同期してオン・オフを繰り返す。 That is, the switching element 19 and the switching element 26 are repeatedly turned on and off in synchronization. When the switching element 19 and the switching element 26 are on, the switching element 20 and the switching element 24 are turned off, and the switching element 19 and the switching element 26 are turned on. When the switching element 20 and the switching element 24 are turned on, the switching element 20 and the switching element 24 are repeatedly turned on and off in synchronization.
　なお、後述するように、スイッチング素子１９とスイッチング素子２０が同時にオンにならないように、またスイッチング素子２４とスイッチング素子２６が同時にオンにならないように、スイッチング素子１９，２４のオン期間とスイッチング素子２０，２６のオン期間は重ならないようにデッドタイムＴｄが設定されている。 As will be described later, the ON period of the switching elements 19 and 24 and the switching element 20 are set so that the switching element 19 and the switching element 20 are not turned on simultaneously, and so that the switching element 24 and the switching element 26 are not turned on simultaneously. 26, the dead time Td is set so as not to overlap.
　さらに詳述すると、スイッチング素子１９，２６がオンしている状態からオフすると、第１のインダクタ７と第１の共振コンデンサ６とスナバコンデンサ２３の共振による緩やかな傾きでスナバコンデンサ２３が放電するため、スイッチング素子１９，２６は零ボルトスイッチング（ＺＶＳ）ターンオフ動作を実現する。また、このときスナバコンデンサ２８が充電され、スナバコンデンサ２３が放電しきると、ダイオード２２，２５がオンし、ダイオード２２，２５がオンしている期間中にスイッチング素子２０，２４のゲートにオン信号を加え待機すると、第１のインダクタ７の共振電流の向きが反転しダイオード２２がターンオフしてスイッチング素子２０，２４に電流が転流し、スイッチング素子２０，２４はＺＶＳ＆零電流スイッチング（ＺＣＳ）ターンオン動作を実現する。 More specifically, when the switching elements 19 and 26 are turned off from the on state, the snubber capacitor 23 is discharged with a gentle inclination due to resonance of the first inductor 7, the first resonance capacitor 6, and the snubber capacitor 23. The switching elements 19, 26 realize a zero volt switching (ZVS) turn-off operation. At this time, when the snubber capacitor 28 is charged and the snubber capacitor 23 is completely discharged, the diodes 22 and 25 are turned on, and an on signal is sent to the gates of the switching elements 20 and 24 during the period in which the diodes 22 and 25 are on. In addition, when waiting, the direction of the resonance current of the first inductor 7 is reversed, the diode 22 is turned off, the current is commutated to the switching elements 20 and 24, and the switching elements 20 and 24 perform the ZVS & zero current switching (ZCS) turn-on operation. Realize.
　次に、スイッチング素子２０，２４がオンしている状態からオフすると、第１のインダクタ７と第１の共振コンデンサ６とスナバコンデンサ２８の共振による緩やかな傾きでスナバコンデンサ２８が放電するため、スイッチング素子２０，２４はＺＶＳターンオフ動作を実現する。また、このときスナバコンデンサ２３が充電され、スナバコンデンサ２８が放電しきると、ダイオード２１，２７がオンし、ダイオード２１，２７がオンしている期間中にスイッチング素子１９，２６のゲートにオン信号を加え待機すると、第１のインダクタ７の共振電流の向きが反転しダイオード２７がターンオフしてスイッチング素子１９，２６に電流が転流し、スイッチング素子１９，２６はＺＶＳ＆零電流スイッチング（ＺＣＳ）ターンオン動作を実現する。以上がインバータ回路４の動作である。 Next, when the switching elements 20 and 24 are turned off from the on state, the snubber capacitor 28 is discharged with a gentle slope due to resonance of the first inductor 7, the first resonant capacitor 6, and the snubber capacitor 28. The elements 20 and 24 realize a ZVS turn-off operation. At this time, when the snubber capacitor 23 is charged and the snubber capacitor 28 is completely discharged, the diodes 21 and 27 are turned on, and an on signal is sent to the gates of the switching elements 19 and 26 during the period in which the diodes 21 and 27 are on. In addition, when waiting, the direction of the resonance current of the first inductor 7 is reversed, the diode 27 is turned off, the current is commutated to the switching elements 19 and 26, and the switching elements 19 and 26 perform the ZVS & zero current switching (ZCS) turn-on operation. Realize. The above is the operation of the inverter circuit 4.
　本実施の形態では、スイッチング素子１９，２６及びスイッチング素子２０，２４は、平滑コンデンサ１８を短絡しないようにデッドタイムＴｄ（例えば、約２μｓ）を設けて、交互にオン・オフさせている。また、図４（ｈ）に示されるように、スイッチング素子１９，２０，２４，２６の駆動（動作）周波数を一定にして、図４（ｇ）に示されるように、通電率（デューティ比）を制御することで高周波電力を制御している。なお、ここでいう「通電率」とは、図７及び図８に示されるように、スイッチング素子１９，２６（あるいはスイッチング素子２０，２４）のオン・オフの１周期に要する時間に対するスイッチング素子１９，２６（あるいはスイッチング素子２０，２４）のオン時間の比として定義している。 In this embodiment, the switching elements 19 and 26 and the switching elements 20 and 24 are alternately turned on / off by providing a dead time Td (for example, about 2 μs) so as not to short-circuit the smoothing capacitor 18. Further, as shown in FIG. 4 (h), the drive (operation) frequency of the switching elements 19, 20, 24, and 26 is made constant, and as shown in FIG. 4 (g), the energization rate (duty ratio). The high frequency power is controlled by controlling. Note that the “energization rate” here means the switching element 19 with respect to the time required for one cycle of ON / OFF of the switching elements 19 and 26 (or the switching elements 20 and 24), as shown in FIGS. , 26 (or switching elements 20, 24).
　図３、図４、図７及び図８を参照しながらさらに詳述すると、図３に示される従来例の場合、図３（ｃ）に示されるように、インバータ回路４には１２０Ｈｚの電圧リプルを含む入力電圧が印加され、図３（ｄ）に示されるように、第１のインダクタ７の電流には電流リプルが発生する。したがって、給電電力は、図３（ｅ）に示されるように変動し、図３（ｆ）に示されるように、負荷１２の入力電流には１２０Ｈｚの電流リプルが発生することになる。 More specifically with reference to FIG. 3, FIG. 4, FIG. 7 and FIG. 8, in the case of the conventional example shown in FIG. 3, the inverter circuit 4 has a voltage ripple of 120 Hz as shown in FIG. As shown in FIG. 3D, a current ripple is generated in the current of the first inductor 7. Therefore, the feed power fluctuates as shown in FIG. 3E, and a 120-Hz current ripple occurs in the input current of the load 12 as shown in FIG.
　一方、本発明においては、図４（ｃ）に示される電圧リプルを含む入力電圧がインバータ回路４に印加されると、給電電力検知部５で検知した第１のインダクタ７の電流（あるいは第１のインダクタ７が発生する磁界）が略一定になるように、スイッチング素子１９，２０，２４，２６の通電率（デューティ比）を制御している。また、このように制御することで、第１のインダクタ７を流れる電流及び給電装置側の給電電力は、図４（ｄ）及び（ｅ）に示されるように、略一定になる。 On the other hand, in the present invention, when the input voltage including the voltage ripple shown in FIG. 4C is applied to the inverter circuit 4, the current (or the first current) of the first inductor 7 detected by the feed power detector 5. The energization rates (duty ratios) of the switching elements 19, 20, 24, and 26 are controlled so that the magnetic field generated by the inductor 7 becomes substantially constant. Moreover, by controlling in this way, the electric current which flows through the 1st inductor 7, and the electric power feeding electric power by the side of an electric power feeder become substantially constant, as FIG.4 (d) and (e) show.
　スイッチング素子１９，２０，２４，２６の通電率制御をさらに詳しく説明する。制御回路１３は、制御回路１４より無線通信で送信される電力指令値を受信すると、給電電力検知部５によって検知する給電電力と、受信した電力指令値とを比較し、電力指令値が得られるように、インバータ回路４のスイッチング素子１９，２０と、スイッチング素子２４，２６と、力率改善回路３のスイッチング素子１６とを駆動する。本実施の形態におけるインバータ回路４は、スイッチング素子１９，２０とスイッチング素子２４，２６との通電率を高くすると給電電力が大きくなるように、該通電率を設定しており、制御回路１３は、電力指令値となるように基本通電率を決定する。制御回路１３はさらに、給電電力検知部５で検知した電流に電流リプルが発生すると、力率改善回路３の出力電圧が高い時に通電率を基本通電率より低く、低電圧時には通電率を基本通電率より高く設定されるように（図４（ｇ）参照）制御する。これによって第１のインダクタ７に流れる電流及び給電電力を略一定にすることができる。基本通電率はこのような給電電力を略一定にする通電率制御を行った際に、給電電力が電力指令値と一致するように選定される。 The energization rate control of the switching elements 19, 20, 24, and 26 will be described in more detail. When receiving the power command value transmitted from the control circuit 14 by wireless communication, the control circuit 13 compares the power supply power detected by the power supply power detection unit 5 with the received power command value to obtain a power command value. Thus, the switching elements 19 and 20 of the inverter circuit 4, the switching elements 24 and 26, and the switching element 16 of the power factor correction circuit 3 are driven. In the inverter circuit 4 in the present embodiment, the energization rate is set so that the power supply power increases when the energization rates of the switching elements 19 and 20 and the switching elements 24 and 26 are increased. The basic energization rate is determined so that the power command value is obtained. Further, when a current ripple occurs in the current detected by the feed power detection unit 5, the control circuit 13 lowers the energization rate below the basic energization rate when the output voltage of the power factor correction circuit 3 is high, and sets the energization rate at the low voltage. Control is performed so that it is set higher than the rate (see FIG. 4G). As a result, the current flowing through the first inductor 7 and the feed power can be made substantially constant. The basic energization rate is selected so that the supply power matches the power command value when the energization rate control is performed so that the supply power is substantially constant.
　本実施の形態における基本通電率は50%で出力最大となり、基本通電率は0～50%の範囲で可変される。従って、通電率制御で電力指令値となる給電電力が得られない場合は、制御回路１３は、駆動周波数を変化させる。給電電力が低い場合には駆動周波数がさらに低く設定され、制御回路１３は基本通電率を再設定する。反対に給電電力が高い場合には駆動周波数がさらに高く設定され、基本通電率も再設定される。このように電力指令値と給電電力が一致するように駆動周波数を変化させて基本通電率を設定することで、より給電電力を電力指令値に一致させることが可能となり、さらに給電電力を略一定にして安定した給電を行うことが可能となる。 ¡In this embodiment, the basic energization rate is 50% and the output is maximum, and the basic energization rate is variable in the range of 0 to 50%. Therefore, when the feed power that becomes the power command value cannot be obtained by the energization rate control, the control circuit 13 changes the drive frequency. When the feed power is low, the drive frequency is set lower, and the control circuit 13 resets the basic energization rate. On the contrary, when the feed power is high, the drive frequency is set higher and the basic energization rate is also reset. In this way, by setting the basic energization rate by changing the drive frequency so that the power command value and the power supply power match, it becomes possible to make the power supply power more consistent with the power command value, and the power supply power is substantially constant. Thus, stable power feeding can be performed.
　すなわち、図７及び図８に示されるように、高入力電力時は低入力電力時に比べ、通電率が大きくなるように設定され（例えば、高入力電力時は５０％、４０％等、低入力電力時は３０％、２０％等）、図４（ｃ）に示されるように、力率改善回路３の出力電圧に電圧リプルがある場合、高電圧時には通電率が小さく、低電圧時には通電率が大きく設定されるように（図４（ｇ）参照）、制御回路１３がスイッチング素子１９，２０，２４，２６を制御することで、第１のインダクタ７に流れる電流及び給電電力を略一定にすることができる。 That is, as shown in FIGS. 7 and 8, the energization rate is set to be higher at high input power than at low input power (for example, 50%, 40%, etc. As shown in FIG. 4C, when the output voltage of the power factor correction circuit 3 has a voltage ripple, the energization rate is small at high voltage and the energization rate at low voltage. Is set to be large (see FIG. 4G), the control circuit 13 controls the switching elements 19, 20, 24, and 26, so that the current flowing through the first inductor 7 and the feeding power are made substantially constant. can do.
　充電動作は、充電開始時に制御回路１４が受電電力検知部１０によって検知するバッテリーの残電圧に応じて充電電流、電圧、電力などの指令値を決定し、無線通信により制御回路１３に送信する。また充電中においても充電電流、電圧、電力などの情報を無線通信により制御回路１３に送信し、制御回路１３は受信した充電電流、電圧、電力などの情報にも基づいて制御する。 In the charging operation, the control circuit 14 determines command values such as charging current, voltage, and power according to the remaining voltage of the battery detected by the received power detection unit 10 at the start of charging, and transmits the command value to the control circuit 13 by wireless communication. Even during charging, information such as charging current, voltage, and power is transmitted to the control circuit 13 by wireless communication, and the control circuit 13 performs control based on the received information such as charging current, voltage, and power.
　また、受電装置でも第２のインダクタ８と第２の共振コンデンサ９を共振させることで、第１のインダクタ７と第２のインダクタ８の間の電力伝送効率を高めることができる。これは第２のインダクタ８のうち、第１のインダクタ７と磁気結合できない漏れインダクタンスによるインピーダンス成分を第２の共振コンデンサ９で打ち消すことにより、２次側のインピーダンスが下がり、電力を伝送しやすくなるとも説明できる。なお、第２の共振コンデンサ９は無くても本発明に影響はない。 In the power receiving device, the power transmission efficiency between the first inductor 7 and the second inductor 8 can be increased by causing the second inductor 8 and the second resonant capacitor 9 to resonate. This is because, when the impedance component due to the leakage inductance that cannot be magnetically coupled to the first inductor 7 among the second inductors 8 is canceled by the second resonance capacitor 9, the impedance on the secondary side is lowered and it becomes easy to transmit power. Can also explain. Even if the second resonant capacitor 9 is not provided, the present invention is not affected.
　制御回路１３は電力指令値を受信完了すると、上述した動作によって電力指令値と給電電力検知部５の検知結果が一致するように力率改善回路３とインバータ回路４を駆動・制御する。 When the reception of the power command value is completed, the control circuit 13 drives and controls the power factor correction circuit 3 and the inverter circuit 4 so that the power command value and the detection result of the feed power detection unit 5 are matched by the above-described operation.
　なお、本実施の形態においては、給電電力検知部５で検知した電流値に基づいてスイッチング素子１９，２０，２４，２６を制御するようにしたが、給電電力検知部５で給電電圧を検知し、検知した電圧値が略一定になるようにスイッチング素子１９，２０，２４，２６を制御することにより、第１のインダクタ７に印加される電圧及び給電電力を略一定にすることができる。 In the present embodiment, the switching elements 19, 20, 24, and 26 are controlled based on the current value detected by the power supply power detection unit 5, but the power supply voltage detection unit 5 detects the power supply voltage. By controlling the switching elements 19, 20, 24, and 26 so that the detected voltage value becomes substantially constant, the voltage applied to the first inductor 7 and the feeding power can be made substantially constant.
（実施の形態２）
　図１の回路構成については、実施の形態１と同じなので、その説明は省略する。 (Embodiment 2)
Since the circuit configuration of FIG. 1 is the same as that of the first embodiment, description thereof is omitted.
　実施の形態１においては、スイッチング素子１９，２０，２４，２６の駆動周波数を一定にして、通電率を制御することで高周波電力を制御するようにしたが、本実施の形態においては、図５に示されるように、通電率を一定にして、スイッチング素子１９，２０，２４，２６の駆動周波数を制御することにより高周波電力を制御している。 In the first embodiment, the driving frequency of the switching elements 19, 20, 24, and 26 is made constant, and the energization rate is controlled to control the high frequency power, but in the present embodiment, FIG. As shown in FIG. 4, the high-frequency power is controlled by controlling the drive frequency of the switching elements 19, 20, 24, and 26 while keeping the energization rate constant.
　本実施の形態においては、駆動周波数を低くすると給電電力が大きくなるように駆動周波数を設定しており（給電電力最大となる周波数よりも高い周波数で動作させる）、給電電力検知部５で検知した電流に電流リプルが発生すると、力率改善回路３の出力電圧が高電圧時には駆動周波数が高く、低電圧時には駆動周波数が低く設定されるように（図５（ｈ）参照）、制御回路１３がスイッチング素子１９，２０，２４，２６の駆動周波数を制御することで、第１のインダクタ７に流れる電流及び給電電力を略一定にすることができる。 In the present embodiment, the driving frequency is set so that the feeding power increases when the driving frequency is lowered (the operation is performed at a frequency higher than the frequency at which the feeding power becomes maximum), and is detected by the feeding power detection unit 5. When a current ripple occurs in the current, the control circuit 13 is set so that the drive frequency is set high when the output voltage of the power factor correction circuit 3 is high and the drive frequency is set low when the output voltage is low (see FIG. 5H). By controlling the driving frequency of the switching elements 19, 20, 24, and 26, the current flowing through the first inductor 7 and the feeding power can be made substantially constant.
　スイッチング素子１９，２０，２４，２６の駆動周波数制御をさらに詳しく説明する。制御回路１３は、制御回路１４より無線通信で送信される電力指令値を受信すると、給電電力検知部５によって検知する給電電力と、受信した電力指令値とを比較し、電力指令値が得られるように、インバータ回路４のスイッチング素子１９，２０と、スイッチング素子２４，２６と、力率改善回路３のスイッチング素子１６とを駆動する。本実施の形態におけるインバータ回路４は、スイッチング素子１９，２０とスイッチング素子２４，２６との駆動周波数を低くすると給電電力が大きくなるように、該駆動周波数を設定しており（給電電力最大となる周波数よりも高い周波数で動作させる）、制御回路１３は、電力指令値となるように基本駆動周波数を決定する。制御回路１３はさらに、給電電力検知部５で検知した電流に電流リプルが発生すると、力率改善回路３の出力電圧が高い時に駆動周波数を基本周波数より高く、低電圧時には駆動周波数を基本周波数より低く設定されるように（図５（ｈ）参照）制御する。これによって第１のインダクタ７に流れる電流及び給電電力を略一定にすることができる。基本駆動周波数はこのような給電電力を略一定にする駆動周波数制御を行った際に、給電電力が電力指令値と一致するように選定される。 The drive frequency control of the switching elements 19, 20, 24, and 26 will be described in more detail. When receiving the power command value transmitted from the control circuit 14 by wireless communication, the control circuit 13 compares the power supply power detected by the power supply power detection unit 5 with the received power command value to obtain a power command value. Thus, the switching elements 19 and 20 of the inverter circuit 4, the switching elements 24 and 26, and the switching element 16 of the power factor correction circuit 3 are driven. In the inverter circuit 4 in the present embodiment, the driving frequency is set so that the feeding power increases when the driving frequency of the switching elements 19 and 20 and the switching elements 24 and 26 is lowered (maximum feeding power). The control circuit 13 determines the basic drive frequency so that the power command value is obtained. Further, when current ripple occurs in the current detected by the feed power detector 5, the control circuit 13 sets the drive frequency higher than the basic frequency when the output voltage of the power factor correction circuit 3 is high, and sets the drive frequency higher than the basic frequency when the output voltage is low. Control is performed so that it is set low (see FIG. 5H). As a result, the current flowing through the first inductor 7 and the feed power can be made substantially constant. The basic drive frequency is selected so that the feed power matches the power command value when such drive frequency control is performed to make the feed power substantially constant.
　このような制御を行う場合、通電率は50%を基本とするが、通電率をより高く設定すると給電電力が大きくなる一方で、通電率を低く設定すると給電電力が小さくなるように設定されている。この通電率を変化させることにより、給電電力と電力指令値をより一致させることができるようになる。 When such control is performed, the energization rate is basically 50%. However, when the energization rate is set higher, the power supply becomes larger, while when the energization rate is set lower, the power supply is reduced. Yes. By changing this energization rate, the supplied power and the power command value can be made more consistent.
負荷における電流リプルは、力率改善回路３の出力電圧すなわち電解コンデンサの電圧が図３のように大小に変化することで発生する。この電圧の大小の幅は給電電力が大きくなると増加するため、本実施の形態のように制御回路１４より無線通信で送られてくる電力指令値が変動するような充電システムでは、負荷における電流リプルも電力指令値に応じて大小に変化することになる。例えば電力指令値が大きくなると、電解コンデンサの電圧変化幅が増加し、負荷の電流リプルも増加することになる。そこで本実施の形態では、駆動周波数を基本周波数より変化させる幅を電力指令値に応じて変化させることで（電力指令値が大きくなると、駆動周波数の変化幅も大きくする）、全電力指令値において負荷電流リプルを改善し、給電電力を略一定になるように動作させる。 The current ripple in the load is generated when the output voltage of the power factor correction circuit 3, that is, the voltage of the electrolytic capacitor, changes as shown in FIG. Since the magnitude of this voltage increases as the feed power increases, in a charging system in which the power command value sent by wireless communication from the control circuit 14 varies as in this embodiment, the current ripple in the load Will also change depending on the power command value. For example, when the power command value increases, the voltage change width of the electrolytic capacitor increases and the load current ripple also increases. Therefore, in the present embodiment, by changing the range in which the drive frequency is changed from the basic frequency in accordance with the power command value (when the power command value is increased, the change range of the drive frequency is increased), the total power command value is The load current ripple is improved and the operation is performed so that the power supply becomes substantially constant.
　また、給電電力検知部５で電流に代えて電圧を検知する場合、給電電力検知部５で検知した電圧に電圧リプルが発生すると、力率改善回路３の出力電圧が高電圧時には駆動周波数が高く、低電圧時には駆動周波数が低くなるように制御回路１３がスイッチング素子１９，２０，２４，２６の駆動周波数を制御することで、電圧リプルが低減し、給電電圧を略一定にすることができる。 In addition, when voltage is detected instead of current by the feed power detection unit 5 and voltage ripple occurs in the voltage detected by the feed power detection unit 5, the drive frequency is high when the output voltage of the power factor correction circuit 3 is high. When the control circuit 13 controls the drive frequency of the switching elements 19, 20, 24, and 26 so that the drive frequency is low when the voltage is low, the voltage ripple is reduced and the power supply voltage can be made substantially constant.
（実施の形態３）
　図１の回路構成については、実施の形態１と同じなので、その説明は省略する。 (Embodiment 3)
Since the circuit configuration of FIG. 1 is the same as that of the first embodiment, description thereof is omitted.
　本実施の形態においては、図６に示されるように、スイッチング素子１９，２０，２４，２６の通電率と駆動周波数の両方を制御することにより高周波電力を制御している。 In the present embodiment, as shown in FIG. 6, the high frequency power is controlled by controlling both the energization rate and the drive frequency of the switching elements 19, 20, 24, and 26.
　すなわち、給電電力検知部５で検知した電流に電流リプルが発生すると、力率改善回路３の出力電圧が高電圧時には通電率が小さくて駆動周波数が高く、低電圧時には通電率が大きくて駆動周波数が低くなるように制御回路１３がスイッチング素子１９，２０，２４，２６を制御することで、電流リプルが低減し、第１のインダクタ７に流れる電流を略一定にすることができる。 That is, when a current ripple is generated in the current detected by the feed power detection unit 5, when the output voltage of the power factor correction circuit 3 is high, the current ratio is small and the drive frequency is high, and when the output voltage is low, the current ratio is large and the drive frequency is high. The control circuit 13 controls the switching elements 19, 20, 24, and 26 so as to reduce the current ripple, thereby reducing the current ripple and making the current flowing through the first inductor 7 substantially constant.
　また、給電電力検知部５で電流に代えて電圧を検知する場合についても同様である。 The same applies to the case where the supply power detection unit 5 detects voltage instead of current.
（実施の形態４）
　図９は、本実施の形態における非接触充電システムの回路図である。
　図９の回路図は、制御回路１３に、力率改善回路３の平滑コンデンサ１８の出力電圧（インバータ回路４の入力電圧）を検知する出力電圧検知部３３が設けられている点で、図１の回路図と相違している。 (Embodiment 4)
FIG. 9 is a circuit diagram of the non-contact charging system in the present embodiment.
9 is that the control circuit 13 is provided with an output voltage detector 33 that detects the output voltage of the smoothing capacitor 18 of the power factor correction circuit 3 (input voltage of the inverter circuit 4). It is different from the circuit diagram.
　本実施の形態においては、出力電圧検知部３３で検知した平滑コンデンサ１８の出力電圧に基づいて、上述したように制御回路１３がスイッチング素子１９，２０，２４，２６を制御（通電率制御及び／又は周波数制御）することで、給電電流あるいは給電電圧を所望の値に制御している。 In the present embodiment, the control circuit 13 controls the switching elements 19, 20, 24, and 26 based on the output voltage of the smoothing capacitor 18 detected by the output voltage detector 33 (energization rate control and / or / (Or frequency control), the feeding current or feeding voltage is controlled to a desired value.
　本実施の形態も、上述した実施の形態１－３と同様、スイッチング素子１９，２０，２４，２６は次のように制御される。 In the present embodiment, the switching elements 19, 20, 24, and 26 are controlled as follows, as in the above-described first to third embodiments.
　（i）通電率制御（駆動周波数は一定）
　この制御では、出力電圧検知部３３で検知した平滑コンデンサ１８の出力電圧が高電圧時には通電率が小さく、低電圧時には通電率が大きくなるように制御回路１３がスイッチング素子１９，２０，２４，２６を制御することで、第１のインダクタ７の電流あるいは電圧及び給電電力を略一定にしている。
　（ii）駆動周波数制御（通電率は一定）
　この制御では、出力電圧検知部３３で検知した平滑コンデンサ１８の出力電圧が高電圧時には駆動周波数が高く、低電圧時には駆動周波数が低くなるように制御回路１３がスイッチング素子１９，２０，２４，２６を制御することで、第１のインダクタ７の電流あるいは電圧及び給電電力を略一定にしている。
　（iii）通電率制御及び駆動周波数制御
　この制御では、出力電圧検知部３３で検知した平滑コンデンサ１８の出力電圧が高電圧時には通電率が小さくて駆動周波数が高く、低電圧時には通電率が大きくて駆動周波数が低くなるように制御回路１３がスイッチング素子１９，２０，２４，２６を制御することで、第１のインダクタ７の電流あるいは電圧及び給電電力を略一定にしている。 (I) Energization rate control (drive frequency is constant)
In this control, the control circuit 13 switches the switching elements 19, 20, 24, and 26 so that the energization rate is small when the output voltage of the smoothing capacitor 18 detected by the output voltage detection unit 33 is high and the energization rate is large when the output voltage is low. By controlling the above, the current or voltage of the first inductor 7 and the feeding power are made substantially constant.
(Ii) Drive frequency control (Conductivity is constant)
In this control, the control circuit 13 switches the switching elements 19, 20, 24, and 26 so that the drive frequency is high when the output voltage of the smoothing capacitor 18 detected by the output voltage detector 33 is high and the drive frequency is low when the output voltage is low. By controlling the above, the current or voltage of the first inductor 7 and the feeding power are made substantially constant.
(Iii) Energization rate control and drive frequency control In this control, the energization rate is small and the drive frequency is high when the output voltage of the smoothing capacitor 18 detected by the output voltage detector 33 is high, and the energization rate is large when the output voltage is low. The control circuit 13 controls the switching elements 19, 20, 24, and 26 so that the drive frequency is lowered, so that the current or voltage of the first inductor 7 and the feeding power are made substantially constant.
　以上のように、本発明に係る非接触充電システムの給電装置は、給電装置の電流・電圧リプルを低減できるばかりでなく、給電装置の部品点数を少なくして給電装置の小型化あるいはコストダウンを図ることができ、給電損失を極力低減することができるので、例えば電気推進車両の受電装置への給電等に有用である。 As described above, the power feeding device of the non-contact charging system according to the present invention can not only reduce the current / voltage ripple of the power feeding device, but also reduce the number of parts of the power feeding device to reduce the size or cost of the power feeding device. Since the power loss can be reduced as much as possible, it is useful for power feeding to a power receiving device of an electric propulsion vehicle, for example.
１　商用電源、　２　第１の整流回路、　３　力率改善回路、
４　インバータ回路、　５　給電電力検知部、
６　第１の共振コンデンサ、　７　第１のインダクタ、
８　第２のインダクタ、　９　第２の共振コンデンサ、
１０　受電電力検知部、　１１　第２の整流回路、
１２　負荷（バッテリー）、　１３　給電装置側の制御回路、
１４　受電装置側の制御回路、　１５　チョークコイル、
１６　第１のスイッチング素子、　１７　第１のダイオード、
１８　平滑コンデンサ、１９　第２のスイッチング素子、
２０　第３のスイッチング素子、　２１　第２のダイオード、
２２　第３のダイオード、　２３　スナバコンデンサ、
２４　第４のスイッチング素子、　２５　第４のダイオード、
２６　第５のスイッチング素子、　２７　第５のダイオード、
２８　スナバコンデンサ、　２９　バイパスコンデンサ、
３０　電流検知部、　３１　電圧検知部、　３２　電力演算部、
３３　力率改善回路の出力電圧検知部。 1 commercial power source, 2 first rectifier circuit, 3 power factor correction circuit,
4 inverter circuit, 5 feed power detector,
6 first resonant capacitor, 7 first inductor,
8 second inductor, 9 second resonant capacitor,
10 received power detection unit, 11 second rectifier circuit,
12 Load (battery), 13 Control circuit on the power feeding device side,
14 power receiving device side control circuit, 15 choke coil,
16 first switching element, 17 first diode,
18 smoothing capacitor, 19 second switching element,
20 third switching element, 21 second diode,
22 third diode, 23 snubber capacitor,
24 4th switching element, 25 4th diode,
26 fifth switching element, 27 fifth diode,
28 snubber capacitors, 29 bypass capacitors,
30 current detector, 31 voltage detector, 32 power calculator,
33 Output voltage detector of power factor correction circuit.

Claims (7)

  1. 受電装置に非接触で給電を行う給電装置であって、
    　力率改善を行って直流電圧を出力する力率改善回路と、
    　前記力率改善回路の出力直流電圧が与えられると、電荷を蓄積する電解コンデンサと、
    　複数のスイッチング素子を有しており、前記電解コンデンサを電源として動作し、交流信号を発生するインバータ回路と、
    　前記インバータ回路からの交流信号が入力されると、前記受電装置に非接触で給電を行う共振コンデンサ及びインダクタと、
    　前記複数のスイッチング素子の通電率及び駆動周波数の少なくとも一方を変化させる給電側制御回路と、
    を備える、給電装置。     A power supply device that supplies power to a power receiving device in a contactless manner,
    A power factor correction circuit for improving the power factor and outputting a DC voltage;
    When an output DC voltage of the power factor correction circuit is given, an electrolytic capacitor that accumulates electric charge;
    An inverter circuit that has a plurality of switching elements, operates using the electrolytic capacitor as a power source, and generates an AC signal;
    When an AC signal from the inverter circuit is input, a resonance capacitor and an inductor that perform power supply in a non-contact manner to the power receiving device,
    A power-feeding-side control circuit that changes at least one of an energization rate and a driving frequency of the plurality of switching elements;
    A power supply apparatus comprising:
  2. 前記インダクタに流れる電流、又は該インダクタの電圧を検知する給電電力検知部をさらに備え、
    　前記給電側制御回路は、前記給電電力検知部が検知した電流あるいは電圧に基づいて、前記複数のスイッチング素子の通電率及び駆動周波数の少なくとも一方を変化させることを特徴とする、請求項１に記載の給電装置。     A power supply detection unit for detecting a current flowing through the inductor or a voltage of the inductor;
    2. The power supply side control circuit according to claim 1, wherein the power supply side control circuit changes at least one of an energization rate and a drive frequency of the plurality of switching elements based on a current or a voltage detected by the power supply power detection unit. Power supply device.
  3. 前記給電側制御回路は、
    　前記受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、前記複数のスイッチング素子の基本通電率を決定し、
    　前記給電電力検知部の出力電圧が高い時には、前記複数のスイッチング素子の通電率を前記基本通電率より低く設定し、低い時には前記複数のスイッチング素子の通電率を基本通電率より高く設定することを特徴とする、請求項２に記載の給電装置。     The power supply side control circuit is:
    Receiving the power command value by wireless communication with the power receiving device side, determining the basic energization rate of the plurality of switching elements so as to be the received power command value,
    When the output voltage of the power supply detection unit is high, the energization rate of the plurality of switching elements is set lower than the basic energization rate, and when the output voltage is low, the energization rate of the plurality of switching elements is set higher than the basic energization rate. The power feeding device according to claim 2, wherein the power feeding device is characterized.
  4. 前記給電側制御回路は、
    　前記受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、前記複数のスイッチング素子の基本駆動周波数を決定し、
    　前記給電電力検知部の出力電圧が高い時には、前記複数のスイッチング素子の駆動周波数を前記基本駆動周波数より高く設定し、低い時には前記複数のスイッチング素子の駆動周波数を基本駆動周波数より低く設定することを特徴とする、請求項２に記載の給電装置。     The power supply side control circuit is:
    Receiving the power command value by wireless communication with the power receiving device side, determining the basic drive frequency of the plurality of switching elements so as to be the received power command value,
    When the output voltage of the power supply detection unit is high, the drive frequency of the plurality of switching elements is set higher than the basic drive frequency, and when the output voltage is low, the drive frequency of the plurality of switching elements is set lower than the basic drive frequency. The power feeding device according to claim 2, wherein the power feeding device is characterized.
  5. 前記給電側制御回路は、前記力率改善回路の出力電圧を検知する出力電圧検知部を有し、前記出力電圧検知部が検知した前記力率改善回路の出力電圧に基づいて、前記複数のスイッチング素子の通電率及び駆動周波数の少なくとも一つを変化させることを特徴とする、請求項１に記載の給電装置。 The power supply side control circuit includes an output voltage detection unit that detects an output voltage of the power factor correction circuit, and the plurality of switching units based on the output voltage of the power factor correction circuit detected by the output voltage detection unit. The power feeding device according to claim 1, wherein at least one of an energization rate and a driving frequency of the element is changed.
  6. 前記給電側制御回路は、
    　前記受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、前記複数のスイッチング素子の基本通電率を決定し、
    　前記出力電圧検知部の出力電圧が高い時には、前記複数のスイッチング素子の通電率を前記基本通電率より低く設定し、低い時には前記複数のスイッチング素子の通電率を基本通電率より高く設定することを特徴とする、請求項５に記載の給電装置。     The power supply side control circuit is:
    Receiving the power command value by wireless communication with the power receiving device side, determining the basic energization rate of the plurality of switching elements so as to be the received power command value,
    When the output voltage of the output voltage detector is high, the conduction rate of the plurality of switching elements is set lower than the basic conduction rate, and when the output voltage is low, the conduction rate of the plurality of switching elements is set higher than the basic conduction rate. The power feeding device according to claim 5, wherein the power feeding device is characterized.
  7. 前記給電側制御回路は、
    　前記受電装置側との無線通信により電力指令値を受信し、受信した電力指令値になるように、前記複数のスイッチング素子の基本駆動周波数を決定し、
    　前記出力電圧検知部の出力電圧が高い時には、前記複数のスイッチング素子の駆動周波数を前記基本駆動周波数より高く設定し、低い時には前記複数のスイッチング素子の駆動周波数を基本駆動周波数より低く設定することを特徴とする、請求項５に記載の給電装置。     The power supply side control circuit is:
    Receiving the power command value by wireless communication with the power receiving device side, determining the basic drive frequency of the plurality of switching elements so as to be the received power command value,
    When the output voltage of the output voltage detector is high, the drive frequency of the plurality of switching elements is set higher than the basic drive frequency, and when low, the drive frequency of the plurality of switching elements is set lower than the basic drive frequency. The power feeding device according to claim 5, wherein the power feeding device is characterized.
PCT/JP2011/001901 2011-01-06 2011-03-30 Non-contact charging system power supply device WO2012093423A1 (en)

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