WO2009113367A1 - コンバータの制御方法及び制御装置 - Google Patents
コンバータの制御方法及び制御装置 Download PDFInfo
- Publication number
- WO2009113367A1 WO2009113367A1 PCT/JP2009/052695 JP2009052695W WO2009113367A1 WO 2009113367 A1 WO2009113367 A1 WO 2009113367A1 JP 2009052695 W JP2009052695 W JP 2009052695W WO 2009113367 A1 WO2009113367 A1 WO 2009113367A1
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- command value
- phase
- potential
- component
- voltage
- Prior art date
Links
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M7/2173—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a biphase or polyphase circuit arrangement
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M7/2176—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only comprising a passive stage to generate a rectified sinusoidal voltage and a controlled switching element in series between such stage and the output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/483—Converters with outputs that each can have more than two voltages levels
- H02M7/487—Neutral point clamped inverters
Definitions
- This invention relates to a technique for controlling a converter, and more particularly to a control technique for a three-level converter.
- a so-called three-level converter that obtains a three-level DC potential from a three-phase rectifier circuit has been proposed.
- the three-level converter is exemplified in the following Patent Documents 1 to 3 and Non-Patent Documents 1 and 2, for example.
- FIG. 16 is a circuit diagram of the three-level converter introduced in FIG. 4 of Patent Document 1.
- a three-phase voltage is connected to one end of each of the inductors 8, 9, 10 that are reactors of the three-width pair.
- the other ends of the inductors 8, 9, and 10 are connected to terminals 14, 15, and 16 through feeder lines 11, 12, and 13, respectively.
- the terminals 14, 15, and 16 function as input terminals of the power supply three-phase diode bridge 17 constituted by the diodes 18, 19, 20, 24, 25, and 26. , 27 to the capacitors 6 and 7.
- Terminals 14, 15, and 16 are connected to a neutral point 33 through bidirectional switches 30, 31, and 32, respectively.
- the switching element 61 has a collector 77 and an emitter 78.
- the switching element 61 has a collector 79 and an emitter 80.
- the switching element 61 is a collector. 81 and an emitter 82.
- FIG. 17 is a circuit diagram of the three-level converter introduced in FIG. A similar circuit is also introduced in FIG. In the circuit, three-phase currents Ia, Ib, Ic flow from the three-phase voltages Va, Vb, Vc side.
- the current Ia becomes the current In flowing to the neutral point n through the diode D12 and the switch S11 or through the diode D13 and the switch S12. Alternatively, the current Ia flows to the capacitor via the diodes D11 and D12 or via the diodes D13 and D14.
- the current Ib becomes the current In through the diode D22 and the switch S21, or through the diode D23 and the switch S22. Alternatively, the current Ib flows to the capacitor via the diodes D21 and D22 or via the diodes D23 and D24.
- the current Ic becomes the current In through the diode D32 and the switch S31 or through the diode D33 and the switch S32. Alternatively, the current Ic flows to the capacitor via the diodes D31 and D32 or via the diodes D33 and D34.
- the switches S11, S22, and S31 and the switches S12, S22, and S32 separately bear the withstand voltages during the period when the phase potential is positive and the period when the phase potential is negative. Charge to 2.
- the withstand voltage of the switches S12 to S32 is almost half as compared with the switching element 61 of the circuit shown in FIG.
- the number of switching elements in the circuit shown in FIG. 17 is doubled compared to the circuit shown in FIG.
- FIG. 18 is a circuit diagram of the three-level converter introduced in FIG.
- a power supply line 48 is connected to the output terminals 23 and 29 via power supply lines 37 and 38, and is given to the control unit 40 together with a power supply line 47 connected to the intermediate point 35, so that a measured value of the output voltage is supplied to the control unit 40.
- the phase voltage of the three-phase power source 5 (phase power source 2, 3, 4) is applied to the control unit 40 from the feeder lines 45 to 46 (coupled as the three-phase feeder line 42).
- An external control signal 41 is also given to the control unit 40 separately.
- Patent Documents 4 and 5 and Non-Patent Documents 3, 4, and 5 are listed as other documents related to the present application.
- Non-Patent Document 1 In order to control switching with respect to the configuration shown in FIG. 16 or FIG. 18, simple control using a hysteresis comparator has been proposed as introduced in Non-Patent Document 1. However, such control requires controlling the current individually for each of the three phases. In addition, it is necessary to detect the sign of the flowing current.
- the invention according to the present application provides a control technique for a three-level converter that does not require detection of the power supply voltage or the polarity of the current flowing through the converter.
- the converter control method includes a rectifier circuit that outputs a low potential (VL), a high potential (VH) higher than the low potential, and an intermediate potential (VQ) between the low potential and the high potential.
- VL low potential
- VH high potential
- VQ intermediate potential
- the first capacitor (205; 7) to which the low potential and the intermediate potential are supplied, and the first capacitor at a connection point (35; n), and the high potential and the intermediate potential are A method of controlling a converter with a second capacitor (204; 6) supplied.
- the rectifier circuit includes first to third potentials at respective other ends of the three-width pair reactors (202; 8, 9, 10) to which three-phase voltages (Vu, Vv, Vw) are applied to the respective one ends.
- a three-phase diode bridge (18, 19, 20, 24, 25, 26; D11, D21, D31, D14, D24, D34) that rectifies (Vr, Vs, Vt) and outputs the low potential and the high potential.
- a switch group (30, 31, 32; S11, S21, S31, S12, S22, S32) that selectively connects the three other ends of the reactor to the connection point.
- the first to third command values (Vr * , Vs * , Vt * ), which are command values for the first to third potentials, are respectively set.
- the switch group When in the predetermined range, the switch group connects the other end corresponding to each command value to the connection point, and the predetermined range is centered on the command value (0) of the intermediate potential.
- a range having a predetermined potential width is adopted for the AC waveform (VK).
- a second aspect of the converter control method according to the present invention is the first aspect, wherein both the amplitude of the AC waveform and the predetermined potential width are changed from the high potential (VH) to the low potential. It is half of the command value (Vdc * ) of the output voltage (Vdc), which is the voltage minus (VL).
- a third aspect of the converter control method is the second aspect, wherein the three-phase current (Iu, Iv, Iw) flowing through the reactor is three-phase / two-phase converted, and the three A first component (Id) whose phase is orthogonal to the phase voltage (Vu, Vv, Vw) and a second component (Iq) in phase with the three-phase voltage are obtained, and a fourth command value is obtained based on the first component.
- (Vid * ) is obtained, and a second component command value (Iq * ) as a command value of the second component is obtained based on a difference between the output voltage (Vdc) and the command value (Vdc * ) of the output voltage.
- a fifth command value (Viq * ) is obtained based on a difference between the second component and the command value of the second component, and the fourth command value and the fifth command value are subjected to two-phase / three-phase conversion.
- the first to third command values (Vr * , Vs * , Vt * ) are obtained.
- a fourth aspect of the converter control method according to the present invention is the third aspect, wherein the first component in a frequency band of three times or more the frequency of the three-phase voltages (Vu, Vv, Vw).
- the fourth command value and the fifth command value are obtained from (Id), the second component (Iq), and the second component command value (Iq * ).
- a fifth aspect of the converter control method according to the present invention is the fourth aspect, wherein the frequency of the AC waveform (VK) is three times the frequency of the three-phase voltage (Vu, Vv, Vw). That's it.
- a sixth aspect of the converter control method according to the present invention is the fifth aspect thereof, and when the fourth command value (Vid * ) is obtained, the one component (Id) is the three-phase voltage (Vu). , Vv, Vw) is corrected by the first harmonic (cos3 ⁇ t) having a frequency three times the frequency of the second frequency (Vq * ), the two components (Iq) are the same as the first harmonic. It is corrected by the second harmonic (sin 3 ⁇ t) whose phase is orthogonal.
- the converter control device includes a rectifier circuit that outputs a low potential (VL), a high potential (VH) higher than the low potential, and an intermediate potential (VQ) between the low potential and the high potential.
- VL low potential
- VH high potential
- VQ intermediate potential
- the first capacitor (205; 7) to which the low potential and the intermediate potential are supplied, and the first capacitor at a connection point (35; n), and the high potential and the intermediate potential are
- a device for controlling a converter comprising a second capacitor (204; 6) supplied.
- the rectifier circuit includes first to third potentials at respective other ends of the three-width pair reactors (202; 8, 9, 10) to which three-phase voltages (Vu, Vv, Vw) are applied to the respective one ends.
- a three-phase diode bridge (18, 19, 20, 24, 25, 26; D11, D21, D31, D14, D24, D34) that rectifies (Vr, Vs, Vt) and outputs the low potential and the high potential.
- a switch group (30, 31, 32; S11, S21, S31, S12, S22, S32) that selectively connects the three other ends of the reactor to the connection point.
- the first to third command values (Vr * , Vs * , Vt * ), which are command values for the first to third potentials, are obtained.
- a voltage command value generation unit (101; 122, 104) generated from a three-phase voltage, and each of the first to third command values has an alternating waveform (VK) centered on the command value (0) of the intermediate potential.
- a switching signal (Srp, Srn, Ssp, Ssn, Stp) that causes the switch group to connect the other end corresponding to each command value to the connection point when it is within a range having a predetermined potential width. , Stn; and a pulse width modulator (102) for generating Sr, Ss, St).
- a second aspect of the converter control device is the first aspect, wherein both the amplitude of the AC waveform and the predetermined potential width are from the high potential (VH) to the low potential. It is half of the command value (Vdc * ) of the output voltage (Vdc), which is the voltage minus (VL).
- a third aspect of the converter control device is the second aspect, in which the three-phase voltage (Vu, Vv, Vw) is derived from the three-phase current (Iu, Iv, Iw) flowing through the reactor. ) And a three-phase / two-phase converter (103) for obtaining a first component (Id) whose phase is orthogonal and a second component (Iq) in phase with the three-phase voltage, and a first component based on the first component. 4 based on the difference between the first command value generation unit (108, 110, 113) for obtaining the command value (Vid * ) and the output voltage (Vdc) and the command value (Vdc * ) of the output voltage.
- a second component command value (Iq * ) that is a command value of the second component is obtained, and a fifth command value (Viq * ) is obtained based on the difference between the second component and the command value of the second component.
- Command value generator (105, 106, 107, 109, 112) and the fourth finger Value and the first to third command values from said fifth command value (Vr *, Vs *, Vt *) further comprises a two-phase / three-phase converter seeking and (104).
- a fourth aspect of the converter control device is the third aspect, and has a frequency band of three times or more the frequency of the three-phase voltages (Vu, Vv, Vw).
- a fifth aspect of the control device for the converter according to the present invention is the fourth aspect, wherein the frequency of the AC waveform (VK) is three times the frequency of the three-phase voltage (Vu, Vv, Vw). That's it.
- a sixth aspect of the converter control device is the fifth aspect thereof, wherein the first command value generation unit converts the first component (Id) into the three-phase voltage (Vu, Vv). , Vw) having a first correction unit (115, 116, 117, 118, 120) that corrects by the first harmonic (cos3 ⁇ t) having a frequency that is three times the frequency of the frequency of Vw), the second command value generation unit includes: A second correction unit (114, 116, 117, 119, 121) that corrects the two components (Iq) with a second harmonic (sin3 ⁇ t) whose phase is orthogonal to the first harmonic is included.
- the pulse width modulation is performed with a larger duty in which the other end of the corresponding reactor is connected to the connection point. Therefore, the potential at the connection point can be set to an intermediate potential.
- the predetermined range to be compared with the first to third command values has a predetermined potential width with respect to the AC waveform centered on the intermediate potential command value. Therefore, in such pulse width modulation, it is not necessary to detect the power supply voltage or the polarity of the current flowing through the converter.
- the power factor can be improved by performing the control to make the reactive power zero.
- the second harmonic of the current flowing through the reactor increases as the capacitance ratio of the first capacitor and the second capacitor deviates from one.
- the second harmonic appears as a third harmonic in the first component and the second component. Therefore, by adopting a band for obtaining the fourth command value and the fifth command value at least three times the frequency of the three-phase voltage, it is possible to reduce the second harmonic of the current flowing through the reactor.
- the third harmonic of the first component and the second component is canceled when the fourth command value and the fifth command value are obtained. . Therefore, it is possible to further reduce the second harmonic of the current flowing through the reactor.
- FIG. 1 is a circuit diagram showing a configuration of a three-level converter to which the present invention is applied and its periphery.
- FIG. It is a circuit diagram which illustrates the composition of a PWM modulation part. It is a graph which shows operation
- FIG. 1 is a circuit diagram showing a configuration of a three-level converter 200 to which the present invention is applied and its periphery.
- Three-level converter 200 is connected to a three-phase power source 201 through a reactor group 202.
- Three-phase voltages exhibiting potentials Vu, Vv, and Vw are output from the three-phase power source 201, and three-phase currents Iu, Iv, and Iw flow in the reactor group 202 corresponding to each.
- Three-level converter 200 receives currents Iu, Iv, and Iw, and generates input potentials Vr, Vs, and Vt corresponding to the currents.
- the reactor group 202 corresponds to the inductors 8, 9, and 10 shown in FIGS. 16 and 18, and the coils (not shown) through which the currents Ia, Ib, and Ic flow in FIG.
- the three-level converter 200 includes a rectifier circuit 203 on the input side and capacitors 204 and 205 on the output side.
- the rectifier circuit 203 applies a high potential VH and an intermediate potential VQ to the capacitor 204, and applies a low potential VL and an intermediate potential VQ to the capacitor 205, respectively.
- the capacitors 204 and 205 correspond to the capacitors 6 and 7 shown in FIGS. 16 and 18 and the capacitor (not shown) charged with the voltage Vd / 2 in FIG.
- the reactor group 202 corresponds to the inductors 8, 9, and 10 shown in FIGS. 16 and 18, and the coils (not indicated) through which the currents Ia, Ib, and Ic flow in FIG.
- the rectifier circuit 203 has a three-phase diode bridge and a switch group.
- the three-phase diode bridge rectifies the potentials Vr, Vs, and Vt and outputs a low potential VL and a high potential VH.
- the switch group includes three ends of the power reactor group 202 on the side opposite to the three-phase power supply 201 (the end of the reactor group 202 on the three-phase power supply 201 side) with respect to the connection point N that connects the capacitors 204 and 205 to each other. The other end when grasped as one end) is selectively connected.
- Examples of the three-phase diode bridge include a set of diodes 18, 19, 20, 24, 25, and 26 shown in FIGS. 16 and 18, and diodes D11, D21, D31, D14, D24, shown in FIG. A set of D34 can be employed.
- the bidirectional switches 30, 31, 32 shown in FIGS. 16 and 18 and the switches S11, S21, S31, S12, S22, S32 shown in FIG. 17 can be adopted.
- the switching signal generation unit 100 includes a voltage command value generation unit 101 and a PWM modulation unit 102.
- the voltage command value generation unit 101 inputs measured values of the potentials Vu, Vv, and Vw, and calculates command values of the input potentials Vr, Vs, and Vt. Since the measurement of the potentials Vu, Vv, and Vw is a well-known technique, in FIG. 1, for simplicity, the potentials Vu, Vv, and Vw are drawn using arrows as if they were being input to the voltage command value generation unit 101. . Other arrows input to the block have the same meaning.
- the PWM modulation unit 102 generates switching signals Sr, Ss, St or switching signals Srp, Ssp, Stp, Srn, Ssn, Stn based on the command values Vr * , Vs * , Vt * of the three-phase voltage.
- the switching signals Sr, Ss, St can be used as the respective gate signals to the bidirectional switches 30, 31, 32 shown in FIGS.
- the switching signals Srp, Ssp, Stp, Srn, Ssn, Stn can be employed as the respective gate signals to the switches S11, S21, S21, S22, S31, S32 shown in FIG.
- the switch group of the rectifier circuit 203 uses the other end side of the reactor group 202 corresponding to each command value as a connection point. Connect to connection point N.
- the switching signal Sr conducts the bidirectional switch 30 and is a terminal connected to the inductor 8. 14 is connected to an intermediate point 35 through a neutral point 33 and a feeder line 34.
- the switching signal Ss causes the bidirectional switch 31 to conduct and connects the terminal 15 to the intermediate point 35.
- the switching signal St causes the bidirectional switch 32 to conduct and connects the terminal 16 to the intermediate point 35.
- switching signals Srp and Srn cause switches S11 and S12 to conduct, respectively, and the end of the U-phase coil opposite to the power source. (Point marked with “U” in the figure: diodes D12 and D13 are connected here) is connected to neutral point n.
- the current Ia flows through one of the switches S11 and S12 according to the comparison result between the potential at the end of the coil and the potential at the neutral point n by the function of the diodes D12 and D13.
- the switching signals Ssp and Ssn cause the switches S21 and S22 to conduct, respectively, and the end of the V-phase coil on the opposite side of the power supply (reference symbol “V” in the figure).
- the points marked “" are connected here to the diodes D22 and D23) to the neutral point n.
- the current Ib flows through one of the switches S21 and S22 according to the comparison result of the potential of the end of the coil and the potential of the neutral point n by the function of the diodes D22 and D23.
- the switching signals Stp and Stn cause the switches S31 and S32 to conduct, respectively, and the end of the W-phase coil on the opposite side of the power supply (indicated by the symbol “W” in the figure)
- the diodes D32 and D33 are connected here) to the neutral point n.
- the current Ic flows through one of the switches S31 and S32 according to the comparison result between the potential of the end of the coil and the potential of the neutral point n by the function of the diodes D32 and D33.
- FIG. 2 is a circuit diagram illustrating the configuration of the PWM modulation unit 102.
- FIG. 3 is a graph showing the operation of the PWM modulation unit 102.
- the intermediate potential VQ is set to 0 for the sake of simplicity. However, the present embodiment and the present invention does not limit the intermediate potential to this value.
- the potential VN is an alternating current that takes a minimum value ( ⁇ Vdc * / 2) and a maximum value Vdc * / 2, and is generated by the signal source 102a.
- the voltage Vdc * / 2 is a command value of the voltage charged in the capacitors 204 and 205.
- the potential VP is set higher than the potential VN by the voltage Vdc * / 2.
- the voltage Vdc * / 2 can be generated by the DC voltage source 102b, for example.
- the predetermined range a range not more than the potential VP and not less than the potential VN is adopted.
- the predetermined range described above is a range having a potential width Vdc * / 4 with respect to the AC waveform VK.
- command value Vr * is greater than potential VP is compared in comparator 102rp, and whether or not it is greater than potential VN is compared in comparator 102rn.
- the command value Vs * is compared in the comparator 102sp to determine whether it is greater than the potential VP, and in the comparator 102sn, it is compared whether it is greater than the potential VN. It is compared whether or not the command value Vt * is greater than the potential VP in the comparator 102tp, and whether or not it is greater than the potential VN in the comparator 102tn.
- Each comparator outputs a logical value “1” if the determination result is affirmative, and outputs a logical value “0” if the determination result is negative.
- the logical values output from the comparators 102rp, 102sp, and 102tp are inverted and output by the inverters 102ri, 102si, and 102ti, respectively.
- the logical values output from the comparators 102rn, 102sn, and 102tn and the logical values output from the inverters 102ri, 102si, and 102ti are the switching signals Srn, Ssn, Stn, Srp, Ssp, and Stp, respectively.
- the switches S12, S22, S32, S11, S21, S31 see FIG. 17
- the PWM modulator 102 may further include AND gates 102rg, 102sg, and 102tg that output the switching signals Sr, Ss, and St, respectively. This is because it is suitable for controlling the bidirectional switches 30, 31, 32 (see FIG. 16 or FIG. 18). Specifically, the AND gate 102rg takes the logical product of the switching signals Srn and Srp to generate the switching signal Sr. Similarly, the AND gate 102sg takes the logical product of the switching signals Ssn and Ssp to generate the switching signal Ss, and the AND gate 102tg takes the logical product of the switching signals Stn and Stp to generate the switching signal St.
- the switching signal Srn has a logical value “1” (high in the graph) corresponding to each of the command value Vr * being “larger” / “smaller” than the potential VN. Potential) / logical value “0” (low potential in the graph).
- the switching signal Srp has a logical value “1” corresponding to each of the command value Vr * being “smaller” / “larger” than the potential VP. / Take "0".
- the pulse width modulation is performed in which the other end is connected to the connection point N. Therefore, the potential at the connection point N can be set to the command value of the intermediate potential VQ.
- the pulse width modulation is performed since the predetermined range compared with the command values Vr * , Vs * , Vt * has a predetermined potential width with respect to the AC waveform VK centered on the command value of the intermediate potential VQ, the pulse width modulation is performed.
- both the amplitude (peak-to-peak) and the predetermined potential width of the AC waveform VK are half Vdc * / 2 of the command value Vdc * of the output voltage Vdc, which is a voltage obtained by subtracting the low potential VL from the high potential VH. It is desirable. This is because it is easy to obtain waveforms of potentials VP and VN that define the upper and lower limits of the predetermined range.
- FIG. 4 is a graph showing a first example of the operation in the present embodiment.
- Deviation when a disturbance of 20 V is forcibly applied to the deviation ⁇ VQ of the intermediate potential VQ (here, the command value of the intermediate potential VQ is considered to be 0, and thus becomes a value obtained by multiplying the intermediate potential VQ by ( ⁇ 1)).
- the behavior of ⁇ VQ and current Iu is shown.
- FIG. 5 is a block diagram illustrating another configuration of the switching signal generation unit 100 and its periphery. Since a technique for obtaining a voltage command based on a current command is known from, for example, Patent Document 5, a description thereof will be kept simple.
- the current detector 206 detects currents Iu, Iv, and Iw flowing from the three-phase power source 201 to the reactor group 202.
- the phase detector 122 detects the phase ⁇ t ( ⁇ : angular frequency, t: time) of the potentials Vu, Vv, Vw output from the three-phase power source 201.
- the three-phase / two-phase converter 103 performs three-phase / two-phase conversion of the currents Iu, Iv, Iw into the d-axis and the q-axis, respectively, and the d-axis current Id as the first component and the q-axis current Iq as the second component, respectively. Ask for.
- the q-axis and the d-axis are orthogonal coordinate axes in a rotating coordinate system that rotates in synchronization with the phases of the potentials Vu, Vv, and Vw, and the q-axis is advanced by 90 degrees relative to the d-axis.
- the q axis is selected to be in phase with the three-phase voltage.
- the adder / subtractor 105 outputs a deviation between the DC voltage Vdc and its command value Vdc * . Based on the deviation, the voltage control unit 106 obtains a q-axis current command value Iq * . The adder / subtractor 107 outputs a deviation between the q-axis current Iq and the command value Iq * . Based on the deviation, the current control unit 109 obtains a q-axis voltage command value Vq * .
- the adder / subtractor 108 outputs a deviation between the d-axis current Id and the command value Id * .
- the current control unit 110 obtains a d-axis voltage command value Vd * based on the deviation.
- the command value Id * is selected to be zero. This is desirable in that the power factor is improved by controlling the reactive power to zero.
- the command values Vq * and Vd * are corrected to command values Viq * and Vid * , respectively, by subtracting the interference terms by the adders / subtractors 112 and 113, respectively.
- Two-phase / three-phase converter 104 command value Viq *, Vid * a two-phase / three-phase conversion to the command value Vr *, Vs *, to generate a Vt *.
- FIG. 6 is a block diagram illustrating the configuration of the interference term generation unit 111 that generates an interference term.
- the interference term generator 111 multiplies the d-axis current Id by ⁇ L (L: a value obtained by two-phase conversion of the inductance of the reactor group 202) and outputs the result to the adder / subtractor 112, and the q-axis current Iq is ( ⁇ L ) And output to the adder / subtractor 113.
- Similar generation of interference terms and compensation of interference terms are also introduced in Non-Patent Document 4, for example.
- the adders / subtracters 108 and 113 and the current control unit 110 can be grasped as a first command value generation unit that calculates the command value Vid * based on the d-axis current Id.
- the adders / subtracters 105, 107 and 112 the voltage controller 106 and the current controller 109 obtain the command value Iq * based on the difference between the voltage Vdc and the command value Vdc *, and the difference between this and the q-axis current is obtained. Based on this, it can be grasped as a second command value generation unit for obtaining the command value Viq * .
- FIG. 7 is a graph showing a second example of the operation in the present embodiment.
- the configuration shown in FIG. 5 is adopted as the switching signal generation unit 100.
- the graph it is shown in the graph that the recovery from the disturbance is remarkably accelerated by adopting the rotating coordinate system. That is, although the deviation ⁇ VQ is reduced to about 5 V and stabilized, only about 0.1 seconds have passed since the disturbance was applied.
- the pulsating current is superimposed on the d-axis current Id and the q-axis current Iq. This is because the intermediate potential VQ is non-equilibrium.
- a technique for reducing the pulsating flow will be described.
- FIG. 8 is a graph showing waveforms of currents Iu, Iv, and Iw when the intermediate potential VQ approaches the low potential PL side and becomes unbalanced, and the horizontal axis indicates the phase angle of the power source. All of these include secondary harmonics.
- FIG. 8 illustrates the second harmonic Iu2 of the current Iu. At the position where the current Iu exhibits a peak, the second harmonic Iu2 also exhibits a peak.
- FIG. 9 is a graph showing normalized currents Id and Iq obtained by three-phase / two-phase coordinate conversion of the currents Iu, Iv and Iw shown in FIG. 8, and the horizontal axis indicates the phase angle of the power source. did.
- a pulsating current is superimposed on the currents Id and Iq, specifically, a third harmonic is superimposed.
- the current obtained by converting the three-phase current to the three-phase / two-phase coordinate conversion includes the third-order harmonic. 3.
- FIG. 10 is a graph showing waveforms of currents Iu, Iv, and Iw when the intermediate potential VQ approaches the high potential PH side and becomes unbalanced, and the phase angle of the power source is adopted on the horizontal axis. All of these include secondary harmonics.
- FIG. 10 illustrates the second harmonic Iu2 of the current Iu. At the position where the current Iu exhibits a peak, the second harmonic Iu2 also exhibits a peak.
- FIG. 11 is a graph showing normalized currents Id and Iq obtained by three-phase / two-phase coordinate conversion of the currents Iu, Iv and Iw shown in FIG. 10, and the horizontal axis indicates the phase angle of the power source. did. Third-order harmonics are superimposed on the currents Id and Iq.
- FIG. 12 is a graph showing a waveform when the intermediate potential VQ approaches the low potential PL side and becomes non-equilibrium.
- Vrr is a potential on the input side of the rectifier circuit 203 while being subjected to pulse width modulation, and a filtered version thereof corresponds to the potential Vr.
- the current Iu is more distorted on the increasing slope than the decreasing slope.
- the d-axis current Id includes a third harmonic as shown in FIG. 9, and the q-axis current Iq is the same.
- FIG. 13 is a graph showing a waveform when the intermediate potential VQ approaches the high potential PH side and becomes non-equilibrium.
- the balanced intermediate potential VQ was set to 350V.
- the current Iu is distorted on the decreasing side with a steeper slope than on the increasing side.
- the d-axis current Id includes a third harmonic as shown in FIG. 11, and the q-axis current Iq is the same.
- FIG. 14 illustrates the configuration of the improved switching signal generation unit 100 and its periphery in a block diagram.
- the configuration shown in FIG. 14 includes adders / subtractors 114, 115, 116, a voltage control unit 117, multipliers 118, 119, and compensation term calculation units 120, 121. It has become the composition.
- the adder / subtractor 116 subtracts the intermediate potential VQ from the intermediate potential command value VQ * (0 V in the example of FIG. 3 and 350 V in the examples of FIGS. 12 and 13) to obtain the deviation ⁇ VQ.
- the voltage control unit 117 outputs a correction command value I0 * . This indicates an absolute value of a value for correcting the current command values Iq * and Id * .
- Both the compensation term calculation units 120 and 121 receive the phase ⁇ t and generate the compensation terms cos (3 ⁇ t) and sin (3 ⁇ t), respectively. These are all third-order harmonic components of a sine wave having an angular frequency ⁇ , and the latter is advanced in phase by 90 degrees compared to the former.
- the correction command value I0 * is multiplied by cos (3 ⁇ t) by the multiplier 118 to generate a correction value ⁇ d.
- the correction command value I0 * is multiplied by sin (3 ⁇ t) by the multiplier 119 to generate a correction value ⁇ q.
- the correction using such third-order harmonics as a compensation term is effective when the band for obtaining the command values Vid * and Viq * is set to three times or more the frequency of the three-phase voltage as described above. Further, since the third harmonics of the two-phase currents Id and Iq are canceled when the command values Vid * and Viq * are obtained, the second harmonics of the three-phase currents Iu, Iv and Iw flowing through the reactor group are further reduced. be able to.
- FIG. 15 is a graph showing a third example of the operation in the present embodiment.
- the configuration shown in FIG. 14 is adopted as the switching signal generation unit 100.
- the pulsation of the two-phase currents Id and Iq is reduced although the disturbance is more serious immediately after the occurrence of the disturbance.
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Rectifiers (AREA)
Abstract
Description
Claims (12)
- 低電位(VL)、前記低電位よりも高い高電位(VH)、及び前記低電位と前記高電位との間の中間電位(VQ)を出力する整流回路(203)と、
前記低電位及び前記中間電位が供給される第1コンデンサ(205;7)と、
接続点(35;n)において前記第1コンデンサと接続され、前記高電位及び前記中間電位が供給される第2コンデンサ(204;6)と
を備え、
前記整流回路は
三相電圧(Vu,Vv,Vw)がそれぞれの一端に印加される三幅対のリアクトル(202;8,9,10)のそれぞれの他端における第1乃至第3の電位(Vr,Vs,Vt)を整流して前記低電位及び前記高電位を出力する三相ダイオードブリッジ(18,19,20,24,25,26;D11,D21,D31,D14,D24,D34)と、
前記接続点に、前記リアクトルの三つの前記他端を選択的に接続するスイッチ群(30,31,32;S11,S21,S31,S12,S22,S32)と
を有するコンバータを制御する方法であって、
前記第1乃至第3の電位のそれぞれについての指令値である第1乃至第3指令値(Vr*,Vs*,Vt*)がそれぞれ所定の範囲内にあるときに、前記スイッチ群はそれぞれの指令値に対応する前記他端を前記接続点に接続し、
前記所定の範囲には前記中間電位の指令値(0)を中心とする交流波形(VK)に対して所定の電位幅を有する範囲を採用する、コンバータの制御方法。 - 前記交流波形の振幅及び前記所定の電位幅のいずれもが、前記高電位(VH)から前記低電位(VL)を引いた電圧たる出力電圧(Vdc)の指令値(Vdc*)の半分である、請求項1記載のコンバータの制御方法。
- 前記リアクトルに流れる三相電流(Iu,Iv,Iw)を三相/二相変換して、前記三相電圧(Vu,Vv,Vw)と位相が直交する第1成分(Id)と、前記三相電圧と同相の第2成分(Iq)とを求め、
前記第1成分に基づいて第4指令値(Vid*)を求め、
前記出力電圧(Vdc)と前記出力電圧の前記指令値(Vdc*)との差に基づいて前記第2成分の指令値たる第2成分指令値(Iq*)を求め、
前記第2成分と前記第2成分の前記指令値との差に基づいて第5指令値(Viq*)を求め、
前記第4指令値及び前記第5指令値を二相/三相変換して前記第1乃至第3指令値(Vr*,Vs*,Vt*)を求める、請求項2記載のコンバータの制御方法。 - 前記三相電圧(Vu,Vv,Vw)の周波数の三倍以上の周波数帯域において、前記第1成分(Id)、前記第2成分(Iq)、前記第2成分指令値(Iq*)から前記第4指令値及び前記第5指令値が求められる、請求項3記載のコンバータの制御方法。
- 前記交流波形(VK)の周波数は、前記三相電圧(Vu,Vv,Vw)の周波数の三倍以上である、請求項4記載のコンバータの制御方法。
- 前記第4指令値(Vid*)を求めるときに前記1成分(Id)は前記三相電圧(Vu,Vv,Vw)の周波数の三倍の周波数の第1高調波(cos3ωt)によって補正され、
前記第5指令値(Viq*)を求めるときに前記2成分(Iq)は前記第1高調波と位相が直交する第2高調波(sin3ωt)によって補正される、請求項5記載のコンバータの制御方法。 - 低電位(VL)、前記低電位よりも高い高電位(VH)、及び前記低電位と前記高電位との間の中間電位(VQ)を出力する整流回路(203)と、
前記低電位及び前記中間電位が供給される第1コンデンサ(205;7)と、
接続点(35;n)において前記第1コンデンサと接続され、前記高電位及び前記中間電位が供給される第2コンデンサ(204;6)と
を備え、
前記整流回路は
三相電圧(Vu,Vv,Vw)がそれぞれの一端に印加される三幅対のリアクトル(202;8,9,10)のそれぞれの他端における第1乃至第3の電位(Vr,Vs,Vt)を整流して前記低電位及び前記高電位を出力する三相ダイオードブリッジ(18,19,20,24,25,26;D11,D21,D31,D14,D24,D34)と、
前記接続点に、前記リアクトルの三つの前記他端を選択的に接続するスイッチ群(30,31,32;S11,S21,S31,S12,S22,S32)と
を有するコンバータを制御する装置(100)であって、
前記第1乃至第3の電位のそれぞれについての指令値である第1乃至第3指令値(Vr*,Vs*,Vt*)を前記三相電圧から生成する電圧指令値生成部(101;122,104)と、
前記第1乃至第3指令値のそれぞれが前記中間電位の指令値(0)を中心とする交流波形(VK)に対して所定の電位幅を有する範囲内にあるときに、前記スイッチ群をしてそれぞれの指令値に対応する前記他端を前記接続点に接続せしめるスイッチング信号(Srp,Srn,Ssp,Ssn,Stp,Stn;Sr,Ss,St)を生成するパルス幅変調器(102)と
を備える、コンバータの制御装置。 - 前記交流波形の振幅及び前記所定の電位幅のいずれもが、前記高電位(VH)から前記低電位(VL)を引いた電圧たる出力電圧(Vdc)の指令値(Vdc*)の半分である、請求項7記載のコンバータの制御装置。
- 前記リアクトルに流れる三相電流(Iu,Iv,Iw)から、前記三相電圧(Vu,Vv,Vw)と位相が直交する第1成分(Id)と、前記三相電圧と同相の第2成分(Iq)とを求める三相/二相変換器(103)と、
前記第1成分に基づいて第4指令値(Vid*)を求める第1の指令値生成部(108,110,113)と、
前記出力電圧(Vdc)と前記出力電圧の前記指令値(Vdc*)との差に基づいて前記第2成分の指令値たる第2成分指令値(Iq*)を求め、前記第2成分と前記第2成分の前記指令値との差に基づいて第5指令値(Viq*)を求める第2の指令値生成部(105,106,107,109,112)と、
前記第4指令値及び前記第5指令値から前記第1乃至第3指令値(Vr*,Vs*,Vt*)を求める二相/三相変換器(104)と
を更に備える、請求項8記載のコンバータの制御装置。 - 前記三相電圧(Vu,Vv,Vw)の周波数の三倍以上の周波数帯域を有する、請求項9記載のコンバータの制御装置。
- 前記交流波形(VK)の周波数は、前記三相電圧(Vu,Vv,Vw)の周波数の三倍以上である、請求項10記載のコンバータの制御装置。
- 前記第1の指令値生成部は、前記第1成分(Id)を前記三相電圧(Vu,Vv,Vw)の周波数の三倍の周波数の第1高調波(cos3ωt)によって補正する第1補正部(115,116,117,118,120)を有し、
前記第2の指令値生成部は、前記2成分(Iq)を前記第1高調波と位相が直交する第2高調波(sin3ωt)によって補正する第2補正部(114,116,117,119,121)を有する、請求項11記載のコンバータの制御装置。
Priority Applications (5)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
AU2009222727A AU2009222727B2 (en) | 2008-03-14 | 2009-02-17 | Method of and apparatus for controlling converter |
EP09720296.4A EP2254232B1 (en) | 2008-03-14 | 2009-02-17 | Converter control method and control apparatus |
KR1020107019923A KR101139645B1 (ko) | 2008-03-14 | 2009-02-17 | 컨버터의 제어 방법 및 제어 장치 |
US12/922,362 US8395918B2 (en) | 2008-03-14 | 2009-02-17 | Method of and apparatus for controlling three-level converter using command values |
CN200980108970.0A CN101971476B (zh) | 2008-03-14 | 2009-02-17 | 变流器的控制方法以及控制装置 |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP2008065888A JP5167884B2 (ja) | 2008-03-14 | 2008-03-14 | コンバータの制御方法及び制御装置 |
JP2008-065888 | 2008-03-14 |
Publications (1)
Publication Number | Publication Date |
---|---|
WO2009113367A1 true WO2009113367A1 (ja) | 2009-09-17 |
Family
ID=41065039
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/JP2009/052695 WO2009113367A1 (ja) | 2008-03-14 | 2009-02-17 | コンバータの制御方法及び制御装置 |
Country Status (7)
Country | Link |
---|---|
US (1) | US8395918B2 (ja) |
EP (1) | EP2254232B1 (ja) |
JP (1) | JP5167884B2 (ja) |
KR (1) | KR101139645B1 (ja) |
CN (1) | CN101971476B (ja) |
AU (1) | AU2009222727B2 (ja) |
WO (1) | WO2009113367A1 (ja) |
Families Citing this family (11)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP4254876B2 (ja) * | 2007-03-30 | 2009-04-15 | ダイキン工業株式会社 | 電源供給回路及びそのpam制御方法 |
CA2868700C (en) | 2012-03-30 | 2016-10-11 | Toshiba Mitsubishi-Electric Industrial Systems Corporation | Power supply apparatus |
CN103630857B (zh) * | 2012-08-27 | 2016-03-02 | 上海联影医疗科技有限公司 | Pin二极管的控制装置和核磁共振设备 |
EP2893628B1 (en) * | 2012-09-05 | 2020-03-04 | ABB Schweiz AG | Interleaved 12-pulse rectifier |
KR101512188B1 (ko) * | 2014-02-11 | 2015-04-22 | 한국전기연구원 | 모듈형 멀티레벨 컨버터의 구동방법 및 구동장치 |
JP6952245B2 (ja) * | 2016-09-30 | 2021-10-20 | パナソニックIpマネジメント株式会社 | 電力変換システム |
US11394295B2 (en) * | 2017-09-27 | 2022-07-19 | Toshiba Mitsubishi-Electric Industrial Systems Corporation | Power supply apparatus |
US11290003B2 (en) * | 2019-06-20 | 2022-03-29 | Toshiba Mitsubishi-Electric Industrial Systems Corporation | Power conversion device |
CN110855164B (zh) * | 2019-11-29 | 2021-04-06 | 深圳市科华恒盛科技有限公司 | 控制方法、系统及终端设备 |
US12107513B2 (en) * | 2020-03-27 | 2024-10-01 | Mitsubishi Electric Corporation | Three-level power converter and method of controlling intermediate potential of direct current power supply unit |
WO2022138038A1 (ja) * | 2020-12-22 | 2022-06-30 | パナソニックIpマネジメント株式会社 | 電力変換制御装置 |
Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPH06233537A (ja) * | 1993-02-01 | 1994-08-19 | Toshiba Corp | 中性点クランプ式コンバータの制御装置 |
JPH09182441A (ja) * | 1995-12-28 | 1997-07-11 | Toshiba Corp | 三相整流装置 |
JPH09238478A (ja) * | 1996-03-04 | 1997-09-09 | Hitachi Ltd | 多重電力変換器およびその制御方法 |
JP2003174779A (ja) * | 2001-09-28 | 2003-06-20 | Daikin Ind Ltd | 電力変換装置 |
JP2004104909A (ja) * | 2002-09-09 | 2004-04-02 | Sanken Electric Co Ltd | 三相スイッチング整流装置 |
Family Cites Families (13)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2754519B2 (ja) | 1993-12-13 | 1998-05-20 | 東洋電機製造株式会社 | 3レベルインバータ装置 |
AT406434B (de) | 1993-12-23 | 2000-05-25 | Ixys Semiconductor Gmbh | Vorrichtung zur umformung eines dreiphasigen spannungssystems in eine vorgebbare, einen verbraucher speisende gleichspannung |
JP3166525B2 (ja) * | 1994-12-28 | 2001-05-14 | 三菱電機株式会社 | 誘導電動機のベクトル制御装置 |
US6031738A (en) * | 1998-06-16 | 2000-02-29 | Wisconsin Alumni Research Foundation | DC bus voltage balancing and control in multilevel inverters |
US6545887B2 (en) * | 1999-08-06 | 2003-04-08 | The Regents Of The University Of California | Unified constant-frequency integration control of three-phase power factor corrected rectifiers, active power filters and grid-connected inverters |
BR9907351A (pt) * | 1999-12-22 | 2001-08-07 | Ericsson Telecomunicacoees S A | Método e circuito de controle para retificador do tipo elevador trifásico de três nìveis |
JP4051875B2 (ja) | 2000-10-31 | 2008-02-27 | 富士電機ホールディングス株式会社 | 整流回路及びその制御方法 |
WO2005043742A2 (en) * | 2003-10-30 | 2005-05-12 | The Regents Of The University Of California | Universal three phase controllers for power converters |
JP4649940B2 (ja) | 2004-10-14 | 2011-03-16 | ダイキン工業株式会社 | コンバータの制御方法及びコンバータの制御装置 |
US7196919B2 (en) * | 2005-03-25 | 2007-03-27 | Tyco Electronics Power Systems, Inc. | Neutral point controller, method of controlling and rectifier system employing the controller and the method |
JP4770639B2 (ja) * | 2006-08-17 | 2011-09-14 | アイシン・エィ・ダブリュ株式会社 | 電気モータ駆動制御方法および装置 |
US20090040800A1 (en) * | 2007-08-10 | 2009-02-12 | Maximiliano Sonnaillon | Three phase rectifier and rectification method |
US7986538B2 (en) * | 2008-06-03 | 2011-07-26 | Hamilton Sundstrand Corporation | Midpoint current and voltage regulation of a multi-level converter |
-
2008
- 2008-03-14 JP JP2008065888A patent/JP5167884B2/ja active Active
-
2009
- 2009-02-17 KR KR1020107019923A patent/KR101139645B1/ko active IP Right Grant
- 2009-02-17 US US12/922,362 patent/US8395918B2/en active Active
- 2009-02-17 WO PCT/JP2009/052695 patent/WO2009113367A1/ja active Application Filing
- 2009-02-17 AU AU2009222727A patent/AU2009222727B2/en active Active
- 2009-02-17 EP EP09720296.4A patent/EP2254232B1/en active Active
- 2009-02-17 CN CN200980108970.0A patent/CN101971476B/zh active Active
Patent Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPH06233537A (ja) * | 1993-02-01 | 1994-08-19 | Toshiba Corp | 中性点クランプ式コンバータの制御装置 |
JPH09182441A (ja) * | 1995-12-28 | 1997-07-11 | Toshiba Corp | 三相整流装置 |
JPH09238478A (ja) * | 1996-03-04 | 1997-09-09 | Hitachi Ltd | 多重電力変換器およびその制御方法 |
JP2003174779A (ja) * | 2001-09-28 | 2003-06-20 | Daikin Ind Ltd | 電力変換装置 |
JP2004104909A (ja) * | 2002-09-09 | 2004-04-02 | Sanken Electric Co Ltd | 三相スイッチング整流装置 |
Also Published As
Publication number | Publication date |
---|---|
AU2009222727B2 (en) | 2012-10-18 |
KR20100107076A (ko) | 2010-10-04 |
EP2254232A4 (en) | 2018-01-03 |
AU2009222727A1 (en) | 2009-09-17 |
JP2009225525A (ja) | 2009-10-01 |
US8395918B2 (en) | 2013-03-12 |
KR101139645B1 (ko) | 2012-05-15 |
JP5167884B2 (ja) | 2013-03-21 |
EP2254232B1 (en) | 2019-11-27 |
EP2254232A1 (en) | 2010-11-24 |
CN101971476B (zh) | 2015-10-21 |
US20110085361A1 (en) | 2011-04-14 |
CN101971476A (zh) | 2011-02-09 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
JP5167884B2 (ja) | コンバータの制御方法及び制御装置 | |
KR101072647B1 (ko) | 전력 변환 장치 | |
AU2009222340B2 (en) | State quantity detection method in power converting apparatus and power converting apparatus | |
CN109874384B (zh) | 直接型电力转换器用的控制装置 | |
JP5907294B2 (ja) | 電力変換装置 | |
US11218107B2 (en) | Control device for power converter | |
TW201330479A (zh) | 電力轉換裝置 | |
WO2017047489A1 (ja) | インバータ基板、接続順序の判断方法、欠相判断方法 | |
JP5888074B2 (ja) | 電力変換装置 | |
JP2012044830A (ja) | 電力変換装置 | |
JP5953881B2 (ja) | 3レベル整流器の制御装置 | |
JP4703251B2 (ja) | 電源装置の運転方法及び電源装置 | |
JP6492031B2 (ja) | 電圧補償装置 | |
JP4924587B2 (ja) | 直接形交流電力変換装置の制御方法 | |
JP2019054569A (ja) | 3レベル電力変換器 | |
JP2022066920A (ja) | 電力変換システム | |
JP2968027B2 (ja) | 電流形インバータの制御装置 | |
JP6337688B2 (ja) | 電力変換装置、発電システムおよび電力変換方法 | |
JPH01298959A (ja) | Pwmコンバータ装置 | |
Misra | Decoupled Vector Control of Grid Side Converter with Less Number of Sensors under imbalanced Grid Conditions | |
JP6201386B2 (ja) | 電流推定装置 | |
CN113039696A (zh) | 电力变换装置 |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
WWE | Wipo information: entry into national phase |
Ref document number: 200980108970.0 Country of ref document: CN |
|
121 | Ep: the epo has been informed by wipo that ep was designated in this application |
Ref document number: 09720296 Country of ref document: EP Kind code of ref document: A1 |
|
ENP | Entry into the national phase |
Ref document number: 20107019923 Country of ref document: KR Kind code of ref document: A |
|
WWE | Wipo information: entry into national phase |
Ref document number: 12922362 Country of ref document: US |
|
WWE | Wipo information: entry into national phase |
Ref document number: 2009720296 Country of ref document: EP |
|
NENP | Non-entry into the national phase |
Ref country code: DE |
|
WWE | Wipo information: entry into national phase |
Ref document number: 2009222727 Country of ref document: AU |
|
ENP | Entry into the national phase |
Ref document number: 2009222727 Country of ref document: AU Date of ref document: 20090217 Kind code of ref document: A |