WO2009084159A1

WO2009084159A1 - Radar apparatus

Info

Publication number: WO2009084159A1
Application number: PCT/JP2008/003692
Authority: WO
Inventors: Yoshihide Uesato; Tomohiro Shinomiya; Katsuyuki Ohguchi; Kunihiro Tanaka
Original assignee: Fujitsu Ten Limited
Priority date: 2007-12-28
Filing date: 2008-12-10
Publication date: 2009-07-09
Also published as: JP2009162521A

Abstract

A radar apparatus wherein even though a pair of propagation paths are provided which propagate transport signals to a pair of transmission antennas spaced by a given distance and even though a switch is provided which distributes the transport signal to the pair of propagation paths, yet the search performance is prevented from degrading. The radar apparatus has an oscillator that outputs a radar signal the frequency of which is lower than the predetermined frequency; a switch that distributes the radar signal outputted by the oscillator to the propagation path pair lead to the transmission antenna pair; and a pair of multipliers that are connected to the pair of propagation paths, respectively, and that convert the frequencies of the radar signals to the predetermined frequency for output to the transmission antenna pair. Thus, the frequencies of the transport signals passing through the switch and along the propagation paths are reduced, whereby the insertion loss of the switch and the propagation loss of the propagation paths can be suppressed, thereby preventing the search performance from degrading due to a decreasing intensity of the transport signals.

Description

Radar equipment

The present invention includes a transmission antenna pair that transmits a radar signal having a predetermined frequency and a reception antenna group that receives the radar signal reflected by a target as a reception signal. The present invention relates to a radar apparatus that detects an azimuth angle at which a target is located by processing a received signal from a group of receiving antennas obtained when performing the operation.

An electronic scanning method is known as a scanning method for an on-vehicle radar device. In the electronic scanning method, the radar apparatus transmits a radar signal from a transmission antenna, and receives the radar signal reflected on the target as a reception signal by a plurality of reception antennas arranged at predetermined intervals. . Then, the radar apparatus detects the azimuth angle at which the target is located using a plurality of received signals having different phases and amplitudes depending on the direction of arrival.

FIG. 1 shows a configuration example of a conventional electronic scan type radar apparatus. In the radar apparatus 2, the voltage controlled oscillator (VCO) 8 outputs a radar signal frequency-modulated by an FM-CW (Frequency Modulated-Continuous Wave) system in accordance with the modulation signal output from the control unit 6. Distributor 1
0 is a radar signal for transmission (hereinafter referred to as a transmission signal) ST, and a radar signal (hereinafter simply referred to as a local signal) SL as a local signal used for processing the received signal. Power distribution to The transmission signal ST is amplified by the amplifier 12_1 and transmitted from the transmission antenna AT_1. When this signal is reflected by the target, it is received as reception signals SR_1, SR_2, SR_3, SR_4 by reception antennas AR_1, AR_2, AR_3, AR_4 arranged at regular intervals. The received signal SR_n (hereinafter, when referring to the plurality of components described above, the subscript is represented by “_n”) is amplified by the amplifiers 14_1, 14_2, 14_3, and 14_4, respectively, and the mixers 16_1, 16_2, 16_3, 16_4.

On the other hand, the local signal SL distributed by the distributor 10 is amplified by the amplifier 12_2 and input to the mixer 16_n. The mixer 16_n mixes the local signal SL and the reception signal SR_n, respectively, and a frequency difference between the local signal SL and the reception signal SR_n, that is, a frequency difference between the transmission signal ST and the reception signal SR_n. Difference signals (hereinafter referred to as beat signals) SB_1, SB_2, SB_3, and SB_4 are output. The bit signal SB_n is sampled by the A / D converters 18_1, 18_2, 18_3, and 18_4 and converted into digital data, and then taken into the control unit 6.

The control unit 6 is mainly composed of a well-known microcomputer having a CPU, a ROM, and a RAM. Further, the controller 6 performs high-speed Fourier transform (FFT) on the data taken in via the A / D converter 18_n. : An arithmetic processing unit (for example, DSP: Digital Signal) for executing Fast Fourier Transform (Fast Fourier Transform) processing
Processor). Here, the controller 6 detects the azimuth angle at which the target is located by performing DBF (Digital Beam Forming) processing on the beat signal SB_n.

In the DBF processing, when the reception signals SR_n from the plurality of reception antennas AR_n arranged at regular intervals are combined, the gain is maximized at the azimuth angle where the directivities of the reception antennas AR_n match. The calculation is performed backward as the arrival direction of the received signal, that is, the azimuth angle where the target is located. That is, the control unit 6 obtains the combined amplitude of the beat signal SB_n generated from the received signal SR_n as a function of the azimuth, and calculates the azimuth corresponding to the maximum value of the combined amplitude. *

Here, as the number of receiving antennas increases, that is, as the number of beat signals to be combined increases, the change curve of the combined amplitude becomes steeper, so the azimuth corresponding to the maximum value of the combined amplitude is more accurate. I can find it well. For this purpose, it is desirable to arrange a large number of receiving antennas.

However, on the other hand, in-vehicle radar devices are required to be downsized due to restrictions on installation space and manufacturing costs. For this reason, there has been proposed a method for realizing a configuration equivalent to a configuration in which a large number of receiving antennas are arranged without increasing the scale of the apparatus by providing a plurality of transmitting antennas instead of arranging a large number of receiving antennas. . A configuration example of such a radar apparatus is shown in FIG.

The radar apparatus 2 shown in FIG. 2 further distributes the power of the transmission signal ST distributed by the distributor 10 and amplifiers 22_1 and 22_2 that amplify the transmission signal ST distributed by the distributor 20, respectively. 1 in that it has a pair of transmission antennas AT_1 and AT_2 that transmit the amplified transmission signal ST.

In this configuration example, the transmission antennas AT_1 and AT_2 are spaced apart from each other by being positioned at both ends of the array of the reception antennas AR_n. Therefore, a difference occurs in the path length from both transmitting antennas to the target according to the interval and the azimuth angle at which the target is located. Therefore, the round trip path length when the transmission signal from the transmission antenna AT_1 is reflected by the target and received by the reception antenna AR_n, and the transmission signal from the transmission antenna AT_2 is reflected by the target and is received by the reception antenna AR_n. Each of the round trip path lengths when received is different. Therefore, in each combination of the transmission antennas AT_1 and AT_2 and the reception antenna AR_n, a plurality of reception signals SR_n having different amplitudes and phases are obtained.

As a result, the positional relationship between the transmitting antenna AT_1 (shown in black) and the receiving antenna AR_n shown in FIG. 3A and the transmitting antenna AT_2 (shown in black) shown in FIG. 3) and the positional relationship between the receiving antennas AR_n and combining the positions of the transmitting antennas that transmit each other in combination, a configuration equivalent to the configuration in which a double number of receiving antennas are arranged as shown in FIG. can get. That is, a configuration equivalent to a configuration including one transmission antenna and eight reception antennas can be realized by two transmission antennas and four reception antennas.

By the way, in the radar apparatus 2 of FIG. 2, since the power of the transmission signal ST is reduced by the power distribution by the distributor 20, the amplifiers 22_1 and 22_2 use the intensity of the distributed transmission signal ST as the expected power. Amplify until. However, in order to compensate for the decrease in strength due to power distribution, a certain degree of amplification is required. Then, noise is also amplified at the same time, and a transmission signal ST having a good S / N ratio cannot be obtained. Therefore, a method for avoiding such a situation by using a switch that alternately outputs the transmission signal ST to the transmission antenna pair AT_1 and AT_2 instead of the distributor 20 has been proposed.

FIG. 4 shows the configuration of the radar apparatus in an example of such a method. This radar apparatus 2 has a switch SW_1 that alternately outputs the transmission signal ST to the transmission paths P_1 and P_2 corresponding to the transmission antennas AT_1 and AT_2, instead of the distributor 20 that distributes the transmission signal ST. This is different from the configuration example of FIG. Here, the switch SW_1 is switch-controlled by the control unit 6.

FIG. 5 shows a configuration of a radar apparatus in still another example. The radar apparatus 2 further includes a switch SW_2 that sequentially switches the reception signal SR_n from the reception antenna AR_n and inputs the signal to one mixer 16, and the beat signal SB output from the mixer 16 is one A / D conversion. 4 is different from the configuration of FIG. Here, the switch SW_2 is switch-controlled by the control unit 6. In this configuration, the number of mixers and A / D converters is reduced by providing the switch SW_2, and the circuit scale of the radar apparatus 2 is further reduced.

Still another example is described in Patent Document 1. The radar apparatus described in Patent Document 1 has a configuration in which a pair of transmission antennas are provided at both ends of a plurality of arranged reception antenna groups, and a radar signal is transmitted from each of them in a time division manner.
JP 2004-198312 A

In the above-described FIG. 2, FIG. 4, FIG. 5, or the example of Patent Document 1, the transmission paths P_1 and P_2 for transmitting the distributed transmission signals are indispensable for a plurality of transmission antennas spaced apart by a fixed interval. . Here, in an on-vehicle radar device, the frequency band of radio waves that can be used is limited to a high frequency band of 76.0 to 77.0 GHz, which is included in the millimeter wave band, due to legal restrictions. Then, when the millimeter-wave band transmission signal is transmitted through the transmission paths P_1 and P_2, the conductor resistance of the transmission path, that is, the sum of the surface resistance of the transmission path due to the skin effect and the internal inductance is the frequency of the transmission signal. It grows according to. Furthermore, when the transmission paths P_1 and P_2 are formed on a dielectric like a microstrip line, dielectric loss also occurs. Therefore, when the transmission signal ST is transmitted to the transmission antenna pair, the strength of the transmission signal ST is reduced due to transmission loss.

Further, when the switch SW_1 that distributes the transmission signal ST output from the VCO 8 to the transmission paths P_1 and P_2 has a large insertion loss according to the frequency of the signal passing through, the switch SW_1 also passes through the switch SW1. The intensity of the transmission signal ST decreases.

And, when the intensity of the transmission signal ST is lowered in this way, there arises a problem that the detection performance as the radar device is lowered.

Therefore, an object of the present invention is to reduce the detection performance even in a configuration including a transmission path pair that transmits a transmission signal to a pair of transmission antennas spaced apart by a predetermined interval and a switch that distributes the transmission signal to the transmission path pair. It is an object of the present invention to provide a radar device that can be prevented.

In order to achieve the above object, a radar apparatus according to a first aspect of the present invention includes a pair of transmitting antennas that transmit a radar signal of a predetermined frequency and a radar that is reflected by a target. A radar that receives a signal as a received signal, and detects a azimuth angle at which the target is located using a received signal obtained by alternately transmitting the radar signal from the transmitting antenna pair. An oscillator that outputs a radar signal having a frequency lower than the predetermined frequency; a radar signal that is output from the oscillator; a first transmission path to the first transmitting antenna; A switch that distributes to a second transmission path to the transmission antenna and a first transmission antenna that is connected to the first transmission path and converts the frequency of the radar signal to the predetermined frequency. Output to the first And a second multiplier that is connected to the second transmission path and converts the frequency of the radar signal to the predetermined frequency and then outputs the second signal to the second transmission antenna. And

In a preferred embodiment of the above aspect, the predetermined frequency is a millimeter wave band, and the frequency of the radar signal output from the oscillator is a microwave band lower than the millimeter wave band. In general, the millimeter wave band is a band of 30 GHz to 300 GHz, and the microwave band is a band of 1 GHz to 1000 GHz. However, in the following description, when simply referred to as a microwave band, a band lower than the millimeter wave band of 30 GHz to 300 GHz, that is, It means a band of 1 GHz to 30 GHz.

The radar device according to the above aspect includes an oscillator that outputs a radar signal having a frequency (microwave band) lower than the predetermined frequency (millimeter wave band), and a first transmission path to the first transmitting antenna. A switch that distributes the second transmission path to the second transmission antenna, and a switch that is connected to the first transmission path and converts the frequency of the radar signal to the predetermined frequency and then the first transmission path. A first multiplier that outputs to the second transmission antenna, and a second multiplier that is connected to the second transmission path, converts the frequency of the radar signal to the predetermined frequency, and then outputs the second signal to the second transmission antenna. And a multiplier. Therefore, even if the switch has a higher insertion loss as the frequency of the radar signal passing therethrough is higher, the switch is used as a radar signal having a frequency (microwave band) lower than a predetermined frequency (millimeter wave band). By passing it through, the insertion loss can be suppressed smaller than when passing through the switch with the radar signal having a predetermined frequency. Further, when the radar signal is transmitted through the first and second transmission paths, the transmission loss increases according to the frequency. However, the frequency of the transmission signal is lowered to reduce the first and second transmission paths. By transmitting, transmission loss can be suppressed small. As described above, since it is possible to prevent a decrease in the intensity of the radar signal transmitted to the transmission antenna via the switch and the transmission path, it is possible to prevent a decrease in the detection accuracy of the radar apparatus.

It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus. It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus different from FIG. It is a figure explaining the positional relationship of a transmitting antenna pair and a receiving antenna group. FIG. 3 is a diagram illustrating a configuration example of a conventional electronic scanning radar apparatus different from FIG. 2. FIG. 5 is a diagram showing a configuration example of a conventional electronic scanning radar apparatus different from FIG. 4. It is a figure explaining the 1st structural example of the radar apparatus in this embodiment. It is a figure explaining the insertion loss of a switch, and the transmission loss of a transmission path. It is a figure which shows the change of the frequency with respect to the time of a transmission / reception signal. It is a figure explaining DBF processing. It is a figure explaining the modification of the 1st example of composition. It is a figure explaining the 2nd structural example of the radar apparatus in this embodiment. It is a figure explaining the modification of the 2nd example of composition. It is a figure explaining the use condition in which a radar apparatus is used as a vehicle-mounted radar apparatus.

Explanation of symbols

2: radar device, AT_n: transmitting antenna, AR_n: receiving antenna, 6: control unit, 8: VCO, SW_n: switch, P_n, T_n: transmission path, 30_n, 32_n: multiplier

Hereinafter, embodiments of the present invention will be described with reference to the drawings. However, the technical scope of the present invention is not limited to these embodiments, but extends to the matters described in the claims and equivalents thereof.

FIG. 6 is a diagram for explaining a first configuration example of the radar apparatus according to the present embodiment. This radar device 2 is different from the conventional configuration shown in FIG. 4 in the following points.

In the first configuration example, the oscillation frequency of the VCO 8 is set to 1 / N (N is a natural number; here, N = 4) of the conventional example shown in FIG. Further, multipliers 30_1 and 30_2 that multiply the frequency of the transmission signal ST distributed by the switch SW_1 by N and output to the transmission antennas AT_1 and AT_2 are connected to the ends of the transmission paths P_1 and P_2. Further, a multiplier 30_3 that similarly multiplies the frequency of the local signal SL by N and outputs it to the mixer 16_n is connected to the transmission path P_4 of the local signal SL.

Here, the VCO 8, the divider 10, the amplifier 12_n, the switch SW_1, the multiplier 30_n, and the amplifier 22_n are modularized to transmit MMIC (Microwave Monolithic IC).
The amplifier 14_n, the mixer 16_n, and the A / D converter 18_n are modularized and configured by a receiving MMIC. Each module is composed of a combination of a high power material (GaAs, SiGe, etc.) and a low power material (Si).

In the first configuration example, the switch SW_1 is configured by a switch circuit using a diode, and is controlled to be switched by the control unit 6. The switch SW_1 has one input terminal and three output terminals. One of the output terminals is connected to the transmission path P_1, the other is connected to the transmission path P_2, and the other is connected to the transmission path P_3. Here, the output to the transmission path P_3 is located in the middle of the outputs to the transmission paths P_1 and P_2, and is output to the transmission path P_1 (or P_2) by connecting the transmission path P_3 to the resistor R. Since the radiated wave of the transmission signal ST is absorbed by the resistor R in the transmission path P_3, it is possible to prevent the generation of noise due to the radiated wave entering the transmission path P_2 (or P_1). As a result, the isolation of the switch SW_1 can be improved. Further, when the insertion loss with respect to the frequency of the signal passing through the switch SW_1 is shown using FIG. 7A, the frequency of the passing signal (horizontal axis) increases, whereas the insertion loss of the passing signal (vertical axis) ) Will increase. Therefore, according to the first configuration example, the transmission signal ST in the microwave band passes through the switch SW_1 instead of the transmission signal ST in the millimeter wave band, so that the strength of the transmission signal ST is reduced due to the insertion loss of the switch SW_1. Can be kept smaller than in the millimeter wave band.

Then, after the frequency of the transmission signal ST is multiplied to a predetermined frequency (for example, 76.5 GHz) in the millimeter wave band by the multipliers 30_1 and 30_2, the transmission signal ST is transmitted from the transmission antennas AT_1 and AT_2. With such a configuration, even if the transmission signal ST is alternately switched by the switch SW_1 and alternately output, the strength of the transmission signal ST can be prevented from decreasing and the transmission signal ST having a desired frequency can be transmitted.

When the frequency of the transmission signal ST increases, the surface resistance of the transmission path P_1, P_2, or P_3 in the skin effect increases in proportion to the square root of the frequency, and the internal inductance increases in proportion to the frequency. The conductor resistance, which is the sum, increases with frequency. Further, when the transmission path P_1, P_2, or P_3 is formed on a dielectric like a microstrip line, dielectric loss also occurs. This is shown in FIG. 7 (B). FIG. 7B shows that the transmission loss (vertical axis) of different types of transmission paths increases as the frequency (horizontal axis) of the transmitted signal increases.

Also in this respect, in the first configuration example, the transmission signal ST in the microwave band is transmitted instead of the transmission signal ST in the millimeter wave band, and this is multiplied to the millimeter wave band just before being transmitted, and then transmitted to the transmission antenna AT_1. , Transmitted from AT_2, or multiplied to the millimeter wave band and input to the mixer 16_n, transmission loss in the transmission paths P_1 and P_2 can be suppressed to be smaller than that in the millimeter wave band. Therefore, since the transmission antennas AT_1 and AT_2 are spaced apart from each other by a certain distance, the strength of the transmission signal ST transmitted through the transmission paths P_1 and P_2 is reduced even when the transmission paths P_1 and P_2 are of a certain length. Can be avoided. Similarly, transmission loss can be suppressed in the transmission path from the VCO 8 to the switch SW_1, and a decrease in the strength of the transmission signal is avoided. Furthermore, a decrease in the strength of the local signal SL input to the mixer 16_n is also avoided. Therefore, it is possible to prevent a decrease in detection accuracy as a radar device.

In the first configuration example, the transmission antennas AT_1 and AT_2 are arranged at both ends of the reception antenna AR_n arranged, which is equivalent to the case where the reception antennas are doubled as shown in FIG. The configuration can be realized. However, the interval between the transmitting antennas AT_1 and AT_2 may not be limited to this. The round-trip path length until the radar signals transmitted from the transmitting antennas AT_1 and AT_2 are reflected by the target and received by the receiving antenna AR_n is different, and therefore, the received signals having different phases and amplitudes are received by all the transmitting antennas. As long as the transmission antennas AT_1 and AT_2 are spaced apart from each other as obtained in combination with the antenna, a configuration equivalent to the case where the number of reception antennas is larger than that with a limited number of reception antennas can be realized. In any case, when the transmission paths P_1 and P_2 having a certain length according to the interval between the transmitting antennas AT_1 and AT_2 are required, the above-described effects can be obtained.

Here, the operation of each unit in the first configuration example will be described. When the FM-CW type triangular wave modulation signal is input from the control unit 6, the VCO 8 has a lower band than the millimeter wave band modulated so that the frequency linearly increases and decreases with time. A microwave band radar signal is generated. For example, in the conventional configuration shown in FIG. 4, if the oscillation frequency of the VCO 8 is set so that the center frequency of the radar signal is 76.5 GHz and the frequency fluctuation width is 100 MHz, the first configuration example The oscillation frequency of the VCO 8 is set so that the center frequency of the radar signal is about 19.1 (= 76.5 / 4) GHz and the frequency fluctuation range is 100 MHz.

When the power of the radar signal is distributed to the transmission signal ST and the local signal SL by the distributor 10, the transmission signal ST is amplified by the amplifier 12_1 and input to the switch SW_1.

The switch SW_1 switches the output destination of the transmission signal ST alternately to the transmission paths P_1 and P_2 in time division in response to the control signal from the control unit 6. Thus, the transmission signal ST output alternately from the switch SW_1 is input to the multipliers 30_1 and 30_2 via the transmission paths P_1 and P_2. The multiplier 30_n multiplies the frequency of the input transmission signal ST to a predetermined frequency of 76.5 GHz in the millimeter wave band and outputs the result. Then, the frequency-multiplied transmission signal ST is amplified by the amplifiers 22_1 and 22_2 and transmitted from the transmission antennas AT_1 and AT_2.

As described above, the control unit 6 alternately switches the output destination of the switch SW_1, so that the transmission signal ST having a predetermined frequency in the millimeter wave band is alternately transmitted from the transmission antennas AT_1 and AT_2.

On the other hand, the local signal SL distributed by the distributor 10 is transmitted through the transmission path P_4, is amplified by the amplifier 12_2, and is input to the multiplier 30_3. Then, the multiplier 30_3 multiplies the frequency of the input local signal ST to the same frequency 76.5 GHz as that of the transmission signal ST, and outputs the result to the mixer 16_n.

When the radar signal transmitted as the transmission signal is reflected by the target, it is received by the reception antenna AR_n as the reception signal SR_n. The received signal SR_n is amplified by the amplifier 14_n and input to the mixer 16_n.

The mixer 16_n mixes the local signal SL having the same frequency as the transmission signal ST and the reception signal SR_n to generate a frequency difference signal (bit signal) SB_n between the transmission signal ST and the reception signal SR_n. .

FIG. 8 shows changes in frequency (vertical axis) with respect to time (horizontal axis) of each signal. As described above, the transmission signal ST is frequency-modulated according to a triangular wave-like modulation signal. Therefore, as indicated by a solid line in FIG. 8A, the frequency of the transmission signal ST gradually increases or decreases linearly with respect to time. The received signal SR_n is delayed by a time ΔT in which the radar signal reciprocates the distance from the target as shown by a dotted line in FIG. 8A, and only by a frequency ΔD corresponding to the relative speed with the target. Doppler shift.

When such a reception signal SR and transmission signal ST are mixed, a beat signal SB_n having a frequency corresponding to the frequency difference between the two is generated as shown in FIG. 8B. When the frequency of the transmission signal ST increases (hereinafter referred to as the frequency increase period), the frequency of the beat signal (hereinafter referred to as “upbeat frequency”) fu and when the frequency of the transmission signal ST decreases (hereinafter referred to as the frequency) The relative distance Rd of the target is obtained by the following equation (1) and the relative velocity Rv is obtained by the equation (2) from the frequency (hereinafter referred to as “downbeat frequency”) fd of the beat signal during the descent period). It is done. Here, C is a radio wave propagation speed (light speed), fm is a modulation frequency of the transmission signal ST, ΔF is a frequency fluctuation range of the transmission signal ST, and fo is a center frequency of the transmission signal ST.

(1) Rd = C · (fu + fd) / (8 · ΔF · fm)
(2) Rv = C · (fd−fu) / (4 · fo) Such a beat signal SB_n is sent from the mixer 16_n via the A / D converter 18_n for each modulation period of the transmission signal ST. Input to the control unit 6.

The control unit 6 performs an FFT process on the beat signal SB_n for each modulation period of the transmission signal ST. Then, the phase and amplitude are detected from each of the beat signals SB_n for each up / down beat frequency. Then, the control unit 6 performs DBF processing using the beat signal SB_n for each up / down beat frequency.

Here, DBF processing will be described with reference to FIG. A reception signal SR_n arriving from a target that is sufficiently far away from the interval (d) of the reception antenna AR_n is considered to be incident on the reception antenna AR_n in parallel. Here, assuming that the received signal SR_n is incident from the azimuth angle θ with respect to the vertical direction of the receiving surface H1 on which the receiving antenna AR_n is arranged, the antenna interval d is between the received signals SR_n between the adjacent receiving antennas AR_n. Phase difference φ (= 2π · d · sin (θ) / λ, where λ is the wavelength of the received signal SR_n).

Here, on the reception surface H1, since the reception signal SR_n has a phase difference φ, there is a portion where the amplitudes of the reception signals SR_n cancel each other. However, all the received signals SR_n have the same phase (hereinafter, the plane H2 is referred to as the same phase plane) in the plane H2 (plane having an angle θ with respect to the reception plane H1) H2 with respect to the arrival direction of the received signal SR_n. Therefore, on the same phase plane H2, the reception signals SR_n intensify each other and the combined amplitude becomes maximum. This means that the angle θ of the in-phase surface H2 corresponds to the azimuth angle at which the reception gain of the receiving antenna AR_n is maximized (that is, the directivity is the same), and the angle θ of the in-phase surface is calculated. Thus, the arrival direction of the received signal SR_n, that is, the azimuth angle at which the target is located is obtained.

Using this fact, the control unit 6 first obtains the combined amplitude of the beat signal SB_n of the upbeat frequency obtained from the received signal SR_n during the frequency rise period. Here, assuming that the amplitude of the beat signal SB_n at each receiving antenna AR_n is E_n, the phase φn of SB_n (n = 2, 3, 4) with reference to the beat signal SB_1 is φn = 2π · d · ( Since (n-1) · sin (θ) / λ, the combined amplitude E of all the bit signals SB_n (n = 1, 2, 3, 4) is obtained by Expression (3).

By the way, in Expression (3), φi = 2π · d · (i−1) · sin (θ) / λ, and therefore, the combined amplitude E is expressed as a function of θ. Therefore, the control unit 6 calculates θ corresponding to the maximum value of the combined amplitude E in such a function. Then, the calculation result corresponds to the azimuth angle at which the target is located during the frequency increase period.

The control unit 6 performs the same processing for the beat signal SB_n having the downbeat frequency in the frequency falling period. Then, the azimuth angle at which the target is located during the frequency drop period is obtained.

Such processing is performed for each frequency increase period and frequency decrease period. Then, the intensity distribution of the beat signal SB_n in the azimuth angle direction and the frequency direction is obtained for each frequency increase period and frequency decrease period.

Next, the control unit 6 detects the frequency at which the intensity distribution in the azimuth direction forms a peak as the up / down beat frequency. At this time, the azimuth angles of the two peaks associated with each other are detected as the azimuth angle at which the target is located.

Further, the control unit 6 specifies the up / down beat frequency by associating peaks having the same intensity or azimuth angle between the frequency increase period and the frequency decrease period, and the above-described equation (1) The relative distance and relative speed of the target are calculated according to (2).

As shown in FIG. 10, the radar apparatus 2 includes a switch SW_3 that alternately outputs a local signal SL transmitted through the transmission path P_4 to a plurality of transmission paths T_1, T_2, T_3, and T_4, and a switch Multipliers 32_1, 32_2, 32_3, and 32_4 that multiply the frequency of the local signal SL that has passed through SW_3 and then input to the mixer may be provided. The output timing from the switch SW_3 is controlled by the control unit 6, and the local signal SL is input to the multiplier 32_n in a time division manner.

Here, after the local signal SL in the microwave band passes through the switch SW_3, the frequency is multiplied to a predetermined frequency in the millimeter wave band by the multiplier 32_n. Therefore, the insertion loss due to the switch SW_3 is suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Also, the loss in the transmission paths P_4 and T_n is suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Therefore, the mixer 16_n can generate the beat signal SB_n using the local signal SL with sufficient strength.

FIG. 11 is a diagram for explaining a second configuration example of the radar apparatus 2 in the present embodiment. The radar apparatus 2 is different from the radar apparatus 2 in the first configuration example shown in FIG. 6 in the following points.

That is, the radar apparatus 2 receives one A / D converter 18 instead of the plurality of A / D converters 18_n and the beat signal SB_n output from the mixer 16_n according to an instruction from the control unit 6. It has switch SW_2 which inputs into A / D converter 18 by turns. With such a configuration, the circuit scale can be reduced as compared with the first configuration example having the plurality of A / D converters 18_n. Therefore, the cost can be reduced.

Note that the frequency of the beat signal SB_n output from the mixer 16_n is a frequency band that is somewhat low because it is the frequency difference between the transmission signal ST and the reception signal SR_n in the millimeter wave band. For this reason, the insertion loss when the beat signal SB_n passes through the switch SW_2 is kept small, and the signal strength of the beat signal SB_n is secured to such an extent that the signal processing in the control unit 6 is not hindered.

As shown in FIG. 12, the radar apparatus 2 uses the switch SW_3 that alternately outputs the local signal SL to a plurality of transmission paths T_n, and the frequency of the local signal SL that has passed through the switch SW_3. A multiplier 32_n that multiplies and then inputs to the mixer may be provided. In this case, the input / output timing of the switch SW_3 is controlled by the control unit 6, and the local signal SL is input to the mixer in a time division manner.

Here, as in the modification of the first configuration example shown in FIG. 10, after the local signal SL in the microwave band passes through the switch SW_3, the frequency is increased to a predetermined frequency in the millimeter wave band by the multiplier 32_n. Since the frequency is multiplied, the insertion loss due to the switch SW_3 is suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Further, the loss due to the transmission paths P_4 and T_n is also suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Therefore, the mixer 16_n can generate the beat signal SB_n using the local signal SL with sufficient strength.

FIG. 13 is a diagram for explaining a usage situation in which the radar device according to the present embodiment is used as a vehicle-mounted radar device. The radar device 2 is mounted in the front front grill of the vehicle 100, and transmits and receives a transmission signal ST and a reception signal SR_n through a bump provided on the bumper or the front grill of the vehicle 100. Then, an azimuth angle at which a target such as a preceding vehicle or an obstacle in front of the vehicle 100 (arrow F) is located, a relative speed with respect to the target, and a relative distance are detected. The target information such as the azimuth angle at which the target detected by the control unit 6 is located as described above, the relative distance to the target, and the relative speed is the vehicle control device (ECU: Electronic Control Unit) 200 of the vehicle 100. Is output.

Based on these detection results, the vehicle ECU 200 controls the actuator such as the throttle and brake of the vehicle 100 to adjust the speed of the vehicle so as to follow the preceding vehicle and perform follow-up running control. In addition, the vehicle ECU 200 outputs a notification to the driver or activates a safety device such as an airbag when the distance from the preceding vehicle or an obstacle approaches a certain distance or more and the probability of a collision increases. Then, the collision handling operation is performed.

Note that the radar device 2 may be installed not only on the front side of the vehicle 100 but also on the side surface and the rear side of the vehicle 100 as a radar device for monitoring the side and the rear.

In the above configuration example, the radar apparatus 2 including four reception antennas AR_1, AR_2, AR_3, and AR_4 is taken as an example, but the number of reception antennas is not limited to this.

Furthermore, the frequency band of the radar signal output from the VCO 8 is not limited to the above-described example, and may not necessarily be the microwave band. Correspondingly, the rate of multiplication by the multiplier 32_n is not limited to the above example, and it is not always necessary to multiply to the millimeter wave band. If the configuration is such that the frequency of the signal passing through the switch or transmission path is lowered and then multiplied, it prevents the transmission signal strength from being reduced due to the insertion loss of the switch or the transmission signal from being reduced due to the transmission loss. It is possible to achieve the operational effects.

As described above, according to the radar apparatus of the present embodiment, even if a transmission path for transmitting a transmission signal to a pair of transmission antennas separated by a predetermined interval or a switch for distributing the transmission signal to the transmission path is used, the detection performance can be improved. Decline can be prevented.

Claims

A transmitting antenna pair spaced by a predetermined interval for transmitting a radar signal of a predetermined frequency; and a receiving antenna group for receiving the radar signal reflected by a target as a received signal; alternately from the transmitting antenna pair In a radar apparatus for detecting an azimuth angle at which the target is located using a received signal obtained by transmitting a radar signal,
An oscillator that outputs a radar signal having a frequency lower than the predetermined frequency;
A switch for distributing a radar signal output from the oscillator to a first transmission path to the first transmission antenna and a second transmission path to the second transmission antenna;
A first multiplier connected to the first transmission path, converting the frequency of the radar signal to the predetermined frequency and then outputting the first transmission antenna, and connected to the second transmission path; And a second multiplier for converting the frequency of the radar signal to the predetermined frequency and outputting the converted signal to the second transmitting antenna.
In claim 1,
The switch has a first insertion loss when the frequency of the radar signal passing through the first frequency is a first frequency, and is larger than the first insertion loss when the frequency is a second frequency higher than the first frequency. 2. A radar apparatus characterized in that an insertion loss of 2 is given to the radar signal.
In claim 1 or 2,
A mixer that outputs a frequency difference signal between the radar signal of the predetermined frequency and the received signal;
A distributor for distributing a radar signal output from the oscillator to a third transmission path to the mixer;
A third multiplier connected to the third transmission path and converting the radar signal to the predetermined frequency and then outputting to the mixer;
A radar apparatus further detecting a relative distance or a relative speed with respect to the target using the frequency difference signal.
In any one of Claims 1 thru | or 3,
The radar apparatus according to claim 1, wherein the predetermined frequency is a millimeter wave band, and a frequency of a radar signal output from the oscillator is a microwave band lower than the millimeter wave band.