WO2003044943A1

WO2003044943A1 - Power amplifier arrangements with predistorter comprising vector modulators

Info

Publication number: WO2003044943A1
Application number: PCT/EP2002/012614
Authority: WO
Inventors: Julian Richard Trinder
Original assignee: Roke Manor Research Limited
Priority date: 2001-11-22
Filing date: 2002-11-12
Publication date: 2003-05-30
Also published as: GB2382480B; GB2382480A; GB0128010D0; AU2002358497A1

Abstract

The invention relates to arrangements intended to improve the linearity of power amplifiers, and especially to arrangements intended to reduce third-order intermodulation distortion in power amplifiers. It provides for linearisation of a power amplifier by second harmonic injection, without the need for either matched group-delay band pass filters or ultra-wideband signal paths, by utilizing vector modulators for two signal sets, one (the main signal) being confined to the fundamental region and the other, the secondary signal, being confined to the second harmonic region. The secondary signal is subjected to baseband processing and is scaled, relative to the primary signal, by a factor dependent upon the ratio of the power amplifier's third- and second-order polynomial coefficients. Preferably, the vector modulators are integrated, with their differential output ports connected together into a single differential load and high-speed digital dividers are also integrated to provide the local oscillator requirements of the vector modulators.

Description

POWER AMPLIFIER ARRANGEMENTS WITH PREDISTORTER COMPRISING VECTOR MODULATORS

This invention relates to power amplifier arrangements, and more particularly to arrangements intended to improve the linearity of power amplifiers, and it relates more especially, though not exclusively, to arrangements intended to reduce third-order intermodulation distortion in power amplifiers.

It has become known in recent years that third-order intermodulation distortion in power amplifiers can be reduced by injecting second- harmonic signals into the amplifier under carefully controlled phase and amplitude conditions. Existing techniques for effecting such injection, however, exhibit certain drawbacks since, in order to avoid applying wideband signals (particularly low frequency, or even d.c. signals) to the power amplifier, which is normally not practicable, they typically employ band pass filters in the injection path. In such circumstances, the direct signal path to the power amplifier also needs to include a band pass filter, in order to match delays imposed on the second-harmonic signals by the band pass filter in the injection path, and difficulties arise in ensuring that the two band pass filters are accurately matched in gain and phase.

The invention thus provides, f om one aspect, a power amplifier arrangement wherein radio frequency (RF) signals for amplification and signals at the second harmonic thereof are each derived from respective vector modulation stages and the respective vector modulated signals are combined for application to the power amplifier, thereby injecting the second-harmonic signals into the power amplifier. Preferably, the respective vector modulation stages are each supplied with signals derived from a common baseband modulator, and it is further preferred that signals derived from the common baseband modulator and intended for application to the vector modulation stage providing said second-harmonic signals are operated upon by a baseband processing function before such application.

In a preferred embodiment of the invention, scaling means are provided to adjust the relative amplitudes of said radio frequency (RF) signals for amplification and said signals at the second harmonic thereof. It is also preferred that the scaling means comprises an amplitude adjustment component disposed to influence the amplitude of said second-harmonic signals. In such circumstances, the scaling means may be disposed to operate either directly upon the said second harmonic signals themselves, or upon the signals derived from the common baseband modulator and intended for application to the vector modulation stage, preferably after they have been operated upon by the aforementioned baseband processing function.

Preferably a further scaling means are provided to permit common adjustment of the overall amplitude of said signals; said further scaling means being disposed so as to operate directly upon the signals derived from said common baseband modulator.

In a further preferred embodiment, each vector modulation stage comprises a respective pair of Gilbert-cell mixers having their differential collector ports connected in parallel. In order that the invention may be clearly understood and readily carried into effect, certain embodiments thereof will now be described, by way of example only, with reference to the accompanying drawings, of which:-

Figures 1 and 2 show, in schematic diagrammatic form, power amplifier arrangements incorporating known techniques for second-harmonic injection,

Figures 3 and 4 show, also in schematic diagrammatic form, power amplifier arrangements in accordance with exemplary embodiments of the invention, and

Figure 5 shows, again in schematic diagrammatic form, an implementation of a power amplifier arrangement in accordance with an example of the invention.

Prior to detailed description of the various power amplifier arrangements shown in the drawings, some basic relationships will be established with reference to a simple bipolar transistor model of a power amplifier. The invention is not in any way limited in its application to power amplifiers of such construction, but the derivative relationships serve to illustrate the dependence of the model on both second-order and third-order non- linearities.

The emitter current in a bipolar transistor can be simply modelled as a function of the base-emitter voltage: IE = Is. exp{VBE/Vτ} , where VT = kT/q.

With base biasing:

VBE = VBIAS + v(t), hence:

IE = Is.exp{VBiAs/Υτ}.exp{v(t)/Vτ}.

The voltage developed across an output load resistor is:

V(t) = ICRL ~ Is.RL.exp{VBiAs/Yτ}.exp{v(t)/Υτ} = Vo.exp{v(t)/Vτ}

Where: Vo = Is.RL.exp{VBiAs/Vτ} .

Using a series expansion:

exp{x} = 1 + x + x²/2! + x³/3! + x⁴/4! + x⁵/5! +

V(t) = Vo.exp{v(t)/Vτ} = Vo[l + v(t)/Vτ +

+ v(t)³/6.Vτ³ + v(t)724.Vτ⁴ + v(t)⁵/120.Vτ⁵ + ] ,

Which is of the general form:

V(t) = go + gl.V(t) + g2.V(t)² + g3.v(t)³ + g4.v(t)⁴ + g5.V(t)⁵ + ....

Where in this case: go = Vo, gi = Vo/v(t), g2 = Vo/2v(t)², g₃ = Vo/6v(t)³, g = Vo/24v(t)⁴, g₅ = Vo/120v(t)⁵, etc.

Referring now to Figure 1, the basic principle of second harmonic injection is illustrated by means of a known technique.

An RF modulator 1 generates source signals which, in an uncorrected arrangement, would be applied to a power amplifier shown in phantom at 2.

In order to improve the linearity of the power amplifier 2, however, a second channel of circuitry, shown generally at 3, is provided. In this channel 3, the second harmomc of the RF signal is generated in a squaring circuit 4, typically a mixer and applied, by way of a band pass filter 5 and a scaling component 6, to a combining circuit 7 which also receives the signals from the RF modulator 1. The output of the combining circuit 7 is applied to the input of the power amplifier 2, thereby injecting into the power amplifier a second-harmonic component of the RF signal derived by way of the channel 3.

The second harmonic scaling factor b2 imposed by the component 6 is typically -1.5(g3/g2), and it is usual also to impose a common scaling factor, bl, directly upon the signals output from the RF modulator 1; this being achieved by means of a conventional component 8 of any convenient kind. In this known arrangement, it is necessary to introduce a band pass filter 9 into the path of the RF signals fed from the modulator 1 to the mixer 7, in order to match the group-delay response of the filter 5 in channel 3 and, as mentioned previously, this leads to difficulties associated with the need for accurate gain and phase matching of the two band pass filters 5 and 9.

If the band pass filters 5 and 9 are omitted, as shown in Figure 2, where components similar to those already described with reference to Figure 1 are given the same reference numbers, the range of signals applied to the power amplifier 2 now extends down to d.c, which is normally not practicable. In the arrangement of Figure 2, incidentally, the second harmonic scaling factor b2 typically reduces to -0.5(g3/g2).

Referring now to Figure 3, there is shown an embodiment of the present invention.

This embodiment of the invention overcomes the difficulties associated with the known arrangements described with reference to Figures 1 and 2 by employing a simple baseband processor and a pair of vector I/Q modulators.

A baseband modulator 10 applies I and Q components Xr and xi respectively to a vector modulator 11 driven by an operating carrier sin(α)o.t). The I and Q signals are, as before, subjected at 12 to a scaling factor bl, and the output of the vector modulator 11 is applied to a combining circuit 13. The scaled I and Q signals Xr and x. are also fed to a baseband processing circuit 14, which operates on them to generate signals y_r= Xr - XΪ² and ys = 2.Xr.Xi for application to a second vector modulator 15 which is driven by an operating carrier eos(2oc>o.t). This generates the required second harmonic component which is scaled in amplitude by the factor b2 at 16 and applied as the second input to the combining circuit 13.

The output of the combining circuit 13 is applied as the input to power amplifier 17, and it will thus be appreciated that, in this embodiment of the invention, the arrangement is such that the wanted modulation is applied to the carrier signal sin(ω₀.t) using a conventional I/Q vector modulator (11). The output of the vector modulator 11 is thus sin(ω₀.t + φ_m), where φ_m represents the wanted modulation.

The second vector modulator (15) is used to generate the second-harmonic zone, utilising a quadrature carrier cos(2ω₀.t). The baseband processor 14 generates, as indicated above, the signals y_r and y., representing the square of the baseband function and the resultant second harmonic zone signal is

For this arrangement, it is found that the second harmonic scaling factor (b2) = +0.75(g3/g2), assuming the polynomial components g to be real.

Referring now to Figure 4, in which components common to the arrangement of Figure 3 carry the same reference numbers, this shows an arrangement addressing the more general case in which the polynomial coefficients g can have complex values, representing both AM- AM and AM-PM effects. The second harmonic zone scaling factor b2 then becomes complex and in such circumstances it is expedient to perform the complex scaling at baseband, i.e. before the vector modulator 15. This is indicated by the application of the signals y_r and y. as described above to the complex scaling component 160 which applies respective scaling factors b2r and b2. (where b2 = +0.75(g3/g_), as before) and outputs signals Zr = b2r.y_r - b2i.yι and z; = b2r.yι + b2i.y_r for application to the power amplifier 17.

The invention provides efficient power amplifier arrangements in which significant hardware simplification is possible.

In particular, a vector modulator can readily be implemented using a pair of Gilbert-cell mixers having their differential collector ports connected in parallel. Local oscillator phasing is normally accomplished utilising highspeed digital dividers or poly-phase networks.

Two vector modulators (four mixers) can thus readily be integrated, with their differential output ports connected together into a single differential load, thus eliminating the need for a separate combining circuit such as that shown at 13 in Figures 3 and 4. Moreover, the local oscillator requirements of the two vector modulators 11 and 15 have a strict harmonic relationship which lends itself to an integrated solution using high-speed digital dividers.

Thus, as shown in Figure 5, a streamlined implementation of the invention provides an arrangement in which a baseband modulator 21 feeds a baseband processor 22 which provides four analogue outputs (xr, Xi, z_r and zi) that feed an integrated dual vector modulator 23 with built-in local oscillator processing, and a power amplifier 24.

The invention provides power amplifier arrangements in which linearisation of the power amplifier can be achieved by second harmonic injection, without the need for either matched group-delay band pass filters or ultra-wideband signal paths. The power amplifier is presented with two signals, one (the main signal) being confined to the fundamental region and the other, the secondary signal, being confined to the second harmonic region. The secondary signal is scaled, relative to the primary signal, by a factor dependent upon the ratio of the power amplifier's third- and second- order polynomial coefficients.

The use of an auxiliary baseband processor and a second vector modulator, which can beneficially, as explained above, be integrated with the first vector modulator, is important to the invention, as it permits creation of the second harmonic zone signal without requiring matched group-delay band pass filters.

Claims

CLAIMS:-

1. A power amplifier arrangement wherein radio frequency (RF) signals for amplification and signals at the second harmonic thereof are each derived from respective vector modulation stages and the respective vector modulated signals are combined for application to the power amplifier, thereby injecting the second-harmonic signals into the power amplifier.

2. An arrangement according to claim 1 wherein the respective vector modulation stages are each supplied with signals derived from a common baseband modulator.

3. An arrangement according to claim 2 wherein signals derived from the common baseband modulator and intended for application to the vector modulation stage providing said second-harmonic signals are operated upon by a baseband processing function before such application.

4. An arrangement according to any preceding claim wherein scaling means are provided to adjust the relative amplitudes of said radio frequency signals for amplification and said signals at the second harmonic thereof.

5. An arrangement according to claim 4 wherein the scaling means comprises an amplitude adjustment component disposed to influence the amplitude of said second-harmonic signals.

6. An arrangement according to claim 5 wherein the scaling means are disposed to operate directly upon the said second harmonic signals themselves.

7. An arrangement according to claim 5 wherein the scaling means are disposed to operate upon the signals derived from the common baseband modulator and intended for application to the vector modulation stage.

8. An arrangement according to claim 7 wherein the scaling means are disposed to operate upon the signals derived from the common baseband modulator, and intended for application to the vector modulation stage, after they have been operated upon by the aforementioned baseband processing function.

9. An arrangement according to any of claims 4 to 8 wherein a further scaling means are provided to permit common adjustment of the overall amplitude of said signals; said further scaling means being disposed so as to operate directly upon the signals derived from said common baseband modulator.

10. An arrangement according to any preceding claim wherein each vector modulation stage comprises a respective pair of Gilbert-cell mixers having their differential collector ports connected in parallel.

11. An arrangement according to claim 10 wherein local oscillator phasing for said mixers is accomplished utilising high-speed digital dividers or poly-phase networks.

12. An arrangement according to claim 10 or claim 11 wherein the vector modulators are integrated, with their differential output ports connected together into a single differential load and wherein high-speed digital dividers are also integrated therewith to provide the local oscillator requirements of the vector modulators.