WO2002039567A2

WO2002039567A2 - Magnetic amplifier ac/dc converter with primary side regulation

Info

Publication number: WO2002039567A2
Application number: PCT/US2001/027789
Authority: WO
Inventors: Seth R. Sanders; J. Mark Noworolski
Original assignee: Munetix, Inc.
Priority date: 2000-11-08
Filing date: 2001-10-31
Publication date: 2002-05-16
Also published as: TW561677B; AU2002212965A1; WO2002039567A3

Abstract

In accordance with the present invention there is provided a single stage ac/dc converter operative to convert a universal ac input voltage to a prescribed dc voltage. The converter comprises a primary side in electrical communication with the input voltage. The primary side has a bridge rectifier and at least one switching device operative to selectively control an input current waveform. In this regard, the voltage is regulated on a dc storage bus of the primary side. In electrical communication with the primary side is a secondary side which has a variable reactor that is operative to selectively control a conduction period thereof. In this respect, the converter is operative to selectively generate the prescribed output voltage by independently controlling the input waveform on the primary side and controlling the conduction period on the secondary side.

Description

MAGNETIC AMPLIFIER AC/DC CONVERTER WITH PRIMARY SIDE REGULATION

BACKGROUND OF THE INVENTION

The present invention generally relates to ac/dc power converters, and more particularly to a single stage ac/dc power converter that provides independent control of the input current waveform and secondary side regulation through the use of an active switching device.

In a conventional ac/dc converter, power is usually delivered to a dc load or multiple dc loads at a constant rate. However, due to the fundamental nature of a single-phase ac source, power drawn from such a source must have a pulsating nature with an average value equal to the output power plus losses incurred by the converter. Accordingly, the ac/dc converter must provide a means for storing and retrieving energy during each half-cycle of the ac line.

In conventional capacitor loaded rectifier-bridge front-end topologies, energy is typically stored in a large electrolytic input capacitor. The average voltage of the input capacitor is dependent upon the line voltage and the load level. The average input capacitor voltage can vary over a wide range when used in universal input applications where the line voltage can vary from 90 Vrms to 265 , Vrms. In this respect, the input capacitor must be rated for a peak voltage of about 400V and be sized to provide adequate voltage support for an input of 90 Vrms under full load conditions. For capacitor loaded rectifier-bridge front-end topologies, the input capacitor is frequently the largest component with a length of about 1.5 inches and a diameter of about 0.75 inches for 60 watt applications. In order to mitigate the need for excessively large input capacitors, two- stage power factor correction (PFC) ac/dc topologies have been created. In these two-stage power factor correction ac/dc converters, the input stage typically consists of a bridge-rectifier followed by a boost or other conventional dc/dc converter topology. The dc/dc converter is controlled to draw a sinusoidal or other desired ac input current waveform, while maintaining the amplitude of the input current at a level which matches the necessary load power and circuit losses. The output of the input stage is fed into a bulk energy storage capacitor which has an average voltage ideally regulated to a prescribed dc level. The bulk energy storage capacitor then provides for the required energy storage in the ac/dc conversion process. By maintaining the average voltage at a prescribed level, the voltage rating and value of the capacitor can be precisely specified in order to achieve a prescribed allowable voltage ripple amplitude that is twice the line frequency. Because the capacitor operates at a prescribed average voltage under all line voltages and load conditions, the size of the capacitor can be significantly smaller than the input capacitor of the aforementioned capacitor-loaded bridge rectifier front end topologies. Furthermore, the size of the bulk energy storage capacitor of the input stage for two-stage PFC ac/dc topologies may be specified to achieve a desired hold-up time, as well as to achieve a tolerable ripple amplitude. In addition to requiring smaller energy storage components, the aforementioned two-stage PFC converters control the harmonic content of the line current. Because the harmonic content is regulated by a number of standards, power factor correction is necessary in numerous applications, especially above a 50-75 watt power range. The two-stage PFC converter also provides flexibility in controlling the input current waveform, while drawing the required average power. The second stage of the PFC converter is operative to provide complete independent regulation of the converter output voltage, or voltages in multiple output applications. However, the main disadvantage of the aforementioned PFC ac/dc converters is the complexity, cost, and inefficiency of the two-stage implementation. Specifically, the two-stage power converter comprises two single stage converters connected in cascade. In this respect, the two-stage power converter contains more components and may suffer from lower efficiency than single stage converters due to the fact that the output power must be processed by each of the two stages.

In order to mitigate the difficulty of two-stage power converters, a number of single-stage ac/dc converters provide input current shaping. However, the main drawback of conventional single-stage converters with input current shaping is the fact that the voltage on the respective main energy storage capacitor varies significantly with line voltage and load condition. Even though the single stage converter may provide adequate input current harmonic regulation, the single-stage converter still suffers from the need to have large energy storage capacitance. A further disadvantage of the single-stage converters is that the single stage effectively combines an input boost converter with an output regulating forward or flyback circuit. In achieving dual functions with a single stage, an extra diode must be placed in the power path thereby degrading efficiency and increasing component count and size. Furthermore, conventional single-stage power converters do not provide high efficiency (i.e., 90%) at full load over universal line voltage ranges.

The present invention addresses the above-mentioned deficiencies in single and two stage power converter design by providing an ac/dc power converter which provides high efficiency with reduced size and part count. More particularly, the present invention provides a single stage ac/dc power converter which provides independent control of the input current waveform and the output voltage, without the need for a second stage.

BRIEF SUMMARY OF THE INVENTION In accordance with the present invention there is provided a single stage ac/dc converter operative to convert a universal ac input voltage to a prescribed dc voltage. The converter comprises a primary side in electrical communication with the ac input voltage. The primary side has a bridge rectifier and at least one switching device operative to selectively generate an input waveform. In electrical communication with the primary side is a secondary side which has a variable reactor that is operative to selectively control a conduction period thereof. In this respect, the secondary side is operative to selectively generate the prescribed output voltage. It will be recognized that the primary side controller may be operative to shape the input current waveform to satisfy international input current harmonic standards thereby effectively implementing power factor correction.

In order to generate the output voltage, the converter includes a controller in electrical communication with the variable reactor and the output voltage. The controller is operative to control the conduction period of the secondary side in response to the output voltage. As such, the secondary side includes a diode connected in series with the reactor. The reactor is operative to control the conduction period of the diode and hence regulate the output voltage. In the preferred embodiment of the present invention, the variable reactor is a magnetic amplifier. A capacitor may be connected in series with the diode such that the voltage across the capacitor is the output voltage.

Similarly, in order to generate the input waveform, the converter includes a bulk capacitor for energy storage. The switching device is operative to control the amount of energy stored within the bulk capacitor and may comprise two switching devices gated complementarity. In the preferred embodiment of the present invention, the converter has a transformer in electrical commumcation with the primary side and the secondary side. The transformer has a primary winding in electrical communication with the primary side and a secondary winding in electrical communication with the secondary side.

In accordance with the present invention, the converter may be operative to generate multiple output voltages. In this respect, the converter will include a second secondary side in electrical communication with the primary side. The second secondary side will be operative to generate a second output voltage which is independent of the first output voltage. In this respect, the second secondary side will include a second variable reactor that is operative to selectively control a conduction period thereof. It will be recognized that the second secondary side will be in electrical communication with second secondary windings of the transformer. Alternatively, the transformer may include a tap that the second secondary side is connected to in order to generate multiple output voltages. Additionally, it will be recognized that the converter may include multiple secondary sides in electrical communication with the primary side in order to generate multiple outputs.

In accordance with the present invention, there is provided a method of converting an ac voltage to an output dc voltage with a converter having a primary side and a secondary side. The method comprises rectifying with the primary side, the ac voltage in order to generate an intermediary rectified line voltage. Next, the intermediary rectified line voltage is regulated on a dc energy storage bus of the primary side. Finally, an output voltage (derived from the regulated dc energy storage bus) is further regulated on the secondary side of the converter with a variable reactor.

In the preferred embodiment of the present invention, the ac voltage is rectified with a bridge rectifier of the primary side and the dc storage bus is regulated by two switching devices and a storage capacitor. In this respect, the storage capacitor, as well as the switching devices, form the dc bus. As previously mentioned, the variable reactor is a magnetic amplifier that is controllable with a respective controller. It will be recognized by those of ordinary skill in the art that multiple outputs may be generated by using multiple secondary sides that are operative to regulate a respective output voltage from the intermediary dc voltage. BRLEF DESCRIPTION OF THE DRAWINGS

These, as well as other features of the present invention will become more apparent upon reference to the drawings wherein:

Figure 1 is a circuit diagram of a power converter constructed in accordance with a first embodiment of the present invention;

Figure 2 is a waveform diagram for the power converter shown in Figure 1; and

Figures 3 - 15 are circuit diagrams for alternative embodiments of the power converter constructed in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Referring now to the drawings wherein the showings are for purposes of illustrating a preferred embodiment of the present invention only, and not for purposes of limiting the same, Figure 1 shows a power converter 10 constructed in accordance with a first embodiment of the present invention. The power converter 10 is a half-bridge flyback, secondary-side magnetic amplifier regulator having a primary side 12 and a secondary side 14. The power converter 10 includes a bridge rectifier BR1 in electrical communication with an input voltage Vac. The bridge rectifier BR1 comprises a full-bridge rectifier and rectifies Vac as is currently known in the art to generate a rectified line voltage. Connected to the output of the bridge rectifier BR1 is an input capacitor Cin and a bulk capacitor

Cbulk. The bulk capacitor Cbulk provides the main energy storage function of the circuit, as will be further explained below, and could be located on either the primary side 12 or the secondary side 14.

Connected in parallel with input capacitor Cin and bulk capacitor Cbulk are MOSFETS Ml and M2 controlled by respective gate voltages Vgsl and Vgs2.

The gate voltages Vgsl and Vgs2 are generated by a conventional controller, as is currently known in the art and shown in Figure 2. As seen in Figure 1, the converter 10 further includes a transformer TI with one input connected between the bulk capacitor Cbulk and a second input connected between the MOSFETS Ml and M2. The transformer TI, as well as capacitor Cbulk and switching devices Ml and M2, form a dc energy storage bus that is operative to produce an intermediary regulated voltage. The transformer TI has a primary coil Vp which couples with a secondary coil Vs. The secondary coil Vs of transformer TI is connected in series to a reactor XI (magnetic amplifier) and diode Dl, as seen in Figure 1. In the preferred embodiment of the present invention, the reactor XI may be a magnetic amplifier which is controlled by a controller K. In this respect, the controller K controls the conductance of the reactor XI . Connected to diode Dl and secondary coil Vs of transformer TI is output capacitor Cout. Output voltage Vout is the voltage across Cout, as seen in Figure 1.

During operation of the power converter 10, the MOSFETS Ml and M2 are gated complementarity, as in conventional asymmetric half-bridge topologies, in order to regulate the average voltage on bulk capacitor Cbulk. The voltage across Cin roughly tracks the rectified line voltage appearing at the output of the bridge rectifier BRl . During each cycle of pulse- width modulation (PWM) action, the conduction time for Ml is controlled to assign the average input current drawn from the bridge rectifier BRl. Accordingly, this action determines the input current waveform for the converter. In order to meet international standards on current harmonics, the input current waveform can be assigned to track the input voltage waveform (i.e., sinusoidal), or other desired wave shapes. The regulation of the input current is achieved with a fast inner PWM control loop. An outer voltage loop is used to generate an average amplitude of the input current waveform, so as to regulate the average voltage on the bulk capacitor Cbulk. As in other power factor correction schemes, the outer loop bandwidth is designed to be slower than twice the line frequency, (e.g. 120 Hz in North America) so as not to distort the input current waveform. When the circuit is loaded with a dc load at Vout, load power is drawn from bulk capacitor Cbulk at a constant rate. However, power is supplied to bulk capacitor Cbulk in a pulsating manner with fundamental frequency at twice the ac line frequency. Thus, die fundamental ac voltage component on Cbulk is at twice the ac line frequency. Capacitor Cbulk thereby operates to provide the main energy storage function in the circuit and is specified to meet the desired ripple amplitude and provide a desired hold-up time. As such, capacitor Cbulk is chosen with a design value for the average voltage and the primary referred output voltage, as determined by the turns ratio of the transformer TI. A practical design voltage on the bulk capacitor Cbulk (for a universal input voltage of between about 90 - 265 Vrms) is in the range of 180V. Accordingly, MOSFETS Ml and M2 can have a rating of about 600V and Cbulk can have a rating of in the range of 200V - 250V.

The duty cycle range for conduction of Ml may be from approximately 0.3 to 0.7. As is evident to those of ordinary skill in the art, flyback designs which center the duty cycle range about 0.5 and which do not involve extremely large or small duty cycles provide for relatively low MOSFET device stresses in terms of the maximum voltage-ampere product ratings.

A second important feature of the present invention is the design of the secondary side which provides independent, fast and accurate output voltage regulation. With the PWM pattern aforementioned, and as seen in Figure 2, for Vgsl and Vgs2, there is provided a fixed, minimum conduction time for MOSFET M2 of approximately 0.3T, where T is the PWM period. The conduction time for the MOSFET M2 is achieved by assigning a maximum duty cycle of Ml of approximately 0.7T. During conduction of M2, the voltage across secondary winding Vs is given by Vbulk/N where Vbulk is the voltage on the bulk capacitor Cbulk and N is the transformer turns ratio for TI. The voltage across secondary windings Vs must be assigned, by design, to exceed the desired output voltage by an amount to be referred to as the secondary-side voltage headroom. Output voltage regulation is therefore achieved by controlling the delay of conduction for diode Dl through appropriate blocking with reactor XI, as shown in Figure 2. Specifically, the current pulse through diode Dl is approximately triangular, with a rising edge duration controlled by reactor XI. The controller K for reactor XI will control the conduction of reactor XI accordingly. The slope of the rise of the current pulse through diode Dl is determined by the ratio of the secondary-side voltage headroom and the total secondary-side referred leakage inductance, including the inductance of the saturated reactor XI. In this respect, the peak secondary current is approximated by:

I_pk=ΔV *T_cond / L_s [1] where ΔV is the secondary-side voltage headroom, L_s is the total secondary side referred leakage inductance (including that of the saturated reactor XI) and T_cond is the conduction interval. The total charge delivered to the output capacitor Cout during each period is thus approximately:

Q=ΔV * T_cond ²/(2 * L_s) [2]

The average output current can be computed as:

I_out = ΔV * T_cond ²/(2 * L_s * T) [3] where T is the PWM period.

Secondary side regulation is achieved in a straightforward manner using the controller K to cause reactor XI to block and achieve the required conduction time T_cond. It will be recognized by those of ordinary skill in the art that conventional controllers for magnetic amplifiers are operative to control the conduction period of reactor XI. Because output voltage regulation is achieved with reactor XI, there is no need for a feedback signal to be transmitted across the main transformer isolation barrier. As such, optocouplers or other such devices are not needed. The power converter 10, shown in Figure 1, is additionally capable of achieving zero-voltage switching wherein MOSFETS Ml and M2 are each turned on with essentially zero volts across their respective drain-source elements thereby resulting in improved efficiency. This situation occurs where Ml is turned off and the magnetizing inductance of the transformer TI is conducting a positive current when referred to the primary windings Vp. With proper design, the transformer magnetizing inductance rings with the drain-source capacitance of Ml and M2 to fully charge Vdsl to Vin + Vbulk while fully discharging Vds2 to zero. At this point, after allowing for an appropriate dead time for this transition, MOSFET M2 can be gated on with Vds2 at zero volts. On the other hand, MOSFET M2 is gated off when the magnetizing current in primary Vp of transformer TI is at a minimum. The minimum may be negative under light or no load conditions, or may remain positive under full or heavy load conditions. In the case of full or heavy load conditions, a large component of secondary side load current is reflected to the primary thereby resulting in a primary side current composed of a moderate magnetizing current and a large negative load current component. Under this heavy load condition, the reflected load current drives the zero-voltage switching transition at the turn off of M2. The ringing transition is dominated by the total primary-side referred leakage inductance ringing with the drain-source capacitances of Ml and M2 in this case. Under light load conditions, the zero- voltage switching transition is driven by the transformer magnetizing current, analogously with that at the turn off of Ml .

Referring now to Figure 3, a second embodiment of a power converter 30 is shown. The power converter 30 has a primary side 32 similar to primary side 12 of the first embodiment 10. At a secondary side 32, the diode Dl has been replaced MOSFET device M3. In this embodiment, the MOSFET device M3 is gated synchronously with the conduction period of the replaced diode Dl in order to reduce conduction losses. Accordingly, MOSFET M3 forms a synchronous rectifier. Gating signals for the MOSFET M3 can be generated from the gate drive from MOSFET M2, the secondary windings Vs of transformer TI, or from any other appropriate source of control.

Referring now to Figure 4, a third embodiment of a power converter 40 is shown. The third embodiment illustrates how multiple independent outputs can be achieved. The primary side 42 of power converter 40 is similar to the primary side 12 of the first embodiment of the power converter 10. However, separate secondary windings Vsl and Vs2 are provided on transformer TI . Each of the secondary windings Vsl and Vs2 are in electrical communication with a respective secondary side 44 and 46. Each of the secondary sides 44 and 46 include respective reactors XI and X2 such that independent control of the output voltage across respective capacitors Coutl and Cout2 can be achieved. As previously described above, by utilizing a respective reactor for each of the secondary sides 44 and 46, it is possible to generate independent output voltages by controlling the conduction period of respective reactors XI or X2. It will be recognized by those of ordinary skill in the art, that multiple secondary sides 44 and 46 may be in electrical communication with the transformer TI such that multiple output voltages may be generated.

A fourth embodiment of a power converter 50 is shown in Figure 5 wherein multiple outputs Coutl and Cout2 are realized by utilizing a tap 58 from the secondary winding of the transformer TI. Specifically, a primary side 52 is similar to the primary side 12 of the first embodiment 10. A secondary side 54 consists of magnetic amplifier XI, diode Dl and capacitor Cl forming an outer loop 55. In order to generate dual voltages, the tap 58 provides voltage to a second magnetic amplifier X2, second diode D2, and a second capacitor Cout2 which form an inner loop 57. A second voltage is developed across Cout2 which is independent of a first voltage developed across Coutl . As previously mentioned, independent control of each reactor XI and X2 permits the generation of two different output voltages across Coutl and Cout2. Additionally, by including multiple taps on the transformer TI, it is possible to generate multiple output voltages with multiple secondary sides. As such, it will be recognized that it is possible to generate multiple independent voltages with the present invention. Referring to Figure 6, a fifth embodiment of a power converter 60 is illustrated. The power converter 60 has a primary side 62 similar to the primary side 12 of the power converter 10. However in the fifth embodiment, the reactor XI has been replaced by a solid state switch such as a bipolar transistor or MOSFET Mx. In this respect, the MOSFET Mx can be switched to provide the necessary voltage regulation across Cout. On the other hand, instead of replacing the magnetic amplifier XI with a solid state switching device, a continuously controlled linear inductor may be used to modulate the output power. The linear inductor controls the conductance of the diode and therefore provides the necessary regulation of the output voltage. In such a case, a controller will be in electrical communication with the linear inductor and will be operative to control the inductance therethrough, as previously explained.

Referring to Figure 7, a sixth embodiment of a power converter 70 is shown. The power converter 70 comprises a primary side 72 and a secondary side 74. The primary side 72 has an input capacitor Cin connected across the output of bridge rectifier BRl . Connected across the input capacitor Cin are switching devices Ml and M2 which are controlled through respective complementary gate voltages Vgsl and Vgs2. As seen in Figure 7, the primary coil Vp of transformer TI is connected between the switching devices Ml and M2 and a bulk capacitor Cbulk. The other side of bulk capacitor Cbulk is attached to ground. In this respect, the primary side 72 forms an input side buck converter. The secondary side 74 is similar to the secondary side 14 of the power converter 10. In this respect the secondary side 74 of power converter 70 is operative to adjust the output voltage Vout with variable reactor XI. In this respect, the switching devices Ml and M2 are used to control the input current waveform and to regulate the voltage on Cbulk while reactor XI provides independent control to regulate output voltage Vout.

A boost converter variation 80 is shown in Figure 8. The converter 80 comprises a primary side 82 and a secondary side 84. The primary side 82 has an input capacitor Cin connected across the output of the bridge rectifier BRl . Connected to one side of the input capacitor Cin is one end of the primary coil Vp of transformer TI . The other end of the primary coil Vp is connected between switching devices Ml and M2. As seen in Figure 8, the bulk capacitor Cbulk is connected between the switching devices Ml and M2. Accordingly a boost converter input is formed.

The secondary side 84 of the boost converter variation 80 has one terminal of the secondary coil Vs of transformer TI connected to one terminal of variable reactor XI and one terminal of variable reactor X2. The other terminals of each variable reactor XI and X2 are connected to respective diodes Dl and D2. Connected to the other sides of the diodes Dl and D2 are series connected capacitors Cl and C2. The secondary side Vs of transformer TI is connected between the capacitors Cl and C2, as seen in Figure 8. Connected in parallel across capacitors Cl and C2 is output capacitor Cout which filters the output voltage Vout. In this respect, the reactors XI and X2 are both operative to shape the output waveform of Vout and capacitor Cout is operative to filter the same. This circuit can also be implemented with a single magnetic amplifier (reactor) by removing either XI or X2.

Referring to Figure 9, a boost flyback converter variation 90 is shown. The converter 90 has a primary side 92 with a boost inductor LI connected between the bridge rectifier BRl and the switching devices Ml and M2. Capacitors Cbulk and Cl are connected across the switching devices Ml and M2. Disposed between the switching capacitors Ml and M2, and capacitors Cbulk and Cl , is the primary coil Vp of transformer TI . In this respect, the topology of the primary side 92 of the converter 90 is a boost flyback. A secondary side 94 of the converter 90 is configured similarly to the secondary side 14 of the converter 10 shown in Figure 1. In this respect, the secondary side 94 is operative to produce an output voltage Vout across output capacitor Cout with reactor XI, diode Dl and secondary windings Vs or transformer TI, as previously explained. Accordingly, switching devices Ml and M2 provide regulation of the primary side 92, while reactor XI provides regulation of the secondary side 94.

Figure 10 shows a flyback-forward converter variation 100 utilizing the present invention. The flyback-forward converter variation 100 has a primary side 102 similar to the primary side 12 of the converter 10. In this respect, switching devices Ml and M2 are operative to shape and generate an input waveform on the primary coil Vp of transformer TI. The converter 100 has a secondary side 104 with one end of the secondary coil Vs of transformer TI connected to the reactor XI which in turn is connected to diode Dl. The secondary side 104 further includes diode D2 connected to diode Dl and the other end of secondary coil Vs of transformer TI. Connected across diode D2 is an LC network having inductor Lout and capacitor Cout with the output voltage Vout being across capacitor Cout. In this regard, the primary side 102 and secondary side 104 form a flyback forward converter variation 100.

Similarly, a forward converter variation 200 is shown in Figure 11 which has an identical primary side 202 to the converter 100 shown in Figure 10. The difference between the converter 100 and the converter 200 is that the polarity of the secondary coil Vs of transformer TI is reversed, as seen in Figure 11. However, the components of the secondary side 204 are identical to the secondary side 104 of converter 100. In this respect, the converter 200 shown in Figure 11 is a forward converter variation. Referring to Figure 12, a boost forward converter variation 300 is shown. The converter 300 has a primary side 302 and a secondary side 304. The primary side 302 is similar to the primary side 82 of the boost converter 80 shown in Figure 8. The secondary side 304 is similar to the flyback forward converter 100 shown in Figure 11. Accordingly, the boost primary side 302 is coupled with a forward secondary side 304 to form the boost forward converter variation shown in Figure 12.

Referring to Figure 13, a boost flyback converter variation 400 is shown utilizing the present invention. The boost flyback converter variation 400 has a primary side 402 which is similar to the boost primary side 302 of the converter 300. A secondary side 404 of the converter 400 is similar to the secondary side 304 of the converter 300, except that the polarity of the secondary windings Vs of transformer TI have been reversed. In this respect, the converter 400 can be considered a boost flyback converter variation. Referring to Figure 14, a boost forward converter 500 off the bulk capacitor Cbulk is shown. A primary side 502 of the converter 500 has an inductor LI is connected to the output of the bridge rectifier BRl and between diodes D2 and D3. The primary winding Vp of transformer TI is connected across the diodes D2 and D3. Additionally, switching device Ml is connected to one terminal of diode D2 and to ground. Connected to one terminal of primary winding Vp and to ground is bulk capacitor Cbulk. The converter 500 has a secondary side 504 which is similar to the secondary side 404 of converter 400. The output voltage Vout is taken across capacitor Cout as seen in Figure 14. Furthermore, a reset circuit (not shown) is needed to reset the transformer during each PWM period.

Referring to Figure 15, a boost flyback converter 600 off the bulk capacitor Cbulk is shown. The boost flyback converter 600 is similar to the boost forward converter 500 except that the polarity of the secondary windings Vs are reversed. In this respect, a primary side 602 is identical to the primary side 502 of the converter 500. However, a secondary side 604 of the converter 600 is different because the polarity of the windings Vs are reversed, as seen in Figure 15.

Additional modifications and improvements of the present invention may also be apparent to those of ordinary skill in the art. Thus, the particular combination of parts described and illustrated herein is intended to represent only certain embodiments of the present invention, and is not intended to serve as limitations of alternative devices within the spirit and scope of the invention.

Claims

1. A single stage ac/dc power converter, comprising: a primary side in electrical communication with an input voltage, the primary side operative to selectively generate an input waveform; a secondary side in electrical communication with the primary side, the secondary side having a variable reactor operative to selectively control a conduction period of the secondary side and thereby generate a prescribed output voltage.

2. The converter of Claim 1 further comprising a controller in electrical communication with the reactor, the controller being operative to control the conduction period in response to the output voltage.

3. The converter of Claim 2 wherein the primary side further comprises a bulk capacitor for energy storage and the secondary side comprises an output capacitor for filtering the output voltage.

4. The converter of Claim 1 further comprising a transformer in electrical communication with the primary side and the secondary side.

5. The converter of Claim 4 wherein the primary side further comprises a bridge rectifier operative to generate a rectified voltage.

6. The converter of Claim 4 wherein the transformer comprises a primary winding in electrical communication with the primary side and a secondary winding in electrical commumcation with the secondary side.

7. The converter of Claim 6 wherein the primary side further comprises at least one switching device to selectively generate the input waveform.

8. The converter of Claim 7 wherein the at least one switching device comprises a pair of switching devices gated complementarity.

9. The converter of Claim 1 further comprising a diode connected in series with the reactor, the reactor controlling the conduction period of the diode to generate the output voltage.

10. The converter of Claim 9 wherein the secondary side further comprises a transformer secondary side, and an output capacitor connected in series with the diode, the reactor, and the transformer secondary side, the voltage across the output capacitor being the output voltage.

11. The converter of Claim 1 wherein the reactor is a magnetic amplifier device.

12. The converter of Claim 11 further comprising a controller in electrical communication with the magnetic amplifier device, the controller being operative to control the conduction period of the magnetic amplifier device.

13. The converter of Claim 1 wherein the reactor is a controlled linear inductor.

14. The converter of Claim 13 further comprising a controller in electrical communication with the controlled linear inductor in order to control the inductance thereof.

15. The converter of Claim 1 further comprising a second secondary side in electrical communication with the primary side, the second secondary side being operative to generate a second output voltage different than the first output voltage.

16. The converter of Claim 15 wherein the second secondary side comprises a second variable reactor operative to selectively control a conduction period of the second secondary side in order to regulate the second output voltage.

17. The converter of Claim 16 further comprising a transformer in electrical communication with the primary side and the second secondary side.

18. The converter of Claim 17 wherein the transformer further comprises first and second secondary windings, the first secondary side being in electrical communication with the first secondary windings and the second secondary side being in electrical communication with the second secondary windings.

19. The converter of Claim 15 wherein the transformer further comprises a secondary winding having a tap and the second secondary side is in electrical communication with the tap.

20. The converter of Claim 15 wherein the second secondary side further comprises a second diode connected in series with the second reactor, the second reactor being operative to control the conduction period of the second diode.

21. The converter of Claim 1 further comprising a plurality of secondary sides, each of the secondary sides in electrical communication with the primary side and operative to generate respective prescribed output voltages.

22. The converter of Claim 1 wherein the primary side is operative to shape the input current waveform according to international input current harmonic standards for power factor correction.

23. A method of converting an ac voltage to an output dc voltage with a converter having a primary side and a secondary side, the method comprising the steps of: a) regulating an intermediary dc voltage on a dc energy storage bus from a rectified line voltage; and b) regulating the output dc voltage from the intermediary dc voltage with a variable reactor of the converter.

24. The method of Claim 23 wherein step (a) is performed by the primary side of the converter and step (b) is performed by the secondary side.

25. The method of Claim 23 prior to step (a) comprises the step of rectifying with the primary side the ac voltage to generate a rectified line voltage.

26. The method of Claim 25 wherein the step of rectifying the ac line voltage comprises rectifying the ac voltage with a bridge rectifier of the primary side.

27. The method of Claim 23 wherein step (a) comprises regulating the intermediary dc voltage with at least one switching device of the primary side.

28. The method of Claim 27 wherein step (a) comprises regulating the intermediary dc voltage with two switching devices.

29. The method of Claim 27 wherein step (a) further comprises regulating the intermediary dc voltage with a storage capacitor of the primary side.

30. The method of Claim 23 wherein the variable reactor of step (b) is a magnetic amplifier device.

31. The method of Claim 23 wherein the converter further comprises a plurality of secondary sides and the method further comprises the step: c) regulating a respective output voltage from the dc voltage with a variable reactor of each one of the respective secondary sides.

32. The method of Claim 23 wherein step (a) comprises regulating an intermediary voltage to shape an input current waveform according to international input current harmonic standards for power factor correction.

33. An ac/dc power converter for converting a universal ac input voltage to a prescribed output voltage, the converter comprising: a primary side in electrical communication with the input voltage, the primary side being operative to rectify the ac voltage to an intermediary dc voltage, and regulate the intermediary dc voltage on a dc energy storage bus; and a secondary side in electrical communication with the primary side, the secondary side having a variable reactor operative to regulate the output voltage from the voltage of the dc energy storage bus.

34. The converter of Claim 33 wherein the primary side comprises at least one switching device and a storage capacitor for regulating the intermediary dc voltage on the dc bus.

35. The converter of Claim 33 wherein the variable reactor of the secondary side is a magnetic amplifier.

36. The converter of Claim 33 wherein the variable reactor of the secondary side is a controlled linear inductor.

37. The converter of Claim 34 further comprising a controller in electrical communication with the linear inductor in order to control the inductance thereof.

38. The converter of Claim 33 further comprising a controller in electrical communication with the variable reactor and the output voltage, the controller being operative to control the conduction period of the secondary side in order to regulate the output voltage.

39. The converter of Claim 38 further comprising a plurality of secondary sides in electrical communication with the primary side, each of the secondary sides having a variable reactor operative to regulate a respective output voltage thereof.

40. The converter of Claim 38 wherein the variable reactors are magnetic amplifiers.

41. The converter of Claim 38 wherein the variable reactors are controlled linear inductors.

42. The converter of Claim 41 wherein each of the secondary sides further comprises a controller in electrical communication with a respective linear inductor to control the inductance thereof.

43. The converter of Claim 33 wherein the primary side is operative to shape the input current waveform of the input voltage according to international input current harmonic standards for power factor correction.