WO1992010918A1 - Signal processor for recreating original audio signals - Google Patents

Abstract

An audio signal processor (10, 12, 14) is disclosed for correcting harmonic and phase inaccuracies caused by the transduction, recording and playback of live audio signals. Operational amplifiers with fast slew rates and large bandwiths are used to implement a simple but effective circuit that can be reduced to a single chip. Signal phase is continuously and automatically shifted as a logarithmic function of frequency to compensate for differences in propagation velocities, thereby reducing the spatial smear that is inherent in conventional loudspeaker designs. Harmonic frequencies present during a live performance are regenerated by the signal processor to restore realism to the reproduced audio signal.

Description

TITLE OF THE INVENTION;

SIGNAL PROCESSOR FOR RECREATING ORIGINAL AUDIO SIGNALS

RELATED APPLICATION DATA;

This application is a continuation-in-part of U.S. patent application Serial No. 627,767, filed by Eldon A. Byrd on December 14, 1990 and entitled "Signal Processor for Recreating Original Stereo Signals".

BACKGROUND OF THE INVENTION;

The present invention relates generally to electronic circuits for audio signals, and is specifically concerned with circuits for restoring realism to stereophonic and monaural audio signals that has been lost during transduction and recording.

In 1969, R.C. Heyser in "Loudspeaker Phase Characteristics and Time Delay Distortion: Part 1", J. Audio En . Soc.. Vol. 17, p. 30, demonstrated that it was possible to obtain not only the amplitude spectrum of sound from a loudspeaker, but also the phase spectrum by an in-place measurement using a microphone. This provided an answer to the question of how well a speaker recreates the original sound field recorded by a microphone. In "Loudspeaker Phase Characteristics and Time Delay Distortion: Part 2", J. Audio Enσ. Soc.. Vol. 17, p. 130, Heyser showed that all loudspeakers that are absorptive and dispersive possess time-delay distortion, due to the fact that acoustic pressure waves do not effectively emerge from the transducer immediately when excited. The wave emerges with a time delay that is a multiple valued function of frequency. As a result, the sonic image is "smeared" in space behind the physical loudspeaker. The sonic effect perceived by the listener is that there is an ensemble of perfect loudspeakers placed at . various positions behind the loudspeaker, the positions being a function of frequency. The time delays are measured in terms of the number of milliseconds required for the sound pressure wave to emerge from the position in space occupied by the loudspeaker.

Theory indicates that acoustic events can be described in at least two ways: in a time domain or a frequency domain, each convertible into the other via a Fourier transformation. The mathematical formulation for this process is well known. The time-domain characterization of an acoustical event is a scalar, while the frequency-domain representation is a complex vector quantity containing amplitude and phase information. The time domain representation can also be expressed as a complex quantity. The scalar portion of the time domain vector represents performance based on impulse excitation; the imaginary part of the vector is the Hubert transform of the scalar.

Loudspeakers and electrical networks which transfer energy from one form to another can be characterized by response to an impulse function, because the impulse response can be manipulated to predict the behavior of the system in response to any arbitrary signal. Fourier transforms work for predictive systems as well as causal systems. However, the group velocity of a set of aduio signals is not related to time delay for all possible systems, and uniform group delay does not insure a distortionless system.

Because the time and frequency domains are two ways of describing the same event, accurate time domain representation cannot be obtained from limited frequency domain information. For example, the time delay of a frequency component passing through a system with nonuniform response cannot be determined with accuracy.

However, a joint time-frequency characterization can be made using first and second order all-pass networks. This is consistent with ordinary human experience. At any frequency there are multiple arrivals of the audio signal at the listener's location as a function of time. These can be viewed as a collection of perfect loudspeakers at various positions in space, each producing single-valued signal arrival times as function of the propagation velocity of each frequency. With conventional loudspeakers and electronics, therefore, a burst of energy from a loudspeaker will create what may be described as a "spatial" smear.

In 1971, R.C. Heyser in "Determination of Loudspeaker Signal Arrival Times: Part I", J. Audio Enσ. Soc.. Vol. 19, No. 9, pp. 734-742 (October 1971), showed that it is not possible to derive a unique time behavior based on incomplete knowledge from a limited portion of the frequency response. It is necessary to know the phase spectrum of the loudspeaker response in order to uniquely determine its impulse response. The general time domain vector contains information relating to the magnitude and partitioning (exchange) of kinetic and potential energy densities of the loudspeaker signals and is given by the equation: h ( t) *f( t) +ig( t)

where f(t) is a time-dependent disturbance and g(t) is its Hubert transform. The exchange ratio of kinetic to potential energy determines the upper bound for the local speed of propagation of the signal from a loudspeaker. Heyser showed that the actual versus virtual distances (i.e., the spatial smear) can be on the order of 12 inches. For recordings made with more than one microphone, the time delay distortion inherent in loudspeakers becomes audible. Live sound is a transient phenomenon. The only available way to compare two non-repetitive audio signals is to record them. For example, this can be done by monitoring transduction of a loudspeaker output signal via a microphone, and then recording the information and comparing it with the electrical signals which drove the loudspeaker. However, since these latter signals were also recorded from the output of microphones, the comparison is in reality between two recordings rather than between a recording and a live sound.

A purely mathematical analysis of a single loudspeaker reveals that there is a direct tie between the frequency response and time smear of information (signal) received by an observer. If the loudspeaker were isolated in an anechoic chamber, the acoustic response of the speaker could be replaced by a number of perfect response loudspeakers, each positioned in its own frequency- dependent location in space behind the physical position of the original imperfect loudspeaker. Time delay distortion is caused by this multiplicity of delayed equivalent sources. Experimental evidence has provided verification of this analysis.

The individual time-frequency components of an audio signal, predicted mathematically, overlap in the time and frequency domains. Therefore, a graphical presentation is not possible, because it is impossible to separate simultaneous arrival times in a single time domain plot. Potential energy (i.e., pressure expressed in dB) and comparisons of input to output signals directly (a measure of distortion) do not completely describe the performance of audio equipment quality such as loudspeakers, microphone, and electrical networks. Total sound energy provides phase distortion information and, although phase is not detectable consciously for simple signals, there are indications that the human hearing mechanism is capable of processing complex functions and perceiving phase information as part of total sound perception.

The square root of the total energy density vector, E, is equal to the sum of the square root of the potential energy vector and the imaginary component of the square root of the kinetic energy vector: Attempts to measure the total energy density at a microphone responding to a remote sound source will only yield part of the total energy density of the source. Thus, at any given moment, a microphone will not directly measure E. Essentially, a microphone compresses a complex spatial, multi-dimensional acoustic signals into a single point in time and space, effectively making the signal two-dimensional as a function of time. However, the information necessary to unravel the entire original signal is contained in the compressed signal and can be retrieved if processed property.

Although the threshold of hearing has been established in terms of vector orientation and frequency of pure tones (see, e.g., L. Sivian and S. White, "On Minimum Audible Sound Fields, J. Acoust. Soc. Am.. Vol. 4, pp. 288-321 (1933)), pure tones have no Fourier transforms. The human hearing mechanism processes total energy density, not just the "minimum audible pressure" associated with a pure audio tone.

The ability to localize direction and distance from a sound source has something to do with the orientation of the ear with respect to the vector components of sound. For pure tones, simply the phase differences between arrival of the signal at the two ears provides a clue to the direction of the source. See Kinsler and Frey, Fundamentals of Acoustics (New York: John Wiley and Sons, 1950), pp. 370-392. Thus, the minimum audible field for binaural hearing varies with amplitude, frequency, and azimuth relative to the source signal.

J. Zwislocki and R. Feld an (1956) "Just Noticeable Differences in Dichotic Phase", J. Acoust. Soc. Am. _f Vol. 28, No. 5, p. 860 (Spetember 1956) pointed out that the ears may not be able to detect phase or time differences above 1300 Hertz and the only directional clues above 1300 Hz are contained in relative intensity differences at the ears. Because Zwislocki and Feldman also state that the same intensity difference can occur at different azimuths, there is little directional information in the higher frequencies that provide accurate positioning. However, as noted above, the hearing mechanism is not limited to pressure stimulation alone.

In reality, the human auditory system binaurally localizes sounds in complex, spherical, three dimensional space using two sensors (ears) that are unlike microphones, a computer (brain) that is unlike any computer constructed by man, and, at a live performance, the eyes. The eyes allow us to "hear" direction by providing a sensory adjunct to the ears for localization of sound in azimuth, distance and height. During reconstruction of a familiar sound, such as a symphony orchestra, the brain remembers instrument placement and correlates this information with auditory clues to provide a more complete sense of the individual orchestra sections and sometimes of the locations of individual instruments. Techniques for localizing sound direction by the ears, neural pathways, and the brain have been termed "psychoacoustics". U.S. Patent 4,817,149, to Myers, points out that the brains of all humans look for the same set of clues (normally provided by the ear structure) concerning the direction of sound even if ear structures differ from person to person.

In addition to direction, the brain will interpret distance as a function of intensity and time of arrival differences. These clues can be provided by reflected sound in a closed environment such as a concert hall, or by other means for sound originating in environments where no reflections occur, such as in a large open field. In a closed environment, there is a damping effect as a function of frequency due to reverberations. When acoustic energy is reflected from a surface, a portion of the energy is lost in the form of heat. Low frequencies tend to lose less energy and are transmitted more readily, whereas high frequencies tend to be absorbed more quickly. This makes the decay time of high frequencies shorter than that of low frequencies. The air itself absorbs all frequencies, with greater absorbtion occurring at high frequencies.

In "Biophysical Basis of Sound Communication" by A. Michelsen (in B. Lewis (ed.), Bioacoustics. A Comparative Approach (London: Academic Press, 1983)), at pages 21-22, the absorption of sound in air is described as a combination of dissapation due to heat and other factors not well understood. In air, the absorbtion coefficient in dB/100 meters is l at about 2 KHz. At about 9 KHz, the signal is down by 10 dB; at 20 KHz it is down by 100 dB; and at 100 KHz (the upper harmonics of a cymbal crash) , it is down by about 1000 dB. Thus, higher harmonics generated by musical instruments are drastically attenuated (in a logarithmic fashion) by even a distance of a few feet when traveling to microphones, and then even more when traveling from speakers to the listener's ears. With conventional stereophonic sound reproduction systems, it is necessary to be equidistant from the speakers in order to experience the proper stereo effect. With earphones, standard stereo provides a strange ping- pong effect coupled with an elevated "center stage" in the middle and slightly above the head. At best, ordinary stereo is an attempt to spread sound out for increased realism, but it is still basically two-dimensional.

In the 1920s Sir Oliver Lodge tested human hearing range out to 100 KHz. It has been suggested that the true range of human hearing is not completely known. However, the outer ear, inner ear (cochlea) , auditory nerve, and human brain are capable of detecting, routing, and processing frequencies in excess of 100 KHz, and possibly to 300 KHz and beyond. However, conscious hearing is limited by the brain to roughly 20 Hz to 20 KHz.

There is no currently accepted theory of how humans actually hear outside the voice range of acoustic signals. Below about 200 Hz, the wavelength of an acoustic pressure wave is too large to enter the ear canal. Experience with low frequency standing waves suggests an interaction with the cochlea or auditory nerve directly. Indeed, standing wave acoustic emitters produce the perception of distortion-free sound throughout the hearing range. Above about 6 Hz, the "volley" theory and active cochlear processes could account for an increase in hearing range beyond 20 KHz. The volley theory is derived from the fact that there is not a single stimulus-response event per nerve; rather, higher frequency stimulation results in a multiplicity of neural firings. The process is one of bifurcation wherein the higher frequencies cause a greater number of neurons to fire. This suggests the possibility of fractal pattern generation. How the brain interprets the volley of information presented to it is unknown, however.

In Auditory Function, edited by G. Edleman, W. Gall, and W. Cowan, (New York: John Wiley & Sons, 1986) , a class of experiments is described which demonstrate acoustic emissions from animal and human ears. The cochlea can function as a generator of acoustic signals which can combine with incoming signals to produce higher frequencies. Both empirical and theoretical studies (indicating that active cochlea processes are necessary for basilar membrane tuning properties) support the concept.

P. Zurek, in "Acoustic Emissions from the Ear - A Summary of Results from Humans and Animals", . Acoust. Soc. Am.. Vol. 78, No. 1, pp. 340-344 (July 1985), indicates that frequency selectivity results from active cochlear processes. When the ear is presented with a non¬ linear pulse, in addition to the stimulus response mechanism, another response with an 8 millisecond (or longer) delay is produced. This phase-shifted signal, generated by the ear, may play a role in the actual way in which we hear musci and other high frequency sounds. When musical instruments produce sound, the various Fourier waveforms are not simply produced independently of each other, but exist in a phase space wherein there are phase interactions among all of the sounds. Even a single string plucked on a harp or struck on a piano will produce phase-related signals and harmonics, not simply frequencies and amplitudes. Thus, the ear must be capable of decoding phase information in order to properly transduce complex sounds such as music. The evoked time-delayed response in the ear is not simply a phase-shifted replica of the original sound, because the higher frequency components are time delayed less (about 8-10 milliseconds) than the lower frequency components of the emission (about 10-15 milliseconds) . Also, the amplitude of the evoken response is non-linear with respect to the stimulus for high stimulus levels, amounting to about 1 dB for every 3 dB increase in the stimulus. The interaction of the stimulus and acoustic emission occurs increasingly with lower and lower levels of input, suggesting that the ear may have a compensation mechanism for low level signals. People with certain types of hearing loss do not product acoustic emissions. At low levels of auditory stimulus, the emissions are almost equal in amplitude to the incoming signal itself, and they occur even for pure tones. The ear can generate continuous signals, and generated signals as high as 8 KHz have been observed.

As noted earlier, the conscious hearing range is roughly between 20 Hz and 20 KHz. Audio equipment has been designed to be optimal within that range. Also, most equipment has been designed to accurately reproduce that which has been transduced and recorded. However, live sound is a transient phenomenon. It is not possible to compare a live sound with anything, because in order to do so, it must be transduced and recorded in some way. It is this fact that forms the motivation for the present invention, and discussion of the prior art that follows. The previous discussion of auditory perception can be summarized as follows:

(a) There are at least six mechanisms involved in the interpretation of the intensity and direction of sound, including:

(1) Relative phase (for low frequencies) .

(2) Relative intensity (for midrange frequencies) .

(3) Relative time of arrival (for high frequencies) .

(4) Angle of the sound vector relative to the ears.

(5) Total energy density of the sound signal.

(6) Active and passive cochlear processes. (b) Arrival time at a listener for a signal emanated by a loudspeaker is a function of the total energy vector and can cause certain frequencies to appear to be located behind the speaker in a spatial smear.

(c) It is not possible to fully describe the time behavior of an audio reproduction system from limited frequency (bandwidth) information.

(d) The quality of audio recording and playback systems is not completely described by input versus output distortion and potential energy variations. (e) The human sound detection, transmission, and analysis system is capable of sensing phase distortion, of sensing frequencies far beyond 20 KHz, and of determining relative distance and direction of sounds.

(g) Live sound is a transient phenomenon which cannot be directly compared with transduced and recorded sound.

The prior art has addressed only some of the problems that are encountered in seeking to reproduce sound that more nearly resembles what the sound was like when it was live. The following patents and commercial audio systems are illustrative: U.S. Patent No. 4,100,371 to Bayliff, describes a two-loudspeaker system with phase difference compensation. Two sources of phase distortion were addressed. One is inherent in crossover networks and provides 90-degree shifts. The other is due to the fact that high and low frequencies emanate from different parts of the speaker cones and can cause 180-degree shifts at 3,000 Hertz. The proposed solution consisted of an active filter and speaker cabinet construction techniques. U.S. Patent No. 4,875,017 to Sakazaki, describes a digital technique for shifting phase of a video signal. U.S. Patent 4,626,796 to Elder, describes a digital technique for shifting audio tones in a cellular telephone. U.S. Patent No. 4,727,586 to Johnson, describes a technique for correcting phase distortion due to differences in frequency propagation times by placing speakers at different locations in their an enclosure. This is similar to the commercial BSR dbx "Soundfield V" system which uses biamplification instead of crossover networks and positions the high-frequency speakers behind the low-frequency driver and at an angle in the enclosure.

U.S. Patent No. 4,866,774 to Klayman, describes a servomechanism to drive potentiometers in a device that provides sums and differences of right and left channels of stereo for directivity enhancement of the stereo image.

U.S. Patent No. 4,218,585 to Carver, describes a three-dimensional sound processor that improves stereo by adding and subtracting inverted portions of the right and left channels via equalizers, passing them through a time delay, and mixing portions of the left channel to the right and vice-versa. Switches control the amount of dimensional effect.

U.S. Patent No. 4,769,848 to Eberbach, describes a complex design for a passive delay network inserted into the high frequency signal path of a crossover network in a loudspeaker. Correction of phase distortion caused by air path differences has been addressed in U.S. Patent Nos. 3,824,343, 3,927,261, and 4,015,089, wherein various schemes are used to physically provide phase compensation. U.S. Patent No. 4,151,369 discloses an active network for four speakers in a "surround sound" configuration.

Japanese Patent Publication No. 54-13321 discloses the combination of a phase delay circuit in the low frequency filter and phase-correcting speaker placement in the enclosure, while Japanese Patent Publication No. 52- 33517 discloses a phase delay circuit in the high frequency filter circuit.

U.S. Patent No. 4,567,607 to Bruney et al., discloses a method and apparatus for improving the accuracy of psychoacoustic images by cross-feeding signals from one channel to the other in an out-of-phase relationship in the 1-5 KHz frequency range, and by increasing the gain for frequencies in the 100-1000 Hertz range (except for signals in the 200-900 Hertz range that are applied to both channels) . The circuit also inverts the phase of low frequencies.

U.S. Patent No. 4,817,149 to Myers, discloses a very complex three-dimensional sound apparatus that imparts front-to-back and elevation cues by selectively boosting and attenuating certain frequencies, azimuth cues by time- delaying part of the signal, and various other cues by adding reverberation.

U.S. Patent No. 4,748,669 to Klayman, discloses a system that is extremely complex for creating three- dimensional sound by generating sum and difference signals from the left and right stereo channels, selectively altering the amplitude of signal components, and recombining the altered signals with the left and right channels to provide direction information for the listener.

Quadraphonic and so-called "surround sound" systems involve four or more speakers. These and other attempts to correct the problems involved in transducing, recording, and playing back sound have created environmental ambience, but have fallen short in re¬ creating the original sound. The "QSound" system developed by Archer Communications of Calgary, Alberta, provides three- dimensional sound but does not improve the basic quality of the sound. Also, it is a relatively complex system that uses computers for sound mixing. The "Natural Sound Digital Sound Field Processor" developed by Yamaha Corporation of Hamamatso, Japan, attempts to recreate the ambience of several types of spaces, such as concert halls, movie theaters, and so on, by adding reverberation, echo, and presence. The signal processing is digital, rather than analog, and hence the processor is relatively complex.

The "Dolby Surround Pro Logic" system developed by Dolby Laboratories claims to recreate a live performance by using four speakers to provide "reflected" sound. The listener is placed in the center of the sound to enhance the dimensionality of video tape sound tracks.

Although the prior art discussed above has attempted to correct some of the problems associated with distortion in audio systems due to phase shifts as a function of frequency, and spatial distortion due to the inherent inaccuracies in standard stereo, these attempts have not completely succeeded in restoring lost realism to recorded sound. At best, some prior art processors create the illusion of ambience. The prior art provides single and, in some cases, multiple corrections to recorded signals. The object of the prior art is, in general, to control the location of sound cues and provide phase correction, not to increase the quality of the sound by putting back in to the signal what was removed by the transduction, recording, and playback systems. As previously pointed out, microphones compress signals that can consist of many fundamental frequencies from different instruments at different spatial locations. These signals also contain complex interactions for the fundamentals and harmonics produced by the same instruments. When cymbals crash, for example, the harmonics produced reach above 100,000 Hertz. As the complex signal develops from these interactions, it can become non-linear and sub-harmonics will be present. At first, it would appear impossible to retrieve or reconstruct a complex signal whose spectral content has been compressed by microphones in both the time and spatial domains. The digital sampling rate of information that is recorded on compact discs and digital audio tapes, for example, results not only in a loss of information, but also in an absolute frequency cutoff that is lower than the upper harmonics produced by some musical instruments. The present invention arises from the recognition that, if the harmonics and subharmonics of recorded sound are allowed to develop from the fundamental frequencies, and if the spectral content of the signal is spatially separated, the original live sound can be recreated.

SUMMARY OF THE INVENTION:

The present invention overcomes the shortcomings of the prior art by causing harmonics and sub-harmonics to develop for all frequencies, by continuously correcting the phase of the signal logarithmically as a function of frequency, by spatially separating the spectral content of the signal, and by dramatically increasing the audio bandwidth of the signal. Preferably, the audio bandwidth is expanded beyond the normal range of human hearing (20- 20,000 Hz) to a range of 0 Hz to more than 2 MHZ. The invention is based, in part, on the recognition that the human hearing mechanism for sensing audio signals (as opposed to electromagnetic and tactile signals) is different from the electronic circuits used to construct amplifiers, microphones, tape recorders, and other types of audio equipment. Thus, when humans hear or sense an audio signal, it is processed differently than standard apparatus attempting to transduce, record, and playback the original sound.

In accordance with one aspect of the present invention, an audio signal processor comprises an input terminal for receiving an audio signal, first and second processing stages for processing the audio signal, and an output terminal for coupling the processed audio signal to an output device. The first and second signal processing stages are arranged in a series or cascade configuration, and each stage functions to generate harmonic frequencies related to fundamental frequencies in the audio signal and to phase shift the fundamental and harmonic frequencies as a function of frequency. The phase shift increases in a negative direction with increasing frequency, so that higher frequency signals to lag the lower frequency signals.

In accordance with another aspect of the present invention, a method for carrying out two-stage processing of an input audio signal comprises the steps of generating harmonic frequencies related to fundamental frequencies in the input audio signal and adding the harmonic frequencies to the input audio signal; phase shifting the fundamental and harmonic frequencies as a function of frequency, with the phase shift increasing in a negative direction with increasing frequency so as to cause higher frequency signals to lag lower frequency signals; generating harmonic frequencies related to fundamental frequencies in the partially processed audio signal resulting from the preceding method steps, and adding the harmonic frequencies to the partially processed audio signal; and phase shifting the fundamental and harmonic frequencies of the partially processed audio signal as a function of frequency, with the phase shift increasing in a negative direction with increasing frequency so as to cause higher frequency signals to lag lower frequency signals.

The present invention can be implemented by means of a relatively simple electrical circuit that can be manufactured and sold at very low cost. The principal components of the circuit can, if desired, be reduced to a single dual inline package (DIP) which can be incorporated into existing types of audio equipment. The invention can be utilized with nearly all existing types of power amplifiers, stereo tuners, and phonographs with preamplifiers, as well as with compact disk (CD) players, digital audio tape (DAT) players, and conventional analog tape recorders and players. All recorded media can be reproduced with a sound that is very close to that of a live performance, and the spatial smear that is perceived with standard loudspeakers is reduced. The invention can be used with any number of audio channels or speakers; the resulting sound will be dimensionalized, to some extent, with even a single speaker. The signal processing that is carried out by the present invention transfers to tape and to virtually any other type of recording medium. Thus, for example, a digital CD output can be processed using the present invention, and the result can be recorded on ordinary stereo audio tape. The present invention restores information that has been lost during digital or analog processing, as well as during the transduction of the original sound, and may be employed at a radio or television broadcasting station to improve the quality of the audio signal received by the listeners.

Further objectives, advantages and novel features of the invention will become apparent from the detailed description which follows. BRIEF DESCRIPTION OF THE DRAWINGS:

The details of several preferred embodiments of the present invention are illustrated in the appended drawings, in which: Fig. 1 is a functional block diagram illustrating the operation of an audio signal processor constructed in accordance with the principles of the present invention;

Fig. 2 is a graph illustrating the manner in which the audio signal processor carries out phase shifting and spatial separation of an audio signal as a function of frequency;

Fig. 3 is a detailed schematic diagram of a first embodiment of an audio signal processor constructed in accordance with the present invention; Fig. 4 is a detailed schematic diagram of one of the operational amplifiers used in the signal processor circuit of Fig. 3;

Fig. 5A is a graph illustrating the phase shift characteristics of the operational amplifier shown in Fig. 4;

Fig. 5B is a graph illustrating the variation in phase shift with frequency that is provided by the phase shift stages used in the audio signal processor of Fig. 3; Fig. 6 is a graph illustrating the bandwidth and frequency response characteristics of the signal processor circuit shown in Fig. 3, based on actual measurements thereof;

Fig. 7 illustrates a second embodiment of an audio signal processor in accordance with the present invention, intended for use with an audio system of the type typically found in an automobile;

Fig. 8 illustrates a third embodiment of an audio signal processor in accordance with the present invention, intended for use in conjuction with a home audio, system; Fig. 9 illustrates a third embodiment of an audio signal processor in accordance with the present invention, also for use with a home audio system;

Fig. 10 illustrates a fourth embodiment of an audio signal processor in accordance with the present invention, intended for use in recording studios, broadcast stations and other professional environments;

Figs. 11A-11D illustrate various methods for connecting the audio signal processor of the present invention to an existing audio system in a home or automobile;

Figs. 12A and 12B depict oscilloscope traces generated by the audio signals at the input and output, respectively, of the audio signal processor of the present invention;

Fig. 13A and 13B depict x-y oscilloscope traces comparing the input and output of the audio signal processor of the present invention; and

Figs. 14A and 14B depict spectrum analyzer traces illustrating the harmonic content restored to an audio signal by the signal processor of the present invention.

Throughout the drawings, like reference numerals should be understood to refer to like parts and components, unless otherwise indicated.

DESCRIPTION OF THE PREFERRED EMBODIMENTS;

The basic functions carried out by the audio signal processor of the present invention are illustrated in the functional block diagram of Fig. 1. At the input 16, an audio input signal is received from an audio source which will typically consist of a playback device such as a phonograph, tape player or compact disk player. However, the audio source may also consist of a receiving device such as an audio or video tuner, or a live output from a microphone or musical instrument transducer. The audio input on line 16 is applied first to a harmonic generator 10, which enhances the signal by adding harmonics and subharmonics of the fundamental frequencies found in the original audio performance. Typically, these harmonics and subharmonics have been lost during the recording and/or transducing process. As will be described in some detail hereinafter, the harmonic generator 10 is preferably implemented by means of one or more operational amplifiers. The output of the harmonic generator 10 is applied to the input of a phase shift and spatial separation network 12. The function of the network 12 is to automatically shift all input frequencies in phase as a logarithmic function of frequency. Thus, all of the audio frequencies appearing at the output of the harmonic generator 10 are phase-shifted, including the newly generated harmonic and subharmonic frequencies. The output of the network 12 is applied to the input of a left and right channel comparator and mixer 14. The function of this portion of the audio signal processor is to mix portions of the right and left channels of a stereophonic audio program, in order to product a dimensional expansion of the stereo image which includes, for example, depth and height cues. Therefore, the output signal which appears on line 18 differs from the input signal at 16 by virtue of having additional harmonic and subharmonic content, by virtue of having phase shifts among various frequency components of the composite audio signal as a logarithmic function of frequency, and by virtue of having the right and left channel information mixed to some extent to produce dimensional expansion of the sound. It should be understood, of course, that the input and output lines 16, 18 each represent two distinct lines (i.e., left and right channel inputs and outputs) in stereophonic embodiments. In monaural applications, the same input signal may be applied to the two input lines 16. It should also be understood that, in actual hardware implementations of the present invention, the functions represented by the blocks 10-14 in Fig. 1 may, at least to some extent, be carried out simultaneously by the same components or groups of components, rather than occurring separately and sequentially as Fig. 1 might suggest.

Fig. 2 is a graph illustrating in a general way the manner in which the phase shift and spatial separation network 12 of Fig. 1 operates. As harmonics are generated by the harmonic generator 10, they are automatically shifted in phase by the network 12 as a function of frequency. This has the effect of creating an increasing time delay for the higher frequencies, as illustrated in Fig. 2. Because higher frequency sounds move faster through the air than lower frequency sounds, the function of the network 12 in Fig. 1 is to correct for the "spatial smearing" created by microphones and loudspeakers. At the same time, the harmonic generator 10 adds back into the signal harmonic information that was lost due to the frequency limitations of the recording equipment, the time and frequency domain compression caused by the microphones used during the recording process, and the high frequency cut-off inherent in any digital equipment used during recording and playback.

Without limiting the invention to any particular theory of operation, it is possible that the spatial separation of different frequency components of the incoming two-dimensional compressed audio signal into time-delayed harmonics and fundamentals allows the human brain more time to process the received information, and to interpret the resulting audio signal as more closely resembling a live performance (in which the fundamentals and harmonics are, in fact, separated spatially) . Also, the possibility exists that the outer ear, inner ear, neural pathway, and brain process signals well beyond 100 KHz, possibly into the gigahertz range. It is well known that microwave frequencies and ultraviolet and infrared frequencies are radiated from living ÷cells at times, and that a small percentage of the population can "hear" microwave frequencies through auditory or cerebral processes that are not completely understood.

Fig. 3 is a detailed schematic diagram of a first embodiment of an audio signal processor constructed in accordance with the present invention. The audio signal processor comprises a total of six operational amplifiers, three for processing the left channel of the audio signal input and three for processing the right channel. Various resistors, capacitors and switches are also provided, as will be described. Referring first to the left channel processing circuit, an input signal from the audio source is applied to the non-inverting input of an operational amplifier 20. The output of the operational amplifier 20 is fed back to its inverting input, so that the operational amplifier 20 functions as a unity gain stage. As such, the operational amplifier 20 serves primarily as a buffer stage to reduce input noise, although in practice some harmonic content may be added to the audio signal in this stage. The output of the operational amplifier 20 is connected through a first resistor 22 to the non-inverting input of a second operational amplifier 30, and through a second resistor 24 to the inverting input of the operational amplifier 30. The node between the resistor 22 and the inverting input of the operational amplifier 30 is connected to ground through a capacitor 26. The output of the operational amplifier 30 is connected to its inverting input through a feedback resistor 28. The value of the feedback resistor 28 is adjusted so that the overall gain of the operational amplifier stage 30 is approximately unity.

The operational amplifier 30, resistors 22, 24 and 28, and capacitor 26 constitute a phase shift circuit 21 which carries out the functions represented by block 12 in Fig. 1. In other words, the individual frequency components of the audio signal at the output of the operational amplifier 20 are phase-shifted as a logarithmic function of frequency, with the phase shift occurring in a negative direction and being greater for higher frequency signals. In effect, therefore, the higher-frequency signals are time-delayed with respect to the lower frequency signals, as illustrated in Fig. 2, and this serves to reduce the spatial smearing effect discussed previously.

In addition to the phase-shifting function, the operational amplifier stage 30 also restores harmonic content to the audio signal. This is believed to result from two separate effects. First, because there is a phase shift between the signals applied to the inverting and non-inverting inputs of the operational amplifier 30, the amplifier is effectively comparing a phase-shifted version of an audio signal to the original signal. This will inherently result in the generation of harmonic frequency components. The second effect has to do with the interconnections between the left and right channels of the audio signal processor, as will be described shortly.

With continued reference to Fig. 3, the output of the operational amplifier 30 is connected to the non-inverting input of a third operational amplifier 32. The output of the operational amplifier 32 is connected directly to its inverting input, as shown. Thus, the operational amplifier 32 serves as a unity gain stage, similar to the initial operational amplifier stage 20. As such, it functions primarily to reduce noise in the audio signal. The output of the operational amplifier 32 constitutes the left channel output of the audio signal processor, as shown.

The right channel of the audio signal processor of Fig. 3 is essentially identical to the left channel that has already been described. The right channel input from the audio source is applied to the non-inverting input of an operational amplifier 36, which is connected in a unity-gain configuration as shown. The output of the operational amplifier 36 at node 37 is connected to the non-inverting input of a second operational amplifier 46 through a resistor 38, and to the inverting input of the amplifier 46 through a resistor 42. A feedback resistor 44 connects the output of the operational amplifier 46 to its inverting input, and a capacitor 40 is connected between the non-inverting input of the amplifier 46 and ground. The circuit 31 comprising the operational amplifier 46, resistors 38, 42 and 44, and capacitor 40 performs phase shifting and harmonic generating functions similar to those performed by the circuit 21 of the right channel, as described above. The output of the operational amplifier 46 is connected to a third operational amplifier 48, which is connected in a unity- gain configuration as shown. The operational amplifier 48 is provided primarily for noise reduction. The output of the operational amplifier 48 constitutes the right channel output of the audio signal processor of Fig.3.

As noted previously in connection with block 14 of Fig. 1, some degree of mixing between the left and right channels is desirable in order to increase the dimensionality of the sound produced by the audio signal processor. In Fig. 3, the mixing is achieved by connecting the non-inverting input of the operational amplifier 30 in the left channel to the inverting input of the operational amplifier 36 in the right channel through a switch 34. In addition, the output of the operational amplifier 20 of the left channel is connected to the non- inverting input of the operational amplifier 46 of the right channel through a second switch 35. The switches 34 and 35 may be combined into a double-pole, single-throw switch as indicated in Fig. 3. When the switches 34, 35 are in the open position, no mixing occurs between the left and right channels of the audio signal processor. When the switch 35 is in the closed position, however, the output of the operational amplifier 20 is applied to the non-inverting input of the operational amplifier 40. This effectively feeds the left channel signal into the right channel. Thus, the circuit 31 also serves as a comparator network for comparing the left and right channel signals and for removing part of the right channel information which is also present in the left channel. In a similar manner, the closing of the switch 34 causes the output of the operational amplifier 36 in the right channel to be applied to the non-inverting input of the operational amplifier 30. As a result, the right channel signal is fed into the left channel, and the circuit 21 functions as a comparator for removing part of the audio information from the left channel that is also present in the right channel. Accordingly, when the switches 34 and 35 are closed, the spatial separation between the right and left channel outputs is enhanced. Monaural signals are unaffected by the comparators 21 and 31, since the audio signals in both channels are identical.

As noted briefly above, the interconnections between the left and right channels in the audio signal processor of Fig. 3 enhance the generation of harmonic and subharmonic frequencies by the operational amplifier stages 30 and 46. For example, by adding the left channel signal to the right channel signal after the right channel signal has been phase-shifted by the circuit 31 and attenuated by the resistor 38, a new left channel signal is created that is rich in harmonics. The harmonics result from the addition, at the non-inverting input of the operational amplifier 46, of the signals present in the non-phase-shifted left channel with the signals present in the phase-shifted right channel. The new signal is no longer coherent in phase with respect to the left or right channels, unless both channels contain the same information. In a similar manner, a new right channel signal with added harmonics is created by adding the right channel signal to the left channel signal at the non-inverting input of the operational amplifier 30. Preferred values for the components used in the audio signal processor of Fig. 3 are provided in Table 1 below. These values are given merely by way of example and are not intended to limit the scope of the present invention.

TABLE 1

Component Value or Type

Resistors 22, 38 100 KΩ, h W

Resistors 24, 28, 42, 44 10 KΩ, h W

Capacitors 26, 40 0.1 μF Mylar

Operational amplifiers National Semiconductor 20, 30, 32, 36, 46, 48 LF147/LF347 or equivalent

Switches 34, 35 (2) SPST or (1) DPST

Fig. 4 is a detailed schematic diagram of one of the operational amplifiers 20, 30, 32, 36, 46 and 48. All of the operational amplifiers are identical, with a frequency bandwidth of 0 Hz to more than 4 MHz and a fast slew rate (greater than 12 volts per microsecond) . Each operational amplifier is provided with the required plus and minus DC supply voltages by a suitable power supply circuit (not shown in Figs. 3 and 4) . The specific type of operational amplifier illustrated in Fig. 4 is a National Semiconductor LF147/LF347 wide-bandwidth JFET input operational amplifier, although other commercially available types of operational amplifiers can be used if desired. Examples include the Analog Devices AD713JN quad operational amplifier, and similar types of operational amplifiers that are manufactured by Motorola and Texas Instruments. Low noise, JFET input operational amplifiers are particularly preferred for use in connection with the present invention, although any operational amplifier with a high slew rate can be used.

Fig. 5A is a graph illustrating the phase shift characteristic of the National Semiconductor LF147/LF347 operational amplifier depicted in Fig. 4. The graph is generic, inasmuch as various resistive and capacitive loads will alter the degree of phase shift as a function of frequency. In the case of the phase shift stages 21, 31 shown in Fig. 3, the variation in phase shift with frequency is illustrated in Fig. 5B. In the general case, the resistors 22 and 38 may be referred to as R_x, resistors 24 and 42 as R₂, and feedback resistors 28 and 44 as R₃. The capacitors 26 and 40 may be referred to as C. Assuming that the values indicated in Table 1 are used, the gain of each of the phase shift circuits 21 and 31, where R₃ = R₂ = Rι/10 and R_XC = T = 10, is unity. The transfer function G(s) for the circuits 21 and 31 is:

_G(s) = ^{1 ~ τ ' s} = ^{1 ~ 10s} 1 + τ ^• s 1 + 10≤r

where s is the Laplacian operator. The phase for frequencies higher than 100 f in Fig. 5B is retarded or delayed by 180° relative to frequencies lower than f/100. Comparing this to Fig. 5A, it will be noted that a 180° phase shift will occur at 100 KHz. This corresponds to the 100 point in Fig. 5B, and hence f is about 1 KHz. As will be described shortly in connection with Figs. 7-10, it may be preferable to provide cascaded phase shift stages in order to enhance the performance of the audio signal processor of Fig. 3. When this is done, the transfer function becomes:

where T is equal for each phase shifter. If the time constant T is not equal in the two stages, a composite non-linear phase shift can occur, creating harmonics and changing the characteristics of the phase shift to tailor the performance of the audio signal processor. Monaural sound processing is enhanced when the two channel time constants τ__ and τ₂ are different, since the resulting differences in the phase shifts and harmonic generation in the left and right channels can simulate or replace the left and right channel separation which characterizes stereophonic recordings. Fig. 5C shows that the net change in phase shift (i.e., delay time) for identical cascaded stages is about 10 milliseconds at 50 Hz and decreases with increasing frequency.

Fig. 6 is a graph showing the frequency response and bandwidth of the output versus input signals for each channel of the audio signal processor of Fig. 3. At 0 Hz, there is a 0.5 dB drop, increasing to about 0 dB from 20 Hz to 400 KHz. This is followed by a roll-off to -3 dB from 400 KHz to about 2 MHz, and then by a sharp drop to about -40 dB at 8 MHz. Substitution of a higher frequency operational amplifier and adjustment of appropriate biasing components will produce a circuit capable of processing higher frequency signals.

Having now described the audio signal processor of Fig. 3 and its individual components in detail, it will be seen that the functions broadly outlined in the block diagram of Fig. 1 have been accomplished. The circuit of Fig. 3 adds harmonic and subharmonic frequencies to the input audio signal to compensate for information lost during transduction and recording. Continuous phase shifting of the fundamental and harmonic frequencies is carried out as a function of frequency in order to reduce the spatial smear that is associated with most types of loudspeakers. In addition, while preserving the original stereo image, the signal processor mixes the right and left channel information in such a way as to restore a sense of dimensionality to the sound. When the audio signal processor of Fig. 3 is used to reproduce recorded music, the realism of the processed sound is commensurate with that of a live performance. Each instrument maintains its individuality and the feeling and emotion of the original performance are recreated.

Although the audio signal processor of Fig. 3 accomplishes the objectives of the present invention, it has been found that cascaded versions of that circuit (i.e., versions in which the outputs shown in Fig. 3 are applied to the inputs of an identical circuit) yield even better performance. In these embodiments, the already processed audio signal is provided with additional harmonic content and phase shifting, and the resulting output signal is markedly improved over the output of the non-cascaded circuit shown in Fig. 3. While circuits containing three or more cascaded stages are also possible, the improvement that is obtained with more than two cascaded stages does not outweigh the cost of providing additional circuit components. Accordingly, audio signal processors comprising two cascaded processing stages are regarded as the most preferred embodiment of the present invention, and four different examples of such a circuit are illustrated in Figs. 7-10, respectively.

Fig. 7 is a detailed schematic diagram of a cascaded audio signal processing circuit which is intended for use in connection with an audio system of the type that is typically found in an automobile. The right channel operational amplifiers 60, 62 and 64 correspond to the operational amplifiers 36, 46 and 48 in Fig. 3, and the resistors 66, 68, 70 and capacitor 71 correspond to the resistors 38, 42, 44 and capacitor 40 in Fig. 3. Similarly, the left channel operational amplifiers 72, 74 and 76 of Fig. 7 correspond to the operational amplifiers 20, 30 and 32 of Fig. 3, and the resistors 78, 80, 82 and capacitor 83 of Fig. 7 correspond to the resistors 22, 24, 28 and capacitor 26 of Fig. 3. In the circuit of Fig. 7, however, the first stage of the right channel which comprises the operational amplifiers 60-64, resistors 66- 70 and capacitor 71 is cascaded with an identical second processing stage which comprises operational amplifiers 84-88, resistors 90-96 and capacitor 97. Similarly, the first processing stage of the left channel which comprises the operational amplifiers 72-76, resistors 78-82 and capacitor 83 is cascaded with an identical second processing stage which comprises operational amplifiers 98-102, resistors 104-108 and capacitor 109.

In the cascaded signal processing circuit of Fig. 7, crossover or mixing between the right and left channels is not provided in the first stage and is permanently hard¬ wired in the second stage. To this end, the output of the operational amplifier 84 in the second stage of the right channel is hard-wired to the non-inverting input of the operational amplifier 100 in the second stage of the left channel. Similarly, the output of the operational amplifier 98 in the second stage of the left channel is hard-wired to the non-inverting input of the operational amplifier 86 in the second stage of the right channel. Although it is possible to provide the mixing or cross- over connections in either stage of the signal processor (or in both stages) , it has been found that the provision of channel mixing in the second stage has the greatest effect on the perceived dimensionality of the resulting audio signal. Since the audio signal processor of Fig. 7 is intended for use in an automobile, the number of switches and indicators has been kept to a minimum. A power switch 110 operates a power supply which provides +8 volts DC and -8 volts DC to each of the operational amplifiers 60-64, 72-76, 84-88 and 98-102. A power-on LED indicates that power is being supplied to the audio signal processor. A second switch 114 serves as a bypass switch for allowing the audio signal processor to be selectively switched into or out of the existing audio system of the automobile. The bypass switch 114, which may be implemented as three ganged single-pole, single-throw switches or as a single 3PST switch, connects the right and left channel inputs to the corresponding right and left channel outputs via the signal processing stages 60-64, 84-88, 72-76 and 98-102 when the switch is in the position shown in Fig. 7. When the switch is in the opposite position, the right and left channel inputs are connected directly to the right and left channel outputs without passing through any of the signal processing stages. In this switch position, the bypass LED 116 is illuminated by current drawn from the +8 volt DC power supply through a current limiting resistor 118 to indicate to the user that the bypass mode has been selected.

The power supply for the audio signal processor of Fig. 7 is specifically designed for use with a +12 volt DC battery of the type that is found in most automobiles. The +12 volt DC terminal 120 is connected via the power switch 110 to a choke coil 122 which serves to reduce ignition noise. The opposite side of the choke coil 122 is connected to two circuit branches, one producing +8 volts DC and the other producing -8 volts DC. The positive circuit branch includes a current limiting resistor 124, a series of smoothing capacitors 126-130, and a positive voltage regulator circuit 132. The output of the voltage regulator 132 is connected to a filter capacitor 134 and provides a +8 volt DC output for the operational amplifiers 60-64, 72-76, 84-88 and 98-102. A current limiting resistor 136 is connected between the voltage regulator output and the power-on LED 112 referred to previously, so that the LED is illuminated whenever the power switch 110 is in a closed position. Referring now to the negative branch of the power supply circuit, the output side of the choke coil 122 is connected through a current limiting resistor 138 to the input of a polarity converting circuit 140. A Zener diode 142 and a smoothing capacitor 144 are connected between the input of the polarity converter 140 and ground, in order to reduce spikes and ripple in the input voltage. The polarity converter 140 has an internal oscillator which requires an external capacitor 146, and also requires a connection 148 to ground. The output voltage from the polarity converter on line 150 has a negative polarity and is applied to the input of a negative voltage regulator 152. Two smoothing capacitors 154, 156 are connected between the line 150 and ground to eliminate voltage transients of either polarity. The output of the negative voltage regulator 152 on line 158 provides -8 volts DC to the operational amplifiers 60-64, 72-76, 84-88 and 98-102. A filter capacitor 160 is connected between line 158 and ground in order to stabilize the output of the regulator 152.

Preferred values for the circuit components of Fig. 7 are provided in Table 2 below. The values of the resistors and capacitors used in the signal processor stages are the same as those of Fig. 3, but the operational amplifiers are designed to operate with +8 and -8 DC supply voltages. As in the case of Fig. 3, the specification of particular component values or types is intended merely by way of example and not by way of limitation.

TABLE 2 (Continued)

Component Value or Type Resistor 118 1 KΩ, h W Switch 110 SPST Choke coil 122 350 μH Resistor 138 50 Ω, h W Capacitor 126 100 μF, electrolytic Resistor 124 5 Ω, h W Capacitor 128 470 μF, electrolytic Capacitors 130, 156 0.1 μF, disk Voltage Regulator 132 Texas Instruments LM7808 or equivalent

Capacitors 134, 146, 154, 10 μF, electrolytic 160

Resistor 136 1 KΩ, h W

Polarity Converter 140 Harris 7660/7660S or equivalent

Voltage Regulator 152 Texas Instruments IM7908 or equivalent

A modified version of the cascaded audio signal processor of Fig. 7, intended for use in connection with home audio systems, is illustrated in Fig. 8. As far as the signal processing stages are concerned, this version is essentially the same as the previous version of Fig. 7, except that the operational amplifiers 60'-64', 72'-76', 84'-88' and 98'-102' operate on +15 and -15 volts DC rather than +8 and -8 volts DC. Also, in order to allow the user to control the overall gain of the audio signal processor, as well as the amount of the processing (i.e., phase shift and harmonic generation) that is carried out on the incoming audio signal, feedback resistors 96' and 108' in the second cascaded signal processing stage of each channel are provided in the form of potentiometers rather than fixed resistors. Preferably, the potentiometers 96' and 108' of the right and left channels are ganged, as shown, so that they can be adjusted simultaneously by the user. The other major difference between the circuit of Fig. 8 and that of Fig. 7 has to do with the nature of the power supply, as will now be described.

Since the audio signal processor of Fig. 8 is intended for use with a home audio system, the power supply is designed to operate from a 120 volt AC power source 162, which will typically consist of a wall outlet. The 120 volt input is applied across the primary side of a step-down transformer 164, and the secondary of the transformer is connected across a filter capacitor 166 through a power switch 168. The node between the power switch 168 and the capacitor 166 is connected to a positive circuit branch through a first diode 170 and to a negative circuit branch through a second diode 172. Referring first to the positive circuit branch, the cathode of the diode 170 is connected to the input of a positive voltage regulator 174. Smoothing capacitors 176 and 178 are connected between the input to the voltage regulator 174 and ground. The output of the voltage regulator 174 is a regulated +15 volt DC level which is connected to the +15 volt power supply input of each of the operational amplifiers 60'-64', 72'-76', 84'-88' and 98'-102'. Two capacitors 180 and 181 are connected between the output of the regulator 174 and ground in order to stabilize the output voltage level and polarity. In addition, a current limiting resistor 182 and an LED 184 are connected in series between the regulator output and ground. When the power switch 168 is in a closed position, the LED 184 is illuminated to provide the user with a power-on indication.

Referring now to the negative branch of the power supply circuit, the anode of the diode 172 is connected to the input of a negative voltage regulator 186. Smoothing capacitors 188 and 190 are connected between the input of the voltage regulator 186 and ground, as shown. The negative voltage regulator 186 converts the negative half- cycles from the diode 172 to a stable -15 volt DC output which is applied to the -15 volt DC inputs of the operational amplifiers 60'-64', 72'-76', 84'-88' and 98'- 102'. Two capacitors 192 and 194 are connected between the regulator output and ground in order to stabilize the output voltage level and polarity. Preferred values for the components used in the audio signal processor of Fig. 8 are provided in Table 3 below. As in the case of the previous Tables, these values are provided simply by way of example and not by way of limitation.

TABLE 3

Component Value or Type

Resistors 66, 78, 90, 100 KΩ, k W

104 Resistors 68, 70, 80, 10 KΩ, k W

82, 92, 106

Potentiometer 96', 108' 0-100 KΩ, k W

Capacitors 71, 83, 97, 0.1 μF, Mylar 109

Operational amplifiers National Semiconductor 60'-64', 72-76', LF147/LF347 or 84'-88', 98'-102' equivalent

Switch 114 3PST

LEDs 116, 184 2 volt nominal TABLE 3 (Continued)

Component Value or Tvoe Resistor 118 1 KΩ, k W Transformer 164 120 volt input, 22.5 volt output, 0.5 ampere secondary

Switch 168 SPST

Diodes 170, 172 1N4002

Capacitors 176, 188 10 μF, electrolytic

Capacitors 178, 181, 0.1 μF, disk 190, 192

Voltage regulator 174 National Semiconductor IM7815 or equivalent

Capacitors 180, 194 470 μF, electrolytic

LED 184 2 volt nominal

Voltage regulator 186 National Semiconductor LM7915 or equivalent

Fig 9 illustrates a modified version of the cascaded audio signal processing circuit of Fig. 8. The embodiment of Fig. 9 is also intended for use in connection with home audio systems, but provides the user with the additional option of employing crossover or channel mixing in the first cascaded signal processing stage while maintaining the hard-wired crossover in the second signal processing stage. To that end, a two-position mode switch 196 is provided between the respective first stages of the right and left channels, as illustrated in Fig. 9. When the mode switch 196 is in the position shown, the output of the operational amplifier 60' in the right channel is tied to the non-inverting input of the operational amplifier 74 in the left channel. Similarly, the output of the operational amplifier 72' in the left channel is tied to the non-inverting input of the operational amplifier 62' in the right channel. In this condition, crossover or mixing between the right and left channels takes place in the first cascaded stage of each channel, and this supplements the mixing that takes place in the second stage of each channel as a result of the hard-wired connections. In order to indicate to the user that the circuit is operating in this mode, the mode switch 196 includes an additional pole which connects an LED 198 to the +15 volt DC power supply through a current limiting resistor 200. For convenience, this mode is referred to as "Mode 2", although the designation of modes is purely arbitrary. When the mode switch 196 is in the position opposite from that shown in Fig. 9, the crossover connections between the first stage operational amplifiers of the right and left channels are broken. Accordingly, no channel mixing takes place in the first stage of the audio signal processor, although channel mixing continues to occur in the second stage as a result of the hard-wired crossover connections. This results in a reduced amount of processing of the audio signal, which may be desired by the user for certain types of audio programs. When the mode switch is in this position, the +15 volt DC supply is disconnected from the LED 198 and is connected to a second LED 202. This causes the LED 198 to be extinguished and the LED 202 to be illuminated, thereby indicating to the user that the audio signal processor is operating in "Mode 1". The mode switch 196 may take a variety of forms, such as a two-position toggle or rocker switch, a two-position rotary switch, or a pair of pushbuttons. The bypass switch 114 may be implemented in a similar manner and may be combined with the mode switch 196 into a single six- pole, three-position switch.

Preferred values for the circuit components in Fig. 9 are provided in Table 4 below. As before, these values are provided merely for the purpose of example, and are not intended to limit the scope of the present invention.

TABLE 4

Component Value or Type

Resistors 66, 78, 90, 100 KΩ, h W 104

Resistors 68, 70, 80, 10 KΩ, h w 82, 92, 106

Potentiometer 96', 108' 0-100 KΩ, h w

Capacitors 71, 83, 97, 0.1 μF Mylar 109

Operational amplifiers National Semiconductor 60'-64', 72'-76', LF147/LF347 or 84'-88', 98'-102' equivalent

Switch 114 3PST

LEDs 116, 184, 198, 202 2 volt nominal

Resistors 118, 200 1 KΩ, k W

Transformer 164 120 volt input, 25.5 volt ouptut, 0.5 ampere secondary

Switch 168 SPST

Diodes 170, 172 1N4002

Capacitors 176, 188 10 μF, electrolytic

Capacitors 178, 181, 0.1 μF, disk 190, 192

Voltage regulator 174 National Semiconductor LM7815 or equivalent

Capacitor(s) 180, 194 470 μF, electrolytic Voltage regulator 186 National Semiconductor I_M7915 or equivalent

Switch 196 3PDT A further embodiment of an audio signal processor in accordance with the present invention is illustrated in Fig. 10. This version is intended for use in recording studios, broadcast stations and other professional environments. The embodiment of Fig. 10 is similar to that of Fig. 9, except that low-noise operational amplifiers 60"-64", 72"-76", 84"-88" and 98"-102" are used and the bypass switch 114, mode switch 196 and LEDs 116, 198 and 202 have been deleted. In addition, a pair of potentiometers 204 and 206 has been interposed in the crossover connections between the respective first stages of the right and left channels, and a similar pair of potentiometers 208, 210 has been interposed between the respective second stages of the right and left channels. The potentiometers 204, 206 and 208, 210 allow the amount of crossover or mixing between the right and left channels to be controlled in a continuous manner, rather than discretely as is the case when a switch is used. Moreover, by providing separate sets of potentiometers 204, 206 and 208, 210 as shown in Fig. 10, the amount of crossover or channel mixing can be controlled separately for each stage of the cascaded signal processing circuit. With appropriate selection of potentiometer values, the amount of crossover can be varied between essentially zero and 100%. If desired, the crossover potentiometers can be ganged by stage (i.e., 204 with 206 and 208 with 210) and/or by channel (i.e., 204 with 208 and 206 with 210).

The audio signal processing circuit of Fig. 10 also differs from that of Fig. 9 in that the feedback resistors 70', 82' associated with the operational amplifiers 62", 74" are implemented as potentiometers, rather than as fixed resistors. This provides the user with additional control over the amount of gain and signal processing that occurs in the first stage of each channel. The feedback resistors 70' and 82' may be ganged, if desired, or controlled individually as shown. The feedback resistors 96', 108' associated with the operational amplifiers 86", 100" are shown as being individually controlled in the circuit of Fig. 10, but may be ganged in the manner shown in Fig. 9 if desired. The final difference between the audio signal processor of Fig. 10 and that of Fig. 9 is that the input resistors 92, 106 to the inverting inputs of the operational amplifiers 86", 100" in Fig. 9 have been replaced by potentiometers 212, 214 in Fig. 10. Each of the potentiometers 212, 214 is placed in series with a fixed resistor 216, 218 in order to insure that some input resistance remains when the potentiometer is adjusted to its zero-resistance value. The use of the potentiometers 212, 214 provides the user with additional control over the amount of signal processing that occurs in the second stage of each channel of the audio signal processor in Fig. 10. By approximately varying the potentiometers 212 and 214 as well as the feedback potentiometers 70', 82', 96' and 108', the audio sound field can be manipulated in order to move the sound image in space. This is accomplished by phase additions and subtractions of the spatially separated harmonic and subharmonic frequencies. Preferred values for the components used in the embodiment of Fig. 10 are provided in Table 5 below. These values are provided merely by way of example and are not intended to limit the scope of the present invention.

TABLE 5 (Continued)

Component Value or Type

Capacitors 71, 83, 97, 0.1 μF, Mylar 109

Operational amplifiers Analog Devices AD713 60^«-64", 72"-76" or equivalent 84"-88", 98"-102"

Transformer 164 120 volt input, 25.5 volt output, 0.5 ampere secondary

Switch 168 SPST Diodes 170, 172 1N4002

Capacitors 176, 188 10 μF, electrolytic

Capacitors 178, 181, 0.1 μF, disk 190, 192

Voltage regulator 174 National Semiconductor LM7815 or equivalent

Capacitors 180, 194 470 μF, electrolytic

LED 184 2 volt nominal

Voltage regulator 186 National Semiconductor IM7915 or equivalent

Potentiometers 204, 206, 0-100 KΩ, h W 208, 210

Potentiometers 212, 214 0-10 KΩ, k W

Resistors 216, 218 4 KΩ, h W

Figs. 11A-11D illustrate several ways in which an audio signal processor of the type contemplated in the present invention can be used in connection with an existing audio system. In Fig. 11A, the preampli'fied line outputs of an audio source 220 such as a compact disk player, a digital or analog tape deck, a tuner, or a video cassette recorder with audio outputs, are connected to the left and right inputs of the signal processor 222. The signal processor 222 may comprise any one of the embodiments shown in Figs. 3 or 7-10, described previously. The left and right channel outputs of the signal processor 222 are connected to the corresponding inputs of an audio amplifier or receiver 224. The left and right channel outputs of the audio amplifier or receiver 224 are connected in a conventional manner to a pair of high-fidelity loudspeakers (not shown) . Fig. 11B illustrates an alternative arrangement in which the signal processor 222 is connected in the tape monitor loop of the audio amplifier or receiver 224 to allow for processing of the audio signal from any source that is connected to the amplifier or receiver 224. In a similar manner, the signal processor 222 can be installed in the processor loop of a recording studio mixer. The signal processor can also be installed on the input side of a broadband limiter at an FM stereo radio station.

Figs, lie and 11D illustrate two possible ways in which the signal processor 222 may be used in connection with an automobile audio system. In Fig. 11C, the preamplified line outputs from an automobile radio 226 are applied to the inputs of the signal processor 222, which is preferably of the type described previously in connection with Fig. 7. The outputs of the signal processor 222 are applied to the line inputs of an audio booster 228. The outputs of the audio booster 228 are connected to the left and right speakers (not shown) installed in the automobile. In Fig. 11D, the amplified speaker outputs of the automobile radio 226 are applied to the audio signal processor 222 through attenuating resistors 230 and 232. The outputs of the signal processor 222 are connected to the line inputs of the audio booster 228, which drives the left and right speakers of the automobile in the same manner as in Fig.

Figs. 12A and 12B depict oscilloscope traces generated by audio signals at the input and output of the audio signal processor illustrated in Fig. 9. These traces were obtained with the mode switch 196 in the position opposite to that shown in Fig. 9 (i.e., with no crossover between the first signal processing stages of the left and right channels) , and with the potentiometers 96' and 108' set at 10 KΩ. The input audio signal was a musical program reproduced on a stereo compact disk (CD) player, and only one output channel of the CD player was used. Fig. 12A illustrates the unprocessed voltage output of the CD player versus time at the input of the signal processor of Fig. 9, and Fig. 12B illustrates the processed output from the signal processor of Fig. 9 during the same time interval. It can be seen that the output signal in Fig. 12B is a more complex waveform than the input signal shown in Fig. 12A. This is a result of the harmonic generation and phase shifting that is carried out by the audio signal processor of Fig. 9.

Figs. 13A and 13B depict x-y oscilloscope traces comparing the input and output of the audio signal processor of Fig. 9. The processor settings were the same as those used in Figs. 13A and 13B, and the same CD player and musical program were employed. In Fig. 13A, the switch 114 of the audio signal processor was set to the bypass position, eliminating any signal processing. The resulting trace on the oscilloscope is a straight line, indicating that there is essentially no phase difference between the audio signals at the input and output of the signal processor. In Fig. 13B, the bypass switch 114 was moved to the opposite position to allow processing of the audio signal. The resulting waveform is a complex Lissajous pattern whose shape continually varies in accordance with the changing phase relationship between the input and output signals. When the musical program consists of a horn or other instrument whose output approximates a pure tone, more conventional types of Lissajous figures are observed.

Figs. 14A and 14B depict spectrum analyzer traces illustrating the harmonic content of the signals present at the input and output of the audio signal processor of

Fig. 9. The same processor settings, CD player and audio program were used as in Figs. 12 and 13. In Fig. 14A, which shows the unprocessed audio spectrum at the input of the signal processor, a normal audio spectrum exists between 0 Hz and 20 KHz, and there is relatively little activity above 20 KHz. In the processed spectrum shown in

Fig. 14B, however, the audio spectrum between 0 Hz and

20 KHz is noticeably richer in harmonics, as indicated by the greater number of peaks and valleys. In addition, the audio bandwidth has been expanded well beyond 20 KHz, as indicated by the peaks and valleys in the area of the power spectrum between 20 KHz and 25 KHz.

Further testing has suggested that the audio signal processor of the present invention may expand the audio bandwidth far beyond the normal range of human hearing.

Harmonics have been observed out to 50 KHz and beyond, and there are indications that the signal processor may generate frequencies as high as 4 MHz. Although a 4 MHz signal is far outside the audible range, the possibility exists that a loudspeaker voice coil excited by a 4 MHz signal may generate electromagnetic waves that affect human auditory perception in a manner that is not yet understood. In view of the preceding description, it can be seen that the present invention produces a significant improvement in sound quality by recreating and spatially separating harmonic frequencies that would otherwise be lost during the transduction, recording and playback processes. In addition, the dimensions of depth and height are added to standard stereo signals, and the spatial smearing that is inherent in conventional loudspeaker designs is reduced.

The audio signal processing circuits shown in Figs. 3 and 7-10 can be manufactured as stand-alone devices that may be used in conjunction with existing types of audio systems. Alternatively, the audio signal processing circuits of Figs. 3 and 7-10 may be incorporated into stereophonic amplifiers, receivers or other types of audio components as an integral part thereof. Equalizers and tone controls are generally not needed for two-speaker systems when the present invention is employed, since the dimensional enhancement of the audio signal has been found to be substantially independent of the acoustics of the space. Although the present invention is of particular utility in enhancing the quality of recorded music and other types of recorded audio programs, the invention can also be used during live performances to enhance the sound of the audio signal after it has been transduced by microphones, and to correct for phase shifts caused by the distance of the listener from the stage or loudspeaker. In other words, the listener can effectively be brought closer to the performance, regardless of the listener's actual location. When the present invention is used to make an original recording, whether on a record, compact disk, audio tape, film, video sound track, laser disk, or computer disk, the recording can later be reproduced on conventional sound reproduction equipment while preserving much of the enhancement provided by the audio signal processor.

The present invention can be used in virtually any type of device or application in which audio recording, reproduction, or transmission is involved. Examples include stereophonic and monaural amplifiers and preamplifiers, automobile radios (including FM, AM and AM stereo) , boat radios, portable tape and compact disk players, portable stereo radios, electronic organs, synthesizers, theater sound systems, citizens band radios, walkie talkies, sound effects generators, electronic keyboards, electrically amplified orchestras and bands, radio and television stations, recording studios, 5 television sets, cellular and other types of telephones, video cassette recorders, radio receivers of all kinds, airline entertainment systems, military and non-military communication systems, public address systems, night clubs and discotheques, and background music for elevators,

]0 skating rinks, shopping malls, stores and so on.

Although the present invention has been described with reference to a number of preferred embodiments thereof, the invention is not limited to the details thereof. For example, although the operational amplifiers

]5 are preferably of the integrated solid state type, discrete solid state devices or even vacuum tubes can be used in higher power applications. In addition, large scale integration (LSI) techniques can be used to reduce most of the circuitry of the audio signal processor to a

20 single chip, or surface mount technology can be used to produce hybrid chips. These and other modifications are intended to fall within the spirit and scope of the present invention as defined in the appended claims.

Claims

WHAT IS CLAIMED IS:

1. An audio signal processor comprising: an input terminal for receiving an audio signal; a first signal processing stage coupled to said input for processing said audio signal by generating harmonic frequencies related to fundamental frequencies in said audio signal and phase shifting said fundamental and harmonic frequencies as a function of frequency, said phase shift increasing in a negative direction with increasing frequency to cause higher frequency signals to lag lower frequency signals; a second signal processing stage coupled to said first signal processing stage in a cascade configuration for further processing said audio signal by generating harmonic frequencies related to fundamental frequencies in said audio signal and phase shifting said fundamental and harmonic frequencies as a function of frequency, said phase shift increasing in a negative direction with increasing frequency to cause higher frequency signals to lag lower frequency signals; and an output terminal for coupling the processed audio signal to an output device.

2. An audio signal processor as claimed in claim 1, wherein each of said first and second signal processing stages comprises: a first unity gain stage for receiving said audio signal; a harmonic generation and phase shift stage coupled to said unity gain stage; and a second unity gain stage coupled to said harmonic generation and phase shift stage.

3. An audio signal processor as claimed in claim 2, wherein said first and second unity gain stages and said harmonic generation and phase shift stages each comprise an operational amplifier.

4. An audio signal processor as claimed in claim 3, wherein said harmonic generation and phase shift stage further comprises an RC circuit for controlling said phase shifting function.

5. An audio signal processor as claimed in claim 1, wherein said cascaded arrangement of first and second signal processing stages forms a first of two channels of a stereophonic audio signal processor, and further comprising a second cascaded arrangement of first and second signal processing stages forming the second channel of said stereophonic audio signal processor.

6. An audio signal processor as claimed in claim 5, further comprising means for coupling said first and second channels to provide audio signal mixing between said channels.

7. An audio signal processor as claimed in claim 6, wherein said coupling means comprises a hard-wired electrical connection.

8. An audio signal processor as claimed in claim 6, wherein said coupling means comprises a switch.

9. An audio signal processor as claimed in claim 6, wherein said coupling means comprises a resistance.

10. An audio signal processor as claimed in claim 9, wherein said resistance is variable.

11. An audio signal processor as claimed in claim 6, wherein said coupling means comprises: first coupling means for coupling the first stage of said first channel to the first stage of said second channel; and second coupling means for coupling the second stage of said first channel to the second stage of said second channel.

12. An audio signal processor as claimed in claim 11, wherein said first coupling means comprises a switch and said second coupling means comprises a hard-wired connection.

13. An audio signal processor as claimed in claim 11, wherein said first and second coupling means each comprise a variable resistance.

14. An audio signal processor as claimed in claim 8 or 12, further comprising a visual indicator for indicating the state of said switch.

15. An audio signal processor as claimed in claim 14, wherein said visual indicator comprises a light emitting device.

16. An audio signal processor as claimed in claim 1, further comprising means for bypassing at least one of said first and second signal processing stages.

17. An audio signal processor as claimed in claim 16, wherein said bypass means comprises a bypass switch for connecting said input terminal to said output terminal.

18. An audio signal processor as claimed in claim 17, further comprising a visual indicator for indicating the state of said switch.

19. An audio signal processor as claimed in claim 18, wherein said visual indicator comprises a light emitting device.

20. An audio signal processor as claimed in claim 3, wherein the operational amplifier in each of said harmonic generation and phase shift stages includes a feedback resistance, said feedback resistance being variable in at least one of said stages.

21. An audio signal processor as claimed in claim 3, wherein the operational amplifier in each of said harmonic generation and phase shift stages includes an input resistance, said input resistance being variable in at least one of said stages.

22. A method for carrying out two-stage processing of an input audio signal, said method comprising the steps of: generating harmonic frequencies related to fundamental frequencies in said input audio signal and adding said harmonic frequencies to said input audio signal; phase shifting said fundamental and harmonic frequencies as a function of frequency, said phase shift increasing in a negative direction with increasing frequency to cause higher frequency signals to lag lower frequency signals, said harmonic generating and phase shifting steps thereby producing a partially processed audio signal; generating harmonic frequencies related to fundamental frequencies in said partially processed audio signal and adding said harmonic frequencies to said partially processed audio signal; and phase shifting the fundamental and harmonic frequencies of the partially processed audio signal as a function of frequency, said phase shift increasing in a negative direction with increasing frequency to cause higher frequency signals to lag lower frequency signals.

23. A method as claimed in claim 22, wherein said method steps are carried out on the right and left channel output signals from a stereophonic audio source.

24. A method as claimed in claim 23, further comprising the steps of mixing at least a portion of said right channel signal with said left channel signal, and mixing at least a portion of said left channel signal with said right channel signal.

25. A method as claimed in claim 24, further comprising the step of varying the amount of mixing between said right and left channel signals.

26. A method as claimed in claim 24, wherein the mixing between said right and left channel signals occurs only during the initial harmonic generating and phase shifting steps carried out on said input audio signal and not during the subsequent harmonic generating and phase shifting steps carried out on said partially processed audio signal.

27. A method as claimed in claim 24, wherein the mixing between said right and left channels occurs during the initial harmonic generating and phase shifting steps carried out on the input audio signal and also during the subsequent harmonic generating and phase shifting steps carried out on said partially processed audio signal.

28. A method as claimed in claim 22, further comprising the step of controlling the amount of harmonic frequency generation and phase shifting that is carried out on at least one of said input audio signal and said partially processed audio signal.

29. A method as claimed in claim 22, wherein at least some of said generated harmonic frequencies lie within the frequency range of 0 Hz to 20 KHz.

30. A method as claimed in claim 22, wherein at least some of said generated harmonic frequencies are higher than 20 KHz.