TWI414147B - For high input voltage, high output current zero voltage switching converter - Google Patents

Abstract

The invention provides a zero-voltage switching converter for high input voltage and high output current, wherein its circuit architecture is a converter composed of a first converter module and a second converter module, which are connected by connecting input ends in series and connecting output ends in parallel, and sharing a resonance element. Since the input ends of the converter are connected in series to have an effect of sharing input voltages, it is suitable for the application of the high input voltage. The switch parasitic capacitor of the first converter module is utilized to generate resonance together with the resonance inductor. Two main switches are operated by interleaved signals having 180 degrees of working phase difference such that all main switches and auxiliary switches can achieve soft switching performance of the zero voltage switching to reduce switching loss and to increase electric power conversion efficiency. Moreover, the output ends of the converter are connected in parallel to have the effect of sharing output current so as to be suitable for the application of high output current. By incorporating with a current-double rectification technique of the first converter module and the second converter module, it is able to generate the effect of ripple cancellation for outputted inductance and current, thereby reducing ripple of output voltage. Accordingly, the converter of the invention is suitable for the application of high input voltage, high output current, high power and high efficiency.

Description

Zero voltage switching converter for high input voltage and high output current

The present invention relates to switched converters for zero voltage flexible switching techniques, and more particularly to applications suitable for high input voltage, high output current, high power, and high efficiency.

In recent years, semiconductor technology has flourished, and power converters for systems such as aerospace, communications, and computers all have high-current power supply specifications, and in response to the miniaturization of electronic products, power converters must have higher power densities. In addition, the demand for high current output will cause power loss and thermal stress of the magnetic component and the semiconductor component, and the output voltage ripple will increase, so that the filter demand must be increased, so the power converter unit must have operation. Performance at high output current and low output voltage chopping.

As the oil source is declining, the awareness of energy conservation is high, the US Environmental Protection Agency has developed the Energy Star, and the 80 PLUS specification is included in the standard, for the AC-DC switching power supply provided to the PC, regardless of the low load ( AC-DC power conversion efficiency must reach 80% under load (20%), medium load (50% load), or high load (100% load). ENERGY STAR is also working with the Computer Energy Conservation Climate Campaign to accelerate the adoption of energy-saving technologies and regulations. Since the 80 PLUS meets the trend of energy saving and environmental protection, the newly introduced switching converters almost all support 80. The PLUS specification is the main selling point, with the characteristics of energy saving and power saving, to obtain the recognition of the European and American consumer market. Therefore, designing high-efficiency power converters to meet the increasingly stringent power supply specifications is the trend of the times.

The relevant academic literature of the prior art is as follows:

[1] B. R. Lin and H. K. Chiang, “Analysis and Implementation of a Soft Switching Interleaved Forward Converter with Current Double Rectifier,” IET Electr. Power Appl., Vol. 1, No. 5, pp. 697-704, 2007.

[2] B. R. Lin, C.-L. Huang, “Analysis and Implementation of A Novel Soft-Switching Pulse-Width Modulation Converter,” IET Power Electronics, 2009, Vol. 2, No. 1, pp. 90-101

[3] T. Qian and B. Lehman, “Dual Interleaved Active-Clamp Forward With Automatic Charge Balance Regulation for High Input Voltage Application,” IEEE Trans. Power Electronics, Vol. 23, No. 1, pp. 38-44, 2008.

[4] T. Jin, K. Zhang, K. Zhang; K. Smedley, “A New Interleaved Series Input Parallel Output (ISIPO) Forward Converter With Inherent Demagnetizing Features,” IEEE Trans. Power Electronics, Vol. 23, No. 2, pp. 888-895, 2008.

The relevant academic documents and disadvantages of the prior art are as follows:

[1] Refer to the fifth figure to propose a parallel architecture interleaved forward converter with double current rectification and flexible switching. Two sets of active clamp forward converters have four switches, using clamp circuit and transformer primary side resonance. The circuit achieves zero voltage switching (ZVS) function to improve efficiency. The switching stress is Vin/1-D, where D is the power switch's duty ratio. When D=0.5, the switching stress is 2Vin, which is not suitable for high input voltage applications. .

[2] Referring to the sixth figure, an interleaved flexible-cut PWM converter with a three-voltage level on the primary side of the transformer and a switching stress of Vin or Vin/2 is proposed, but the circuit contains six power switches.

[3] Referring to Figure 7, a double-interleaved active clamp forward converter with automatic charge balance adjustment is suitable for high input voltage applications. It must use two sets of additional windings, so the transformer fabrication is complicated. In the four switches on the primary side, two high voltage stress switches are required, and two lower voltage stress switches. Since this converter lacks the design of the resonant inductor, zero voltage switching (ZVS) cannot be guaranteed. It is mentioned in the article that the leakage inductance of the transformer will cause voltage surge on the switch, resulting in high voltage stress of the switch. It is a disadvantage to additionally add a snubber circuit.

[4] Referring to the eighth figure, combined with the characteristics and advantages of the interleaved two-switch forward converter and half-bridge converter, an interleaved series input parallel output forward converter is proposed, which has the advantages of natural flux reset. Suitable for high Input voltage, high output current, and high power applications. The maximum voltage stress of the switch is Vin. However, the switch of this converter is hard switching and does not have flexible switching performance, so the large switching loss is a disadvantage.

Therefore, based on the background and motivation of the above research, the present invention will design a converter suitable for high input voltage, low voltage stress of power switching elements, high output current, output low voltage chopping, and high efficiency.

Therefore, in view of the shortcomings of the existing converter technology, the present invention aims to provide a converter suitable for high input voltage, high output current, high power and high efficiency, and the circuit structure is composed of the first converter module. The second converter module can be realized by connecting the input terminal in series and the output terminal in parallel, and using a converter composed of a resonant component, using the parasitic capacitance of the power switch and the resonant inductor, so that all the main switch and the auxiliary switch can reach The soft-cutting performance of zero-voltage switching reduces switching losses and improves power conversion efficiency. With the double-current rectification technology, it produces the effect of chopping cancellation of the output inductor current, which can reduce the chopping of the output voltage.

In order to achieve the above object, the present invention provides a zero-voltage switching converter suitable for a high input voltage and a high output current, comprising: an input terminal; a first converter module including a first input capacitor C _i1 , a first a main switch S _m1 , a first auxiliary switch S _a1 , a first switch body diode D _m1 and a second switch body diode D _a1 , a first switch parasitic capacitance C _m1 and a second switch parasitic a capacitor C _a1 , a first DC blocking capacitor C _B1 , a first transformer T ₁ , a first magnetizing inductance L _m1 , a first leakage inductance L _{l 1} , and a first output side rectifying diode D ₁₁ And a second output side rectifying diode D ₁₂ , a first output inductor L ₁₁ and a second output inductor L ₁₂ and a resonant inductor L _r , wherein the first main switch S _m1 and the first switch body 2 The pole body D _m1 and the first switch parasitic capacitance C _m1 are connected in parallel, and the first auxiliary switch S _a1 is connected in parallel with the second switch body diode D _a1 and the second switch parasitic capacitance C _a1 , the first input capacitor C _i1 end electrically connected to one end of the resonant inductor L _r, said first input capacitor C _i1 One end is electrically connected to the first end of the main switch S _m1 and said input end of said first main switch S _m1 is electrically connected to the other end of the first auxiliary switch S _a1 and the first end of DC blocking capacitor C _B1 one end of said first DC blocking capacitor C _B1 and the other end is electrically connected to the first end of the magnetizing inductance L _m1, L _m1 and the other end of the first magnetizing inductance electrically connected to the first end of _a leakage inductance _{L l,} the The other end of the first leakage inductance L _{l 1} is electrically connected to the other end of the resonant inductor L _r , and the first transformer T ₁ is connected in parallel with the first magnetizing inductance L _m1 , the first output side rectifying diode D ₁₁ of the female terminal based electrically connected to the first transformer T ₁ and the first end of the output side of the output end of the inductor L _11, L the cathode terminal ₁₂ of the second output lines electrically connected to the inductance of the first transformer T ₁ the other end of the output side of the inductor L ₁₂ and a second output end, the output of the first inductor L ₁₁ and the other end is electrically connected to the other end ₁₂ of the second output inductance L, the output side of the first rectifying diode D ₁₁ of The anode end is electrically connected to the anode end of the second output side rectifying diode D ₁₂ ; The converter module includes a second input capacitor C _i2 , a second main switch S _m2 , a second auxiliary switch S _a2 , a third switch body diode D _m2 and a fourth switch body diode D _A2 , a third switch parasitic capacitance C _m2 and a fourth switch parasitic capacitance C _a2 , a second DC blocking capacitor C _B2 , a second transformer T ₂ , a second magnetizing inductance L _m2 , a second leakage inductance L _{l 2,} a third output-side rectifier diode D ₂₁ and a fourth output side of the rectifying diode D _22, a third output inductor L ₂₁ and a fourth output inductor L ₂₂ and the resonant inductor L _r, The second main switch S _m2 is connected in parallel with the third switch body diode D _m2 and the third switch parasitic capacitance C _m2 , the second auxiliary switch S _a2 and the fourth switch body diode D _a2 , and the fourth The switching parasitic capacitance C _a2 is connected in parallel. One end of the second input capacitor C _{i2 is} electrically connected to one end of the resonant inductor L _{r and} one end of the first input capacitor C _i1 , and the other end of the second input capacitor C _i2 is electrically connected. the other end of the second main switch S _m2 and said input end of said second main switch and the other end is electrically S _m2 Connected to the end of the second auxiliary switch S _a2 and the second end of DC blocking capacitor C _B2, the second DC blocking capacitor C _B2 and the other end electrically connected to the second end of the magnetizing inductance L _M2, the second magnetizing inductance L the other end is electrically connected to the second _m2 leakage inductance _{L l 2} at one end, said second leakage inductance _{L l 2} and the other end is electrically connected to the other end of the resonant inductor L _r and the leakage inductance _{L l} of said first end phase ₁ Electrically connected, the second transformer T ₂ is connected in parallel with the second magnetizing inductance L _m2 , and the cathode end of the third output side rectifying diode D ₂₁ is electrically connected to the output of the second transformer T ₂ . a side end and a third output inductor L ₂₁ end, the cathode end of the fourth output side rectifying diode D ₂₂ is electrically connected to the other end of the output side of the second transformer T ₂ and the fourth output inductor L ₂₂ end, The anode end of the third output side rectifying diode D ₂₁ is electrically connected to the anode end of the fourth output side rectifying diode D ₂₂ and connected to one end of the output load R, and the third output inductor L ₂₁ The other end is electrically connected to the other end of the fourth output inductor L ₂₂ and the first output inductor L The other end _11, an output capacitor C _o one end and the other end of the output load R, the output capacitor C _o is electrically connected to the other end ₁₁ of the male terminal D and the output side of the second output side of the first rectifier diode rectifying diodes D anode end of the body _12.

The first converter module and the second converter module described above are combined with a resonant inductor L _r .

The first input capacitor C _i1 and the second input capacitor C _i2 are voltage dividing capacitors, and the input voltage input to the input terminal is V _in /2.

The third switching parasitic capacitance C _m2 and the fourth switching parasitic capacitance C _{a2 described above} are used as resonant capacitors.

The first switching parasitic capacitance C _m1 and the second switching parasitic capacitance C _{a1 described above} are used as resonant capacitors.

From the above, the circuit architecture and the implementation description of the present invention show that the following advantages are obtained compared with the prior art circuit architecture:

1. The converter of the present invention has the characteristics of zero voltage switching performance (ZVS), reduces switching loss, and improves power conversion efficiency.

2. The circuit series input architecture of the present invention has a voltage sharing function, and when the input voltage is high, the switch has a lower voltage stress.

3. The circuit parallel output architecture of the present invention has a current sharing function and is suitable for high output current applications.

4. The present invention utilizes interleaved (PWM) operation and current doubler rectifier characteristics to have current chopping cancellation, which can reduce the chopping of the output voltage.

Therefore, the innovative circuit topology of the present invention is indeed novel and progressive compared to the circuit architecture of the prior art.

First, referring to the first figure, the present invention is a zero-voltage switching converter suitable for high input voltage and high output current, comprising: an input terminal; a first converter module (11), including The first input capacitor C _i1 , a first main switch S _m1 , a first auxiliary switch S _a1 , a first switch body diode D _m1 and a second switch body diode D _a1 , a first switch parasitic a capacitor C _m1 and a second switching parasitic capacitance C _a1 , a first DC blocking capacitor C _B1 , a first transformer T ₁ , a first magnetizing inductance L _m1 , a first leakage inductance L _{l 1} , a first An output side rectifying diode D ₁₁ and a second output side rectifying diode D ₁₂ , a first output inductor L ₁₁ and a second output inductor L ₁₂ and a resonant inductor L _r , wherein the first main switch S _m1 is connected in parallel with the first switch body diode D _m1 and the first switch parasitic capacitance C _m1 , and the first auxiliary switch S _a1 is connected in parallel with the second switch body diode D _a1 and the second switch parasitic capacitance C _a1 the first input end of capacitor C _i1 is electrically connected to the one end of the resonant inductor L _r, said first input The other end is electrically connected to the first capacitance C _i1 main switch S _m1 said input end and one end and the other end of said first main switch electrically connected to the S _m1 S _a1 one end of the first auxiliary switch and the first DC blocking capacitor C _B1 end, the other end of the first DC blocking capacitor C _B1 is electrically connected to the first end of the magnetizing inductance L _m1, L _m1 and the other end of the first magnetizing inductance electrically connected to the first leakage inductance _{L l The first} end of the first leakage inductance L _{l 1} is electrically connected to the other end of the resonant inductor L _r , and the first transformer T ₁ is connected in parallel with the first magnetizing inductance L _m1 , the first output side The cathode end of the rectifier diode D ₁₁ is electrically connected to one end of the output side of the first transformer T ₁ and one end of the first output inductor L ₁₁ , and the cathode end of the second output inductor L ₁₂ is electrically connected to the first end The other end of the output side of the transformer T ₁ and the second output inductor L ₁₂ , the other end of the first output inductor L ₁₁ is electrically connected to the other end of the second output inductor L ₁₂ , the first output side rectifying diode D electrically connected to the male member ₁₁ of the second output-side terminal based rectifier diode D the anode terminal ₁₂ A second converter module (12), includes a second input capacitance C _i2, a second main switch S _m2, a second auxiliary switch S _a2, a body diode of the third switch and a fourth D _m2 The switch body diode D _a2 , a third switch parasitic capacitance C _m2 and a fourth switch parasitic capacitance C _a2 , a second DC blocking capacitor C _B2 , a second transformer T ₂ , and a second magnetizing inductance L _m2 a second leakage inductance L _{l 2} , a third output side rectifying diode D ₂₁ and a fourth output side rectifying diode D _{22 ,} a third output inductor L ₂₁ and a fourth output inductor L ₂₂ and the resonant inductor L _r, wherein the second main switch and the third switch S _m2 body diode D _m2, a third switch connected in parallel with the parasitic capacitance C _m2, the second auxiliary switch S _a2 and two fourth switch body diode D _a2, a fourth switch connected in parallel with the parasitic capacitance C _a2, the second end of input capacitor C _i2 is electrically connected to one end of the resonant inductor L _r and the first end of input capacitor C _i1, the second input capacitor The other end of the C _i2 is electrically connected to one end of the second main switch S _m2 and the other end of the input end, and the second main switch S _The other end of the _m2 is electrically connected to one end of the second auxiliary switch S _{a2 and} one end of the second DC blocking capacitor C _B2 , and the other end of the second DC blocking capacitor C _B2 is electrically connected to one end of the second magnetizing inductance L _m2 , L _m2 and the other end electrically connected to the second magnetizing inductance leakage inductance _{L l 2} of the second end, the second leakage inductance _{L l 2} and the other end is electrically connected to the other end of the resonant inductor L _r and the first leakage inductance _One end of the L _{l 1} is electrically connected, the second transformer T ₂ is connected in parallel with the second magnetizing inductance L _m2 , and the cathode end of the third output side rectifying diode D ₂₁ is electrically connected to the second One end of the output side of the transformer T ₂ and one end of the third output inductor L ₂₁ , and the cathode end of the fourth output side rectifying diode D ₂₂ is electrically connected to the other end of the output side of the second transformer T ₂ and the fourth output The anode end of the third output side rectifying diode D ₂₁ is electrically connected to the anode end of the fourth output side rectifying diode D ₂₂ and is connected to one end of the output load R, the first end of the inductor L ₂₂ L ₂₁ and the other end three output inductor ₂₂ are electrically connected to the other end of the fourth output inductor L, the first L ₁₁ and the other end of the output inductor, an output capacitor C _o one end and the other end of the output load R, the other end of the output capacitor C _o is electrically connected to the anode terminal D ₁₁ and a second output of the first output-side rectifier diode side Anode end of rectifier diode D ₁₂

The present invention has the following advantages:

(1). Input series: In the case of high input voltage, the voltage withstand voltage and cost consideration of the component, and the need to reduce the withstand voltage of the switch are the key points of the converter circuit topology design. Since the input end of the converter uses two input capacitors [first input capacitor Ci1, second input capacitor Ci2] and is connected in two series, it has the function of sharing the input voltage Vin of the input terminal, so that each group of conversions The voltage sources of the [first converter module (11) and the second converter module (12)] are respectively Vin/2, which can reduce the switching withstand voltage requirement. After analysis, the converter's switching voltage stress is only Vin/2, so the metal oxide half field effect transistor (MOSEFT) with lower rated voltage can be selected, which has a smaller on-resistance R _DS (ON), which can reduce the conduction loss. Improve efficiency.

(2). Output Parallel: In high current output applications, a single converter often causes power loss and thermal stress problems of the magnetic and semiconductor components. The output of the converter is connected in two parallel modes, which share the output current and can disperse the power loss and thermal stress of the magnetic component transformer, the output inductor and the semiconductor component, and is applied to high output current.

(3). Interleaved operation: When the high current output, the output voltage ripple is easy to increase, the converter uses two sets of interleaved operation, plus the characteristics of the double current rectifier, so that the output inductor current is chopped in reverse. Reduce the output capacitor current chopping, greatly reduce the volume of the output filter capacitor, and reduce the output voltage ripple.

(4). Zero voltage switching: using the switching parasitic capacitance of the converter and the resonant inductor (including the transformer leakage inductance) to generate resonance, so that the converter has zero voltage switching (ZVS) function, reducing switching loss; DC blocking capacitors can also recover leakage inductance energy, avoid voltage surge caused by leakage inductance, and protect switching components from high voltage stress.

It can be seen from the above description that the converter of the present invention is suitable for applications with high input voltage, high output current, high power and high efficiency.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT A set of specifications is: a converter with an input voltage of 400V, an output voltage of 12V, an output power of 240W, and a switching frequency of 100 kHz to verify the correctness of the theoretical analysis and the converter. Excellent characteristics, and design of the voltage regulator controller, reduce the influence of input voltage and load variation on the output voltage, achieve steady state requirements and fast transient response.

Please refer to the driving signal and the crossover voltage of the first main switch Sm1 and the second auxiliary switch Sa2 when the waveform of the second A waveform is 240 W, and it can be seen that before the first main switch Sm1 and the second auxiliary switch Sa2 turn on, the switch The crossover voltage has dropped to zero to achieve zero voltage switching performance (ZVS) operation.

Please refer to the waveform of the second B diagram for the driving signal and the voltage across the second main switch Sm2 and the first auxiliary switch Sa1 to achieve zero voltage switching performance (ZVS) operation.

Please refer to the second A picture and the second B picture. It can be seen that when the input voltage Vin=400V, the voltage stress of the switch is only Vin/2=200V (half of the input voltage), and the voltage stress of the switch is Vin/1-D or Vin, therefore the converter of the invention does have the advantage of low voltage stress.

Please refer to the waveform in the third figure. The inductor currents iL ₁₁ and iL _{12 of the} double current rectifier at the load of 240 W [ie, the inductor current flowing through the first output inductor L ₁₁ and the second output inductor L _{12 respectively} ] are inverted. Therefore, the output current i _o1 =iL ₁₁ +iL ₁₂ can cancel the current ripple and reduce the chop size.

Please refer to the waveform of the fourth figure to illustrate the interleaved operation of the main switch, so that the total output current io=io ₁ + io ₂ can cancel the oscillating of io ₁ and io ₂ again, and reduce the current chopping of the output capacitor. And the two sets of converters do share half of the total output current (10A).

However, the above description is only one of the preferred embodiments of the present invention, and the scope of the patent application and the content of the specification according to the present invention are simply equivalently changed and replaced. All should still fall within the scope of protection covered by the scope of the patent application of the present invention.

(1)‧‧‧ converter

(11)‧‧‧First converter module

(12)‧‧‧Second converter module

The first figure is a circuit architecture diagram of the present invention.

The second A diagram is a zero voltage switching (ZVS) waveform diagram of the first main switch Sm1 and the second auxiliary switch Sa2 of the present invention.

The second B diagram is a zero voltage switching (ZVS) waveform diagram of the second main switch Sm2 and the first auxiliary switch Sa1 of the present invention.

The third figure is a chopping cancellation performance diagram of the inductor current of the current doubler rectifier of the present invention.

The fourth figure is the interlaced operational current chopping cancellation performance diagram of the present invention.

The fifth figure is a circuit diagram of the related academic literature of the prior art.

The sixth drawing is a circuit diagram of the related academic literature of the prior art.

The seventh figure is a circuit diagram of the related academic literature of the prior art.

The eighth figure is a circuit diagram of the related academic literature of the prior art.

(1)‧‧‧ converter

(11)‧‧‧First converter module

(12)‧‧‧Second converter module

Claims (5)

A zero-voltage switching converter suitable for high input voltage and high output current comprises: an input terminal; a first converter module comprising a first input capacitor C _i1 , a first main switch S _m1 , a first An auxiliary switch S _a1 , a first switch body diode D _m1 and a second switch body diode D _a1 , a first switch parasitic capacitance C _m1 and a second switch parasitic capacitance C _a1 , a first straight a flow blocking capacitor C _B1 , a first transformer T ₁ , a first magnetizing inductance L _m1 , a first leakage inductance L _{l 1} , a first output side rectifying diode D ₁₁ and a second output side rectifying diode a body D ₁₂ , a first output inductor L ₁₁ and a second output inductor L ₁₂ and a resonant inductor L _r , wherein the first main switch S _m1 and the first switch body diode D _m1 , the first switch parasit The capacitors C _m1 are connected in parallel, the first auxiliary switch S _a1 is connected in parallel with the second switch body diode D _a1 and the second switch parasitic capacitance C _a1 , and one end of the first input capacitor C _{i1 is} electrically connected to the resonant inductor L _{The other} end of the first input capacitor C _i1 is electrically connected to the first main switch S _m1 And at one end of the input end, the other end of the first main switch S _m1 is electrically connected to one end of the first auxiliary switch S _{a1 and} one end of the first DC blocking capacitor C _B1 , the first DC blocking capacitor C _B1 and the other end is electrically connected to the first end of the magnetizing inductance L _m1, the first magnetizing inductance L _m1 and the other end is electrically connected to the first end of the leakage inductance _{L l 1,} the first leakage inductance _{L l 1} and the other end is electrically Connected to the other end of the resonant inductor L _r , the first transformer T ₁ is connected in parallel with the first magnetizing inductance L _m1 , and the cathode end of the first output side rectifying diode D ₁₁ is electrically connected to the first An output side of the transformer T ₁ and an end of the first output inductor L ₁₁ , and a cathode end of the second output inductor L ₁₂ is electrically connected to the other end of the output side of the first transformer T ₁ and the second output inductor L ₁₂ The other end of the first output inductor L ₁₁ is electrically connected to the other end of the second output inductor L ₁₂ , and the anode end of the first output side rectifying diode D ₁₁ is electrically connected to the second output An anode terminal of the side rectifying diode D ₁₂ ; a second converter module including a second input power a capacitor C _i2 , a second main switch S _m2 , a second auxiliary switch S _a2 , a third switch body diode D _m2 and a fourth switch body diode D _a2 , a third switch parasitic capacitance C _m2 And a fourth switching parasitic capacitance C _a2 , a second DC blocking capacitor C _B2 , a second transformer T ₂ , a second magnetizing inductance L _m2 , a second leakage inductance L _{l 2} , a third output side rectification a body D ₂₁ and a fourth output side rectifying diode D ₂₂ , a third output inductor L ₂₁ and a fourth output inductor L ₂₂ and the resonant inductor L _r , wherein the second main switch S _m2 and The third switch body diode D _m2 and the third switch parasitic capacitance C _m2 are connected in parallel, and the second auxiliary switch S _a2 is connected in parallel with the fourth switch body diode D _a2 and the fourth switch parasitic capacitance C _a2 . One end of the second input capacitor C _{i2 is} electrically connected to one end of the resonant inductor L _{r and} one end of the first input capacitor C _i1 , and the other end of the second input capacitor C _i2 is electrically connected to one end of the second main switch S _m2 The other end of the second main switch S _m2 is electrically connected to one end of the second auxiliary switch S _a2 and the other end The second end of the DC blocking capacitor C _B2, the other end of said second DC blocking capacitor C _B2 is electrically connected to the second end of the magnetizing inductance L _m2, L _m2 the other end is electrically connected to the second second magnetizing inductance leakage inductance _{L l 2} at one end, said second leakage inductance _{L l 2} and the other end is electrically connected to the other end of the resonant inductor L _r and electrically connected to the first end of the leakage inductance _{L l 1,} the second transformer T ₂ In parallel with the second magnetizing inductance L _m2 , the cathode end of the third output side rectifying diode D ₂₁ is electrically connected to one end of the output side of the second transformer T ₂ and one end of the third output inductor L ₂₁ . The cathode end of the fourth output side rectifying diode D ₂₂ is electrically connected to the other end of the output side of the second transformer T _{2 and} the end of the fourth output inductor L ₂₂ , and the third output side rectifying diode D the anode terminal ₂₁ based anode terminal electrically connected to the output side of the fourth rectifying diode D ₂₂ and the load R connected to an output end, L ₂₁ and the other end of the output of the third inductor are electrically connected to the fourth output inductor The other end of L _22, L ₁₁ and the other end of the first output inductor, an output capacitor C _o and the output end The other end of the load R, the other end of the output capacitor C _o is electrically connected to the anode terminal and the anode terminal D ₁₁ of the output side of the second rectifier rectifying output side of the first diode of the diode D _12. The high voltage input, high output current zero voltage switching converter according to claim 1, wherein the first converter module and the second converter module are combined with a resonant inductor L _r . The zero-voltage switching converter with high input voltage and high output current as described in claim 1, wherein the first input capacitor C _i1 and the second input capacitor C _i2 are voltage dividing capacitors, so that the input terminal The input voltage input is V _in /2. The zero-voltage switching converter for high input voltage and high output current as described in claim 1, wherein the third switching parasitic capacitance C _m2 and the fourth switching parasitic capacitance C _a2 are used as resonant capacitors. The zero-voltage switching converter for high input voltage and high output current as described in claim 1, wherein the first switching parasitic capacitance C _m1 and the second switching parasitic capacitance C _a1 are used as resonant capacitors.