TW202425513A - Control device for power conversion apparatus - Google Patents

Abstract

A control device for a power conversion apparatus, includes the following elements. A driving circuit is connected to a control terminal of a power module, for controlling a voltage of the control terminal of the power module. A Miller switch transistor provides a bypass transmission path to conduct a Miller current generated when switching of a power semiconductor element of the power module. A Miller switch control circuit controls the Miller switch transistor to be turned-on or turned-off, and conducts the Miller current through the Miller switch transistor when the Miller switch transistor is turned-on. When the power module operates at a normal negative voltage, the adjustable power source controls a voltage difference between the control terminal and a second terminal of the power module.

Description

Control devices for power conversion equipment

The present disclosure relates to a control device for a power conversion device, and more particularly to a control device for suppressing the Miller effect of a power module in the power conversion device.

Power conversion equipment can be used as a conversion medium between power supply (grid) and renewable energy (or load). In response to power supply equipment or load equipment with different power specifications, power conversion equipment can convert power into specific power and specific forms of electricity. The power module in the power conversion equipment includes power semiconductor components, which have the function of switching switches. In order to improve the performance of power conversion equipment, the power module must have a higher switching speed. In current technology, power semiconductor components of different materials have been introduced and set in the power module of the power conversion equipment. For example, power semiconductor components of silicon carbide (SiC) are introduced to adapt to equipment with power specifications of 1kW to 10MW. Power semiconductor components made of silicon carbide material can operate at a switching speed of 10kHz~100kHz. For example, power semiconductor devices using gallium nitride (GaN) are suitable for devices with power specifications ranging from 100W to 10kW. Power semiconductor devices made of gallium nitride material can operate at a higher switching speed of 100kHz~10MHz.

Although the power module of the power conversion device uses power semiconductor components made of specific materials to achieve a higher switching speed, the parasitic effects of the power semiconductor components may reduce the performance of the power conversion device. Among them, the more serious parasitic effects are caused by the parasitic capacitance between the conductors inside the component. For example, the parasitic capacitance of the metal oxide semiconductor transistor (i.e., MOS transistor) inside the power semiconductor component includes: the gate-source parasitic capacitance Cgs, the gate-drain parasitic capacitance Cgd (or "Miller capacitance") and the drain-source parasitic capacitance Cds of the transistor. The parasitic effect caused by the Miller capacitance Cgd is called the "Miller effect". When the transistor is driven, the gate-source parasitic capacitance Cgs of the transistor reaches the critical voltage Vth of the transistor, and the transistor enters the turned-on state, then the current value of the drain-source current Ids increases, and the transistor enters the saturation region. Due to the Miller effect, the gate-source voltage Vgs of the transistor remains at a constant voltage value within a time period (this time period is called the "Miller platform") and no longer increases. When the drain-source current Ids reaches the maximum current value, and the drain-source voltage Vds begins to drop, the Miller capacitance Cgd continues to be charged, and the gate-source voltage Vgs rises to the voltage value of the driving voltage. At this time, the transistor enters the resistance region (triode region), and the drain-source voltage Vds drops to a lower voltage value, completing the conduction of the power semiconductor element. From the above, the duration of the Miller effect (i.e., the length of the time interval of the Miller platform) will increase the conduction time of the power semiconductor element, thereby increasing the switching loss of the power semiconductor element and reducing the conversion efficiency of the power conversion equipment.

Furthermore, when the power conversion device has a higher switching speed and the maximum rated value of the positive gate voltage/negative gate voltage of the transistor is different, the Miller effect between the gate and source of the transistor may deteriorate, and the amplitude of the glitch voltage in the gate-source voltage Vgs will also increase. Please refer to FIG. 1A, which shows a circuit diagram of a portion of a power module 610 in a known power conversion device. The power module 610 includes a power semiconductor element of a half-bridge topology, and the power module 610 can perform a switching switch to provide a power conversion function. In the half-bridge topology, the power module 610 includes complementary upper and lower arm components, the upper arm component includes a transistor Q11, and the lower arm component includes a transistor Q12. The transistor Q11 has a parasitic capacitor Cgd11 between the gate g11 and the drain d11, a parasitic capacitor Cgs11 between the gate g11 and the source s11, and a parasitic capacitor Cds11 between the drain d11 and the source s11. Similarly, there is a parasitic capacitance Cgd12 between the gate g12 and the drain d12 of the transistor Q12, a parasitic capacitance Cgs12 between the gate g12 and the source s12, and a parasitic capacitance Cds12 between the drain d12 and the source s12. During the switching period of the gate-source voltage Vgs11 of the transistor Q11 of the upper arm element, the Miller capacitance Cgd12 of the transistor Q12 of the lower arm element injects current into the gate, causing a voltage drop on the impedance of the gate path, causing a glitch voltage to appear on the gate-source voltage Vgs12 of the transistor Q12.

Please refer to FIG. 1B, which shows a timing diagram of the voltage change of the gate-source voltage Vgs11 of the transistor Q11 and the gate-source voltage Vgs12 of the transistor Q12 of the power module 610 of FIG. 1A. The normal positive voltage of the gate-source voltage Vgs11 of the transistor Q11 is +15V, and its maximum rated value is +20V, and the normal negative voltage of the gate-source voltage Vgs11 is -15V, and its minimum rated value is -20V. The normal positive/negative voltage and the maximum/minimum rated values of the gate-source voltage Vgs12 of the transistor Q12 are also the same. At time points t11, t12, t13, and t14, it can be observed that the Miller effect causes a glitch voltage to appear in the gate-source voltage Vgs12 of the transistor Q12. The gate-source voltage Vgs12 of the transistor Q12 may exceed the minimum rated value after the glitch voltage is superimposed. For example, at time points t12 and t14, the gate-source voltage Vgs12 reaches a voltage value of -20V or lower after the glitch voltage is superimposed, exceeding the minimum rated value of -20V. Furthermore, when the gate-source voltage Vgs12 plus the glitch voltage exceeds the critical voltage Vth12 of transistor Q12, transistor Q12 will be turned on. At this time, transistor Q11 of the upper arm element and transistor Q12 of the lower arm element are turned on at the same time, causing a short circuit in the power module 610 and the risk of circuit element breakdown.

In response to the above technical problems caused by the Miller effect, technicians in this technical field are committed to improving the circuit design of power conversion equipment to effectively suppress the Miller effect and accurately control the gate voltage of power semiconductor devices.

According to one aspect of the present disclosure, a control device for a power conversion device is provided. The control device includes the following elements. A power module having a first end, a second end and a control end, the power module including a power semiconductor element. A driving circuit connected to the control end of the power module and used to control the voltage of the control end of the power module. A Miller switching transistor connected to the control end of the power module, the Miller switching transistor providing a bypass transmission path to conduct the Miller current generated when the power semiconductor element is switched. The Miller switch control circuit is connected to the gate of the Miller switch transistor and the drain of the Miller switch transistor. The Miller switch control circuit controls the Miller switch transistor to be in an on or off state. When the Miller switch transistor is in an on state, the Miller switch transistor conducts the Miller current. The adjustable power supply is connected to the second end of the power module. When the power module operates at a normal negative voltage, the adjustable power supply controls the voltage difference between the control end and the second end of the power module.

Other aspects and advantages of the present disclosure will become apparent by reading the following drawings, detailed description, and claims.

The technical terms in this specification refer to the customary terms in this technical field. If this specification explains or defines some terms, the interpretation of these terms shall be subject to the explanation or definition in this specification. Each embodiment of this disclosure has one or more technical features. Under the premise of possible implementation, a person with ordinary knowledge in this technical field can selectively implement part or all of the technical features in any embodiment, or selectively combine part or all of the technical features in these embodiments.

Please refer to FIG. 2, which shows a circuit diagram of a control device 1000a in a power conversion device of a comparative example. The control device 1000a includes a control chip 400, a driving circuit 500, and a power module 600. The power module 600 acts as a switching switch, and the power semiconductor element in the power module 600 is a transistor Q5 (i.e., the transistor Q5 acts as a power switching transistor). The Miller current Im1 caused by the gate-drain parasitic capacitance Cgd5 (i.e., the Miller capacitance) of the transistor Q5 may flow into the driving circuit 500. The Miller current Im1 flows through the transistor Q2 of the driving circuit 500. In this comparative example, in order to suppress the Miller current Im1 from flowing into the driving circuit 500, a transistor Q6 and a logic circuit 10 are added to the control device 1000a. The transistor Q6 acts as a Miller switch transistor, and the logic circuit 10 controls the operation of the transistor Q6, so that the transistor Q6 acts as a "Miller switch" (or "Miller clamp switch") to suppress the Miller effect. The transistor Q6 is connected between the gate g5 and the source s5 of the transistor Q5 of the power module 600, and is connected to the source of the transistor Q2 of the driving circuit 500. Transistor Q6 can provide a bypass transmission path for Miller current to prevent Miller current Im1 from directly flowing into transistor Q2 of driving circuit 500.

However, in the comparative example of FIG. 2, the Miller clamping voltage of the Miller switch (i.e., transistor Q6) cannot be accurately controlled, nor can the driving time of the Miller switch be accurately controlled. In view of this, the present disclosure proposes an improved control mechanism for the Miller switch, which is intended to accurately control the Miller clamping voltage of the Miller switch and the driving time of the Miller switch.

FIG. 3 is a circuit diagram of a control device 1000b of an embodiment of the present disclosure in a power conversion device. The control device 1000b of the embodiment of FIG. 3 can accurately control the Miller clamping voltage of the Miller switch and the driving time of the Miller switch. The control device 1000b includes a Miller switch control circuit 100, an adjustable power supply 200, a control chip 400, a driving circuit 500 and a power module 600. The control chip 400 controls the driving circuit 500 to drive the power module 600. The control circuit Miller switch control circuit 100 and the adjustable power supply 200 accurately control the Miller clamping voltage of the Miller switch (i.e., transistor Q6) and the driving time of the Miller switch.

The electrical connection method of the Miller switch control circuit 100, the adjustable power supply 200, the control chip 400, the driver circuit 500 and the power module 600 is described as follows. The driver circuit 500 includes a transistor Q1 and a transistor Q2. The transistor Q1 and the transistor Q2 form an inverter. The gates of the transistors Q1 and Q2 are connected to the input terminal 51 of the driver circuit 500, and the drains of the transistors Q1 and Q2 are connected to the output terminal 52 of the driver circuit 500. The input terminal 51 of the driver circuit 500 is connected to the input terminal 11 of the Miller switch control circuit 100 and the output terminal 41 of the control chip 400. The output terminal 52 of the driving circuit 500 is connected to the control terminal 63 of the power module 600 via the resistor _Rg. In addition, the high potential terminal 53 of the driving circuit 500 (i.e., the source of the transistor Q1) is connected to the peripheral circuit 350, which is, for example, a protection circuit with a short circuit protection function. The low potential terminal 54 of the driving circuit 500 (i.e., the source of the transistor Q2) is connected to the adjustable power supply 200.

The power module 600 includes a power semiconductor element as a power switch having a switching function. When the power conversion device provides power to devices with different power specifications, the power module 600 performs the switch in response to the different powers. When switching, the power module 600 is in a turned-on state or a turned-off state in response to the voltage received by the control terminal 63. When the power module 600 is turned on, the first terminal 61 of the power module 600 can be connected to the second terminal 62. The power semiconductor element of the power module 600 is, for example, a transistor (not shown in FIG. 3 ), the drain of the transistor is connected to the first terminal 61, the source of the transistor is connected to the second terminal 62, and the gate of the transistor is connected to the control terminal 63. The first terminal 61 of the power module 600 can be connected to the peripheral circuit 350. The second terminal 62 of the power module 600 is connected to the output terminal 22 of the adjustable power supply 200. The control terminal 63 of the power module 600 is connected to the output terminal 52 of the driving circuit 500 via the resistor _Rg , and the control terminal 63 is connected to the Miller switch control circuit 100 via the transistor Q6.

Transistor Q6 acts as a Miller switch, which provides a bypass transmission path for Miller current Im1 of the transistor of power module 600, and Miller current Im1 is conducted to Miller switch control circuit 100 via transistor Q6. The gate of transistor Q6 is connected to output terminal 12 of Miller switch control circuit 100, and the source of transistor Q6 is connected to output terminal 52 of driving circuit 500 via resistor Rg. In addition, the source of transistor Q6 is connected to control terminal 63 of power module 600 to be connected to the gate of the transistor of power module 600. Output terminal 42 of control chip 400 is connected to first terminal 61 of power module 600 via peripheral circuit 350.

FIG. 4 is a detailed circuit diagram of the Miller switch control circuit 100, the adjustable power supply 200, the drive circuit 500 and the power module 600 in the control device 1000b of FIG. 3. As shown in FIG. 4, the semiconductor power element of the power module 600 includes a transistor Q5. The gate g5, the drain d5 and the source s5 of the transistor Q5 are respectively connected to the control terminal 63, the first terminal 61 and the second terminal 62 of the power module 600. The gate g5 and the drain d5 of the transistor Q5 have a parasitic capacitance Cgd, the drain d5 and the source s5 have a parasitic capacitance Cds, and the gate g5 and the source s5 have a parasitic capacitance Cgs.

The adjustable power supply 200 includes a resistor R1, a capacitor C1 and a diode ZD. The resistor R1 is connected in parallel to the capacitor C1. The first end of the resistor R1 and the first end of the capacitor C1 are connected to the cathode of the diode ZD. The second end of the resistor R1 and the second end of the capacitor C1 are connected to the ground terminal GND1, so the potential of the second end of the resistor R1 and the second end of the capacitor C1 is the ground voltage. The anode of the diode ZD receives a first constant voltage V1-. The first constant voltage V1- is lower than the ground voltage of the ground terminal GND1, and the first constant voltage V1- is a negative voltage value. The diode ZD is, for example, a zener diode, and the current flowing through the diode ZD is the current i_ZD. The first end of the resistor R1, the first end of the capacitor C1 and the cathode of the diode ZD are connected to the low potential end 54 of the driving circuit 500. The potential of the first end of the resistor R1, the first end of the capacitor C1 and the cathode of the diode ZD is the adjustable voltage Vadj. The second end of the resistor R1 and the second end of the capacitor C1 are connected to the output end 22 of the adjustable power supply 200 and to the second end 62 of the power module 600. In operation, the diode ZD is used to adjust the voltage value of the adjustable voltage Vadj. The capacitor C1 has a voltage stabilizing function, which is used to stabilize the voltage value of the adjustable voltage Vadj at the first end of the capacitor C1. The resistor R1 has a current limiting function, which is used to limit the current i_ZD of the diode ZD.

Miller switch control circuit 100 includes transistor Q3, transistor Q4, resistor R2, capacitor C2 and diode D1. Transistors Q3 and Q4 are power switches, and transistors Q3 and Q4 form an inverter. The gates of transistors Q3 and Q4 are commonly connected to the input terminal 11 of Miller switch control circuit 100, and the drains of transistors Q3 and Q4 are commonly connected to node N34, which is the output terminal of the inverter formed by transistors Q3 and Q4. The source of transistor Q3 is connected to the ground terminal GND1, and the potential of the source of transistor Q3 is the ground voltage. The source of transistor Q4 is connected to the first node N1, and the potential of the source of transistor Q4 is the first constant voltage V1-. Resistor R2 is connected in series to capacitor C2. A first end of capacitor C2 is connected to node N34, and a second end of capacitor C2 is connected to a first end of resistor R2. The second end of resistor R2 and the cathode of diode D1 are connected to output terminal 12 of Miller switch control circuit 100. The anode of diode D1 is connected to a first node N1, and the first node N1 has a first constant voltage V1-, that is, the anode of diode D1 receives the first constant voltage V1-. Input terminal 11 of Miller switch control circuit 100 is connected to output terminal 41 of control chip 400, and output terminal 12 of Miller switch control circuit 100 is connected to the gate of transistor Q6 (that is, Miller switch). The source of transistor Q6 is connected to the gate g5 of transistor Q5, and the drain of transistor Q6 receives the first constant voltage V1-. In operation, diode D1 provides a loop transmission path for transistor Q4, and diode D1 has a discharge function to discharge transistor Q4. Capacitor C2 has a voltage stabilization function, which is used to stabilize the voltage value of the potential of node N34.

The driving circuit 500 includes a transistor Q1 and a transistor Q2, and the transistors Q1 and Q2 form an inverter. The gates of the transistors Q1 and Q2 are connected to the input terminal 51 of the driving circuit 500 (i.e., the input terminal of the inverter) and to the output terminal 41 of the control chip 400. The drains of the transistors Q1 and Q2 are connected to the output terminal 52 of the driving circuit 500 (i.e., the output terminal of the inverter) and to the control terminal 63 of the power module 600 via the resistor Rg. The source of the transistor Q2 is connected to the low potential terminal 54 of the driving circuit 500 and to the first terminal of the capacitor C1, the first terminal of the resistor R1 and the cathode of the diode ZD. The source of transistor Q1 is connected to the high potential terminal 53 of the driving circuit 500 and receives the second constant voltage V1+. The voltage value of the second constant voltage V1+ is higher than the first constant voltage V1-. The second constant voltage V1+ is higher than the ground voltage of the ground terminal GND1, and the second constant voltage V1+ is a positive voltage value.

Figures 5A and 5B are schematic diagrams of the operation of the Miller switch control circuit 100 and the adjustable power supply 200 of Figure 4. Figure 6 is a waveform diagram showing the on/off state changes of transistors Q1 and Q2 of the driving circuit 500, transistors Q3 and Q4 of the Miller switch control circuit 100, transistor Q6 as a Miller switch, and transistor Q5 of the power module 600, as well as the gate-source voltage Vgs5 of the transistor Q5 and the voltage change of the cross-voltage VC2 of the capacitor C2. Please refer to FIG. 5A and FIG. 6 . The gates of transistors Q1, Q2, Q3 and Q4 are all connected to the output terminal 41 of the control chip 400 . Therefore, transistors Q1, Q2, Q3 and Q4 are switched between the on state and the off state according to the voltage Vout of the output terminal 41 of the control chip 400 .

The transistor Q5 of the power module 600 is similar to the transistor Q12 of the lower arm element of FIG. 1A (the upper arm element of the power module 600 is not shown in the figure). Between the time point t0 and the time point t1, the voltage Vout of the output terminal 41 of the control chip 400 is a high voltage value, the transistor Q1 is turned on, the transistor Q2 is turned off, the transistor Q3 is turned off, and the transistor Q4 is turned on. Therefore, the transistor Q1, the resistor Rg, the transistor Q5, the resistor R1 and the diode ZD form a conductive transmission path P1. Transistor Q5, as a lower arm element of the power module 600, operates at a normal positive voltage between time point t0 and time point t1 in response to the high voltage value of the voltage Vout at the output terminal 41 of the control chip 400. For example, the gate g5 of transistor Q5 receives the second constant voltage V1+ through the resistor Rg in the transmission path P1 and the transistor Q1, that is, the voltage of the gate g5 of transistor Q5 is equal to the second constant voltage V1+. In addition, the source s5 of transistor Q5 is connected to the ground terminal GND1 through the transmission path P1, and the voltage of the source s5 of transistor Q5 is equal to the ground voltage of the ground terminal GND1 (the ground voltage is, for example, a zero voltage value). Accordingly, the gate-source voltage Vgs5 of the transistor Q5 is equal to the second constant voltage V1+.

On the other hand, transistor Q4, diode D1, resistor R2 and capacitor C2 form a conductive transmission path P2. Between time point t0 and time point t1, since the voltage across capacitor C2 VC2 is zero voltage, transistor Q6 is disconnected, and transmission path P2 is a loop.

Next, referring to FIG. 5B and FIG. 6, between time point t1 and time point t3, the voltage Vout at the output terminal 41 of the control chip 400 is reduced to a low voltage value, and the transistor Q5 operates at a normal negative voltage. In response to the low voltage value of the voltage Vout, the transistor Q1 is disconnected, the transistor Q2 is turned on, the transistor Q3 is turned on, and the transistor Q4 is disconnected. Therefore, the transistor Q2, the resistor Rg, the transistor Q5, the resistor R1, and the diode ZD form a conductive transmission path P3.

In the first stage (i.e., between time point t1 and time point t2) when transistor Q5 operates at a normal negative voltage, the voltage VC2 across capacitor C2 gradually increases and is higher than the zero voltage value, so transistor Q6 is turned on. Transistor Q3, capacitor C2, resistor R2, and transistor Q6 form a conductive transmission path P4. The gate g5 of transistor Q5 receives the first constant voltage V1- through the turned-on transistor Q6, and the source s5 of transistor Q5 is still connected to the ground terminal GND1 through the transmission path P3 (the voltage of the source s5 of transistor Q5 is still zero voltage), so the gate-source voltage Vgs5 is equal to the first constant voltage V1-. From the above, between the time point t1 and the time point t2, the Miller switch control circuit 100 controls the transistor Q6 to be turned on, so that the gate-source voltage Vgs5 of transistor Q5 is controlled to be equal to the first constant voltage V1-, so as to suppress the spike voltage generated by the gate-source voltage Vgs5.

The time that the Miller switch control circuit 100 controls the transistor Q6 to be turned on (i.e., the time difference Δt between time point t1 and time point t2) is called the "Miller switch driving time t _miller " of the gate-source voltage Vgs5 of the transistor Q5, or can be called the "Miller clamping time". The Miller switch driving time t _miller is equal to the time difference Δt between time point t1 and time point t2, as shown in formula (1): (1)

"Vth6" in formula (1) is the critical voltage of transistor Q6 as a Miller switch, "V1+" is the second constant voltage, "R_2" is the resistance value of resistor R2, and "C_2" is the capacitance value of resistor C2. The resistor R2, capacitor C2 and diode D1 in the Miller switch control circuit 100 form a charge-discharge loop. The Miller switch control circuit 100 disclosed in the present invention can adjust the Miller switch driving time t _miller in response to different requirements (for example, in response to power conversion devices with different power specifications), and the Miller switch driving time t _miller can be changed by adjusting the resistance value R_2 of the resistor R2 and the capacitance value C_2 of the capacitor C2.

Next, in the second stage (i.e., between time point t2 and time point t3) during which transistor Q5 operates at a normal negative voltage, the cross voltage VC2 of capacitor C2 gradually increases, and the cross voltage VC2 is higher than the critical voltage Vth6 of transistor Q6, so transistor Q6 is disconnected. Since transistor Q6 is disconnected, the gate g5 of transistor Q5 receives the adjustable voltage Vadj of the cathode of diode ZD via the transmission path P3. From the above, between time point t2 and time point t3, the Miller switch control circuit 100 controls transistor Q6 to be off, so that the gate-source voltage Vgs5 of transistor Q5 is controlled to be equal to the adjustable voltage Vadj; the gate-source voltage Vgs5 at this time is called "Miller clamping voltage V _miller ", as shown in formula (2): (2)

V _ZD in equation (2) is the reverse bias of the diode ZD. The adjustable power supply 200 disclosed herein can adjust the Miller clamp voltage V _miller according to different requirements, by adjusting the negative bias V _ZD of the diode ZD (or selecting a different diode _ZD with a different negative bias V ZD) to change the voltage value of the adjustable voltage Vadj, thereby changing the Miller clamp voltage V _miller of the gate-source voltage Vgs5. When the transistor Q5 of the power module 600 operates at a normally negative voltage, the adjustable power supply 200 is used to control the voltage difference between the control terminal 63 and the second terminal 62 of the power module 600 (i.e., the gate-source voltage Vgs5 of the transistor Q5) so that the voltage difference between the control terminal 63 and the second terminal 62 is an appropriate Miller clamping voltage V _miller .

From the above, in the first stage (i.e., between time point t1 and time point t2) during the period when transistor Q5 operates at a normal negative voltage, the gate-source voltage Vgs5 of transistor Q5 is controlled to be equal to the first constant voltage V1-. In the second stage (i.e., between time point t2 and time point t3), the gate-source voltage Vgs5 of transistor Q5 is controlled to be equal to the adjustable voltage Vadj. Accordingly, when transistor Q5 operates at a normal negative voltage, the gate-source voltage Vgs5 has a voltage transition state of two stages.

Then, a periodic operation is performed. Between time point t3 and time point t4, the voltage Vout of the output terminal 41 of the control chip 400 is increased to a high voltage value, and the transistor Q5 operates at a normal positive voltage. Between time point t4 and time point t6, the voltage Vout of the output terminal 41 of the control chip 400 is reduced to a low voltage value again, and the transistor Q5 operates at a normal negative voltage.

FIG. 7 is a comparison diagram of the voltage variation of the gate-source voltage Vgs5 of the transistor Q5 of the power module 600 of the present disclosure and the voltage variation of the gate-source voltage Vgs12 of the transistor Q12 of the conventional power module 610 of FIG. 1A . The voltage variation of the gate-source voltage Vgs12 of the transistor Q12 of the known power module 610 is represented by the dotted line portion in FIG. 7. From the dotted line portion, it is observed that the glitch voltage caused by the Miller effect of the transistor Q12 is superimposed on the gate-source voltage Vgs12, causing the gate-source voltage Vgs12 to increase or decrease sharply from the original normal negative voltage of -15V. Among them, the gate-source voltage Vgs12 that decreases sharply after the superimposed glitch voltage may be lower than the minimum rated value of the negative voltage of -20V, resulting in damage to circuit components.

In contrast, the gate-source voltage Vgs5 of the transistor Q5 of the power module 600 of the present disclosure is controlled to be equal to the first constant voltage V1- within the range of the Miller switch driving time t _miller (i.e., within the time difference Δt shown in FIG. 7 ). Furthermore, after the Miller switch driving time t _miller , the gate-source voltage Vgs5 increases the voltage difference ΔV to become the Miller clamping voltage V _miller (the Miller clamping voltage V _miller is equal to the adjustable voltage Vadj).

According to the comparison of FIG. 7 , the glitch voltage caused by the Miller effect of the conventional power module 610 causes the gate-source voltage Vgs12 of the transistor Q12 to increase or decrease sharply. In contrast, in the power module 600 disclosed in the present invention, the gate-source voltage Vgs5 of the transistor Q5 can be controlled to be between the first constant voltage V1- and the Miller clamping voltage V _miller , effectively suppressing the Miller effect.

FIG. 8A is a schematic diagram of the control circuit 700 disclosed herein applied to a power conversion device 2000. The power conversion device 2000 includes a power source 2100, a power module 620, an energy storage element 2200, a load 2300, and a control circuit 700. The power source 2100 is, for example, a large fixed grid device. The energy storage element 2200 is, for example, a small (even portable) battery device. The power module 620 is disposed between the power source 2100 and the energy storage element 2200, and the power module 620 serves as a switching switch. In response to the different power specifications of the power source 2100 , the energy storage element 2200 , and the load 2300 , the power module 620 converts the power provided by the power source 2100 to generate specific power and specific form of power and provide it to the energy storage element 2200 .

The function of the control circuit 700 is the same as that of the Miller switch control circuit 100 and the adjustable power supply 200 of the embodiment of FIGS. 3 and 4. The control circuit 700 is connected to the power module 620. When the power module 620 is in operation, the power semiconductor elements in the power module 620 may be short-circuited or malfunction due to the Miller effect. The control circuit 700 performs Miller clamp control on the power semiconductor elements in the power module 620 to suppress the Miller effect, for example, controlling the gate-source voltage of the power semiconductor elements in the power module 620 to an appropriate Miller clamp voltage V _miller , and controlling the Miller switch to operate at an appropriate Miller switch driving time t _miller .

Please refer to FIG. 8B, which is a block diagram of an embodiment of the power conversion device 2000 of FIG. 8A. The power module 620 of this embodiment includes six power switch elements, and the control circuit 700 includes six units 710-760. Each power switch element of the power module 620 has the function of switching a switch, and each power switch element of the power module 620 is controlled by a corresponding one of the units 710-760 of the control circuit 700. For example, the units 710 and 720 of the control circuit 700 respectively control a group of power switch elements of the power module 620, and the group of power switch elements serves as the upper arm element and the lower arm element of the power module 620. Similarly, units 730 and 740 of the control circuit 700 respectively control another set of power switch elements of the power module 620, and units 750 and 760 of the control circuit 700 respectively control yet another set of power switch elements of the power module 620.

Although the present invention has been disclosed in detail with preferred embodiments and examples, it is understood that these examples are intended to be illustrative rather than restrictive. It is expected that a person with ordinary knowledge in the art can think of various modifications and combinations, which fall within the spirit of the present invention and the scope of the attached patent application.

1000b,1000a:control device 2000:power conversion device 2100:power supply 2200:energy storage element 2300:load 10:logic circuit 100:Miller switch control circuit 200:adjustable power supply 350:peripheral circuit 400:control chip 500:drive circuit 600,610,620:power module 700:control circuit 710~760:unit 11,51:input end 12,22,41,42,52:output end 61:first end 62:second end 53:high potential end 54:low potential end 63:control end C1,C2,C3:capacitors R1,R2,Rg:resistors ZD,D1,D2:diodes i_ZD:currents Vout,Vadj:voltages P1,P2,P3,P4:transmission paths Vth6:critical voltages Q1,Q11,Q12,Q2,Q3,Q4,Q5,Q6:transistors g11,g12,g5:gates d11,d12,d5:drains s11,s12,s5:sources Cgd11,Cgs11,Cds11,Cgd12,Cgs12,Cds12:parasitic capacitances Cgd,Cgs,Cds:parasitic capacitances Cgd5 gate-drain parasitic capacitance N34: node Vgs5, Vgs11, Vgs12: gate-source voltage Vds5: drain-source voltage VC2: cross voltage △V: voltage difference △t: time difference t11, t12, t13, t14: time point V1-: first constant voltage V1+: second constant voltage GND1: ground terminal Im1: Miller current

FIG. 1A is a circuit diagram of a portion of a power module in a known power conversion device. FIG. 1B is a timing diagram of the voltage change of the gate-source voltage of two transistors in the power module of FIG. 1A. FIG. 2 is a circuit diagram of a control device in a power conversion device of a comparative example. FIG. 3 is a circuit diagram of a control device of an embodiment of the present disclosure in a power conversion device. FIG. 4 is a detailed circuit diagram of a Miller switch control circuit, an adjustable power supply, a drive circuit, and a power module in the control device of FIG. 3. FIG. 5A and FIG. 5B are schematic diagrams of the operation of the Miller switch control circuit and the adjustable power supply of FIG. 4. FIG. 6 is a waveform diagram showing the on/off state change of the transistor of the driving circuit, the transistor of the Miller switch control circuit, the transistor as the Miller switch, and the transistor of the power module, as well as the gate-source voltage of the transistor and the voltage change of the cross-voltage of the capacitor. FIG. 7 is a comparison diagram of the voltage change of the gate-source voltage of the transistor of the power module disclosed in the present invention and the voltage change of the gate-source voltage of the transistor of the known power module of FIG. 1A. FIG. 8A is a schematic diagram of the control circuit of the present invention applied to a power conversion device. FIG. 8B is a block diagram of an embodiment of the power conversion device of FIG. 8A.

1000b: Control device

100: Miller switch control circuit

200:Adjustable power supply

400: Control chip

500:Drive circuit

600: Power module

11,51: Input port

12,22,41,42,52: output port

61: First end

62: Second end

63: Control terminal

53: High potential end

54: Low potential end

C3: Capacitor

Rg: resistance

Q1, Q2, Q6: transistors

V1+: Second constant voltage

GND1: ground terminal

Im1: Miller current

Claims (11)

A control device for a power conversion device, comprising: a power module having a first end, a second end and a control end, the power module comprising a power semiconductor element; a driving circuit connected to the control end of the power module, used to control the voltage of the control end of the power module; a Miller switching transistor connected to the control end of the power module, the Miller switching transistor providing a bypass transmission path to conduct a Miller current generated when the power semiconductor element is switched; A Miller switch control circuit is connected to a gate of the Miller switch transistor and a drain of the Miller switch transistor. The Miller switch control circuit controls the Miller switch transistor to be turned on or off. When the Miller switch transistor is turned on, the Miller switch transistor conducts the Miller current; and An adjustable power supply is connected to the second end of the power module. When the power module operates at a normal negative voltage, the adjustable power supply controls a voltage difference between the control end and the second end of the power module. A control device as described in claim 1, wherein the power semiconductor element of the power module is a power switching transistor, a drain, a source and a gate of the power switching transistor are respectively connected to the first end, the second end and the control end of the power module, a parasitic capacitance is provided between the gate and the drain of the power switching transistor, and the Miller current is related to the parasitic capacitance. A control device as described in claim 2, wherein the driving circuit includes a first transistor and a second transistor, the first transistor and the second transistor form an inverter, an output end of the inverter is connected to the gate of the power switching transistor, and the driving circuit is used to control the voltage of the gate of the power switching transistor. A control device as described in claim 1, wherein the adjustable power supply comprises: a Zener diode, a cathode of the Zener diode is connected to the driving circuit, the cathode of the Zener diode has an adjustable voltage, and the potential of an anode of the Zener diode is equal to a first constant voltage, and the first constant voltage is a negative voltage value; a first capacitor connected between the cathode of the Zener diode and the second end of the power module; and a first resistor connected between the cathode of the Zener diode and the second end of the power module, Wherein, when the power module operates at the normal negative voltage, the voltage difference between the control end and the second end of the power module is equal to the first constant voltage or the adjustable voltage. A control device as described in claim 4, wherein: During a first stage during which the power module operates at the normal negative voltage, the voltage difference between the control terminal and the second terminal of the power module is equal to the first constant voltage, and During a second stage during which the power module operates at the normal negative voltage, the voltage difference between the control terminal and the second terminal of the power module is equal to the adjustable voltage. A control device as described in claim 4, wherein the adjustable voltage is related to a negative bias of the Zener diode. A control device as described in claim 4, wherein the first capacitor is used to stabilize the voltage value of the adjustable voltage, and the first resistor is used to limit the current of the Zener diode. A control device as described in claim 1, wherein the Miller switch control circuit comprises: a third transistor and a fourth transistor, the third transistor and the fourth transistor forming an inverter, an input terminal of the inverter connected to the drive circuit, and the fourth transistor connected to the drain of the Miller switch transistor; a second capacitor and a second resistor connected between an output terminal of the inverter and the gate of the Miller switch transistor; and a first diode connected to the second resistor and connected to the drain of the Miller switch transistor. A control device as described in claim 8, wherein: When the power module operates at a normal positive voltage, the third transistor is in an off state, the fourth transistor is in an on state, and the Miller switch transistor is in an off state; and the second capacitor, the second resistor, the first diode and the fourth transistor form a loop transmission path. A control device as described in claim 8, wherein: During a first phase during which the power module operates at the normal negative voltage, the third transistor is in an on state, the fourth transistor is in an off state, and the Miller switch transistor is in an on state; and The duration of the first phase is related to the cross-voltage of the second capacitor. A control device as described in claim 10, wherein: The potential of a source of the fourth transistor is equal to a first constant voltage, and the potential of an anode of the first diode is equal to the first constant voltage, and the first constant voltage is a negative voltage value; and In the first stage during which the power module operates at the normal negative voltage, the voltage difference between the control terminal and the second terminal of the power module is equal to the first constant voltage.

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