SE429704B - SET FOR CODE SIGNALS IN AN FM RADIO SYSTEM AND THE RECEIVER AND TRANSMITTER IN A RADIO RADIO SYSTEM FOR EXECUTION OF THE SET - Google Patents

Description

7802031-0 at CCIR (Conite Coasultatif International des Radiocommunications). In this system, the code signal is transmitted by means of a suitable subcarrier above the frequency spectrum of the stereo information signal. This subcarrier is frequency modulated with the near code signal, which through a digital code contains information about, for example, the name of the program, the position of the transmitter, the nature of the program and the channel number, so that e.g. the following information consisting of sixteen characters is received: hed1 Roerm KL 25 The receivers in such a system are provided with a decoder, which decodes the binary code signal from the received signal, and uses it to, for example, fully or partially optically reproduce the information thus transmitted. so that the user can immediately see which transmitter his receiver is tuned to. Alternatively, it is possible to arrange the receiver in such a way that during the occurrence of a preset code a part of the receiver or the tape recording or reproducing apparatus is switched on or off. If the code contains a special code, which was transmitted for traffic reporting, the code can in particular be used to connect the reproducing part of a car radio receiver or to stop a tape reproducing device which is in operation.

This known radio signaling system has been tested in practice with the following values: The subcarrier frequency was 66 kHz and the frequency sweep 1 kHz so that due to the binary information the frequency was varied between 65 kHz and 67 kHz.

The code used was the 6-bit ASCII code with 16 characters for information.

The amplitude of the modulated code signal was so selected that 1 kHz, i.e. 1.33% of the total 75 kHz frequency sweep available for the FM modulation of the main wave was occupied by the code signal. The relatively small amplitude (1 kHz) of this signal has been chosen because experiments have shown that a larger amplitude can cause interference noise in some FH receivers.

However, it was found that the necessarily small amplitude of the modulated code signal and its relatively high frequency (66 kHz) results in a poor signal-to-noise ratio. In order to recover the code signal error-free, the receiver requires a filter with a high ground value and with a good goodness factor and a good temperature stability. In addition, it was found that despite the use of such an expensive high-order filter, the decoding of the code signal does not occur flawlessly for antenna voltages below 10 / uV (at 60 Ohm), while the average FM receivers still provide acceptable mono reception. at these antenna voltages. An object of the invention is to provide a method for code signaling in an FM broadcasting system, which allows a substantially error-free decoding of the code signal at received antenna voltages, at which an acceptable mono reception is not or hardly possible, while the signal reception in existing receivers are not disturbed or hardly disturbed, and filters of high order and thus expensive filters in the receiver according to the invention can be dispensed with. For this purpose, the method according to the invention is characterized in that said further subcarrier is a harmonic to an undertone of the stereopilot signal, which does not coincide with a harmonic of the stereopilot signal and is derived on the transmitter side from the same frequency source as the stereopilot signal.

The term binary phase modulation here, as usual, refers to a phase modulation in which the phase of the subcarrier is shifted 180 ° by the binary code signal. This provides a modulated signal with a completely suppressed carrier.

By using phase modulation of the subcarrier with the binary code signal (phase shift keying) instead of frequency modulation (frequency shift keying), an improvement in the signal-to-noise ratio is obtained. However, in contrast to the demodulation of the frequency modulated subcarrier, the demodulation of the phase modulated subcarrier requires an unmodulated ("pure") subcarrier. This subcarrier is not present in the binary phase modulated code signal, because the subcarrier itself is suppressed and only the sidebands are transmitted. However, generating this subcarrier on the receiving side can be done by squaring the incoming binary phase modulated signal, resulting in a dual frequency carrier, and then filtering this dual frequency carrier and recovering the carrier carrier from the original frequency from the dual frequency carrier. through a divider, which divides the frequency by two.

With this method, in poor signal-to-noise conditions, the dual-frequency carrier must be generated from a signal with a high content of noise. If e.g. and s.k. phase-locked loop is used for this purpose, this snow can be used by using a low-pass filter with a low cut-off frequency in the loop in such a way that the phase of the voltage-controlled oscillator in the phase loop is not modulated too much by noise, but on the other hand such a low-pass filter phase-locked loop retraction area to such an extent that a voltage-controlled oscillator with a very stable idle frequency is required. In practice, this can only be achieved with a crystal controlled oscillator.

By the present invention in which binary phase modulation of the subcarrier of the code signal is used in combination with a frequency ratio between the stereopilot signal and said other carrier, which ratio is fixed on the transmitter side, a system is obtained in which decoding can take place without high word filter means. and which is relatively unrecognizable for poor signal-to-noise conditions. 7802031-0 The subcarrier for the transmitter identification signal can now be recovered by much simpler means, since the stereopilot signal is modulated with a much larger frequency sweep (10% of the total frequency sweep of 75 kHz) on the main carrier than the transmitter identification signal itself (1.33% of the total the frequency sweep of 75kHz). In a receiver in a plant according to the invention, phase errors can be generated due to different delay times for the modulated code signal and for the stereopilot signal in the tuner's tuning unit and in the intermediate frequency section. Phase multiples are also generated because the subcarrier frequency (Åik) of the modulated code signal is so selected that it is equal to a "sub" harmonic of the stereopilot signal QJ); this means that àfb = n / m, af, where m and n are integers but n is not divisible-with m. The frequency division required in this case in the transmitter and in the receiver can generate these phase diversity.

According to a further aspect of the invention, an automatic phase correction device is used in a receiver according to the invention, which can set the phase of the unmodulated wave required for detecting the modulated code signal relative to the modulated code signal itself. This phase correction device is controlled from a phase detector which compares the phase of the modulated source signal with the phase of the unmodulated wave; obtained from the stereopilot signal, and depending on the result of this comparison, the phase correction device corrects all phase errors. However, since the carrier itself is missing in the binary modulated code signal, this cannot be done without further measures.

A first method of solving this difficulty is to use frequency doubling of the modulated code signal, which gives an unmodulated carrier with twice the subcarrier frequency. This subcarrier with double subcarrier frequency is fed to one input of the phase detector, while a wave which also has double subcarrier frequency, which is generated by frequency multiplication and / or division of stereopilot signal, is fed to the other input.

A second method consists in using a phase inverter in one of the input lines of the phase detector or in its output line, which phase inverter is controlled by the demodulated output signal from the synchronous detector. It is obvious that in both cases this leads to a fixed ambiguity in the detection of the code signal.

This fixed ambiguity is not disturbing if a code is used, which is insensitive to such ambiguity, e.g. and s.k. differentialkbd. This is a code in which the two binary states are not transmitted by two fixed states of the subcarrier but by the occurrence or not of a phase transition from one phase to the other and vice versa.

The frequency of the subcarrier can e.g. be selected to be between the third and fifth harmonics of the stereo pilot signal. If you select it during the third harmonic, the subcarrier comes too close to the spectrum of the stereo information signal and if you select it over the fifth harmonic, the risk of interference due to adjacent transmitters increases. 5 7su2os1-o Furthermore, interference may occur in a number of stereo receivers between the subcarrier for the transmitter identification and the second harmonic of the 38 kHz signal required for the stereo detection, which corresponds to the fourth harmonic of the pilot signal. For this purpose, the sub-curve for the code signaling shall not be too close to this fourth harmonic.

Due to the non-linear phase characteristic of the receiver frequency section of the receiver, an interference product with a frequency equal to the difference frequency between the subcarrier and the stereopilot signal occurs in the multiplex signal.

This interference product, after detection with the voltage of 38 kHz, may cause audible noise if the subcarrier is too close to the third harmonic of the stereo pilot signal.

The said non-linear phase characteristic further causes noise in the range of all the harmonics of the stereopilot signal. All these considerations lead to the choice that the subcarrier of the code signal should not coincide with a whole harmonic of the stereopilot signal. Accordingly, it should preferably be a "base" harmonic to the subcarrier of the code signal, removing the phase ambiguity which then occurs in the receiver in the manner described above.

On the basis of the above, the subcarrier should preferably be halfway between two harmonics to the stereopilot signal, e.g. at 7/2 or 9/2 times the stereo pilot signal. The invention has been investigated with a carrier frequency of 7/2 times the pilot frequency; for the sake of clarity, an embodiment with a subcarrier frequency of I6 / 5 times a pilot frequency is described.

A further improvement of the method of code signaling according to the invention, in which while maintaining a reliable transmission of the code information a reduced risk of interference in the receivers is possible, is characterized in that the additional subcarrier with the modified code signal is located in at least one of the two levels of the frequency range. , which is divided into two parts of the stereo signal, between the upper limit of the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modified stereo information signal, and that the modulated code signal has an amplitude which preferably deviates the main carrier. 0.25 kHz.

This action has the following effects. 1. Since the subcarrier for the kcd signaling is now at a distance from the higher harmonics to the 38 kHz stereo detection signal, these higher harmonics cannot produce audible noise in the existing receivers. 2. Since the subcarrier for the code signaling is now located much lower in the frequency range of the multiplexed signal, the signal-to-noise ratio is much more favorable.

Consequently, the modulated code signal may have even less amplitude than was possible for a subcarrier with e.g. 65.5 kHz. By way of comparison, in the presently preferred embodiment, the modulated code signal need only occupy approximately 0.25 kHz of the maximum frequency sweep of 75 kHz in order to obtain a 78D2031 ~ 0 e reliable code signaling. In the case of a subcarrier of 66.5 kHz, approximately 1 kHz is required. The much smaller subcarrier amplitude naturally reduces the risk of interference noise caused by other components of the multipix signal. A further reduction of the risk of noise in remote receivers, in particular in mono reception, can be achieved in accordance with a further feature of the invention if a subcarrier, the phase of which is binary modulated with the code signal, lies in each of the two halves. of said frequency range, which is divided into two parts of the stereopilot signal, and if the two subcarriers modulated by the code signal have equal amplitudes and such a phase relative to the stereopilot signal that together with the stereopilot signal they form a signal obtained by quadrature modulation of the stereopilot signal with a subcarrier which is derived from the stereopilot signal, which subcarrier itself is binary phase modulated by the code signal.

In such a signal, each of the binary phase modulated subcarrier signals can be considered as a sideband in a dual sideband signal with the stereopilot signal as the carrier. The stereo pilot signal is quadrature modulated by a modulation signal, which itself is binary phase modulated by the code signal. The modulation signal has a frequency which is equal to the difference between the frequency of the stereopilot signal and a subcarrier. A plant according to this further feature of the invention, which has been investigated in practice, has closest to the stereopilot signal with a frequency f of 19 kHz a first subcarrier of 16.625 kHz (7/8 f), the phase of which is binary modulated by the code signal, and a second subcarrier at 21,375 kHz (9/8 fp) whose phase is binary modulated by the code signal. In the case of equal amplitudes of the two subcarriers and a correct relative phase relationship between the subcarriers and the stereopilot signal, the three signals together form a stereopilot signal, which is quadrature modulated by a subcarrier signal of 1/8 f, which is itself binary phase modulated by the code signal. For this purpose, a subcarrier phase must be before the stereopilot signal which is offset 900 the same amount as the other subcarrier is after this 900 phase shifted stereopilot signal, in other words the resultant of the two modulated subcarriers has a 900 phase shift relative to the stereopilot signal. The sum of the stereo pilot signal and the two subcarriers forms a pilot signal whose amplitude is substantially constant. In particular, since amplitude variations in the pilot signal give rise to distortion products due to the non-linear phase characteristic of the intermediate frequency section of the receivers, the measure described above gives a further reduction in distortion.

Within the scope of the invention, it is alternatively possible to give the subcarrier modulated with the code signals such a phase that the resultants always coincide (00 or 1800) with the stereopilot signal. The two subcarriers which act as aid bands to the stereo pilot signal thereby cause an amplitude modulation of the stereo pilot signal with a carrier signal which is itself binary phase modulated by the code signal. When the stereopilot signal, which is double-side band squared or amplitude modulated by the two subcarriers, or the stereopilot signal which is single-side band phase and amplitude modulated by a subcarrier, is supplied to the stereo decoder already in a radio receiver, strongly of the stereopilot signal filter present in such receivers, relative to the stereopilot signal itself that interference with the stereo detector does not occur in the main. Such a disturbance would be much greater with a direct phase or amplitude modulation of the stereopilot signal with the code signal.

Of course, this disturbance also becomes greater the closer to the stereopilot signal the subcarriers are (eg at 11/12 fp and / or 13/12 fp). At an excessive distance between the subcarrier and the stereopilot signal, on the other hand, the subcarrier is located too close to the frequency spectrum of the audio information signal or the spectrum of the modulated stereo information signal. Based on these considerations, a distance of 1/8 fp between the subcarrier (respectively subcarriers) and the stereo pilot signal is preferred.

Since the subcarrier frequency in the system according to the present embodiment is relatively close to the frequency of the stereopilot signal, it is preferable to first convert in the receiver for receiving such signals the modulated subcarrier (7/8 fp and / or 9/8 fp) with the stereopilot signal to an intermediate frequency ( 1/8 fp), which is in a harmonic relationship to the serial pilot signal and which is equal to the difference between the subcarrier frequency and the stereo pilot frequency. The synchronous detection of the code signal can then be performed at this lower frequency in a manner corresponding to that described previously.

The invention is explained in more detail by means of examples with reference to the drawings, where Fig. 1 shows a block diagram of an embodiment of a transmitter in a first realization of the plant according to the invention, Fig. 2 shows the frequency spectrum of the multiplexed signal generated on the transmitter side. Fig. 3 shows a block diagram of a first embodiment of a receiver according to the invention, Fig. 4 shows a block diagram of a second embodiment of a receiver according to the invention, Fig. 5 shows a block diagram of a first embodiment of a receiver according to the invention. and Fig. 5a shows block diagrams for a transmitter in a second embodiment of the plant according to the invention, Fig. 6 shows the frequency spectrum of the multiplex signal generated on the transmitter side and obtained on the receiver side after FH demodulation according to the second embodiment of the plant, Figs. 7 shows a block diagram of an embodiment of a receiver for reception of a signal according to Fig. 6 and Fig. 8 shows a block diagram of a second embodiment of a receiver for receiving a signal according to Fig. 6.

The transmitter according to Fig. 1 comprises a source for left-hand audio signals 1 and a source for right-hand audio signals 2. Each of these audio signals is fed via a pre-phase network 5 and 4, respectively, and via a low-pass filter 5 and 0, respectively, with a limit-free 15 kHz to a summing circuit 7 and to a subtraction circuit 8. Thereafter, the sum signal L + R derived from the summing circuit is fed to an input 9 of a multiplexer 10. The difference signal LR from the subtraction circuit 8 is modulated in a balanced modulator 11 on a stereo subcarrier of for example 38 kHz and the thus obtained modulated stereo information signal, which consists of two side beams with suppressed stereo subcarrier, is fed via a bandpass filter 12 to a second input 13 on the multiplexer 1.0. In addition, the transmitter in Fig. 1 comprises a stable oscillator 14, e.g. a crystal oscillator, which generally emits a wave of 19 kHz, which is used as a stereopilot signal. This stereo pilot signal is fed to a third input 15 of the multiplexer 10.

The stereopilot signal of the oscillator 14 is also fed to a so-called phase locked loop 16 which includes a phase detector 16a, a low pass filter 16b, a voltage controlled oscillator 16c and a frequency division divider with two 16d- The phase locked loop 16 is used to generate a subcarrier whose frequency (38 kHz) is equal to twice the stereopilot frequency and which is locked to the etereopilot signal. The function of such a phase locked loop is known. The 38 kHz output signal from the oscillator 16c is converted in the divider 16d to a 19 kHz signal, which is compared in the phase detector 16a with the 19 kHz oscillator 14 pilot signal. The output voltage from the phase detector 16a is filtered in the low-pass filter 16b and supplied as control voltage to the oscillator 160.

Via a phase shifting device 17, the output signal of 38 kHz is supplied from the phase-locked loop 16 as a stereo subcarrier to the modulator 11 for modulating the signal L-R. The phase shift device 17 is used to provide the subcarrier with the internationally prescribed phase relative to the stereo pilot signal of 19 kHz.

A second phase read loop 18 connected to the 19 kHz oscillator includes a phase detector 18a, a low pass filter 18b, a voltage controlled oscillator 18c and a 16 divider 18d. The phase locked loop 18 operates in the same manner as the loop 16 and emits an output signal, which is locked to the stereopilot signal, at 304 kHz, i.e. 16 times the pilot signal frequency. Thereafter, the 304 kHz signal is reduced from the phase-locked loop 18 in a divider 19 for division by five to 60.8 kHz and the latter signal is fed as a subcarrier for the transmitter identification signal to the carrier input of a balanced modulator 20. The modulation input of this modulator is connected to a schematically shown arrangement 21 for generating an appropriate binary code containing the transmitter identification information, e.g. a code of the kind defined in the introduction.

The modulator 20 can e.g. be a ring modulator or a long-tailed "dua1" circuit or any other known modulator which under the influence of the bits diverted from the device 21 rotates the phase of the signal of 60.8 kHz from the divider 19 18Û0_ Signa_ 9 7802031-0 kHz which is phase modulated in this way is fed via a bandpass filter 22 with a bandwidth of approximately 4 kHz to a fourth input 23 of the multiplexer 10. The multiplexer combines the signals at the inputs 9,13,15 and 23 and outputs them combined into a FBI transmitter not shown.

For further explanation, Fig. 2 shows the frequency spectrum of the signal obtained at the output of the multiplexer. Between 0 and 15 kHz there is the sum signal L + R which is applied through the input 9, at 19 kHz the stereopilot signal which is applied via the input 15 appears, between 23 and 53 kHz the signal LR occurs which is modulated at 38 kHz and which is applied through input 13 and at 60.8 kHz the approximately 4 kHz wide transmitter identification signal applied through input 23 appears.

It is observed that the relative amplitude conditions usually deviate more from each other than what is shown for the sake of clarity. In general, stereopilot signal selection is about 9 times smaller than components L + R and L-R, while the amplitude of the transmitter identification signal is preferably selected to be about 10 times smaller than the amplitude of the stereopilot signal.

The receiver according to Fig. 3 comprises a tuning unit 24, an intermediate frequency amplifier 25 and an FM detector 26. The multiplex signal composed of the components shown in Fig. 2 appears at the output of this FM detector. In the case of a stereo receiver, this multiplex signal is fed to a stereo decoder 27, which emits left and right audio signals which are applied via audio amplifiers 28 and 29 to a left and a right speaker 30 and 31, respectively.

To demodulate the transmitter identification signal, the multiplex signal is fed to a 19 kHz bandpass filter 32 for the stereopilot signal and a 60.8 kHz bandpass filter 33 for the transmitter identification signal. The stereopilot signal filtered out by the filter 32 is further filtered and its frequency is multiplied by a phase locked loop 34 which includes a phase detector 34a, a low pass filter 34b, a voltage controlled oscillator 340 and a frequency divider 34d for division by 1:32. Its function is the same as the function of the phase-locked loops 16 and 16 in Fig. 1. The output voltage from the phase-locked loop 34, which has a frequency of 32 x 19 = 608 kHz, is then reduced to 121.6 kHz in a divider 35 for division. with five, then passes through a controllable phase shift device 36, the function of which will be explained later, then divided into a divider 37 for division by two to 60.8 kHz and finally fed to a first input 38 on a synchronous demodulator 39.

The 60.8 kHz phase modulated transmitter identification signal coming from the bandpass filter 33 is fed via a 45 ° phase shift device 40 to a second input 41 of the synchronous demodulator 39. The synchronous detection of the 60.8 kHz phase modulated transmitter identification signal at the input 41 by means of the the unmodulated voltage of 60.8 kHz at the input 38 provides at the output of the synchronous demodulator 39 the demodulated binary transmitter identification signals. This binary coder 78023031-0 W signal is passed through a low pass filter 42, then it is formed into rectangular pulses in a pulse shaper 43 and fed to a decoder 44. This decoder converts the binary transmitter identification signal into signals suitable for driving a "display". "45.

For correct synchronous detection in the demodulator 39, the unmodulated voltage at the input 38 must have the correct phase ratio relative to the modulated signal applied to the input 41. In general, this correct phase ratio is not ensured for the following reasons: 1. Due to insufficient linear phase characteristic of intermediate frequency At 25 kHz, the stereo pilot signal at 19 kHz and the transmitter identification signal at 60.8 kHz can be subjected to mutually different delay times. 2. The input filters 32 and 33 can cause unwanted phase shifts. 3. Due to the frequency division of the divider 19 in the transmitter, the phase of the transmitted transmitter identification signal of 60.8 kHz is no longer unambiguously determined relative to the transmitted stereopilot signal. A similar phase diversity is caused by the frequency divider 35 'in the receiver. To eliminate these phase problems, the circuit of Fig. 3 comprises the above-mentioned adjustable phase shift device 36. This phase shift device is controlled via a low-pass filter 46 from a phase detector 47. The phase detector 47 has two inputs 48 and 49, of which the input 48 is connected to the phase shift output. the input device 36, while the input 49 is connected to the output of a device 50 which generates a modulated wave with the dual frequency (ie 121.6 kHz) of the phase modulated signal from the filter 33. For this purpose, the device 50 has a non-linear characteristic with a term of even power and includes e.g. a squaring circuit or a full-wave rectifier.

Since the phase-locked loop 34 multiplies the stereopilot signal by a factor of 2 more than is necessary for the synchronous detection, the frequency of the voltage applied to the input 48 of the phase detector is equal to twice the carrier frequency.

Determination of the phase by the phase detector 47 is thus performed at twice the carrier frequency and the result of the measurement is used to compensate for the above-mentioned undesired phase shifts by means of the controllable phase shift device 36. It is noted that the frequencies of the two signals fed to the phase detector 47 always are equal to each other why no capture problems can occur. The phase errors that are corrected vary only slowly and the low-pass filter 46 can consequently have a very low cut-off frequency (eg 10 Hz). Due to this low cut-off frequency, rapid phase variations that can be generated due to the noise in the transmission path 33-40-41 of the transmitter identification signal can be effectively suppressed. By the stated measures it is possible for the synchronous detector 39 to generate an unmodulated voltage with the correct frequency and the correct phase and which is noise-free to a sufficient degree. As a result of the phase control being corrected by means of the 36,45,47 phase at twice the carrier frequency, the phase relationship at the inputs of the synchronous detector 39 is still not unambiguous (1800 fixed-wave ambiguity). However, when using a (differential) code which is insensitive to this, a correct transmission of the code signal can nevertheless be ensured.

In practice, the automatic phase control by means of the phase control device 36 always functions so that the two input signals to the phase detector 47 are offset 90 ° in phase relative to each other. In addition, it is desirable that the mutual phase ratio between the input signals of the synchronous detector be 0 or 1800. If the divider 57 for division by two is designed so that the zero crossings in its output voltage coincide with the zero crossings in its input voltage and if the frequency doubler 50 is made circuit in which the peaks of the input voltage coincide with the peaks of the output voltage, this preferred phase relationship is automatically achieved. In other cases an additional phase correction may be required in one of the input lines of the detectors 47 and 39, e.g. a 900 correction at twice the carrier frequency or a 450 correction at the carrier frequency itself. The phase shift device 40 for 450 phase shift was used for this purpose. It is observed that several variants of the circuit according to Fig. 3 are conceivable within the scope of the invention. For example, it is possible to replace the divider 57 for division by two with a frequency doubler in the input line 48 of the phase detector 47. The frequency multiplication factor of the phase-locked loop 34 should then be a factor of 2 less. Alternatively, it is e.g. possible to include the phase control device 36 in the output line of the filter 33.

If, instead of 16/5 times the stereo pilot signal 7/2 times this pilot signal, for example, were to be selected as the carrier frequency of the transmitter identification signal, the divisor of the divider 34d may be equal to 14 and the divisor of the divider 35 may be equal to 2.

It is then easier to make the divider 54d divisor equal to 7, whereby the divider 35 can be omitted.

In the embodiment according to Fig. 4, the units corresponding to units in Fig. 3 have been provided with the same reference numerals as in this figure.

While in the embodiment according to Fig. 3 the phase implementation for controlling the phase correction device 56 is performed at the double carrier frequency, this comparison is performed with the embodiment according to Fig. 4 at the carrier frequency itself. For this purpose, the frequency doubling circuit 50 and the divider 37 for frequency division by 2 are omitted and the divisor in the divider 34d is reduced to 16.

Via the phase shift device 36, the divider 35 now emits an unmodulated carrier of carrier frequency (60.8 kHz) to the phase detector d? input 48.

A phase inverter 51 (balanced modulator) is connected in the input line to the input 49 of the phase detector 47. The phase inverter 51 is controlled by the output signal from the synchronous detector 59 or alternatively by the output signal of the pulse shaper 45. Each time the transmitter identification signal phase changes 1800 to the code signal, this produces a transient in the output signal of the pulse shaper 43, which gives a phase reversal of the phase inverter 51 so that at the input_49 the original phase reversal is eliminated. The input 49 of the phase detector 47 is thus fed with the transmitter identification carrier of 60.8 kHz from which the original phase modulation has been removed. In the phase detector 47, the phase of this unmodulated carrier is compared in relation to the voltage at the input 48 and all phase errors are compensated away by the phase shift device 36 via the low-pass filter 46. "Instead of being connected in the line to the input 49, the phase inverter 51 may also be the supply line to the input 48 of the phase detector 47.

The carrier of 60.8 kHz which is applied via the phase shift device 36 is then phase modulated by the binary code signal in the same way as the transmitter identification signal itself has been modulated. In this case, the phase detector 47 again emits an output voltage which can be used for phase correction.

A third possibility is to connect the phase inverter to the output line of the phase detector 47 either before or after the filter 46. The phase detector 47 emits the binary code signal itself, but when the phase inverter changes status for each signal transient in this signal, the phase inverter output becomes a DC voltage.

A 90 ° phase shift device 52 in the input line 38 of the synchronous detector has a similar function to the 450 phase shift device 40 in Fig. 2. Alternatively, the phase shift device 52 may be connected in the input line 41 of the synchronous detector 39 or in one of the input lines of the phase detector.

A further analysis of the circuit according to Fig. 4 shows that the whole phase correction system has two stable control states where the phase difference of the signal at the input detector 49 of the faad detector in relation to the signal at the input 48 can be + 90 ° or -900. The detection of the binary code signal by means of the synchronous detector 59 consequently has the same ambiguity as in the receiver according to Fig. 5.

The circuits shown in Figs. 3 and 4 do not require resonant circuits which meet high selectivity requirements, since a large part of the required selectivity can be realized at low frequency, i.e. of low-pass filter- (34b, 46,42). Consequently, the bandpass filters 32 and 33 only need to have moderate goodness factors (approximately 20). Subsequent investigations have even shown that the stereopilot filter 32 can be dispensed with completely. In some cases it is also possible to obtain from the stereo decoder 27 a stereo pilot signal which has already been filtered. The input of the phase-locked loop 34 is then connected to a suitable point in the stereo decoder 27. The functional units shown in Figs. 1,5 and 4 are known per se and thus require no further explanation.

The transmitter according to Fig. 5 comprises a stereo multiplex encoder 101 to which the sources 102 and 103 for left and right audio signals are connected and a oscillator 104 for 19 kHz which generates a stereopilot signal iè- The encoder_101 composes in a similar manner, as described with reference to Fig. 1, the standard multiplex signal of the applied signals, which standard multiplex signal comprises the audio frequency sum signal L + R, the stereo information signal LR modulated on a suppressed carrier of the double pilot frequency and the stereopilot signal f itself.

It is assumed that the steropilot signal derived from the oscillator 104 has the same phase as the pilot signal in the composite signal.

The stereo pilot signal is fed directly to a second contact b and in addition via a 90 ° phase shift device 105 to a first contact a in a switch 106.

The main contact c of the switch 106 is connected to a first input 107 of a linear modulator 108. When the switch 106 is in the position shown in the drawing, a stereopilot signal is thus fed which is phase shifted 900 relative to the stereopilot signal in the composite signal thereof. entrance. In the second position of the switch, the input 107 of the modulator 108 receives the stereopilot signal in phase with the pilot signal in the composite signal.

In addition, the stereopilot signal is fed via a pulse shaper 109 to a frequency divider 110 which makes a square wave of 1/8 x the stereopilot frequency (2.575 kHz). A bandpass filter 111 tuned to this frequency filters out the fundamental frequency so that a sinusoidal voltage of 1/8 x the pilot frequency appears at the second input 112 of the modulator 108.

The modulator 108 is a linearly balanced modulator which generates the sum and difference frequencies (fp-fp / 8 and fp + fp / 8) of the two applied sine signals, while the frequencies (fp and fp / 8) which were originally supplied are initially missing. - the signal.

The output signal of the modulator 108 is then supplied directly to a first contact a 1 a three-position switch 113 and also via a bandpass filter 114, which is tuned to 16,625 k fl x (7/8 fp), to a second contact b and via a bandpass filter 115. which is tuned to 21,375 kHz (9/8 fp) to a third contact in the three-position switch 115. The main contact d of the three-position switch 113 is connected to a first input 116 on a linearly balanced modulator 117. A device 120 which provides the binary code signal containing the transmitter identification information is connected via a trapezoidal waveform shaper 19 to the second input 11 of the modulator 117. The trapezoidal waveform shaper 119 reduces the content of higher frequency components so that the code signal supplied to the modulator covers a limited frequency range (up to about 600 Hz).

In the third position (c) of the switch 113, the 9/8 fp subcarrier passed through the filter 115 is phase modulated binary in the modulator 117 with the code signal from the device 120. In the second position (b) of the switch 115, the subcarrier 7aß2os1-o is phase modulated. , k the path of 7/8 fn, passed through the filter 114, binary of the code signal. In the first position (Ä) of the switch 113, both subcarriers (7/8 fp and 9/8 fp), which originate from the modulator 108, are phase modulated binary by the code signal. Finally, the output signal of the modulator 117 is added to the sterec multiplex signal from the encoder 101 in a summing step 121; all this in such a way that the amplitude of the added subcarrier and the subcarriers, respectively, is substantially (eg 30 times) smaller than the amplitude of the stereopilot signal contained in the multiplex signal. Finally, the output signal of the summing step 121 is fed to an ET transmitter (not shown).

In neither (a) of ae do both switches 106 and 113 contain the complete transmitted signal closest to the stereopilot signal get the subcarriers fp + 1 / Sfp and get-1 / Bfp which are both binary phase modulated by the code signal. The resultant of the two subcarriers is always phase shifted 90 Relative to the stereopilot signal so that the stereopilot signal with the two subcarriers as sidebands forms a signal which is quadrature modulated, so that the stereopilot signal is amplitude modulated to a very limited extent. The modulation signal itself is a subcarrier of 1/8 fp which is binary phase modulated by the code signal. In the second position (b) of the switch 106, the resultant of the two subcarriers is in phase (or 1800 out of phase) with the stereopilot signal so that the stereopilot signal with the two subcarriers as sidebands forms a signal which is amplitude modulated but not phase modulated with the modulated signal on 1 / 8fp.

In the second and third positions of the switch 113, respectively, only the lower and the upper sideband, respectively, are added to the stereopilot signal in the multiplex signal.

Adjusting the switch 106 results in a 900 phase shift of the single sideband relative to the stereopilot signal but this has little practical significance.

It is obvious that the block diagram "in Fig. 5 refers to an experimental transmitter which is suitable for examining which system is the best in practice. In its final version the transmitter only needs to be adapted to a system and can therefore be realized on a simpler Thus, a transmitter in which only a modulated carrier wave of, for example, 7/8 fp or 9/8 fp is added to the multiplex signal (see Fig. Ea) may comprise a phase-locked loop 122 which derives a signal of 7 fp or 9 ~ I from the stereopilot signal, further an 8-divider 123 for emitting a pulse-shaped signal of 7/8 fp or 9/8 fp, and a bandpass filter 124 for converting the pulse-shaped signal into a sinusoidal signal of 7/8 fp or 9 / B fp, which sinusoidal signal is fed to the first input 116 of the modulator 117.

In a final transmitter for a system with two modulated subcarriers, the elements 106,113,114,115 in Fig. 5 can be omitted. The oscillator 104 can then be connected directly or via the phase shift device 105 to the first input 107 of the modulator 108 and the output from 108 be directly connected to the first input 106 of the modulator 117, instead of first mixing the pilot signal fp with the signal fp / 8 and then modulate the result with the code signal, it is also possible to modulate the signal fp / 8 with the code signal and then mix it with the stereopilot signal or to modulate the stereopilot signal fp with the code signal and then mix it with the signal fp / 8. Fig. 6 shows the frequency spectrum of the signal emitted by the summing step 121.

The figure shows, calculated from the sound frequency information signal at 0-15 kHz at 19 kHz the stereopilot signal, at 23 kHz to 53 kHz (not visible) the stereo information signal modulated at 38 kHz and at 16.625 and 21.375 kHz the two binary phase modulated subcarriers each having approximately one bandwidth 1200 Hz. It is observed that the amplitudes of the signal components differ considerably much more from each other than is shown in the drawing for the sake of clarity. In practice, the signal components L + R and L-R can be approximately 9 times larger than the stereopilot signal, while the two subcarrier signals e.g. can be 30 times smaller than the stereo pilot signal.

The receiver according to Fig. 7 is particularly suitable for a system in which only a binary phase-modulated subcarrier was transmitted, e.g. at 7/8 fp (16.625 kHz). The conventional receiver elements, such as the high frequency, intermediate frequency and low frequency steps, are not shown in Fig. 7. The composite signal derived from the receiver's frequency discriminator is fed to a bandpass filter 125 which is tuned to the subcarrier frequency of 16.625 kHz and may have an effective goodness factor of e.g. 15. This filter transmits the modulated subcarrier frequency as well as the stereopilot signal itself, which, even if it coincides with a flank of the filter, is still considerably larger than the subcarrier signal. After being amplified in an amplifier 126, the two signals are fed to a first input 127 of a dual function multiplier stage 128.

Primarily, the step 128 operates as a phase detector in a phase locked loop which further comprises a low pass filter 129, a voltage controlled oscillator 130 for 38 kHz and a divider 131 for division by 2, the latter device supplying a square voltage of 19 kHz back to a second input 132 on the multiplication stage 128. This phase-locked loop locks in on the received stereo pilot signal and consequently emits at the output of the divider 131 a square voltage of 19 kHz which is synchronized with the received stereo pilot signal. The low pass filter 129 used to prevent the phase locked loop from being affected by signal components other than the stereo arrow signal may e.g. have a cut-off frequency of 500 Hz and a slope of the filter characteristic above this cut-off frequency of 6 dB / octave.

Second, the multiplication step 128 operates as a 16.625 kHz (7/8 fp) modulated subcarrier mixer step. This subcarrier is mixed with the square wave of 19 kHz (fp) at the input 132, which results in a binary phase modulated intermediate frequency signal of 2.375 kHz (1/8 fp), which is passed through the pass filter 133 such as e.g. . can have a cut-off frequency of 3 kHz and a high-frequency slope of 20 aB / omav. Instead of a 19 kHz voltage controlled oscillator, a 38 kHz voltage controlled oscillator 130 was used followed by a divider 131 for division by 2, since in general a divider for division by 2 emits a more symmetrical square wave than a voltage controlled oscillator. Consequently, step 128 is controlled by a perfectly symmetrical square voltage so that input signal components around the smooth harmonics to 19 kHz, in particular around 38 kHz, do not affect the output signal of step_128. Detection of signal components around 57 kHz in step 128 is sufficiently prevented by the filter 125 which has a suitable attenuation for these signal components.

Accordingly, by means of elements 128, 123, 130 and 131, a filtered stereopilot signal is obtained at the output of the divider 131 and a converted binary modulated subcarrier at the output of step 128. However, it is given that these functions can also be performed by other suitable filter and converter arrangements.

The square wave of 19 kHz from the divider 131 is divided into another divider 134 for division maa s to a square voltage with a frequency of 2.375 kHz (1/8 fp). A binary phase modulated carrier signal of 2.375 kHz therefore occurs at the output of the filter 133 and an unmodulated square edge voltage of 2.375 kHz derived from the stereopilot signal at the output of the divider 134. The modulated carrier signal can now be demodulated synchronously by the method described with reference to Figures 3 and 4. The actual detection is performed in a synchronous detector 135 with a first input 136, where the modulated signal is applied via an amplifier 37, while the unmodulated voltage is applied to a second input 138 via a controllable phase shift device 139 The detected code signals are filtered in a low-pass filter 140, which has, for example, a cut-off frequency of 350 Hz and a slope of the filter characteristic of 20 dB / octave, and are then converted into square pulses by means of a pulse shaper 141 and then fed to a decoder. 142 which converts the binary transmitter identifier thus obtained signal to signals suitable for supplying a load 143. The load 143 may differ depending on the information contained in the code. If the code contains information about the received transmitter and / or the received program, the load 143 may consist of a reproducing device for reproducing this information, so that e.g. the usual reconciliation scale can be dispensed with. Alternatively, the load 143 may include an automatic transmitter search circuit, so that the receiver is only tuned to the stations transmitting a given type of program, e.g. classical music. If the code contains a time indication, the device 143 can e.g. connect or disconnect a tape recorder connected to it at a preset time. If the code is a semaphone signal, the device 143 is constituted by a semaphone receiver.

The phase shift device 139 serves to eliminate all phase errors that can be generated between the 2.375 kHz modulated signal and the 2.375 kHz unmodulated fi transverse voltage. These phase errors can occur at the divider 10 in the transmitter and the divider 134 in the receiver or due to delay time differences in the different filters, e.g. in the filter 125. For this phase control, the binary phase modulated signal of 2.375 kHz is converted by means of a squaring device 141 and a pulse shaper 145 into a square voltage of 4.75 kHz. The unmodulated voltage from the phase shift device 139 is converted to a square voltage of 4.75 kHz by means of a frequency multiplier 146. The two square voltages of 4.75 kHz are compared in a phase detector 147 which generates a control signal of these square voltages, which after filtering in a low-pass filter 148 and gain in an amplifier 149 are fed to the control input 150 of the controllable phase shift device 139. The phase shift device 139 thus ensures that the square voltage of 2.375 kHz and the signal of 2.375 kHz, which are fed to the synchronous detector, have the same phase (or are 180 ° phase shifted).

The fixed ambiguity that still exists can again be ignored by using a code that is insensitive to this. The phase shift device 139 as well as the phase shift device 36 in Figs. 3 and 4 can e.g. consist of two monostable circuits arranged in cascade, the time constant of the first circuit being regulated by the control signal and the time constant of the second circuit being equal to half a period of the signal to be delayed, the first circuit being started by the incoming signal and the second circuit of the rear flank from the first circuit. Such a phase shift device can phase shift the signal almost 3600, which is more than sufficient since the signal is to be shifted 1800.

A preferred embodiment of a receiver for receiving signals containing two binary phase modulated subcarriers on each side of the stereopilot signal, the stereopilot signal being squared to the resultant of the two subcarriers, may be of the same shape as in Fig. 7. it being understood that the filter 125 is not tuned to a subcarrier but to the stereo arrow signal, while the bandwidth of the filter must be large enough to transmit the two subcarriers. On the other hand, the off-band attenuation, especially for signals around 57 kHz, must be high enough to prevent interference.

In a receiver for receiving signals with a binary phase modulated subcarrier on each side of the stereopilot signal, the resultant of the two subcarriers being in phase with the stereopilot signal, it is not possible to use the phase detector (128) in the phase read loop. also to convert the signal because the phase detector and the mixer stage then have to be controlled with stereopilot signals which are offset 900 relative to each other. Fig. 8 shows a possible embodiment of such a receiver. in which figure corresponding elements have been given the same reference numerals as in Fig. 1, the output signal from the amplifier 126 is then fed to the phase detector 128 and to a second detector 151, which serves as a mixer stage. In this mixer step, the input signal is mixed with a square voltage of 19 kHz which is derived by frequency division with a divider 152 for division with two from the oscillator 150 for 38 kHz.

The two parts 131 and 152 are tuned so that they emit voltages of 19 kHz, the capacitors 90 of which are relatively smooth 90 relative to each other.

Claims (18)

7862031-0 patent requirements. A method of code signaling in an FM broadcasting system, wherein a multiplex signal which is frequency modulated on a main carrier is transmitted on the transmitter side, which multiplex signal comprises: an audio frequency information signal (L + R), in stereo transmission, a stereo information signal (L - R) which is modulated on a suppressed subcarrier (38 kHz), and a stereopilot signal (fp = 19 kHz) whose frequency is between the frequency spectra of the sound frequency information signal (L + R) and the modulated stereo information signal (L - R) and which serves to demodulate the stereo information signal (L - R) ), and a binary code signal which is modulated on a further subcarrier located outside said frequency spectra and having an amplitude which does not cause the main carrier to deviate by more than 1 kHz, characterized in that said further subcarrier is an harmonic to an undertone of the stereopilot signal, which does not coincide with a harmonic to the stereopilot signal and is diverted on the transmitter side from sa mma frequency source as the stereopilot signal and that the code signal is binary phase modulated on this subcarrier. 2. A method according to claim 1, characterized in that the frequency of said further subcarrier is in the middle between two harmonics of the stereopilot signal. 3. A method according to claim 1, characterized in that the further subcarrier with the modulated code signal is located in at least one of the two halves of the frequency range, which is divided into two by the stereopilot signal, between the upper limit of the frequency spectrum of the audio frequency information signal and the lower the limit of the frequency spectrum of the modulated stereo information signal and that the modulated code signal has an amplitude which causes the main carrier to deviate less than 1 kHz, preferably 0.25 kHz. 4. A method according to claim 3, characterized in that a subcarrier which is binary phase modulated by the code signal is located in each of the two halves of the frequency range, which is divided into two by the stereopilot signal. 5. A method according to claim 4, characterized in that the two subcarriers modulated by the code signal have equal amplitudes and such a phase relative to the stereopilot signal that they together with the stereopilot signal form a signal which is generated by quadrature modulation of the stereo pilot signal with a subcarrier derived from the stereopilot signal, this subcarrier being binary phase modulated by the code signal. 6. A method according to claim 3, characterized in that the subcarrier, which is modulated with the code signal, is at a distance from the stereopilot signal which is equal to 1/8 x the frequency of the stereopilot signal. A receiver (24-52; 125-152) in a broadcasting system for performing the method of code signaling according to any one of claims 1-6, comprising a frequency discriminator (26) for demodulating the received main carrier, characterized by a synchronous demodulator (39; 135) having first (38; 138) and second (41; 136) inputs and one output, a first transmission path (32, 34-37, 52; 128-131, 134, 139) connected to the frequency discriminator (26) to supply the binary phase modulated code signal to the first input (38, 138) of the synchronous demodulator (39; 135), a second transmission path (33, 40; 133, 137) connected to the frequency discriminator (26) for supplying an unmodulated wave, which is synchronized with the stereopilot signal, to the second input (41; 136) of the synchronous demodulator (39; 135) and to the output circuit (42, 140) of the demodulated binary code signal. Receiver according to claim 7, characterized by a device (46, 51; 144-149) which is connected between the two transmission paths (32, 34-37, 52; 128-131, 134, 139 and 33, 40, respectively; 133, 137) to generate a control signal depending on the relative phase between the carrier of the binary phase modulated signal applied to the first input (38) of the synchronous demodulator (39) and the unmodulated wave supplied to the synchronous demodulator ( 39) second input (41), and an electronically controllable phase shift device (36, 139) connected in one of the two transmission paths (32, 34-37, 52; 128-131, 134, 139 and 33, 40, respectively; 133, 137) and controlled by said control signal to control said relative phase. Receiver according to claim 8, characterized in that the device (46-50) for generating a control signal comprises a phase detector (47) with first (48) and second (49) inputs and one output and a frequency doubling circuit. (50), which is connected between a connection point in the second transmission path (33, 40) and the second input (49) of the phase detector (47), and that a connection point in the first transmission path (32, 34-37 ) is connected to the first input (48) of the phase detector (47) and that the output of the phase detector (47) controls the electronically controllable phase shift device (36). Receiver according to claim 8, characterized in that the device (46-49, 51) for generating a control signal comprises a phase detector (47) with first (48) and second (49) inputs and an output, wherein it the first input (48) is connected via a first connection to a connection point (36, 52) in the first transmission path (32, 34-36, 62) and the second input (49) is connected via a second connection to a connection point (33 , 41) in the second transmission path (33), while the output of the phase detector (47) controls the electronically controllable phase shift device (36) via a third connection, and a phase converter (51) which is connected in one of said connections and controlled by the output signal of the synchronous demodulator (39) Receiver according to claim 9 or 10, characterized in that the electronically controllable displacement device (36) is connected in the first transmission path (32, 34-37, 52) before said connection point in the first transmission path (32, 34-37, 52). Receiver according to claim 7, characterized in that a filter and conversion device (128-133, 151, 152) is connected to the output of the frequency discriminator (26) for filtering out the stereopilot signal (fp) and for using the filtered out stereopilot signal (fp) converting the subcarrier which is binary phase modulated by the code signal into an intermediate frequency carrier which is binary phase modulated by the code signal and has a frequency equal to the frequency distance between the subcarrier and the stereopilot signal, that said first transmission path (134, 139) being connected output on the filter and conversion device (128-133, 151, 152) and includes one or more frequency dividers (134) for feeding an unmodulated intermediate frequency carrier synchronized with the filtered out stereopilot signal to the synchronous demodulator (135) first input (138) and that an output of the filter and conversion device (128-133, 151, 152) is connected to the second transmission path (133, 137) for feeding the binary phase modulated intermediate frequency wave to the second input (136) of the synchronous demodulator (135). 7802031-0 22 Receiver according to claim 12, characterized in that said filter and conversion device (128-133) comprises a phase-locked loop formed with a voltage-controlled oscillator (130), a filter (129) and a phase detector (128). for filtering out the stereopilot signal (fp), the output signal from the frequency discriminator (26) being fed to a first input (127) on the phase detector (128) and the filtered out stereopilot signal (fp) derived from the voltage controlled oscillator (130) being fed to a second input (132) on the phase detector (128) and that the phase detector (128) in the phase locked loop also functions as a converter for the modulated subcarrier because said second transmission path (133-137) is connected to the output of the phase detector (128). . Receiver according to claim 12, characterized in that a bandpass filter (125) tuned to the subcarrier, which is modulated with the code signal, is connected between the output of the frequency discriminator (26) and the filter and conversion device (128-133, 151, 152). ) input, a slope of the filter characteristics of this bandpass filter (125) conducts the stereopilot signal. A transmitter in a broadcasting system for performing the code signaling method according to any one of claims 1-6, comprising a device for generating an audio frequency information signal, at a stereo transmitter, a stereo information signal modulated on a suppressed stereo subcarrier, and an oscillator for generating a stereopilot signal whose frequency is between the frequency spectra of the audio frequency information signal and the modulated stereo information signal, characterized by a source (21; 120) of binary code signals and a modulation signal generator (18-20) connected to this source (21). 120) and to the oscillator (14-104) for generating a subcarrier which is binary phase modulated with the code signal, which subcarrier is an harmonic of an undertone of the stereopilot signal which does not coincide with a harmonic of this pilot signal and is outside said frequency spectra. Transmitter according to claim 15, characterized in that the modulation signal generator (18-20) generates a subcarrier which is binary phase modulated with the code signal and is located in at least one of the two halves of the frequency range, which is divided into two of the stereopilot signal , between the upper limit of the frequency spectrum of the audio frequency information signal and the lower limit of the frequency spectrum of the modulated stereo information signal. 7802031-0 za Transmitter according to claim 15, characterized in that the modulation signal generator (18-20) comprises a subcarrier generator (18, 19) for generating said subcarrier and a modulator (20) connected to the subcarrier generator (18, 19) and the binary code signal source (21) for binary phase modulating the subcarrier with the code signals. Transmitter according to claim 15, characterized in that the modulation signal generator (105-117) comprises a frequency generator (109-111) connected to the oscillator (104) for generating a mean frequency carrier with a frequency equal to the frequency distance the subcarrier and the stereo signal, and first and second modulators (108, 117) each having first and second inputs and one output, the output of the first modulator (108) being connected to the first input of the second modulator (117) (116). , while of the two inputs (107, 112) of the first modulator (108) and the second input (118) of the second modulator (117) one input is connected to the oscillator (104), a second input (112) to the frequency generator (109). -111) and a third input (118) to the binary code signal source.