RU2816455C2 - Unit for aerosol generation device - Google Patents

Abstract

FIELD: electrical engineering; induction heaters for an aerosol generation device.

SUBSTANCE: technical result consists in creating an induction heater that allows a changing current from a constant voltage source to produce induction heating. Achieved by the fact that the induction heater has an induction heating circuit, where the circuit has an inductive element for inductively heating a current-collecting structure for heating an aerosol-forming material; a capacitive element; and a switching device that, in use, switches between the first state and the second state to receive a varying current from a constant voltage source flowing through the inductive element to cause inductive heating of the current-collecting structure.

EFFECT: induction heater that allows a changing current from a constant voltage source to produce induction heating.

22 cl, 18 dwg

Description

Field of technology to which the invention relates

The present invention relates to an apparatus for an aerosol generation device.

State of the art

Smoking products such as cigarettes, cigars, etc. During their use, tobacco is burned to produce tobacco smoke. Attempts have been made to provide alternatives to these products that burn tobacco by creating products that release the compounds without burning. Examples of such products are heating devices that release compounds by heating, but not burning, the material. The substance may be, for example, tobacco or other non-tobacco products, which may or may not contain nicotine.

The essence of the invention

According to a first aspect of the present invention, there is provided an apparatus for an aerosol generation apparatus, the apparatus comprising: a resonant induction heating circuit, the circuit comprising an inductive element for inductively heating a current-receiving structure to heat an aerosol-forming material to thereby produce an aerosol; capacitive element; and a switching device that, in use, switches between the first state and the second state to receive varying current from the constant voltage source and flow it through the inductive element to cause inductive heating of the current-receiving structure; and a control device, wherein the control device is configured to switch the switching device from a first state to a second state in response to a first voltage condition detected in the circuit, and the control device is configured to switch the switching device from a second state to a first state in response to the second voltage condition detected in a circuit.

The first voltage condition may indicate the amount of magnetic energy that is stored in the inductive element due to direct current flowing through the inductive element.

The switching device in the first state can allow direct current to flow through the inductive element and thereby store magnetic energy in the inductive element, and the switching device in the second state can prevent direct current from flowing through the inductive element, so that if the switching device is in the second state, it can oscillate between the inductive element and the capacitive element.

The control device may include a voltage comparator, and the voltage comparator may be configured to detect the first voltage condition by comparing a voltage indicative of the amount of direct current flowing through the inductive element with a control voltage.

The circuit may also contain a resistor, and the voltage, indicating the amount of direct current flowing through the inductive element, depends on the voltage across the resistor.

The control voltage can be changed to control the power supplied to the circuit.

The apparatus may include a controller configured to sense power supplied to the circuit and compare the power supplied to the circuit to a target power, and control a control voltage based on the result of comparing the power supplied to the circuit to the target power.

The second voltage condition may indicate that, from the moment the switching device was switched from the first state to the second state, a predetermined fraction of a cycle of current oscillation between the inductive element and the capacitive element has been completed.

The control device may include a zero voltage detector, and the zero voltage detector may be configured to detect a second voltage condition, and the second voltage condition may be a zero voltage condition or a near zero voltage condition at the switching device, which when detected by the zero voltage detector indicates that from the moment the switching device was switched from the first state to the second state, half a cycle of current oscillation between the inductive element and the capacitive element was completed.

The control device may include a flip-flop that can be switched between two states, and the state of the switching device may depend on the state of the flip-flop, and the control device may be configured to detect a first voltage condition and a second voltage condition, as well as a change in the state of the flip-flop to change state of the switching device.

The control device may include a comparator configured to detect a first voltage condition, and the flip-flop may be configured to receive a first input from the comparator if the comparator detects a first voltage condition, and a second input from a zero voltage detector if the zero voltage detector detects a second voltage condition.

The switching device may comprise a field effect transistor.

The control device may be configured to switch the state of the FET by selectively applying a voltage to the gate terminal of the FET.

The inductive element and the capacitive element can be arranged in parallel to each other in an induction heating circuit.

The inductive element, the capacitive element and the switching device may be located in the first resonator section, and the device may also include a second resonator section containing a second inductive element, a second capacitive element and a second switching device, wherein: the second inductive element is configured to inductively heat the current collecting structure for heating the aerosol-forming material to thereby produce an aerosol; the second switching device, in use, switches between the first state and the second state to receive a varying current from the constant voltage source and flow it through the second inductive element to cause inductive heating of the current collecting structure; and the control device may be configured to do the following when the first resonator section is active: switch the first switching device from a first state to a second state in response to detecting a first voltage condition in a circuit of the first resonator section; and switch the first switching device from the second state to the first state in response to detecting a second voltage condition in the first section of the resonator; and the control device may be configured to do the following when the second resonator section is active: switch the second switching device from a first state to a second state in response to detecting a first voltage condition in a circuit of the second resonator section; and switch the second switching device from the second state to the first state in response to detecting a second voltage condition in the second resonator section.

The apparatus may include a controller configured to selectively activate the first resonator section and the second resonator section such that only one of the first resonator section and the second resonator section is active at any given time.

In accordance with the second aspect of the present invention, there is provided an aerosol generating apparatus comprising an apparatus in accordance with the first aspect of the present invention.

The device may be a tobacco heating device, also known as a non-combustion heating device.

According to a third aspect of the present invention, there is provided a control device for controlling an induction heating circuit of an aerosol generation device, wherein: the control device is configured to control a switching device in the induction heating circuit to, when in use, cause the switching device to switch between a first state and a second state to obtain alternating current from a direct current source and pass it through an inductive element in an induction heating circuit, causing induction heating of the current collector structure; and wherein the control device is configured to switch the switching device from a first state to a second state in response to a first voltage condition detected in the circuit, and the control device is configured to switch the switching device from a second state to a first state in response to a second voltage condition voltage detected in the circuit.

The control device may be configured to: detect a first voltage condition and a second voltage condition in the induction heating circuit; and in response to detecting the first voltage condition, switch the switching device from the first state to the second state; and in response to detecting the second voltage condition, switch the switching device from the second state to the first state.

According to a fourth aspect of the present invention, there is provided an aerosol generation system comprising an aerosol generating device according to the third aspect and an article containing an aerosol-forming material to be heated by the device in use, thereby producing an aerosol.

The aerosolized material may be tobacco material.

According to a fifth aspect of the present invention, there is provided an apparatus for an aerosol generation device, comprising: an induction heating circuit comprising an inductive element for inductively heating a current-receiving structure to heat an aerosol-forming material to thereby produce an aerosol; capacitive element; and a switching device that, in use, switches between the first state and the second state to receive varying current from the constant voltage source and flow it through the inductive element to cause inductive heating of the current-receiving structure; and a controller configured to: measure a constant voltage and a constant current supplied to the induction heating circuit by a constant voltage source, and determine from the measured direct voltage and direct current a power supplied to the circuit; and control the switching of the switching device based on a comparison of the determined power supplied to the circuit with the target power.

The controller may be configured to control power supplied to the circuit by controlling the switching of the switching device.

The switching device in the first state can allow direct current to flow through the inductive element and thereby store magnetic energy in the inductive element, and the switching device in the second state can prevent direct current from flowing through the inductive element, so that if the switching device is in the second state, it can oscillate between the inductive element and the capacitive element.

The controller may be configured to control the power supplied to the circuit by controlling the amount of direct current that builds up in the inductive element before the controller switches the state of the switching device from a first state to a second state.

The apparatus may comprise a control device configured to detect that the inductive element has been allowed to receive a predetermined amount of direct current by detecting a first voltage condition in the circuit.

The control device may include a comparator configured to detect the first voltage condition by comparing a voltage in the circuit with a control voltage, and the controller may be configured to adjust the control voltage to control power supplied to the circuit.

The control voltage may be a result of a time-varying voltage output by the controller, wherein the time-varying voltage has a duty cycle, and the controller is configured to adjust the control voltage by adjusting the duty cycle of the time-varying voltage.

The controller may be configured to control the power supplied to the circuit by determining the power supplied to the circuit during a first time interval and adjusting the power supplied to the circuit during a subsequent time interval based on a comparison of the determined power supplied to the circuit at during the first predetermined time interval, and the target power.

The controller may be configured to control the power supplied to the circuit over a usage session comprising multiple predetermined intervals by comparing the determined power supplied to the circuit during one predetermined interval with a target power.

The first predetermined time interval and/or the subsequent predetermined time interval may have a duration of from 1/80 s to 1/20 s, or about 1/64 s.

The controller may be configured to increase the power supplied to the circuit during a subsequent time interval if the power supplied during the first predetermined time interval is less than the target power.

The controller may be configured to determine power supplied to the circuit based on a measured voltage indicative of the current received from the constant voltage source over a predetermined interval.

The voltage, indicative of the current received from the constant voltage source, may be substantially constant during the first predetermined interval.

The controller may be configured to control the power supplied during a subsequent time interval by adjusting the control voltage to a predetermined amount.

The controller may be configured to set a control voltage equal to a first value for a first predetermined interval, wherein the first value is less than a control voltage value corresponding to the target power.

The target power may be a target power range, such as a range of 10 to 30 W, or a range of 15 to 25 W, and the controller may be configured not to adjust control of the switching device if the determined supplied power is within the target power range.

The controller may be configured to adjust the target power throughout a device use session.

The controller may be configured to monitor the temperature of the susceptor structure and reduce the target power at a time during a use session when the temperature of the susceptor structure has reached a predetermined target temperature.

The target power may remain constant such that the controller is configured to control the switching device to maintain substantially constant power supplied to the circuit if the voltage supplied by the constant voltage source changes.

The inductive element, the capacitive element and the switching device may be located in the first resonator section, and the device may also include a second resonator section containing a second inductive element, a second capacitive element and a second switching device, wherein: the second inductive element is configured to inductively heat the current collecting structure for heating the aerosol-forming material to thereby produce an aerosol; the second switching device, in use, switches between the first state and the second state to receive a varying current from the constant voltage source and flow it through the second inductive element to cause inductive heating of the current collecting structure; and the controller may be configured to selectively activate the first resonator section and the second resonator section such that only one of the first resonator section and the second resonator section is active at any given time, and the controller may be configured to: measure a constant voltage and a constant current, supplied to the first resonator section or the second resonator section, and determine from the measured direct voltage and direct current the power supplied to the circuit; and control the switching of the switching device of the active section of the resonator based on a comparison of a certain power supplied to the circuit with a target power.

The controller may be configured to determine the power supplied to the circuit during a use session by determining the power supplied to the first resonator section or the second resonator section.

The controller may be configured, in a first part of a use session, to determine the power supplied to the circuit by determining the power supplied to the first resonator section, and in a second part of the use session, to determine the power supplied to the circuit by determining the power supplied to the second resonator section .

According to a sixth aspect of the present invention, there is provided an aerosol generating apparatus comprising an apparatus according to the first aspect of the present invention.

The device may be a tobacco heating device, also known as a non-combustion heating device.

According to a seventh aspect of the present invention, there is provided a method for an apparatus controller for an aerosol generating device, the apparatus comprising: an induction heating circuit comprising an inductive element for inductively heating a current-receiving structure to heat an aerosol-forming material to thereby produce an aerosol; capacitive element; and a switching device that, in use, switches between a first state and a second state to receive a varying current from the constant voltage source and flow it through the inductive element to cause inductive heating of the current-receiving structure; and controller; wherein the method comprises: measuring a constant voltage and a constant current supplied to the induction heating circuit by a constant voltage source, and determining from the measured direct voltage and direct current the power supplied to the circuit; and controlling switching of the switching device based on a comparison of the determined power supplied to the circuit with the target power.

In accordance with an eighth aspect of the present invention, there is provided a set of machine-readable instructions that, when executed, perform a method in accordance with the third aspect.

In accordance with a ninth aspect of the present invention, there is provided a computer-readable storage medium comprising a set of instructions in accordance with the fourth aspect.

According to a tenth aspect of the present invention, there is provided an aerosol generation system comprising an aerosol generating device according to the second aspect and an article containing an aerosol-forming material to be heated by the device in use, thereby producing an aerosol.

In the aerosol generation system, the aerosol generating material comprises tobacco material.

According to other examples of the present invention, there is provided an apparatus for an aerosol generating device, comprising: a heating circuit comprising: a heating device configured to, in use, heat an aerosol-forming material to thereby produce an aerosol; and a power supply for supplying power to the heating circuit for heating the aerosol-forming material; and a controller configured to: determine power supplied to the heating circuit to heat the aerosol-forming material; control the power supplied to the heating circuit based on a comparison of the determined supplied power with the target power, and wherein the controller is configured to adjust the target power throughout a session of use of the device.

The controller may be configured to reduce the target power during a usage session. The controller may be configured to reduce the target power from a first value during a first portion of a use session to a second value during a second portion of a use session. The heating device may comprise one or more heating zones, and the first portion of a use session may comprise the following: the device applies power to substantially increase the temperature of the one or more heating zones. The second part of the use session may include the following: the device supplies power to substantially maintain the temperature of the heating zones. The first target power value can be from 15 W to 23 W. The second target power value can be from 9 W to 13 W. The target power may be a range. The first target power value may be approximately 20 W or may be in the range of 20 W to 21 W. The second target power value may be approximately 12 W or may be in the range of 12 W to 13 W.

The controller may be configured to reduce the supplied power if the supplied power exceeds the target power, and to reduce the supplied power if the supplied power is less than the target power. If the target power is a range, then the controller may be configured to not adjust the supplied power if it is determined that the supplied power is within the target range. The heating device may include one or more resistive heating elements or one or more inductive heating elements. One or more heating elements may be provided to heat one or more heating zones. The determined power may be the power supplied to one or both heating elements.

According to an eleventh aspect of the present invention, there is provided an apparatus for an aerosol generation device, comprising: an induction heating circuit comprising a first inductive element and a second inductive element, the first inductive element and the second inductive element being configured to inductively heat a current-receiving structure to heat an aerosol-generating material. , thereby obtaining an aerosol; and a controller for controlling activation of the first inductive element and the second inductive element, wherein: the controller is configured to selectively activate the first inductive element and the second inductive element such that only one of the first inductive element and the second inductive element is active at any time; and the controller is configured to determine predetermined intervals at which to activate the first inductive element and the second inductive element.

The controller may be configured to determine at predetermined intervals which of the first inductive element and the second inductive element to activate by determining once for each of the multiple predetermined intervals which of the first inductive element and the second inductive element is to be activated for the first predetermined interval from several predefined intervals.

The pantograph may comprise a first pantograph zone and a second pantograph zone, and the first inductive element may be configured to heat the first pantograph zone, and the second inductive element may be configured to heat the second pantograph zone, and the controller may be configured to determine which of the first pantograph element and the second induction element must be activated based on which of the first pantograph zone and the second pantograph zone is to be heated during the first predetermined interval.

The controller may be configured to determine which of the first pantograph zone and the second pantograph zone needs to be heated during a first predetermined interval based on comparing the measured temperature of the first pantograph zone to the first target temperature and comparing the measured temperature of the second pantograph zone to the second target temperature.

The controller may be configured to: determine whether a temperature value of the first pantograph zone is below the first target temperature; determining whether the temperature value of the second pantograph zone is below the second target temperature; activate the first inductive element during a first predetermined interval if the controller determines that the temperature of the first pantograph zone is lower than the first target temperature and the temperature of the second pantograph zone is not lower than the second target temperature; activate the second inductive element during the first predetermined interval if the controller determines that the temperature of the second pantograph zone is lower than the second target temperature, and the temperature of the first pantograph zone is not lower than the first target temperature; and activate one of the first inductive element and the second inductive element during a first predetermined interval if the controller determines that the temperature of the first pantograph zone is lower than the first target temperature, and the temperature of the second pantograph zone is lower than the second target temperature.

The controller may be configured to activate one of the first inductive element and the second inductive element for each predetermined interval of one or more intervals subsequent to the first predetermined interval if the measured temperature of the first pantograph zone remains below the first target temperature, and the measured temperature of the second the pantograph zone remains below the second target temperature for one or more predetermined intervals subsequent to the first predetermined interval, such that the first inductive element and the second inductive element are alternately active for each predetermined interval.

Preset intervals can range from 1/80 sec to 1/20 sec or approximately 1/64 sec in duration.

The circuit may comprise: a first resonator section comprising a first inductive element, a first capacitive element, and a first switching device that, in use, switches between a first state and a second state to receive a varying current from a constant voltage source and flow it through the first inductive element to cause an induction heating the current-collecting structure, and a second resonator section comprising a second inductive element, a second capacitive element, and a first switching device that, in use, switches between a first state and a second state to receive a varying current from a constant voltage source and flow it through the second inductive element to cause induction heating of the current-receiving structure; and the controller may be configured to selectively activate the first resonator section and the second resonator section such that only one of the first resonator section and the second resonator section is active at any time to selectively activate the first inductive element and the second inductive element.

The circuit may include control means configured to control the first switching device and the second switching device.

The control means may comprise a first driver for controlling the first switching device and a second driver for controlling the second switching device, and the controller is configured to activate the first resonator section by selectively applying a signal to the first driver and activating the second resonator section by selectively applying a signal to the second driver.

The control means may be configured to switch the active resonator section switching means from a first state to a second state in response to the control means detecting a first voltage condition in the active resonator section.

The control means may be configured to switch the active resonator section switching means from a second state to a first state in response to the control means detecting a second voltage condition in the active resonator section.

The first voltage condition may indicate the amount of magnetic energy that is stored in the active inductive element due to direct current flowing through the active inductive element.

The second voltage condition may indicate that from the moment the resonator active section switching device was switched from the first state to the second state, a predetermined portion of the current oscillation cycle between the inductive element and the capacitive element of the resonator active section has been completed.

According to a twelfth aspect of the present invention, there is provided an aerosol supply device comprising an apparatus according to the first aspect of the present invention.

The device may be a tobacco heating device, also known as a non-combustion heating device.

According to a thirteenth aspect of the present invention, there is provided a method for an apparatus controller for an aerosol generation apparatus, the apparatus comprising: an induction heating circuit comprising a first inductive element and a second inductive element, the first inductive element and the second inductive element being configured to inductively heat a current-receiving structure so as to heat the aerosol-forming material to thereby produce an aerosol; and a controller, wherein the controller is configured to control activation of the first inductive element and the second inductive element for heating the current collector structure; wherein the method comprises the following: selectively activating the first inductive element and the second inductive element, so that at any time only one of the first inductive element and the second inductive element is active; and determining predetermined intervals at which to activate the first inductive element and the second inductive element.

In accordance with a fourteenth aspect of the present invention, there is provided a set of machine-readable instructions that, when executed, perform the method in accordance with the third aspect.

In accordance with a fifteenth aspect of the present invention, there is provided a computer-readable storage medium comprising a set of instructions in accordance with the fourth aspect.

According to a sixteenth aspect of the present invention, there is provided an aerosol generation system comprising an aerosol generating device according to the second aspect and an article containing an aerosol-forming material to be heated by the device in use, thereby producing an aerosol.

Alternatively, the aerosol-forming material comprises tobacco material.

Other features and advantages of the invention will become apparent from the following description of preferred embodiments of the invention, given by way of example with reference to the accompanying drawings.

Brief description of drawings

In fig. 1 is a front view of an example of an aerosol generation device;

in fig. 2 is a front view of the aerosol generating apparatus shown in FIG. 1, with outer cover removed;

in fig. 3 is a cross-sectional view of the aerosol generation device shown in FIG. 1;

in fig. 4 is a sectional view of the aerosol generation device shown in FIG. 2;

in fig. 5A is a cross-sectional view of an example of a heating assembly in an aerosol generating apparatus;

in fig. 5B is an enlarged view of a portion of the heating assembly shown in FIG. 5A;

in fig. 6 is a schematic diagram of an example of an induction heating circuit for the aerosol generating apparatus shown in FIG. 1-5B;

in fig. 7A is a schematic representation of the current passing through an inductor for the example induction heating circuit shown in FIG. 6;

in fig. 7B is a schematic representation of the voltage across a current sensing resistor for the example induction heating circuit shown in FIG. 6;

in fig. 8 is a schematic representation of the voltage across the switching device of the circuit shown in FIG. 6;

in fig. 9 is another schematic diagram of an example induction heating circuit for the device shown in FIG. 1-5B;

in fig. 10-13 show various parts of an example control device for the example induction heating circuit shown in the previous figures;

in fig. 14 is a flowchart of an example method of controlling aspects of an induction heating circuit for an illustrative example of an induction heating circuit;

in fig. 15 is a flowchart of another example of a method for controlling aspects of an induction heating circuit for an illustrative example of an induction heating circuit; And

in fig. 16 shows a schematic representation of the susceptor temperature and the target power that must be supplied to heat the susceptor during a session of use of the aerosol generation device.

Detailed Description of the Invention

As used herein, the expression "aerosol-forming material" includes substances that release volatile components when heated, typically in the form of an aerosol. The aerosol-forming material includes any tobacco-containing substances and may, for example, include one or more of the following: tobacco, tobacco derivatives, ground tobacco, reconstituted tobacco, or tobacco substitutes. The aerosol-forming material may also include other non-tobacco products, which may or may not contain nicotine, depending on the product. The aerosol-forming material may, for example, be in solid, liquid, gel, or wax form. The aerosol-forming material may also, for example, be a combination or mixture of materials. The aerosol-forming material may also be known as "smoking material".

A device is known that heats an aerosol-forming material to vaporize at least one component of the aerosol-forming material, typically to form an aerosol that can be inhaled without burning or igniting the aerosol-forming material. Such a device is sometimes described as an “aerosol generating device”, “aerosol providing device”, “non-combustion heating device”, “tobacco product heating device” or “tobacco heating device” and the like. Likewise, there are also so-called electronic cigarettes, which typically vaporize an aerosol-forming material in liquid form, which may or may not contain nicotine. The aerosol-forming material may be provided in the form of or as part of a rod, cartridge or cassette or the like that can be inserted into the device. The heater for heating and evaporating the aerosol-forming material may be configured as a "permanent" part of the device.

The aerosol supply device may receive an article containing an aerosol-forming material for heating. "Product" in this context is a component that includes or contains, in use of a device, an aerosol-forming material that is heated to vaporize the aerosol-forming material, and, optionally, other components used. The user may insert the article into the aerosol delivery device before it is heated to produce an aerosol, which is then inhaled by the user. The product, for example, can be of a predetermined or strictly defined size, that is, it can be placed in the heating chamber of the device, which is sized to receive the product.

Described herein are examples of an apparatus for an aerosol generation system configured to produce an aerosol by inductively heating an aerosol-generating material. In examples, the device includes an inductive element, which may be an inductor, for heating the current collecting structure. The current-receiving structure is configured to heat the aerosol-forming material to form an aerosol. In examples, the device allows the generation of alternating current through an inductive element from a direct voltage source such as a battery. To provide alternating current from a constant voltage source, a switching device is provided that can switch between a first state and a second state. To enable the switching device to switch between a first state and a second state, a control device is provided that is configured to switch the switching device from a first state to a second state in response to a first voltage condition detected in the circuit. The control device is also configured to switch the switching device from a second state to a first state in response to detecting a second circuit voltage condition.

Accordingly, an induction heating apparatus is proposed for an aerosol generation system that generates an aerosol for inhalation by a user, which allows varying current from a constant voltage source to produce induction heating. A control device that actuates the switching device in response to detected voltage conditions in the circuit allows the circuit to "self-oscillate" to produce an alternating current, as will be described in detail below. The examples provided in this document may provide for the creation of alternating current without the use of a special component, such as an inverter, to create the alternating current. In some examples, at least one of the first and second voltage conditions is at least partially dependent on the resonant frequency of the circuit, and as such, the self-oscillating mode of the circuit takes into account the resonant frequency of the circuit, without having to determine the resonant frequency by a controller or the like.

In some examples, the control device detects voltage conditions on the circuit and controls the switching device accordingly. In some examples, the apparatus also includes a controller that controls switching of the switching device using a control device. The controller, for example, may determine the power supplied to the circuit and control the switching device based on such determinations of the power supplied to the circuit. Additional features and an example of an apparatus for an aerosol generation device will become apparent from the following description. An example of an aerosol generation apparatus will now be described in detail.

In fig. 1 shows an example of an aerosol supply device 100 for generating an aerosol from an aerosol-forming medium/material. In general terms, device 100 can be used to heat a removable article 110 containing an aerosol generating medium to produce an aerosol or other respirable medium that is inhaled by the user of device 100.

The device 100 includes a housing 102 (in the form of an outer cover) that surrounds and houses various components of the device 100. The device 100 has an opening 104 at one end through which the article 110 can be inserted to be heated by a heating unit. In use, article 110 may be fully or partially inserted into a heating assembly, where it may be heated by one or more components of the heating assembly.

The device 100 in this example includes a first end member 106 that includes a cover 108 that is movable relative to the first end member 106 to cover the opening 104 when the product 110 is not in place. In fig. 1, cover 108 is shown in an open configuration, however, cover 108 may change to a closed configuration. For example, the user may slide the cover 108 in the direction of arrow "A".

The device 100 may also include a user-operable control 112, such as a button or switch, that, when pressed, activates the device 100. For example, the user may turn on the device 100 using the switch 112.

The device 100 may also include an electrical component, such as a jack/port 114, that can receive a cable for charging the battery of the device 100. For example, the jack 114 may be a charging port, such as a USB charging port. In some examples, jack 114 may be used in addition to or as an alternative to transmitting data between device 100 and another device, such as a computing device.

In fig. 2 shows the device 100 shown in FIG. 1, with the outer cover 102 removed. The device 100 detects the longitudinal axis 134.

As shown in FIG. 2, a first end member 106 is located at one end of the device 100, and a second end member 116 is located at the opposite end of the device 100. The first and second end members 106, 116 together at least partially define the end surfaces of the device 100. For example, the bottom surface of the second end member member 116 at least partially defines the bottom surface of the device 100. The edges of the outer cover 102 may also define a portion of the end surfaces. In this example, cover 108 also defines a portion of the top surface of device 100. In FIG. 2 also shows a second circuit board 138 associated with control element 112.

The end of the device closest to the opening 104 may be known as the proximal end (or end closer to the mouth) of the device 100 because it is closest to the user's mouth when in use. In use, the user inserts the article 110 into the opening 104, operates the user control 112 to begin heating the aerosol-forming material, and draws in the aerosol generated in the device. This causes the aerosol to pass through the device 100 along a flow path to the proximal end of the device 100.

The other end of the device furthest from the opening 104 may be known as the distal end of the device 100 because it is furthest away from the user's mouth when in use. When the user draws in the aerosol generated in the device, the aerosol passes from the distal end of the device 100.

The device 100 also includes a power source 118. The power source 118 may be, for example, a battery, such as a rechargeable battery or a non-rechargeable battery. Examples of suitable batteries include, for example, a lithium battery (such as a lithium-ion battery), a nickel battery (such as a nickel-cadmium battery), and an alkaline battery. The battery is electrically coupled to the heating assembly to supply electrical power when needed and under the control of a controller (not shown) to heat the aerosol-forming material. In this example, the battery is connected to a central support 120, which holds the battery 118 in place.

The device also includes at least one electronic module 122. Electronic module 122 may include, for example, a printed circuit board (PCB). Circuit board 122 may support at least one controller, such as a processor and memory. Circuit board 122 may also include one or more electrical paths for electrically interconnecting various electronic components of device 100. For example, battery terminals may be electrically connected to circuit board 122 so that power can be distributed throughout device 100. Receptacle 114 may also be electrically connected to the battery via electrical tracks.

In the example apparatus 100, the heating assembly is an induction heating assembly and includes various components for heating the aerosol-forming material article 110 through an induction heating process. Induction heating is the process of heating an electrically conductive object (for example, a pantograph) using electromagnetic induction. An induction heating assembly may include an inductive element, such as one or more induction coils, and a device for passing a varying electrical current, such as alternating electrical current, through the inductive element. A changing electric current in an inductive element creates a changing magnetic field. The alternating magnetic field penetrates through a pantograph suitably positioned relative to the inductive element, creating eddy currents within the pantograph. The pantograph has electrical resistance to eddy currents and hence the flow of eddy currents against this resistance causes heating of the pantograph due to Joule heating. In cases where the pantograph contains a ferromagnetic material such as iron, nickel or cobalt, heat may also be generated by magnetic hysteresis losses in the pantograph, that is, the changing orientation of magnetic dipoles in the magnetic material as a result of their alignment with the changing magnetic field. With induction heating, compared to heating by heat transfer, for example, heat is generated inside the pantograph, which ensures rapid heating. In addition, there is no need for any physical contact between the induction heater and the current collector, allowing greater freedom in design and application.

The induction heating assembly of the example device 100 includes a current collector structure 132 (referred to herein as a "susceptor"), a first inductor 124, and a second inductor 126. The first and second inductors 204, 206 are made of electrically conductive material. In this example, the first and second inductors 124, 126 are made of winding wire/cable wound in a spiral to form helical inductors 124, 126. Winding wire consists of many individual wires that are individually insulated and twisted together to form a single wire. Winding wires are designed to reduce skin effect losses in the conductor. In the example device 100, the first and second inductors 124, 126 are made of copper winding wire having a substantially circular cross-section. In other examples, the winding wire may have a different cross-sectional shape, such as rectangular.

The first inductor 124 is configured to generate a first alternating magnetic field to heat the first portion of the pantograph 132, and the second inductor 126 is configured to generate a second alternating magnetic field to heat the second portion of the pantograph 132. Here, the first section of the pantograph 132 is referred to as the first pantograph zone 132a. , and the second section of the pantograph 132 is called the second pantograph zone 132b. In this example, the first inductor 124 is adjacent to the second inductor 126 in a direction along the longitudinal axis 134 of the device 100 (ie, the first and second inductors 124, 126 do not overlap). In this example, the current collector structure 132 includes one pantograph containing two zones, however, in other examples, the current collector structure 132 may contain two or more pantographs. In some examples there are more than two heating zones. Each zone can be formed by corresponding parts of one single pantograph of a current-collecting structure or separate pantographs of a pantograph. The ends 130 of the first and second inductors 124, 126 are connected to the printed circuit board 122.

It will be appreciated that the first and second inductors 124, 126 may, in some examples, have at least one characteristic that is different from each other. For example, the first inductor 124 may have at least one different characteristic than the second inductor 126. More specifically, in one example, the first inductor 124 may have a different inductance value than the second inductor 126. Referring to FIG. . 2, the first and second inductor coils 124, 126 are of different lengths such that the first inductor 124 is wound on a smaller section of the pantograph 132 than the second inductor 126. Thus, the first inductor 124 may contain a different number of turns than the second inductor 126 (as long as the spacing between the individual turns is substantially the same). In yet another example, the first inductor 124 may be made of a different material than the second induction coil 126. In some examples, the first and second induction coils 124, 126 may be substantially identical.

In this example, the inductors 124, 126 are wound in the same direction. That is, both the first inductor 124 and the second inductor 126 are left-handed spirals. In another example, both inductors 124, 126 may be right-handed coils. In yet another example (not shown), the first inductor 124 and the second inductor 126 are wound in opposite directions. This can be useful when the inductors are turned on at different times. For example, the first inductor 124 may first operate to heat the first section of the article 110, and at a later date the second inductor 126 may operate to heat the second section of the article 110. Winding the coils in opposite directions helps reduce the current induced in the inactive coil when in use in combination with a certain type of control scheme. In one example, inductors 124, 126 are wound in different directions (not shown), wherein the first inductor 124 may be a right-handed helix and the second inductor 126 may be a left-handed helix. In another such embodiment, the first inductor 124 may be a left-handed helix and the second inductor 126 may be a right-handed helix.

The current collector 132 in this example is hollow and therefore defines a container into which the aerosol-forming material is placed. For example, article 110 may be inserted into pantograph 132. In this example, pantograph 132 is tubular with a circular cross-section.

Device 100 in FIG. 2 also includes an insulating element 128, which may be generally tubular and at least partially surround the current collector 132. The insulating element 128 can be made of any insulating material, such as plastic. In this particular example, the insulating element is made of polyetheretherketone (PEEK). The insulating element 128 may help isolate the various components of the device 100 from the heat generated in the current collector 132.

The insulating element 128 may also fully or partially support the first and second inductors 124, 126. For example, as shown in FIG. 2, the first and second inductors 124, 126 are located around the insulating element 128 and are in contact with the radially outer surface of the insulating element 128. In some examples, the insulating element 128 does not abut the first and second inductors 124, 126. For example, there may be a small gap between the outer surface of the insulating element 128 and the inner surface of the first and second inductors 124, 126.

In a specific example, pantograph 132, insulating element 128, and first and second inductors 124, 126 lie on the same central longitudinal axis of pantograph 132.

In fig. 3 is a partial cross-sectional side view of device 100. In this example, the outer cover 102 is again missing. The circular cross-sectional shape of the first and second inductors 124, 126 is visible in FIG. 3 more clearly.

The device 100 also includes a support 136 that engages one end of the pantograph 132 to hold the pantograph 132 in place. The support 136 is connected to the second end member 116.

The device 100 also includes a second cover/cap 140 and a spring 142 disposed toward the distal end of the device 100. The spring 142 allows the second cover 140 to be opened to provide access to the pantograph 132. For example, a user may open the second cover 140 to clean the pantograph 132 and /or support 136.

The device 100 also includes an expansion chamber 144 that extends from the proximal end of the pantograph 132 to the opening 104 of the device. Located at least partially within the expansion chamber 144 is a retaining clip 146 that abuts the article 110 and holds it in the device 100. The expansion chamber 144 is connected to the end member 106.

In fig. 4 is an exploded view of the device 100 shown in FIG. 1, again without outer cover 102.

In fig. 5A shows a cross-section of a portion of the device 100 shown in FIG. 1. In FIG. 5B is a close-up view of the area of FIG. 5A. In fig. 5A and 5B show article 110 placed in pantograph 132, with article 110 sized such that the outer surface of article 110 is adjacent to the inner surface of pantograph 132. This provides the most efficient heating. Article 110 of this example includes aerosol generating material 110a. The aerosol generating material 110a is located within the pantograph 132. The article 110 may also include other components such as a filter, wrapping materials, and/or a cooling structure.

In fig. 5B shows that the outer surface of the pantograph 132 is spaced from the inner surface of the inductors 124, 126 by a distance 150, measured in a direction perpendicular to the longitudinal axis 158 of the pantograph 132. In one specific example, the distance 150 is from about 3 mm to 4 mm, from about 3 mm to 3.5 mm or approximately 3.25 mm.

In fig. 5B also shows that the outer surface of the insulating element 128 is spaced from the inner surface of the inductors 124, 126 by a distance 152 measured in a direction perpendicular to the longitudinal axis 158 of the pantograph 132. In one specific example, the distance 152 is approximately 0.05 mm. In another example, the distance 152 is substantially 0 mm such that the inductors 124, 126 abut and touch the insulating element 128.

In one example, the pantograph 132 has a wall thickness 154 of about 0.025 mm to 1 mm, or about 0.05 mm.

In one example, pantograph 132 has a length of from about 40 mm to 60 mm, from about 40 to 45 mm, or about 44.5 mm.

In one example, the insulating element 128 has a wall thickness 156 of about 0.25 mm to 2 mm, 0.25 mm to 1 mm, or about 0.5 mm.

As discussed above, the heating assembly in the example apparatus 100 is an induction heating assembly containing various components for heating the aerosol-forming material article 110 through an induction heating process. Specifically, the first inductor 124 and the second inductor 126 are used to heat the respective first 132a and second 132b areas of the susceptor 132 to heat the aerosol-forming material and produce an aerosol. Below with reference to FIG. 6-12 will describe in detail the operation of the device 100 using the first and second inductors 124, 126 to inductively heat the current collector structure 132.

The induction heating unit of the device 100 includes an LC circuit. An LC circuit has an inductance L provided by the inductive element and a capacitance C provided by the capacitor. In the device 100, the inductance L is provided by the first and second inductors 124, 126, and the capacitance C is provided by several capacitors, as will be described below. An induction heater circuit containing an inductance L and a capacitance C may in some cases be represented as an RLC circuit containing a resistance R provided by a resistor. In some cases, the resistance is provided by the ohmic resistance of the parts of the circuit connecting the inductive element and the capacitor, and therefore the circuit does not necessarily need to include a resistor as such. In such circuits, electrical resonance can be observed, which occurs at a certain resonant frequency when the imaginary parts of the impedances or the total conductance of the circuit elements cancel each other out.

One example of an LC circuit is a series circuit in which an inductive element and a capacitor are connected in series. Another example of an LC circuit is a parallel LC circuit in which an inductive element and a capacitor are connected in parallel. Resonance occurs in an LC circuit because the collapsing magnetic field of an inductive element generates an electric current in its windings, which charges the capacitor, while the discharging capacitor provides an electric current, which creates a magnetic field in the inductive element. When a parallel LC circuit is driven at its resonant frequency, the dynamic impedance of the circuit is maximum (since the reactance of the inductive element is equal to the reactance of the capacitor) and the current in the circuit is minimum. However, for a parallel LC circuit, the inductive element and capacitor connected in parallel act as a current multiplier (effectively multiplying the current in the circuit and therefore the current passing through the inductive element). Thus, if the RLC or LC circuit is allowed to operate at its resonant frequency for at least some of the time that the circuit is operating to heat the current collector, then effective and/or efficient induction heating can be achieved by providing the greatest value of the magnetic field penetrating into the pantograph.

The LC circuit used in the device 100 to heat the current collector 132 may use one or more transistors to act as a switching device, as will be described below. A transistor is a semiconductor device for switching electronic signals. A transistor usually contains at least three terminals for connection to an electronic circuit. A field-effect transistor (FET) is a transistor in which the effect of an applied electric field can be used to change the effective conductance of the transistor. The field effect transistor may include a body, a source terminal S, a drain terminal D, and a gate terminal G. A field-effect transistor contains an active channel containing a semiconductor through which charge carriers, electrons or holes, can flow between source S and drain D. The conductivity of the channel, that is, the conductivity between drain D and source S, is a function of the potential difference between the gate terminals G and the source S, for example, created by a potential applied to the G terminal of the gate. In enrichment mode FETs, the FET may be turned off (i.e., essentially preventing current from flowing through it) when there is essentially zero voltage between gate G and source S, and can be turned on (i.e., essentially allow current to flow through it) when there is a substantially non-zero voltage between gate G and source S.

One type of transistor that may be used in the circuitry of device 100 is an n-channel (or n-type) field-effect transistor (n-FET). An n-FET is a field-effect transistor whose channel consists of an n-type semiconductor in which electrons are the majority carriers and holes are the minority carriers. For example, n-type semiconductors may contain an intrinsic semiconductor (such as silicon) doped with donor impurities (such as phosphorus). In n-channel FETs, the drain terminal D is at a higher potential than the source terminal S (i.e., there is a positive drain-to-source voltage, or in other words, a negative source-to-drain voltage). To turn on the n-channel FET (that is, allow current to flow through it), the gate G terminal is applied with a switching potential that is higher than the potential at the source terminal S.

Another type of transistor that may be used in device 100 is a p-channel (or p-type) field-effect transistor (p-FET). A p-FET is a field-effect transistor whose channel consists of a p-type semiconductor in which holes are the majority carriers and electrons are the minority carriers. For example, p-type semiconductors may contain an intrinsic semiconductor (such as, for example, silicon) doped with acceptor impurities (such as, for example, boron). In p-channel FETs, the source terminal S is at a higher potential than the drain terminal D (i.e., there is a negative drain-to-source voltage, or in other words, a positive source-to-drain voltage). To turn on a p-channel FET (that is, allow current to flow through it), a switching potential is applied to the gate terminal G that is lower than the potential at the source terminal S (and which may, for example, be higher than the potential at the drain terminal D ).

In examples, one or more field-effect transistors used in device 100 may be metal-oxide-semiconductor field-effect transistors (MOSFETs). A MOSFET is a field-effect transistor whose gate terminal G is electrically isolated from the semiconductor channel by an insulating layer. In some examples, the gate terminal G may be metal and the insulating layer may be an oxide (such as, for example, silicon dioxide), hence the name "metal-oxide-semiconductor". However, in other examples, the gate may be made of materials other than metal, such as polysilicon, and/or the insulating layer may be made of materials other than oxide, such as other dielectric materials. Such devices, however, are generally referred to as metal-oxide-semiconductor field-effect transistors (MOSFETs), and it should be understood that the term field-effect transistors or MOSFETs as used herein should be interpreted to include such devices.

The MOSFET may be an n-channel MOSFET (or n-type transistor), where the semiconductor is n-type. An n-channel MOSFET (n-channel MOSFET) can operate in the same way as described above for an n-channel field effect transistor. As another example, the MOSFET may be a p-channel MOSFET (or p-type transistor), where the semiconductor is p-type. A p-channel MOSFET (p-MOSFET) can operate in the same way as described above for a p-channel field-effect transistor. An n-type MOSFET typically has a lower source-drain resistance than a p-type MOSFET. Therefore, in the on state (i.e. when current is flowing through it), n-type MOSFETs generate less heat compared to p-type MOSFETs, and therefore can consume less power during operation than MOSFETs p-type. In addition, n-type MOSFETs typically have shorter switching times (i.e., the characteristic response time from a change in the switching potential applied to the gate's G pin to switching current flow through the MOSFET) compared to p-type MOSFETs. -type. This may allow for higher switching speeds and improved switching control.

Now with reference to FIG. 6, a circuit for induction heating by the device 100 will be described. In FIG. 6 shows a simplified schematic diagram of a portion of the induction heating circuit 600 of the aerosol generation device 100. In fig. 6 shows a portion of an induction heating circuit 600 that includes a first inductor 124 for heating the first susceptor zone 132a when alternating current flows through the first inductor 124. The first pantograph zone 132a is shown in FIG. 6 as having an inductive element and a resistive element to show how the current collector 132 is inductively coupled to the first inductor 124 and heated by generating eddy currents. It should be noted that device 100 also includes a second inductor 126, which is not shown in FIG. 6. The second inductor 126 is also part of the induction heating circuit 600 and is controlled to heat the second pantograph region 132b, as will be described below. However, for the sake of clarity, circuit 600 will first be described with reference to features that are shown in FIG. 6.

Circuit 600 includes a first resonator section 601, a DC voltage source 118 for supplying a DC voltage to the first resonator section 601, and a control device for controlling the circuit 600. The first resonator section 601 includes a first inductor 124 and a switching device including a first field-effect transistor 608, and the control device is configured to switch the field effect transistor 608 between the first state and the second state in response to a voltage detected in the circuit 600, as will be described in more detail below, to drive the first inductor 124. The circuit 600, excluding the current collector 132, is located on a circuit board 122 of the device 100, with an inductor 124 connected to the circuit board 122 at a first end 130a and a second end 130b.

The first resonator section 601 includes a first capacitor 606 and a second capacitor 610, both positioned in parallel with the first inductor 124 such that when the first resonator section 601 is allowed to resonate, alternating current flows between the first capacitor 606 and the second capacitor 610 and through the inductor 124. As mentioned above, the first field effect transistor 608, in this example an n-channel MOSFET, is configured to operate as a switching device in the first resonator section 601.

It should be noted that in other examples, resonator section 601 may contain only one capacitor, such as at the location of the first capacitor 606 or at the location of the second capacitor 610. In other examples, resonator section 601 may contain any other number of capacitors, such as three or more capacitors. For example, the first capacitor 606 and/or the second capacitor 610 may be replaced by two or more capacitors arranged in parallel with each other. As will be well understood, resonator section 601 has a resonant frequency that depends on the inductance L and capacitance C of resonator section 601. The number, type, and arrangement of capacitors in resonator section 601 may be selected based on the power levels to be used in circuit 600 and the desired frequency of operation of circuit 600. For example, it should be understood that individual capacitors and the arrangement of said capacitors may be considered to have equivalent series resistance (ESR), as well as the limit of the current-carrying ability of said capacitors. Such features may be taken into account when determining the arrangement of capacitors to provide capacitance in the resonator section 601. For example, depending on the required power levels and frequency of operation, it may be advantageous to have multiple capacitors in parallel to provide higher capacitance or lower ESR. In this example, the first and second capacitors 606, 610 are C0G ceramic capacitors, each having a capacitance of about 100 nF. In other examples, a capacitor and/or other types of capacitors with different capacitance values are used, for example, capacitors with different capacitance values can be used in accordance with the considerations set forth in this paragraph.

The first resonator section 601 is supplied with a constant voltage from a constant voltage source 118, which, for example, as described above, is the voltage supplied by a battery. As shown in FIG. 6, DC voltage source 118 includes a positive terminal 118a and a negative terminal 118b. In one example, a constant voltage source 118 supplies a constant voltage of about 4.2 V to the first resonator section 601. In other examples, the DC voltage source 118 may supply, for example, a voltage of 2 to 10 V, or about 3 to 5 V.

The controller 1001 is configured to control the operation of the circuit 600. The controller 1001 may include a microcontroller, such as a microprocessor unit (MPU), containing multiple inputs and outputs. In one example, controller 1001 is an MPU model STM32L051C8T6. In some examples, the DC voltage source 118 provided in circuit 600 is provided with an output from controller 1001, which itself is powered by a battery or other power source.

The positive terminal 118a of the DC voltage source 118 is electrically connected to the first node 600A. In the example, the constant voltage source 118 is connected to the node 600A through a controller 1001, which receives power from the constant voltage source 118 and supplies the voltage supplied by the constant voltage source to the components of the device, including the circuit 600. The first node 600A is electrically connected to the first end 606a of the first capacitor 606 and with the first end 130a of the first inductor 124. The second end 130b of the first inductor 124 is electrically connected to the second node 600B, which in Fig. 6 is represented at two electrically equivalent points in the circuit diagram. The second node 600B is electrically connected to the drain terminal 608D of the FET 608. In this example, the second node 600B is also electrically connected to the first end 610a of the second capacitor 610. Continuing around the circuit, the source terminal 608S of the first FET 608 is electrically connected to the third node 600C. The third node 600C is electrically connected to ground 616 and in this example to the second end 610b of the second capacitor 610. The third node 600C is electrically connected through a current sensing resistor 615 to the fourth node 600D, and the fourth node 600D is electrically connected to the negative terminal 118b of DC voltage source 118 current, which, like the positive terminal, is supplied through controller 1001 in the example.

It should be noted that in examples where the second capacitor 610 is not present, the third node 600C may have only three electrical connections: to the first FET source terminal 608S, to ground 616, and to current sensing resistor 615.

As mentioned above, the first field effect transistor 608 acts as a switching device in the first resonator section 601. The first field effect transistor 608 may be in a first state, that is, an "ON" state, and a second state, that is, an "OFF" state. As those skilled in the art will appreciate, when an n-channel FET is in the "OFF" state (ie, when the appropriate control voltage is not applied to its gate), it effectively acts as a diode. In fig. 6, the diode functionality exhibited by the first field effect transistor 608 in the off state is represented by the first diode 608a. That is, when the field effect transistor 608 is in the off state, the first diode 608a operates, largely preventing current from flowing from the drain terminal 608D to the source terminal 608S, but allowing current to flow from the source terminal 608S to the drain terminal 608D if the diode 608a is accordingly forward biased. An n-channel field effect transistor is in the on state when the appropriate control voltage is applied to its gate, so that there is a current-carrying path between its drain D and source S. Thus, when the first field effect transistor 608 is in the on state, it acts as a closed switch in the first resonator section 601.

As mentioned above, circuit 600 can be viewed as including a first resonator section 601 and an additional control device. The control device includes a comparator 618, a zero voltage detector 621, and a trigger 622 and is configured to detect a voltage in the first resonator section 601 and control the first field-effect transistor 608 in response to the detected voltage. This control of the first field effect transistor 608 by the control device will now be described in more detail.

Electrically connected to the second node 600B is a zero voltage detector 621 that is configured to detect a voltage, that is, a voltage equal to or near 0 V relative to a ground voltage, at a point in circuit 600 to which the zero voltage detector 621 is connected. The zero voltage detector 621 is configured to output a signal to control switching state of the field effect transistor 608. That is, the zero voltage detector 621 is configured to output a signal to a flip-flop 622. The flip-flop 622 is an electrical circuit that can be adjusted between two stable states. The flip-flop 622 is electrically coupled to the first gate driver 623, which is configured to apply a voltage to the first FET gate terminal 608G depending on the state of the flip-flop. That is, the first gate driver 623 is configured to supply a corresponding voltage to the first FET gate terminal 608G to switch the FET 608 to an ON state when the flip-flop is in one state, but is configured not to supply a corresponding voltage to maintain the FET 608 in the on state when flip-flop 622 is in another state. For example, the first gate driver 623 may be configured to apply an appropriate gate-source voltage to the first FET gate 608G to turn on the FET 608 when the flip-flop 622 is in the "1" state, and the first gate driver 623 may be configured not to supply gate-source voltage when flip-flop 622 is in the "0" state. Thus, the state of the trigger means 622 controls whether the first field effect transistor 608 is turned on or off.

In this example, the zero voltage detector 621 and the first control device gate driver 623 are configured to receive corresponding signals 1011, 1021 from the controller 1001, with which the controller 1001 can initiate and control the operation of the circuit 600, as will be described in more detail below.

At the fourth node 600D, a control voltage line 619 is electrically connected. Control voltage line 619 is electrically connected to fifth node 600E through resistor 617a, and fifth node 600E is electrically connected to voltage comparator 618, hereinafter referred to as comparator 618. Fifth node 600E is electrically connected to the positive terminal of comparator 618. The negative terminal of comparator 618 is connected to ground 616 In this example, the comparator 618 is configured to output a signal based on a comparison of the voltage at the fifth node 600E with the ground voltage. The output of comparator 618 is sent to flip-flop 622. Control voltage 1031 is supplied in this example from controller 1001 to control voltage line 618 through second resistor 617b.

As mentioned above, the comparator 618 is electrically connected to provide an output to the flip-flop 622. The flip-flop 622 is configured such that an output signal from the comparator 618 can change the state of the flip-flop 622 and thereby cause the first driver 623 to change the state of the first field-effect transistor 608.

The operation of the example circuit 600 will now be described in more detail in the context of the first resonator section 601 activated by the controller 1001 such that the first inductor 124 is driven to heat the first pantograph zone 132a.

To begin, the first field effect transistor 608 is in the off state and thus acts as a diode 608a, preventing current from flowing through the inductor 124. The controller 1001 initiates operation of the circuit 600 to heat the first pantograph zone 132a, causing the field effect transistor 608 to switch from an off state to an on state. In this example, the controller initiates operation of circuit 600 by providing a RUN signal 1011 to zero voltage detector 621. Thus, flip-flop 622 changes state and causes the first gate driver 623 to apply a signal to the FET gate terminal 608G, thereby switching the FET to the on state.

Once FET 608 is switched to the on state, what may be called a self-oscillating heating cycle of circuit 600 begins. FET 608, now in the on state, acts as a closed switch, allowing DC current to begin flowing from the positive terminal 118a of the DC voltage source through the first inductor 124 and return to the negative terminal 118b of the constant voltage source through a current sensing resistor 615. The first inductor 124 counteracts this initial increase in current, as is well known, creating a back emf in accordance with Faraday's and Lenz's laws. In the on state, the voltage between the drain terminal 608D and the source terminal 608S is substantially zero.

In fig. 7A is a schematic graphical representation of the current flowing through the first inductor 124 as a function of time t, starting from the moment the FET 608 is turned on at time t0. From time t0 in the inductor 124, a direct current begins to increase from zero at a rate that depends on the inductance L1 of the inductor 124, the resistance of the circuit 600 to the direct current and the constant supply voltage. In one example, current sensing resistor 615 has a resistance of about 2 mΩ, while inductor 124 has a DC resistance of 2 to 15 mΩ, or 4 to 10 mΩ, or in this example, about 5.2 mΩ. This increase in current in the inductor corresponds to the inductor 124 storing magnetic energy, and the amount of magnetic energy that can be stored by the inductor 124 depends on its inductance L1, as will be understood.

In fig. 7B shows a simplified representation of the voltage across current sensing resistor 615 as a function of time t, again from the time t0 at which FET 608 was turned on. Shortly after FET 608 is turned on, a voltage appears across inductor 124, which is the back emf generated inductor 124 because the inductor counteracts the increase in current. Therefore, at this time, the voltage across the current sensing resistor 615, as shown in FIG. 7B is small because almost all of the voltage difference provided by the DC source 118 falls across the inductor 124. Then, as the current through the inductor 124 increases and the back emf of the inductor 124 decreases, the voltage across the current sensing resistor 615 increases. This can be seen as the occurrence of a negative voltage across current sense resistor 615, as shown in FIG. 7B. That is, the voltage across the current sensing resistor 615 becomes increasingly negative during the time that the FET 608 is in the on state.

Since an increasing negative voltage across the current sensing resistor 615 corresponds to an increasing current flowing through the inductor 124, the magnitude of the voltage across the current sensing resistor 615 is indicative of the current flowing through the inductor 124. While the field effect transistor 608 remains in the on state, the current flowing through the inductor 124 and the voltage across the current sensing resistor 615 increase substantially linearly to the respective maximum values Imax, Vmax (which depend on the DC voltage supplied by the DC source 118, and the DC resistance of circuit 600) for a constant time depending on the inductance L1 and the DC resistance of circuit 600. It should be noted that since the current flowing through the inductor 124 changes after time t0, some inductive heating of the pantograph 132 may occur while the direct current flowing through the first inductor 124 increases.

The circuit 600 is configured such that the control device determines the amount of energy that is stored in the first inductor 124 during the time that the field effect transistor 608 is in the on state, and the controller 1001 can control it. That is, the controller 1001 controls the amount of direct current (and therefore the amount of magnetic energy) that can build up in the inductor 124, as will be described below.

As described above, control voltage 1031 is supplied to control voltage line 619. In this example, control voltage 1031 is a positive voltage, and the voltage applied to the positive terminal of comparator 618 (ie, the voltage at the fifth node 600E) at any time depends on the value of the control voltage 1031 and the voltage at the fourth node 600D. When the negative voltage across the current sensing resistor 615 reaches a certain value, it cancels the positive control voltage 1031 at the fifth node 600E and produces a voltage of 0 V (ie, ground voltage) at the fifth node 600E. In this example, resistor 617a has a value of 2k. Resistor 617b represents an effective resistance for controller 1001 of 70 kΩ. The voltage at the fifth node 600E reaches 0 V when the negative voltage across the current sensing resistor 615 is the same value as the control voltage 1031.

The comparator 618 is configured to compare the voltage at its positive terminal with the voltage of the ground 616 connected to its negative terminal, and output a signal as a result. In one example, the comparator is a standard FAN156 component that can be obtained from On-Semiconductor. Accordingly, when the voltage at the fifth node 600E reaches 0 V, the comparator 618 receives a 0 V signal at its positive terminal, and the comparison result obtained by the comparator 618 is that the voltage at the positive terminal is equal to the voltage at the negative terminal. Therefore, the comparator 618 provides a signal to the flip-flop 622 and causes the FET 608 to turn off. Essentially, the turn-off of the FET 608 depends on the voltage detected in the circuit 600. Specifically, in this example, if the comparator 618 by comparing the voltage at its terminals detects, that the negative voltage across the current-sensing resistor 615 has reached the same value as the control voltage 1031, which occurs at time t1, the field-effect transistor 608 is turned off. In fig. 7A, the direct current flowing through the inductor 124 at time t1 when the field-effect transistor 608 is turned off is denoted by I1.

When the field effect transistor 608 is turned off at time t1, the field effect transistor 608 is switched from operating as a closed switch to operating as a diode 608a in the resonator section 601, and to effectively operating as an open switch for supplying power from the DC source 118. At time t1, the direct current path through inductor 124 to ground 616 is interrupted by field effect transistor 608. This causes the current flowing in the first inductor 124 to drop (not shown in FIG. 7A), and inductor 124 counteracts this change in current, generating an induced voltage. Accordingly, the current begins to oscillate between the inductor 124 and the capacitors 606, 608 at the resonant frequency of the first resonator section 601.

Likewise, the voltage across the inductor 124 and thereby between the drain terminal 608D and the source terminal 608S of the first FET begins to oscillate at the resonant frequency of the first resonator section 601. When the current and voltage across the inductor 124 begin to fluctuate, inductive heating of the pantograph 132 occurs. Thus, switching the FET 608 to the off state releases the magnetic energy stored in the inductor 124 at time t1 to heat the pantograph 132.

In fig. 8 shows a plot 800 of the voltage across the first FET 608, starting with the FET 608 being in the on state from time t0 to t1. During the time shown in FIG. 8, the first field effect transistor 608 is turned off and on twice.

Voltage waveform 800 includes a first portion 800a between times t0 and t1 when the first FET 608 is turned on, and a second portion 800b to 800d when the first FET 608 is turned off. At time 800e, the FET 608 is turned on again and a third section 800f begins, which is equivalent to the first section 800a, while the first FET 608 remains on and the above-described process of increasing the DC current flowing through the inductor 124 is repeated. In fig. 8 also shows a fourth region 800g when the first FET 608 is turned off again to allow the voltage across the FET 608 to oscillate, and a fifth region 800h when the first FET 608 is subsequently turned on again.

The voltage across the first field effect transistor 608 is zero when the first field effect transistor 608 is turned on at sections 800a, 800f and 800h. When the first FET 608 is turned off, as shown by portions 800b-800d as well as portion 800g, the first induction coil 124 uses the energy stored in its magnetic field (the magnetic field resulting from the DC current buildup when the first FET 608 was turned on) to induce a voltage that resists a drop in current flowing through the first inductor coil 124 resulting from turning off the first field effect transistor 608. The voltage induced in the first inductor coil 124 causes a corresponding voltage change across the first field effect transistor 608. During this voltage change, the first the inductor 124 and the capacitors 606, 610 begin to resonate with each other in accordance with the sine wave. The voltage indicated by voltage line 800 first increases (see, for example, 800b) as the induced voltage in the first inductor 124 increases to counteract the drop in current due to turning off the first field effect transistor 608, reaches a peak (see, for example, , 800c), and then, as the energy stored in the magnetic field of the first inductor 124 decreases, decreases back to zero (see, for example, 800d).

The varying voltage 800b-800d and 800g produces a corresponding varying current (not shown), and since during the time that the first FET 608 is turned off, the capacitors 606, 610 and the first inductor 124 act as a resonant LC circuit, the overall impedance the combination of the first inductor 124 and capacitors 606, 610 is minimal during this time. Therefore, it is understood that the maximum amount of changing current flowing through the first inductor 124 will be relatively large. This relatively large changing current correspondingly causes a relatively large change in the magnetic field in the first induction coil 124, which causes the sensing element 132 to generate heat. The period of time during which the voltage across the first field effect transistor 608 varies as shown by portions 800b-800d and portion 800g in this example depends on the resonant frequency of the first resonant portion 601.

Referring now to FIG. 6 and fig. 8, circuit 600 is configured such that when first FET 608 is turned off and the voltage across first FET 608 decreases back to 0 V, zero voltage detector 621 detects this voltage parameter and provides a signal to flip-flop 622, which causes first FET 608 to switch back to the on state. That is, in response to this voltage parameter detected in the first resonator section 601, the field effect transistor 608 switches from an off state to an on state. The zero voltage detector 621 can be considered to detect a voltage indicating that a predetermined portion of the current oscillation cycle between the inductive element and the capacitive element has completed since the field effect transistor 608 is turned off. That is, the zero voltage detector 621 detects that a half-cycle of the current (and voltage) oscillation is resonant frequency of the first resonator section 601 was terminated by a zero voltage detector 621 detecting that the voltage across the FET 608 had returned to 0 V or nearly 0 V.

In some examples, zero voltage detector 621 may detect when the voltage across the first FET 608 has returned to level 801 or below, and as such may provide a signal causing the state of the FET 608 to switch before the voltage across the FET 608 reaches exactly 0 V. As shown in Fig. 8, operation of the zero voltage detector 621 limits the voltage fluctuations in the resonator section 601 after one half cycle and thus results in a substantially half-sinusoidal voltage profile across the first FET 608. Additional details of the operation of the zero voltage detector 621 will be described below with reference to FIG. 9.

When the first field effect transistor 608 is turned on again at point 800e, the direct current driven by the direct current source 118 again builds up across the first inductor 124. The first inductor 124 can then again store energy in the form of a magnetic field, which will be released when the first FET 608 next turns off to initiate resonance in the first resonator section 601. When the first field effect transistor 608 is cycled on and off in this manner, the above process is continuously repeated to heat the sensing element 132.

It should be noted that the above-described increase in current flowing through the inductor 124 described with reference to FIG. 7A and 7B occurs both when FET 608 is initially turned on in response to RUN signal 1011 from controller 1001 and when FET 608 is subsequently turned on when zero voltage is detected by zero voltage detector 621. In the first case, in response to the RUN signal 1011, the current in the inductor 124 rises substantially linearly from 0. In the second case, when the field effect transistor 608 is turned on again in response to the detection of zero voltage at point 800e, some excess current circulates in the circuit 600 ( for example, from previous on and off cycles of the field effect transistor 608). When the FET 608 is turned on again after detecting zero voltage, the recirculating current creates an initial negative current through the FET 608. Then, while the FET 608 remains turned on, the current through the FET 608 and inductor 124 increases substantially linearly from the initial negative current value, created by the recirculating current. As the current through inductor 124 increases, the voltage across current sensing resistor 615 correspondingly becomes increasingly negative, as described above.

In examples, turning on and off of field effect transistor 608 may occur at a frequency of from about 100 kHz to 2 MHz, or from about 500 kHz to 1 MHz, or about 300 kHz. The frequency with which the field effect transistor 608 turns on and off depends on the inductance L, the capacitance C, the DC supply voltage supplied by the source 618, and further, the extent to which current continues to recirculate through the resonator section 601, and the effect load of the current collector 132. For example, if the DC supply voltage is 3.6 V, the inductance of the inductor 124 is 140 nH, and the capacitance of the resonator section 601 is 100 nF, then the time that the FET 608 remains on may be about 2700 ns, and the time to complete a half cycle of oscillation when FET 608 is turned off may be about 675 ns. These values correspond to about 20 W of power supplied from the DC voltage source 118 to the resonator section 601. The above-mentioned amount of time that the FET 608 remains on depends on the amount of current that is recirculated in the circuit because, as described above, this recirculating current causes an initial negative current through the inductor when the FET 608 is turned on. It should also be noted that the time it takes for the current to reach a value that causes FET 608 to turn off also depends at least in part on the resistance of inductor 124, however this has a relatively minor effect on time compared to the effect of the inductance of resonator section 601. The half-cycle completion time (675 ns in this example) depends on the resonant frequency of the resonator section 601, which is affected not only by the inductance and capacitance values of the inductor 124 and capacitors

606, 610 respectively, but also the effective resistance provided by loading the inductor 124 by the current collector 132.

Thus far, circuit 600 has been described in terms of its operation to heat pantograph 132 using a single inductor, first inductor 124, and thus only the portion of circuit 600 used by device 100 has been described. However, as described above, the device 100 also includes a second inductor 126 for heating the second zone 132b of the pantograph 132. In FIG. 9 shows a simplified diagram of a circuit 600 including a second inductor 126 in addition to the first inductor 124.

As shown in FIG. 9, in addition to the features described with reference to FIG. 6-8, circuit 600 includes a second resonator section 701 including a second inductor 126, a third capacitor 706, a fourth capacitor 710, and a second field-effect transistor 708 having a drain terminal 708D, a source terminal 708S, and a gate terminal 708G. In addition, circuit 600 includes a second gate driver 723 configured to apply a gate-source voltage to a second FET gate terminal 708G. The controller 1001 is not shown in FIG. 9, but controller 1001 controls circuit 600 in the manner described with reference to FIG. 6-8, and is further configured to supply a control signal 1012 to the second gate driver 723. Certain feature reference numbers of the circuit 600 already described with reference to FIG. 6 have been omitted from FIG. 9 for clarity.

As described above, the first inductor 124 is configured to heat the first zone 132a of the pantograph 132, and the second inductor 126 is configured to heat the second zone 132b of the pantograph 132. The second inductor 126, the third and fourth capacitors 706, 710, and the second field-effect transistor 708 are arranged to forming the second resonator section 701 in the same manner as the first inductor 124, the first and second capacitors 606, 610 and the first field-effect transistor 608 are arranged to form the first resonator section 601. In one example, the third and fourth capacitors 706, 710 are also C0G capacitors and may have a capacitance of about 100 nF. The second inductor 126 in one example has a DC resistance of about 8 mΩ. In the active state, the second resonator section 701 operates to heat the current collector 132 in the same manner as described above for the first resonator section 601, and this description will not be repeated here.

It should be understood that the amount of DC resistance of the inductors 124, 126 will affect the efficiency of the circuit 600 since a higher DC resistance will result in higher resistive losses in the inductor 124, 126 and thus it may be desirable to minimize the inductor resistance DC, for example, by changing the number of windings or the cross-section of the inductors 124, 126. In addition, it should be understood that the AC resistance of the inductor 124 increases compared to the DC resistance due to the skin effect. Thus, the use of winding wire in the examples reduces the skin effect and thus reduces the AC resistance and corresponding resistive losses in the inductors 124, 126. As an example, where the first inductor 124 has a DC resistance of about 5 mΩ and the second inductor 126 has a DC resistance of about 8 mΩ, and the circuit operates at a frequency of about 300 kHz, the particular arrangement of the winding wire forming the coils produces an effective resistance inductance coils 124, 126 are approximately 1.14 times higher than their DC resistance value.

The node 700A in the second resonator section 701 is equivalent to the first node 600A of the first resonator section 601 and is electrically connected to the first node 600A and thus to the positive terminal 118a of the DC source 118. Node 700C is in an equivalent position in the second resonator section 701 as is the third node 600C of the first resonator section 601, and node 700C is similarly connected to ground 616.

It is important to note that circuit 600 is configured to operate under the control of controller 1001 such that only one of the resonator sections 601, 701 is active at any given time. Examples of this work will be described in more detail below.

During activation of one of the resonator sections 601, 701, the zero voltage detector 621 is configured to detect zero voltage in the active resonator section 601, 701 and thereby control switching of the corresponding field effect transistors 608, 708 of the active resonator section 601, 701. The zero voltage detector 621 monitors when the corresponding FET 608, 708 of the active resonator section 601, 701 is turned on again (eg, at point 800e), and an example of this will now be described in more detail with reference to FIG. 8-10.

In circuit 600, zero voltage detector 621 is configured to detect zero voltage at a second node 600B of the first resonator section 601 or an equivalent node 700B of the second resonator section 701. When the first resonator section 601 or the second resonator section 701 is active, the zero voltage detector 621 detects that the voltage across that FET 608, 708 returns to zero whenever the corresponding FET 608, 708 is turned off (e.g., point 800e in FIG. 8) or close to zero, such as below level 801. In response to this detection, zero voltage detector 621 outputs a signal to change the state of flip-flop 622. Then, the corresponding operating gate driver 623 outputs a gate-source voltage to switch the corresponding FET back to on state.

The first small-signal diode 725 connects the zero-voltage detector 621 to the second node 600B of the first resonator section, and the second small-signal diode 726 connects the zero-voltage detector 621 to the equivalent node 700B of the second resonator section 701. Specifically, the anodes of the first small-signal diode 725 and the second small-signal diode are connected to the input of the zero-voltage detector 621 through a common node 701B, while the cathodes of the diodes 725, 726 are connected, respectively, to nodes 600B, 700B.

Now, in a specific example, the operation of the zero voltage detector 621 will be described with reference to FIG. 10, which shows zero voltage detector 621 and trigger 622. FIG. 10, the components making up the zero voltage detector 621 are enclosed within a dotted rectangle. Assembly 701B is shown connected to the anodes of the first and second small signal diodes 725, 726. FIG. 13, a RUN signal 1011 can also be seen from controller 1001 to zero voltage detector 621.

The zero voltage detector 621 in this example includes an inverter gate U103 having an input 2 from node 701B and an output 4 connected to the input of flip-flop 622. The inverter gate U103 is powered from pins 5 and 3, and capacitor C108 isolates pin 5 from ground. The 621a logic power supply voltage of 2.5V in this example is applied to input 5 and through pull-up resistor R111 to input 2 of inverter gate U103. Logic power 621a in this example is supplied from controller 1001. Inverter gate U103 is configured to act as an OR gate for RUN signal 1011 and a zero voltage detection signal from node 701B. That is, the inverter valve U103 is configured to receive a low logic level signal in the form of a RUN signal 1011 from the controller 1001 to initiate operation of the circuit 600a. The RUN signal 1011 may be provided by an open drain signal contact of the controller 1001. The inverter gate U103 is also configured to receive a logic low signal from node 701B when the first or second signal diode 725, 726 is forward biased due to one of the nodes 600B, 700B have zero or close to zero voltage, as will be explained below. If one or both of these low logic level signals are applied to inverter gate input 2, inverter gate U103 inverts the received signal and outputs a logic high signal to flip-flop 622.

When the first inductor 124 operates to heat the sensing element 132, the second field effect transistor 708 remains turned off. If the second FET 708 remains off, then the second small signal diode 726 is either unbiased or reverse biased depending on the logic power supply voltages, and the DC supply 118 is the voltage at the cathode end (closest to node 700B) of the second small signal diode 726 is either substantially the same as or higher than the voltage at the anode end (closest to the zero voltage detector 621) of the second small signal diode 726.

During operation of the first resonator section 601, when the first field effect transistor 608 is turned off and the voltage thereon changes, as shown by reference numerals 800b-d in FIG. 8, the first small signal diode 725 is reverse biased. At the end of this voltage change, when the voltage reaches 0 V, as indicated by point 800e, or close to 0 V (eg, level 801 or below), the first small-signal diode 725 becomes positively biased. Accordingly, when the first small-signal diode 725 is forward biased at point 800e, the signal supplied to input 2 of inverter gate U103 becomes a logic low signal because a voltage drop is generated from logic signal 621a across resistor R111. Thus, as soon as inverter gate U103 inverts this logic low signal, output 4 of inverter gate U103 supplies a logic high signal.

Although the above description describes the operation of the zero voltage detector 621 in terms of controlling switching of the first field effect transistor 608, it is understood that the zero voltage detector 621 operates in the same manner when using the second small signal diode 726 instead of the first small signal diode 725 to drive the second field effect transistor 708.

Still with reference to FIG. 10, flip-flop 622 contains a CLK clock input, a reset/RST input, and a Q output. Flip-flop 622 also contains additional D and VCC inputs for power supply, in this example the flip-flop receives the same 2.5 V supply voltage 621a from controller 1001 , as inverter valve U103. The clock input CLK is connected to output 4 of the inverter gate U103 and is configured to receive a signal from it. When output 4 of inverter gate U103 is switched from a low logic level to a high logic level (due to the fact that input 2 of inverter gate U103 receives a detected zero voltage condition or RUN signal 1011, as described above), the clock input CLK of flip-flop 622 receives a rising signal. a logic high edge that "clocks" flip-flop 622 and makes the flip-flop's Q output state high. The flip-flop 622 includes an additional input /RST configured to receive a signal from the output of the comparator 618, with which the comparator 618 can change the state of the flip-flop 621 so that the output of the flip-flop Q is low. The Q flip-flop output is coupled to the first and second gate drivers 623, 723 and upon receiving a high output signal from the Q flip-flop output, the active gate driver 623, 723 (due to receiving signal 1021, 1022 as described above) supplies the gate driver signal to its corresponding field effect transistor 608, 708.

In one specific example, flip-flop 622 may be switched at half the voltage of logic power supply 621a, that is, 1.25 V. This means that the positive bias voltage of the first small-signal diode 725 and the drain voltage of the first FET 608D must add up to 1. 25 V to turn on the first FET 608. Therefore, in this example, the first FET 608 turns on when the voltage at its drain 608D is 0.55 V, and not exactly 0 V. It should be noted that ideally for maximum efficiency, switching should occur at 0V across the FET 608. This zero voltage switching advantageously prevents the first FET 608 from discharging the capacitors 606, 610, and thus losing the energy stored in the capacitors 606, 610.

In fig. 11 shows in more detail the first and second gate drivers 623, 723 and their connection to the gates 608G, 708G of their respective field-effect transistors 608, 708. Each of the gate drivers 623, 723 has an IN input that is configured to receive a signal dependent on the signals 1021 , 1022 heater activations supplied from the controller 1001. In addition, the signals received by the IN inputs of the gate drivers 623, 723 depend on whether the flip-flop output signal Q is high. The IN inputs are connected to the Q output of the 622 flip-flop through the corresponding resistors R125, R128, each of which has a value of 2 kOhm in this example.

Each gate driver 623, 723 has two additional inputs VDD and XREF, with each VDD input receiving a 6V supply voltage from controller 1001 and XREF receiving a 2.5V logic voltage, which in this example is the same logic voltage supplied by controller 1001 for trigger 622 and inverter valve U103. The VDD inputs of the first and second gate drivers 623, 723 are connected to a 6V supply voltage, and the VDD inputs are connected to ground through two buffer capacitors C120, C121. Each gate driver 623, 723 also has a GND pin connected to ground, with the VDD and GND pins operating to supply power to the gate drivers 623, 723. In this example, each capacitor C120, C121 has a value of 1 µF. Gate drivers 623, 723 are configured to output gate drive voltages from respective OUT terminals. The OUT outputs of the gate drivers 623, 723 are connected respectively to the gates of the FETs 608G, 708G through resistors R114, R115, each of which has a resistance of 4.99 ohms in this example.

Each gate driver 623, 723 is configured to receive a signal at its IN input to activate the gate driver only at the time the flip-flop's Q output is driven high and receives a heater activation signal 1021, 1022 from the controller 1001. At the controller 1001 an open drain signal terminal may be provided that is configured to provide signals 1021, 1022.

In the examples, turning on the circuit 600 to heat one of the resonator sections 601, 701 is performed by the controller 1001, which first initiates the desired one of the gate drivers 623, 723 using the corresponding heater on signal 1021, 1022. The controller 1001 then supplies a RUN signal 1011 to the zero voltage detector 621. The duration of the START signal 1011 must be shorter than the half-cycle period of the active resonator section 601, 701 (this period may be referred to as the "resonant flyback period"). This allows the circuit to properly begin to self-oscillate in response to a detected zero voltage condition. In another example, the order of the START signal 1011 and the corresponding heater enable signal 1021, 1022 may be reversed such that the START signal 1011 is first applied to set the flip-flop output Q to a logic high level, and then one of the heater enable signals 1021, 1022 is applied. to start self-oscillation of the resonator section 601, 701 corresponding to the heater to which the signal 1021, 1022 is applied.

To continue with a more detailed description of the control device for controlling circuit 600, FIG. 12 shows a portion of the control device containing comparator 618 and associated components. In fig. 12, the positive terminal 118a of the DC power supply 118 is shown connected to a node 1500A, which is connected to nodes 600A, 700A of the first and second resonator sections 601, 701, respectively. The negative terminal 118b of the DC power supply is connected to node 1500B, which is equivalent to node 600D shown in FIG. 6. Node 1500B is connected to ground 616 through a current-sensing resistor 615. Capacitors C111, C112, C115, and C116 are connected in parallel between nodes 1500A and 1500B, each of 100 µF in this example, providing buffering between nodes 1500A and 1500B.

In fig. 12 shows in more detail the components associated with operating the comparator 618 to detect that the current flowing through the active inductor 124 or 126 has reached a predetermined level. As described with reference to the previous figures, comparator 618 operates to compare a voltage indicative of the amount of direct current flowing in the active inductor (124 or 126) with control voltage 1031 emanating from controller 1001. Comparator 618 receives power through input 6, which connected via 100 ohm resistor R116 to a 2.5 V logic power supply, in this example supplied by controller 1001, and the same logic signal as signal 621a received by flip-flop 622. Comparator power input 6 is connected to ground via 10 nF capacitor C119 . The other pin of comparator 618 is connected directly to ground.

In some examples, controller 1001 is a microprocessor containing a timer (not shown) to generate a signal that generates control voltage 1031. In this example, control voltage 1031 is generated using a pulse width modulated signal PWM_DAC generated by controller 1001. The timer of controller 1001 generates a pulse-width modulated square wave signal, for example of about 2.5 V and a frequency of about 20 kHz, and having a specific duty cycle. The pulse width modulated PWM_DAC signal is filtered by 10 nF capacitors C127 and C128, two 49.9 kΩ resistors R121, R123, and a 10 kΩ resistor R124 to provide a substantially constant control voltage 1031 at the frequency at which controller 1001 controls the control voltage 1031 (for example, about 64 Hz in the examples). To adjust the control voltage 1031, the controller 1001 in the examples is configured to adjust the duty cycle of the pulse width modulated signal PWM_DAC applied to the circuit 600. Thus, components located between the PWM_DAC input and the positive terminal of the comparator 618 effectively provide the control voltage 1031 with using the pulse width modulated signal and adjusting the magnitude of the control voltage 1031 by adjusting the duty cycle of the pulse width modulated signal. Thus, control voltage line 619 shown in FIG. 6 and 9 can be replaced with these components. However, in other examples, control voltage 1031 may be generated by a substantially constant voltage supplied, for example, by controller 1001. In such examples, some or all of the components shown in FIGS. 12, for filtering the PWM_DAC signal may be missing.

Node 1500B, which is connected to the positive input of comparator 618, as mentioned above, is equivalent to node 600D of circuit 600. In FIG. 12 it can be seen that, as described with reference to the simplified diagram shown in FIG. 6, node 1500B is connected through resistor 617a to the positive input of comparator 618. Essentially, the operation of comparator 618 is the same as described above: receiving an input signal at its positive terminal, which depends on the control voltage 1031 and the voltage across the current sensing resistor 615. When the voltage is on The positive terminal of comparator 618 reaches the ground voltage, the /FF RST signal is output through resistor R118 to the /RST input of flip-flop 622 to change the state of flip-flop 622 and thereby turn off the active FET 608/708.

In fig. 13 shows additional components for a specific example of a control device for circuit 600. The components shown in FIG. 13, define a current sensing device 1300 to provide a signal I_SENSE indicating the amount of current consumed from the DC voltage source 118 during operation of the circuit 600. From this signal, the controller 1001 can determine the current consumed from the voltage source 118 and can use it in conjunction with a value voltage supplied by DC voltage source 118 to determine the value of power supplied to circuit 600. In some examples, as will be described below, the determined value of power may be used by controller 1001 to control circuit 600.

Input 1301 into current measuring device 1300 is made through resistor R120 shown in FIG. 12. Thus, the input is connected to node 1500B through resistor R120 and receives a voltage indicative of the voltage across current sensing resistor 615. Current measuring device 1300 operates as a low voltage side current measuring device for circuit 600. In this regard, current measuring device 1300 operates as a low voltage side current measuring device for circuit 600. In this regard, current measuring device 1300 contains op amp U110 operating from 3.8V supplied to input 5 of op amp U110 (part type TS507) configured to sense low side current using current sensing resistor 615 as will be understood. Transistor U109 with built-in bias resistor (part type RN4986) switches the 2.5 V supply from controller 1001 to 3.8 V for op amp U110. The power line from transistor component U109 is connected to ground through a 10 nF capacitor C132. In addition, a 1 kΩ resistor R130 is connected between the positive input of op amp U110 and ground, and a 412 kΩ resistor R129 is connected between the 2.5 V input from controller 1001 and the positive input of comparator U110. The negative terminal of op amp U110 receives a voltage dependent on the voltage across current sense resistor 615. Series resistor R131 and capacitor C133 provide filtering of the voltage signal received at input 1301. An additional resistor R133 (in this example with a resistance of 97.6 kΩ) and A 10 nF capacitor C134 is connected in parallel between the negative input of op-amp U110 and the output of op-amp U110 so that the op-amp operates in closed-loop mode.

The position of current sensing resistor 615, which as noted above is a 2 mΩ resistor in one example, in a circuit allows multiple parameters to be measured with a single current sensing resistor, which can allow good efficiency. That is, the position of the current-sensing resistor 615 in the circuit allows it to measure: the peak current of the FET, which can be used, for example, in controlling the power of an induction heating circuit; average value of current from the battery, which can be used when monitoring battery discharge and when setting inductive power; and an average value of current into the battery, which can be used, for example, when monitoring battery charge.

Operational amplifier U110 operates to output a voltage signal I_SENSE to controller 1001, which indicates current through current sensing resistor 615 and thus allows controller 1001 to sense the current drawn from DC voltage source 118 through circuit 600.

It should be noted that, taking into account the first and second field-effect transistor 608, 708 and the topology of the circuit 600, the phasing of the first and second inductors 124 and 126 relative to each other can be selected such that if the first inductor 124 is operating, flow through the second inductor coil 126 of a current sufficient to significantly heat the pantograph 132, and if the second inductor coil 126 is operating, then the first inductor coil 124 is prevented from flowing in a current sufficient to significantly heat the pantograph 132.

As described above, the first 608 and second 708 FETs actually act as diodes 608a, 708a when they are turned off, and thus can conduct current if they are positively biased (that is, the FETs are not ideal switches). Accordingly, in the examples, circuit 600 is configured such that when the first 124 or second 126 inductor is operated to heat the pantograph 132, the voltage induced across the other inoperative inductor does not cause a positive bias of the self-diode of the FET associated with that inoperative inductor. , but on the contrary shifts it in the opposite direction.

The effect of the above-described control device configured to control the switching devices 608, 708 of the circuit 600 in response to detected voltage conditions is that when the first resonator section 601 is active or the second resonator section 701 (that is, its gate driver 623, 723 is activated controller 1001), then this section of the resonator “auto-oscillates”, while the other section remains inactive. That is, the switching of the respective FETs 608, 708 in the resonator section 601, 701 is repeated at high frequency because the first voltage condition (detected by the comparator 618) causes the FET to switch from an on to an off state, and the second voltage condition (detected by a zero voltage detector 621 ) causes the field effect transistor to switch from off to on.

The controller 1001 is configured to control the induction heating circuit 600 of the device 100 such that only the first inductor 124 or the second inductor 126 is active at any time. The controller 1001 is configured to determine, at a predetermined frequency, which of the first coils 124 or the second 126 to activate.

In examples, during use of device 100, controller 1001 determines at a predetermined frequency, that is, once for each of a plurality of predetermined time intervals, which resonator section, the first 601 or the second 701, to activate. In one example, each time the controller 1001 determines which resonator section, the first 601 or the second 701, to activate, the controller 1001 may decide to activate that resonator section to heat the pantograph 132 for the next predetermined time interval. That is, if the predetermined frequency (which may be referred to as the "clipping frequency") is, for example, 64 Hz, then the controller 1001 may determine at predetermined intervals of 1/64 second which resonator section 601, 701 should be activated within next 1/64 second period until the controller makes the next determination of the resonator section 601, 701 at the end of the next 1/64 second period. In other examples, the interruption frequency may be, for example, from 20 Hz to 80 Hz, or accordingly, the predetermined intervals may have a duration of from 1/80 to 1/20 sec. To determine which inductor 124, 126 should be activated during a predetermined interval, the controller 1001 determines which pantograph zone 132a, 132b should be heated during this predetermined interval. In the examples, the controller 1001 determines which pantograph zone 132a, 132b should be heated with reference to the measured temperature of pantograph zones 132a, 132b, as will be described below.

In fig. 14 is a flowchart of an example method for determining which of the two resonator sections 601, 701 should be activated within a certain predetermined interval. In this example, the controller 1001 determines which of the resonator sections, the first 601 or the second 701, to activate during a predetermined interval based on the current temperature T1 of the first pantograph zone 132a heated by the first inductor 124 and the current temperature T2 of the second pantograph zone 132b heated the second inductor 126. In one example, the current temperatures T1 and T2 of the first 132a and second 132b pantograph zones may be measured by corresponding thermocouples (not shown) attached to each pantograph zone 132. The thermocouples provide an input signal to the controller 1001, allowing the controller 1001 to determine the temperatures T1, T2. In other examples, another suitable means may be used to determine the respective temperatures of pantograph zones 132a, 132b.

At block 1051, controller 1001 determines the current temperature value T1 and compares it with the target temperature value target1 for the first zone 132a to be heated by the first inductor 124. The target temperature target1 of the first zone 132a has a value that may change during a session of use of the device utilizing circuit 600. For example, a temperature profile may be defined for the first zone that defines values for target1 throughout a session of use of the device 100.

At block 1052, controller 1001 performs the same operation as for the first inductor 124 at block 1051, and determines whether, at this time, the current temperature T2 of the second zone 132b is lower than the target temperature target2 for the second zone 132b or not. Again, the target temperature of the second zone 132b may be determined by a temperature profile defining target2 values during a use session. The temperature of the second zone 132b may, like the first zone 132a, be measured by any suitable means, such as a thermocouple.

If both responses in block 1051 and block 1052 are “no”, that is, the temperature of both pantograph zones 132a, 132b are currently at or above their respective target temperatures target1, target2, then controller 1001 determines that neither the first one needs to be activated , nor the second resonator section 601, 701 for the next predetermined interval.

If the answer in block 1051 is “no” and the answer in block 1052 is “yes”, that is, the temperature of the first zone 132a is at or above its target temperature target1, but the temperature of the second zone 132b is below its target temperature target2, then the controller 1001 determines that the second resonator section 701 should be activated to heat the second zone 132b during the next predetermined interval.

If the answer in block 1051 is "yes" and the answer in block 1052 is "no", that is, the temperature of the first zone 132a is below its target temperature target1, and the temperature of the second zone 132b is at or above its target temperature target2, then the controller 1001 determines that the first resonator section 601 should be activated to heat the first zone 132a for the next predetermined interval.

If the answer in block 1051 is "yes" and the answer in block 1052 is "yes", that is, the temperature of both the first zone 132a and the second zone 132b are below the respective target temperatures target1, target2, then the controller 1001 proceeds to block 1053. In the block 1053, controller 1001 effectively operates to alternately turn on the first resonator section 601 and the second resonator section 701 for each predetermined interval during which both zones 132a, 132b remain below their respective target temperatures.

In one example, to alternately activate the first 601 and second 701 sections of the resonator, at block 1053, the controller 1001 in some examples determines whether an even number of predetermined intervals have passed since the start of the session. If an even number of predetermined intervals have passed since the start of the session, then the controller 1001 determines that the first resonator section 601 should be activated for the next interval. If an odd number of predetermined intervals have passed since the start of the session, then the controller 1001 determines that the second resonator section 701 should be activated for the next interval. It should be understood that in other examples, the controller 1001 may instead activate the second resonator section 701 if an even number of intervals have passed, and the first resonator section 601 if an odd number of intervals have passed.

In some examples, circuit 600 is configured such that once one of the resonator sections 601, 701 is activated by receiving a signal 1021 or 1022 on one of the gate drivers 623, 624, that resonator section 601/701 continues to operate, i.e. self-oscillate until deactivated by controller 1001, for example by applying another signal to the gate driver of this resonator section 601/701. Thus, after determining which of the resonator sections 601, 701 to activate during a given interval, the controller 1001, in order to initiate this activation, can deactivate one of the resonator sections 601, 701 that was active during the previous interval.

To illustrate an example block 1053, where the method 1050 shown in FIG. 14 are performed at 1/64 second intervals if controller 1001 determines that both zones 132a, 132b are below their respective target temperatures target1, target2, and an even number of 1/64 second intervals have passed since the start of the device 100 session. , then the controller 1001 activates the first resonator section 601 for the next 1/64 second interval while the second resonator section 701 becomes inactive, which in the examples requires the controller 1001 to deactivate the second resonator section 701. If after this next 1/64 second interval the temperatures of both zones 132a, 132b remain below their respective target temperatures target1, taget2, then during the next 1/64 second interval the controller 1001 activates the second resonator section 701 while the first section The resonator 601 becomes inactive, which in the examples requires the controller 1001 to deactivate the second resonator section 701. For each interval in which the temperatures of both zones 132a, 132b remain below their respective target temperatures, this alternation between activation of the first 601 and second 701 resonator sections continues.

In general, method 1050 results in the two inductors 124, 126 never being active simultaneously. If it is determined that both inductors 124, 126 require activation to bring the temperature of the respective zones 132a, 132b to a set temperature, then the controller 1001 alternately energizes the inductors 124, 126 at a predetermined frequency to bring the temperatures of both zones 132a, 132b to their respective target temperature. Therefore, during a use session, both pantograph zones 132a, 132b may be at such a temperature as to form an aerosol at a certain point in the use session, but at such a point in the use session, the activation of the inductors 124, 126 for heating their respective pantograph zones 132a, 132b may be switched with a certain frequency, for example 64 Hz. It can be seen that, for example, during a use session period comprising a plurality of intervals during which the temperature of the first zone 132a is substantially below its target temperature and the temperature of the second zone 132b is at or above its target temperature, power may be supplied by method 1050 into the first resonator section 601 for almost 100% of this period. However, during a period of use that includes multiple intervals during which the temperatures of both zones 132a, 132b are below target temperatures, each inductor may be energized for approximately 50% of that period.

In the examples, the controller 1001 is also configured at predetermined intervals, which in the examples coincide with the predetermined intervals at which the method 1050 is performed to determine the power supplied to one of the resonator sections 601, 701 from the DC source 118.

As described above with reference to FIG. 9-11, in particular, to control which of the resonator sections, the first 601 or the second 701, is active at any time, the controller 1001, also transmitting the RUN signal 1001 to initiate operation of the circuit 600, is configured to selectively transmit the first signal 1011 heater operation signal to the first gate driver 623 to activate the first resonator section 601, or a second heater operation signal 1012 to the second gate driver 723 to activate the second resonator section 701.

For example, if controller 1001 initiates operation of circuit 600 and controller 1001 transmits a first heater operation signal 1011, then circuit 600 operates as described above to activate the first inductor 124 to heat the first pantograph zone 132a. If controller 1001 transmits a second heater operation signal 1012, then circuit 600 activates second inductor 126 to heat second pantograph zone 132b. If the controller 1001 does not transmit either the first heater operation signal 1011 or the second heater operation signal 1012, then neither of the inductors 124, 126 is activated, and the pantograph 132 is not heated.

The controller 1001 is configured to control the power supplied from the DC voltage source 118 to the induction heating circuit 600 of the current collector 132 based on a comparison of the measurement of the power supplied to the circuit 600 and the target power. The controller 1001 is configured to control the power supplied to the circuit 600 by controlling the switching device of the circuit 600, that is, by controlling the switching of the field-effect transistors 608, 708. The controller 1001 can control the switching of the field-effect transistors 608, 708 by setting the control voltage 1031, which determines the constant a current that can build up in the inductor 124, 126 corresponding to the FET 608, 708 before switching the FET 608, 708 to the off state.

In fig. 15 illustrates an example of a method 1100 performed by controller 1001 to control power supplied to circuit 600. At step 1101, controller 1001 determines the power P supplied from DC source 118 to circuit 600. For example, controller 1001 may determine the average value of power supplied to circuit 600. 600 during the previous predefined interval. In the examples, the power P supplied to circuit 600 during an interval can be determined by measuring the voltage and DC current passing through a given one of the resonator sections 601, 701. The controller 1001 may then determine the product of voltage and DC current flowing through a given one of the resonator sections 601, 701 to determine the power P supplied to that resonator section 601, 701.

In the examples, the determined power P is the average power supplied from the DC source 118 during a predetermined interval, which can be determined by performing the product of the average DC voltage at the DC source 118 and the average DC current consumed from the DC source 118 over the previous interval.

In the example device 100, the DC source 118 is a battery that is connected to the controller 1001, and the controller 1001 then supplies the voltage of the DC source 118 to the circuit 600. The controller 1001 is configured to determine the DC voltage supplied by the battery 118. The current drawn from battery 118 is determined by the operation of the current measuring device 1300. The controller 1001 determines the constant voltage and constant current once for each 1/64 second interval. The DC voltage can be considered virtually constant during this short period of time. However, the current changes at a rate depending on the speed at which the circuit is turned on and off. As described above, in some examples this frequency is approximately 300 kHz. The current measuring device 1300, as described above with reference to FIG. 13 produces the I_SENSE signal, which is filtered to remove this signal at about 300 kHz. Thus, the average DC current for a 1/64 s interval is obtained by measuring this filtered I_SENSE signal, and the I_SENSE measurement is taken just before the end of the 1/64 s interval to allow the signal from the filter to settle. Thus, the controller 1001 obtains the DC voltage and DC current measurements for an interval of 1/64 s and can calculate the product of these values to obtain a certain power P. This determined power P can be thought of as the average of the power supplied by the DC source 118 during the interval duration 1/64 s.

At block 1102, the delivered power P determined at block 1101 is compared with the target power. If the determined power P is the average power over a predetermined interval, then the target power is the target average power over the same interval. In one example, the target power is the target for the average power supplied over a predetermined interval, and may be 10 to 25 W, or 15 to 23 W, or about 20 W. In this example, the target power is in the range, for example, 20-21 W or 15-25 W. The controller 1001, respectively, at block 1102, may compare the delivered power value P determined in block 1101 with the target range and determine whether the delivered power is below the range, within the target range, or above the target range. For example, if the target range is 20-21 W, then in block 1102, controller 1001 determines whether P < 20 W, or 20 W ≤ P ≤ 21 W, or P > 21 W.

Based on a comparison of the supplied power P with the target range, the controller 1001 determines whether and how the power should be adjusted for the next predetermined interval to bring the actual power supplied to the active inductor 124 or 126 during the next predetermined interval to the target power range . That is, if the supplied power P is below the target range, then the controller 1001 decides to increase the power supplied to the circuit 600 during the next predetermined interval. If the supplied power P is above the target range, then the controller 1001 decides to reduce the power supplied to the circuit 600 during the next predetermined interval. If the supplied power P is below the target range, then the controller 1001 decides not to regulate the power supplied to the circuit 600 for the next predetermined interval.

Due to the configuration of circuit 600 described above, the supplied power P for a given predetermined interval depends on the value of control voltage 1031 for that interval. Taking the example of a single 1/64 second interval for which the first resonator section 601 is active, that 1/64 second interval contains a plurality of repeating cycles comprising portions 800a to 800e of the voltage curve 800 and their repetitions. For each cycle, during the time period t1 to t0, the resonator section 601 is allowed to resonate, and since the FET 608 is turned off during this period, no power is consumed from the DC source 118 through the first resonator section 601. Thus, substantially all of the power consumed from the DC source 118 during a given 1/64 second interval to power the resonator section 601 is consumed during the period between t0 and t1 while the inductor 124 is energized, then is present while the field effect transistor 608 is turned on. The time between t1 and t0 is determined by the resonant frequency of the first resonator section 601. This resonant frequency may remain substantially constant for at least a given 1/64 second interval (although may vary over the period of operation of circuit 600 due to dependence on coil and current collector temperature and battery voltage). The duration of time from t0 to t1 is determined by the value of the control voltage 1031, as well as the DC voltage supplied by the DC source 118, and the resistance and inductance of the first resonator section 601 (the same applies to the second resonator section 701). That is, for a given DC supply voltage, the control voltage 1031 sets the current I1, which can rise in the inductor 124 between times t0 and t1, but if the DC supply voltage decreases, the time required for the setpoint I1 to rise increases. Thus, the average power delivered over a 1/64 second interval depends on the value of control voltage 1031.

Therefore, in the examples, in order to control the power supplied to the circuit 600 during the next interval, the controller 1001 sets the control voltage value 1031 for the next interval. In examples, for a given DC source 118, during a predetermined interval during which one of the resonator sections 601, 701 is active, a greater positive value of control voltage 1031 results in a greater value of power P supplied to circuit 600. Therefore, in such examples, if controller 1001 determines that the delivered power P for the last interval was above the target range, then controller 1001 reduces control voltage 1031 for the next interval. If controller 1001 determines that the delivered power P for the last interval was below the target range, then controller 1001 increases control voltage 1031 for the next interval. And if the controller 1001 determines that the supplied power P for the last interval was above the target range, then the controller 1001 leaves the control voltage 1031 unchanged for the next interval.

It should be noted that in one example of the above method 1100, the supplied power P determined in block 1101 is the power supplied to a particular one of the resonator sections 601, 701. For example, the power P can be determined by measuring the voltage across the first resonator section 601 and the direct current flowing through the first resonator section 601. In such an example, the power P supplied to the first resonator section 601 is used to control the control voltage 1031. It should also be noted that for a given control voltage 1031, in some examples, the power supplied to each of the inductors 124, 126 when the respective sections 601, 701 resonators are active, may be different. This may be, for example, because the inductors 124, 126 have different inductance or DC resistance values, or the capacitances of the two resonator sections 601, 701 are not equal. Therefore, in this example, during a predetermined interval, a target power outside the target range may be supplied to the second resonator section 701, but since the control voltage 1031 is adjusted based on the power P supplied to the first resonator section 601, in this example, the controller 1001 may not adjust control voltage 1031.

For example, for a given value of control voltage 1031, controller 1001 may determine at block 1101 that the first resonator section 601 has been supplied with an average power of 20 W with a target voltage for a given interval in this example of 20-21 W. At step 1102, controller 1001 determines that the applied voltage was within the target range, and accordingly, controller 1001 decides not to adjust control voltage 1031. Note that for the next predetermined interval, controller 1001 determines (using example method 1050) that it should the second resonator section 701 be activated rather than the first resonator section 601. For a given value of control voltage 1031 in this example, 22.5 W is supplied due to differences in the electrical properties of the first 601 and second 701 resonator sections. However, in this example, at block 1102, the controller 1001 compares the last measured value of the power P supplied to the first resonator section 601, and therefore, at block 1103, decides not to adjust the control voltage 1031. Essentially, in the example method 1100, the actual power supplied to the circuit 600 may be outside the target range. However, this may allow the power supplied to the inductors 124, 126 to be controlled by measuring only the power P supplied to one of the resonator sections 601, 701. This may provide a simple and useful solution for maintaining the power supplied to circuit 600 within an acceptable range if, for example, the resonator sections 601, 701 and their components have approximately the same electrical properties.

As mentioned above, in some examples, the DC source 118 is a battery with a voltage of about 2 to 10 V, or 3 to 5 V, or, in one example, about 4.2 V. In some examples, the DC voltage generated DC voltage source 118 may vary, such as decrease, during operation of circuit 600. For example, if DC voltage source 118 is a battery, then the battery may initially supply a voltage of 4.2 V, but the voltage supplied by the battery may decrease as the battery discharges . Therefore, after a predetermined period, the constant voltage source 118 may supply, for example, 3.5 V instead of the initial 4.2 V.

As described above, at a given supply voltage, the value of the control voltage 1031 controls the amount of current that can build up in the active inductor 124/126 before turning off the corresponding FET 608/708. Power is supplied from a DC voltage source 118 to “energize” the active inductor 124/126, allowing the DC current to build up when the FET 608, 708 is turned on. As also described above, the time t1 for the current to rise to a value that causes the FET 608/708 to switch depends on the DC voltage source. Therefore, for example, if the voltage supplied by the DC source 118 decreases, the rate at which the current in the inductor 124 rises decreases, resulting in a decrease in the power P supplied to the circuit 600.

An example method 1100 may maintain a target power even when the voltage supplied from the DC source 118 changes. That is, since the actual supplied power P is detected and used to control the control voltage 1031, the controller 1001 may operate to maintain the target power by adjusting the control voltage 1031. For example, if the battery charge level is depleted, the controller 1001 measures that the power P, supplied to circuit 600 at a given control voltage 1031 has decreased and acts to increase the power P supplied to circuit by increasing control voltage 1031. As such, the target power level may be maintained while the battery used to power circuit 600 is discharged . This is advantageous because maintaining a given power level can ensure optimal operating efficiency of the induction heating circuit 600. For example, maintaining a substantially constant power supply allows for constant heating of the aerosol material 110a regardless of the supply voltage. Likewise, example method 1100 provides for substantially constant power regardless of other varying factors in the circuit that may affect the amount of power delivered, such as varying load on circuit 600 imposed by pantograph 132 as the temperature of pantograph 132 increases. This provides a consistently good experience for the consumer, for example by providing a constant time to first puff, that is, a constant time between activation of the device 100 and its readiness to dispense an aerosol for inhalation by the user.

In another example, the measured power value P on which control of the control voltage 1031 is based changes during a usage session. For example, during a particular use session, for the first portion of the use session (eg, the first ~60 s of the use session), the temperature profile may be such that primarily the first inductor 124 is active and the second inductor 126 is inactive. For this first part of the use session, it may be appropriate to base control of the control voltage 1031 on measurements of the power supplied to the first resonator section 601. However, later in the session, again for example due to the temperature profile for the session, it may be that the second inductor 126 is primarily active while the first inductor 124 is active for a lesser amount of time. Thus, for the second portion of a use session (eg, after ~60 s), it may be useful to control the control voltage 1031 based on measurements of the power supplied to the second resonator section 701. The controller 1001 may accordingly switch from controlling the control voltage 1031 based on measurements of the power supplied to the first resonator section 601 to controlling the control voltage 1031 based on measurements of the power supplied to the second resonator section 701. In this way, the target power can be more accurately maintained throughout the entire use session because, for example, the control voltage 1031 is set based on a comparison of the actual power supplied to the active inductor 124, 126 with the target power range.

In some examples, where controller 1001 at block 1103 determines that power needs to be adjusted, controller 1001 may adjust control voltage 1031 in predetermined steps. For example, the controller 1001 may be configured to adjust the control voltage 1031 by a predetermined amount over a predetermined period of time. If, at block 1102, controller 1001 determines that the supplied power P is below the target power range, then controller 1001 may increase control voltage 1031 by a predetermined number of volts for the next predetermined interval. Conversely, if, at block 1102, controller 1001 determines that the supplied power is above the target power range, then controller 1001 may increase control voltage 1031 by a predetermined amount for the next predetermined interval.

In the example described above with reference in particular to FIG. 12, control voltage 1031 is generated using a pulse modulated signal PWM_DAC. The PWM_DAC signal, as described above, is a square wave at 2.5 V. The duty cycle of the PWM_DAC signal is controlled by the controller 1001, which sets a value from 0 to 800 for the PWM_DAC duty cycle, this value corresponds to a duty cycle of 0% at 0 and 100% at 800 The PWM_DAC signal, after filtering, provides a substantially constant control voltage 1031, and therefore the duty cycle settings of the PWM_DAC signal 0 to 800 ensure that the control voltage 1031 has a value of 0 to 2.5 V. In this example, the controller 1031 can adjust the duty cycle setting PWM_DAC signal to a set value, for example 8 out of 800, or leave the setting unchanged for each predefined interval. In another example, the controller 1001 may cause the control voltage 1031 to be adjusted by some other means, and if the controller 1001 determines that the control voltage 1031 should be adjusted, the controller 1001 may adjust the control voltage 1031, for example, to a level of 1%, or 2%, or 5 % of the maximum control voltage value 1031 for the next preset interval.

In some examples, when controller 1001 initiates operation of circuit 600, such as to begin a session of use of device 100 containing circuit 600, control voltage 1031 is set to a predetermined initial value. In one example, a control voltage value 1031 (eg, setting the duty cycle of the PWM_DAC signal that produces that control voltage value 1031) that corresponds to a target power level is determined during tuning of circuit 600. That is, the power supplied to circuit 600 can be determined (eg, measured or theoretically determined) for a range of control voltage values 1031, for example, to construct a calibration curve. The control voltage 1031 value corresponding to the target power can then be determined. In one example, DC source 118 may supply 4.2 V, and to achieve a target power of 20 W, controller 1001 may determine a PWM_DAC signal duty cycle setting value of approximately 344 out of 800 during calibration.

In one example, controller 1001 is configured to set control voltage 1031 to an initial value that is based on this determined value of control voltage 1031.

For example, the initial duty cycle value PWM_DAC, which determines the control voltage 1031, can be set to half a certain value corresponding to the target power. For example, if the duty cycle setting for control voltage 1031, which is determined to correspond to the target power, is 344 out of 800, then controller 1001 may begin the session with the setting set to 152 out of 800 and increase the setting by a predetermined amount with each advance. at specified intervals until the measured power P is within the target range. This may result in the delivered power being well below the target power at the start of a use session, and then the delivered power may increase (by controller 1001 increasing the control voltage 1031) until it reaches the target power range. This initial increase in delivered power may provide increased operational safety for circuit 600 by preventing the pantograph from overheating early in the session and allowing circuit 600 to respond to the actual applied power determined by controller 1001.

In one example, the predetermined interval is the same predetermined interval that the controller 1001 uses in the method 1050 of determining which of the first 124 and second 126 inductors to activate. In one such example, as mentioned above, the predetermined intervals have a duration of 1/64 sec. The duration of the predetermined interval (or, equivalently, the interrupt frequency) can be selected to provide a favorable time interval during which the controller can monitor the circuit and adjust parameters accordingly. For example, a chopping frequency of 64 Hz or in the range of approximately 10-100 Hz may be used. At these example chopping frequencies, controller 1001 can measure the temperature rise of pantograph zones at a fast enough rate that it can decide to stop heating a particular inductor 124, 126 before the temperature of pantograph zones 132a, 132b of pantograph 132 can also increase well above the target temperature. . Likewise, the examples given for the chopping frequency may provide a preferred frequency at which the control voltage 1031 can be adjusted to provide adequate control of the power supplied to the inductors 124, 126 within a safe target range.

In an example of the method of operation of the circuit 600, the target power for use by the controller 1001 in controlling the power supplied to the circuit 600 is predetermined based on the characteristics of the scheduled usage session. For example, the target power range can be adjusted during a usage session.

In fig. 16 shows a schematic example of a target temperature profile target1 for part of a usage session, which in this example is the target temperature for a single pantograph zone 132a. In this example, initially in the first portion 1201 of the use session, the temperature of the first zone 132a is substantially below its target temperature target1. In this first portion 1201, circuit 600 operates to bring the temperature of first zone 132a to target temperature target1. In such an example portion of a usage session, the target power P1 may have a range of values, for example, 20-21 W. The target power during the first portion 1201 of the session may be relatively high to quickly bring the temperature of the pantograph 132, and therefore the aerosol material 110a, to a temperature suitable for generating an inhalable aerosol by the user.

As the use session continues, the first zone 132a substantially reaches its target temperature target1. The second portion 1202 of the usage session may be set to begin shortly after the first zone 132a reaches its target temperature target1. For example, for this use session portion 1202, the first zone 132a may substantially have a predetermined target temperature target1, such as 250°C, and may be held at the target temperature target1 in accordance with method 1050. Likewise, although not shown in FIG. 16, the temperature of the second zone 132b may be maintained equal to its own target temperature setpoint target2 using method 1050 (and the target temperature target2 of the second zone 132b may define a temperature profile different from the profile determined by target1).

The portion 1202 in a use session after the first zone 132a has substantially reached the set temperature target1 may be characterized by the controller 1001 operating to maintain the temperature of the first zone 132a (or both zones 132a, 132b) rather than to bring the temperature of the first zone 132a to the target value target1, as in the first portion 1201. Thus, during portion 1202 of the use session, relatively little power may be required to be supplied to the pantograph zone 132a to maintain the target temperature target1, compared to the power required to bring the zone temperature 132a pantograph to target temperature target1. In the second usage portion 1202, it may be advantageous to reduce the value of the target power P1 compared to its value in the portion 1201. In one example, the target power level P1 may be reduced from 20-21 W in the portion 1201 to about 15 W in the usage portion 1202 . In another example, the target power P1 may be reduced to about 12-13 W or to about 9 W. Reducing the target power P1 in this manner may be advantageous in some examples because by using a lower power level, energy loss in the circuit can be reduced and thus efficiency can be increased.

For the third part 1203 of the use session, the value of the target temperature target1 is 0, that is, the first inductor 124 is not activated. At this point, the target power P1 may also be reduced to 0 if the use session has come to an end, or, if the second inductor 126 is still activated, then the target power P1 may remain at a non-zero value as long as the second inductor 126 is active. Accordingly, the target power may take into account the temperature profile of both zones 132a, 132b at any point during the use session. If, for example, part of a usage session requires a significant increase in temperature in one of the zones, then a relatively high target power may be required. Conversely, for portions of a usage session where significant heating of any zone 132a, 132b is not required, a relatively low power target may be used.

As mentioned above, using lower power levels during certain periods of a usage session may provide the advantage that energy savings can be achieved over the duration of the session. For example, if the target power level is reduced from 20-21 W in the first period to about 15 W in the second period, then in some examples, energy savings of about 5-10% can be achieved by reducing energy losses in circuit 600 when operating at lower power. . In one example, during a typical session lasting about 260 seconds, maintaining the target power at about 20 W for the entire duration of the session could result in energy consumption of about 1000 J. However, reducing the target power to about 15 W after the first zone 132a first reaches set temperature, and maintaining the target power level at 15 W for the remainder of the session, essentially the same duration, can result in energy consumption of 900 to 950 J. In the examples, almost all of the energy used by the device is due to the energy supplied for heating the current collector 132. The power consumption of electrical components other than the heating circuit, such as LED indicators and microcontroller, can be less than 0.1 W, and in some examples less than 0.01 W.

It should be understood that the method of determining the delivered power and comparing it with a target power and adjusting the delivered power based on this comparison may be applied to example heating circuits for an aerosol generating device other than those described in the examples provided herein. Moreover, the principles of the method for adjusting the target power throughout a use session may also be applied to other example circuits, such as an example circuit that uses a resistive heating element to heat the aerosolized material rather than an inductive element. That is, from the above description it is clear that in some examples, energy savings, for example about 5-10%, can be achieved by reducing the target power supplied to the heating device for heating the aerosolized material. Particularly in the examples, this reduction in target power may be preferably implemented once the target temperature has been reached for a given point in the use session, such that the example apparatus operates to maintain, rather than increase, the temperature of a heating device, such as a resistive heating element.

Certain methods described herein may be implemented by non-volatile computer program code that may be stored on a non-volatile storage medium. For example, in some examples, controller 1001 may include a non-transitory computer-readable storage medium containing a set of computer-readable instructions stored thereon, and a processor for performing the method described herein by controller 1001. Controller 1001 may include one or more processors. For example, in some examples, as described above, the controller 1001 is a programmable microprocessor. The controller 1001 may include a storage medium containing a set of machine-readable instructions, such as computer code, that when executed by the controller 1001 causes the method described herein to be performed.

It should be noted that although a circuit containing two inductors has been described above, the aspects described above, for example for controlling the power supplied to an induction heating circuit, can be applied to circuits with a different number of coils, for example with one or more with two coils. In addition, although this document describes an inductive circuit containing inductors, the aspects described herein can be equally applied to an inductive circuit using other types of inductive elements having inductance and suitable for creating an alternating magnetic field for heating current-receiving structure.

Although some of the example circuits described in this document use silicon FETs for some switching functions, other suitable components can be used in place of the FET. For example, components containing wide bandgap materials such as silicon carbide, SiC, or gallium nitride, GaN, can be used. Such components in some examples may be field effect transistors, but in other examples they may be high electron mobility transistors (HEMTs). Such components may be faster and may have higher breakdown voltages than silicon FETs, which may be preferable in some examples.

The above embodiments are to be understood as illustrative examples of the invention. Other embodiments of the invention are provided. It should be understood that any feature described with respect to any one embodiment may be used alone or in combination with other features described, and may also be used in combination with one or more features of any other embodiment or any combination of any other embodiments. . In addition, equivalents and modifications not described above may also be used without departing from the scope of the invention, which is defined in the accompanying claims.

Claims (41)

1. An induction heating device for an aerosol generation device, containing: an induction heating circuit, the circuit comprising: an inductive element for inductively heating the current-receiving structure for heating the aerosol-forming material, thereby producing an aerosol; capacitive element; And a switching device configured, in use, to switch between a first state and a second state to receive a varying current from a constant voltage source and flow it through the inductive element to cause inductive heating of the current collecting structure; And a control device, wherein the control device is configured to switch the switching device from a first state to a second state in response to a first voltage condition detected in the circuit, and the control device is configured to switch the switching device from a second state to a first state in response to the second condition by voltage detected in the circuit. 2. The induction heating apparatus according to claim 1, wherein the first voltage condition indicates the amount of magnetic energy that is stored in the inductive element due to direct current flowing through the inductive element. 3. Induction heating device according to claim 1 or 2, in which: The switching device is configured in a first state to cause direct current to flow through the inductive element and thereby store magnetic energy in the inductive element, and in a second state to prevent direct current from flowing through the inductive element, so that if the switching device is in the second state, then current can oscillate between the inductive element and the capacitive element. 4. The induction heating apparatus of claim 3, wherein the control device comprises a voltage comparator, and the voltage comparator is configured to detect the first voltage condition by comparing a voltage indicating the amount of direct current flowing through the inductive element with a control voltage. 5. Induction heating device according to any one of paragraphs. 2-4, in which the circuit also contains a resistor and a voltage indicating the amount of direct current flowing through the inductive element depends on the voltage across the resistor. 6. The induction heating apparatus according to claim 4 or 5, wherein the control voltage is variable to control the power supplied to the circuit. 7. The induction heating apparatus of claim 6, comprising a controller configured to detect power supplied to the circuit and compare the power supplied to the circuit with the target power, and control a control voltage based on the comparison of the power supplied to the circuit with the target power . 8. Induction heating device according to any one of paragraphs. 2-7, in which the second voltage condition indicates that from the moment the switching device is switched from the first state to the second state, a predetermined fraction of the current oscillation cycle between the inductive element and the capacitive element has been completed. 9. The induction heating apparatus of claim 8, wherein the control device comprises a zero voltage detector and the zero voltage detector is configured to detect a second voltage condition, wherein the second voltage condition is a zero voltage condition or a near zero voltage condition at the switching device , which, when the detector detects zero voltage, indicates that from the moment the switching device switches from the first state to the second state, half a cycle of current oscillation between the inductive element and the capacitive element has been completed. 10. Induction heating device according to any one of paragraphs. 1-9, in which the control device includes a trigger for switching between two states, wherein the state of the switching device depends on the state of the trigger, and the control device is configured to detect a first voltage condition and a second voltage condition, as well as a change in the state of the trigger to change the state switching device. 11. The induction heating device according to claim 9, wherein the control device comprises a trigger for switching between two states, wherein the state of the switching device depends on the state of the trigger, and contains a comparator configured to detect the first voltage condition, wherein the control device is configured to the ability to detect a first voltage condition and a second voltage condition, as well as a change in the state of the flip-flop to change the state of the switching device, and the flip-flop is configured to receive first input data from the comparator if the comparator detects the first voltage condition, and second input data from the zero detector voltage if the zero voltage detector detects a second voltage condition. 12. Induction heating device according to any one of paragraphs. 1-11, in which the switching device includes a field-effect transistor. 13. The induction heating device according to claim 12, in which the control device is configured to switch the state of the field-effect transistor, selectively, causing voltage to be applied to the gate terminal of the field-effect transistor. 14. Induction heating device according to any one of paragraphs. 1-13, in which the inductive element and the capacitive element are arranged parallel to each other in an induction heating circuit. 15. Induction heating device according to any one of paragraphs. 1-14, in which an inductive element, a capacitive element and a switching device are located in a first resonator section, and the device also includes a second resonator section containing a second inductive element, a second capacitive element and a second switching device, wherein: the second inductive element is configured to inductively heat the current receiving structure to heat the aerosol-forming material, thereby producing an aerosol; the second switching device is configured, in use, to switch between the first state and the second state to receive a varying current from the constant voltage source and flow it through the second inductive element to cause inductive heating of the current collecting structure; And The control device is configured with the ability, if the first section of the resonator is active: switch the first switching device from a first state to a second state in response to detecting a first voltage condition in the first section of the resonator; And switch the first switching device from the second state to the first state in response to detection of a second voltage condition in the first section of the resonator; And, The control device is configured to also, if the second section of the resonator is active: switch the second switching device from the first state to the second state in response to detection of a first voltage condition in the second section of the resonator; And switch the second switching device from the second state to the first state in response to detection of a second voltage condition in the second section of the resonator. 16. The induction heating apparatus of claim 15, comprising a controller configured to selectively activate the first resonator section and the second resonator section such that only one section, the first resonator section or the second resonator section, is active at any given time. 17. An aerosol generation device containing an induction heating device according to any one of claims. 1-16. 18. The aerosol generating device according to claim 17, which is a tobacco heating device, also known as a non-combustion heating device. 19. A control device for controlling the induction heating circuit of the aerosol generation device, wherein: the control device is configured to control the switching device in the induction heating circuit to, when in use, cause the switching device to switch between a first state and a second state to receive alternating current from a direct current source and pass it through the inductive element in the induction heating circuit, causing induction heating of the current collector designs; and wherein the control device is configured to switch the switching device from a first state to a second state in response to a first voltage condition detected in the circuit, and the control device is configured to switch the switching device from a second state to a first state in response to a second voltage condition detected in the circuit in the chain. 20. The control device according to claim 19, which is also configured to: detect a first voltage condition and a second voltage condition in the induction heating circuit; And in response to detection of the first voltage condition, switch the switching device from the first state to the second state; And in response to detection of the second voltage condition, switch the switching device from the second state to the first state. 21. An aerosol generation system comprising an aerosol generating device as claimed in claim 17 or 18 and an article containing an aerosol-forming material for heating the material by said device in use, thereby producing an aerosol. 22. The aerosol generation system of claim 21, wherein the device is the aerosol generation device of claim 18, wherein the aerosol generating material comprises tobacco.