RU2336626C2 - Method of heterodyne signal penetration control in direct conversion methods - Google Patents

Abstract

FIELD: physics, radio.

SUBSTANCE: invention concerns radio communication and can be applied in multiband wireless communication devices. System and method of heterodyne (HD) frequency generation in a receiver or transmitter with zero intermediate frequency (IF) are claimed. Method of HD frequency generation in multiband wireless communication device involves receiving first signal from voltage-controlled generator (VCG) with VCG frequency within first frequency band; division of the first signal frequency by programmed value N to obtain second signal with downconverted frequency, with programmed value N change partially based on master signal; and first signal is mixed with second signal to obtain output signal with HD frequency within the second frequency band determined by the first frequency band and programmed value N.

EFFECT: reduced heterodyne signal penetration allowing receiver or transmitter to operate in several radio communication modes and bands and to comply with relevant technical specifications.

50 cl, 8 dwg

Description

The invention relates generally to radio communications. In particular, the invention relates to systems and methods for direct conversion transceivers.

The great development of communication technology has largely happened due to the increased capabilities of radio devices. Radio devices use radio waves to carry out long-distance communications without the physical limitations inherent in wired systems. Information, such as speech, data, or paging information, is carried by radio waves transmitted in certain frequency bands. The distribution of available frequency spectra is regulated so that multiple users can communicate without interference.

Information transmitted from a source point to a destination rarely comes in a format ready for transmission by radio. Typically, a transmitter receives an input signal and formats it for transmission in a given frequency band. An input signal, also called a modulating signal, modulates the carrier in the desired frequency band. For example, a radio transmitter receiving an audio input signal modulates the carrier frequency with an input signal.

The corresponding remote receiver tuned to the same carrier frequency as the transmitter must receive and demodulate the transmitted signal. That is, the remote receiver must recover the modulating signal from the modulated carrier. The modulating signal may be directly presented to the user or subjected to further processing before being presented to the user.

Transceivers are radio devices that combine both a transmitter and a receiver. Transceivers provide almost instant two-way communication. Examples of transceivers are: duplex radio communication, portable duplex radio stations, duplex subscriber receivers paging system and radiotelephones.

To assess the effectiveness of the design of the receiver, the following essential Q-factors are of primary importance. Sensitivity determines the ability of the receiver to detect a weak signal. The sensitivity of the receiver must be such that it can detect the minimum distinguishable signal (MRS) from the background noise. The noise is arbitrary fluctuations in voltage and current. MRS is a measure of the sensitivity of a receiver, including the frequency band of a given system. On the other hand, the selectivity of the receiver characterizes the protection provided to the receiver from extra-channel interference. The higher the selectivity, the better the receiver will be able to suppress unwanted signals.

Sensitivity reduction - a decrease in the overall sensitivity of the receiver due to artificial or natural external radio interference (GRP). Sensitivity reduction occurs when a very strong interfering signal overloads the receiver and makes it difficult to detect weaker signals. The sensitivity response of a receiver determines its ability to operate successfully under the influence of strong sources of interference, such as active intentional radio interference.

Noise is another important indicator of receiver performance. The noise performance is deteriorating, i.e. increases in each subsequent cascade in the receive channel. To provide an acceptable noise response, a gain or attenuation technique can be applied to the receiver. Noise, along with distortion, determines the ratio of the sum of the signal, noise and distortion to the sum of noise and distortion in decibels and characterizes the performance of the receiver in the presence of noise.

Distortion - the presence of unwanted signals at the output of the devices in the RF channel of the receiver. Distortion may include harmonic distortion, intermodulation distortion, and cross-modulation distortion. Harmonic distortion occurs when the desired input signal is so strong that non-linear distortion occurs in the receiver, and they are usually measured on the output signal of the frequency band of the modulating signals as a function of the frequency shift from the desired signal and as a function of the desired signal power. Distortions caused by cross-modulation occur when the amplitude-modulated component from the transmitter (for example, the MSDRC radio telephone) is transferred to another carrier (active intentional radio interference) on the output signal of the device (output signal of a low-noise amplifier). The most common type of distortion is intermodulation distortion (IMD).

Intermodulation distortion is the result of two or more signals that mix with each other and produce additional unwanted distortions in the frequency band of the signal. In the case of two input signals, intermodulation components arise in the sum and difference of integer multiple of the initial frequencies. That is, in the case of two input signals with frequencies f ₁ and f _{2, the} components of the output frequency can be expressed as mf ₁ ± nf ₂ , where m and n are integers ≥1. The order of the intermodulation component is the sum of m and n. The "two-tone" third-order components (2f ₁ -f ₂ and 2f ₂ -f ₁ ) can occur at frequencies near the desired signals or which are interference signals, and therefore their easy filtering is impossible. Higher-order intermodulation components have lower amplitudes and, as such, present less difficulty. Second order intermodulation interference-generating components can be formed in the frequency band of the modulating signal if the frequency range is within half the bandwidth of the signal.

FIG. 1 shows a graph of the levels of the components of the IMR of the fundamental frequency, of the second and third orders, depending on the level of the input signal. The theoretical points at which second and third order levels cross the fundamental frequency are known as a second order intersection point (TP2) or a third order intersection point (TP3). TP2 of the receiver is a point of intersection of the second order of the input signal level. TP3 is the third-order intersection point of the input signal level. The third-order intersection point and the noise characteristic of the receiver are directly related to the dynamic range of the receiver. The dynamic range determines the range of signals that the receiver can process within the specified operating parameters of the receiver; those. the range in which the receiver can give an accurate output with an acceptable ratio of the sum of the signal, noise and distortion to the sum of noise and distortion. In particular, for a baseband receiver, such as an analog-to-digital converter, the dynamic range can be represented as an imaginary free dynamic range: from the device’s minimum noise level to the maximum signal, before the restriction occurs.

The leakage of the local oscillator (HD) signal occurs when the HD signal leaks into the input signal of the receiver. This leakage can be transmitted by the antenna of the transceiver in the form of spurious emissions, which can cause interference with other devices. In addition, HD leakage can be reflected back to the receiver itself and can reduce its sensitivity if it is not eliminated before demodulation.

Leakage of a signal of active intentional radio interference occurs when a signal of such interference leaks into the input or output signal of the main signal of any device in the receiver. This leakage can be mixed with the signal of the active intentional radio interference and therefore produce unwanted signals such as levels of the constant component of the signal proportional to the amplitude modulation component of the signal of the intentional radio interference. Amplitude-modulated signals of active intentional radio interference can be at any frequency within the received frequency band.

The low-frequency flicker noise (woofer / m) is due to defects in the emitter junction of the transistors. Although flicker noise and other noises of this kind are usually weak, they must be eliminated in the receiver in order to maintain signal integrity in the frequency band of the modulating signals.

Isolation is the ratio (in decibels) of the power level applied to one terminal of the device to the received power level at the same frequency that occurs on the other terminal. Decoupling - the reciprocal of the allocation, is one of the indicators of the quality factor of the components of the receiver. The isolation shows the extent to which the energy input to the output terminal is reflected back to the input source. In order to achieve a low level of HD leakage and leakage of active intentional radio interference, a very effective isolation is desirable.

The 1 dB non-linear distortion point in an amplifier is a measure of the output power level when the amplifier gain is 1 dB lower than the gain at a low signal level. The saturation point of an amplifier is a measure of the maximum output power of an amplifier. This characteristic is shown in FIG. one.

The above mentioned characteristics and signal features must be taken into account when designing radio communication devices. In general, recently in the radio communications Code Division Multiple Access (MSDCS) has prevailed - extended-spectrum or broadband communication in which radio signals travel over a very wide frequency band. Many modulation standards are based on the MDCDC technique, such as the MDCDC (IS-95 and CDMA2000) and WCDMA (IMT2000). All of these airborne modulation or interfacing standards are valid in many radio frequency bands, including Cellular (Cellular in Japan, Cellular in the USA), SES (Private Communications System in the American and Korean frequency bands) and ICES (International Telecommunication Union). Other modulation standards include the following standards: FM (frequency modulation, IS-19), GSM (Global System for Mobile Communications), US-TDMA (IS-136), GPS (Global Satellite Radio Definition System), Wireless LAN (802.11), Bluetooth

Frequency bands are distributed in different communication modes. For transceivers, U.S. Emergency Response Systems accepts the 1930-1990 MHz frequency band and transmits 1850-1910 MHz on the corresponding frequency band. Cellular US reception frequency band: 869-894 MHz, and the corresponding transmission frequency band: 824-849 MHz. Similarly, the transmit and receive frequency bands are allocated for Cellular Japan, MESC and Korea's Emergency Situations.

Communication standards set out the specifications that must be followed by radio communications devices. For example, specifications must be respected for spurious emissions, sensitivity, intentional active interference (two-tone intermodulation and one-tone sensitivity reduction) and residual sideband.

On an international basis, and even within the countries themselves, radio communications have not yet been standardized. The current level of technology recognizes that a transceiver that can operate in more than one band, or in more than one mode, has increased functional mobility. In particular, dual-band handsets operate on two frequency bands. For example, a dual-band handset handset of the MDCRC can operate in the 800 MHz (Cellular USA) and 1.9 GHz (USEF frequency bands) bands. If the base stations operating on these two bands use the MSDKRC standard, then the mobile device with the double-band handset of the MSDKKK standard can be served by one or the other, or both of these base stations. In addition, the double-band handset of the MSDKRK / FM standard can operate in both modes: MSKDKK and FM. But with today's variety of modulation standards and corresponding frequency bands, dual-mode and dual-band phones can provide subscribers with only limited compatibility with the world's current communication systems.

FIG. 2 shows a schematic block diagram of a conventional dual receiver in which the frequency is down-converted. Receiver 101 has a superheterodyne receiver architecture. In particular, the received RF signal 11 is supplied via the RF signal path and undergoes preliminary processing (cascade 1). The pre-processed RF signal 13 is first converted with decreasing frequency into a signal 15 having an intermediate frequency (IF) (cascade 2). The IF signal 15 is then again downconverted to a modulating signal 17, which includes a “in-phase” (I) and “quadrature” (Q) phase component (cascade 3). The I- and Q-components of the modulating signal differ in phase by 90 °. The I- and Q-components are then sent to the other components of the receiver 101, for example, to the baseband processor (cascade 4), for further processing. Similarly, in a dual upconverter, the analog I and Q modulating signals are first upconverted to an IF signal and then the IF signal is upconverted to a transmitted RF signal.

FIG. 3 illustrates a receiver 101 in more detail. Receiver 101 has several inherent advantages. For example, this design provides good sensitivity and selectivity, extended dynamic range of the signal, flexible frequency planning and reduced dynamic range and current consumption for receiver elements 101 after IF filters 70. In addition, phase and amplitude matching between I- and Q-channels 106, 107 can be facilitated since the IF signal is in the lower frequency range. Due to these advantages, the receiver 101 is very suitable for multi-mode and multi-band applications, and it is possible to process the received RF signals - modulated in several modes and transmitted in several frequency bands.

To ensure several ranges and several modes of operation, the receiver 101 must contain some components specifically for these modes. For example, in a multi-band receiver, a separate RF signal path is typically required for each frequency band. In a multi-mode receiver, separate paths of the baseband frequency band may be required for each mode, depending on the requirements that the dynamic range of active intentional interference must adhere to.

In conventional receivers, such as receiver 101, the IF signal path typically consists of amplifiers, filter circuits, and automatic gain control (AGC). Therefore, the receiver 101 can eliminate noise originating outside the signal band and active deliberate interference and can compensate for changes in signal power and receiver gain. In a multi-mode receiver, IF signals are filtered for a specific mode. Therefore, the receiver 101 has one IF filter 70 per mode. For example, a receiver in a dual-mode telephone has two IF SAWs (surface acoustic wave filters). For a receiver operating in the MSDKRK 1X, MSDKRK 3x, WCDMA, GSM, FM, Bluetooth, and Global Satellite Radio Detection modes: in the IF signal path, four to six SAW filters and 1 discrete high-pass filter may be required.

The need for an IF filter for each mode is a significant disadvantage of the receiver 101. Each IF filter increases the cost of the receiver, the number of important parts, and increases the area of the receiver circuit board. Since each IF filter can have large losses, an IF preamplifier or AGC may also be required. A voltage-controlled IF generator (VCO) and a phase locked loop (PLL) system 65 are also required to generate a local oscillator (GP) frequency, which is introduced into the IF mixer 60. Other disadvantages of receiver 101: the need for a switching matrix or several IF amplifiers and AGC modules; the need for a low-loss RF bandpass filter (PF) to reduce unwanted sideband noise and the presence of additional IF mixers. Therefore, the IF cascade of a dual receiver with downconversion increases the cost, complicates the design and increases the PCB area in these receivers.

FIG. 4 shows a block diagram of a direct downconverting receiver 200 with zero IF. In receivers with direct conversion with decreasing frequency, the received RF signal 201 is directly converted with decreasing frequency into a modulating signal 225. Similarly, in a transmitter with direct conversion with increasing frequency, or with zero IF, the modulating signal is directly converted with increasing frequency into a transmitted RF signal. At receiver 200, the received RF signal is mixed with the frequency of the HD to obtain a modulating signal. Since it does not have an IF signal path, the receiver 200 eliminates the cost increase, the increase in PCB area and power consumption due to IF components, such as SAW IF filters, high-pass filters and discrete filters, preamplifier, AGC, IF mixers, and VCO and PLL IF. In this case, less inter-detail and temperature fluctuations occur.

The design of the receiver 200 allows greater signal processing, for example filtering channel selectivity, in the analog or digital region of the modulating signal via integrated circuits, thereby increasing the uniformity of the RF and analog components of the receiver 200. Since the AGC is digital, calibration can be simplified or calibration is not required at all . For some operating modes, such as the Global Satellite Radio Detection System, Bluetooth and GSM, the 200 receiver may not need an RF filter, as this filter is primarily intended to reduce cross-modulation in such MDSCC modes as Cellular and SSC. But for the Global Satellite Radio Detection System mode, an RF filter may be required if the modulated signals of this system are simultaneously received with other modulated signals.

Despite the advantages mentioned, direct down-conversion in radio telephones has not been widely used. The reason is that it is very difficult to solve the main structural problems of the receiver while ensuring the proper dynamic range for the receiver. Structural tasks for receivers such as receiver 200 are to provide high gain and low noise characteristics, high TP3 and TP2, and low power consumption. A multi-mode and multi-band receiver may require a very wide dynamic range. Accordingly, it is even more difficult to solve these structural problems for this receiver.

In particular, the leakage of the main signal and leakage of the signal of the active intentional radio noise to the main outputs of the mixer I and Q cause significant difficulties in receivers with direct conversion with decreasing frequency. For Cellular and CES modes, the requirements for spurious emissions are particularly stringent. Therefore, a higher level of isolation is needed. In addition, in a receiver with direct conversion with decreasing frequency, the leakage of the main signal reflected back to the receiver itself, as well as the leakage of active intentional radio noise to the output of the main mixers I and Q, can be processed by the direct conversion circuit with decreasing frequency. Therefore, an undesired input constant zero bias voltage may occur at the output of the mixer together with an undesirable modulating signal, which may also contain spectral components of the frequency band of the modulating signals. Accordingly, the input constant zero bias voltage must be eliminated in order to provide a sufficiently high signal to noise ratio.

In the MDRC, the sensitivity is checked by a signal set to a level at which a certain error rate of the data block appears. According to the IS-98 standard, the device under test must comply with a sensitivity level of -104 dBm (signal power), while the specified coefficient must be less than 0.5%. Intermodulation testing involves setting the signal level to -101 dBm (3 dB higher than in the sensitivity test) with two offset intervals relative to the RF signal (-43 dBm / interval for offsets creating in-band distortion components, or usually ± 900 and ± 1700 kHz), with a data block error rate of less than 1%. Depending on the frequency band, the tested power levels and frequency shifts for active intentional radio interference may be different. To test a one-interval decrease in sensitivity, the level of active intentional radio interference in the RF output of I and Q mixers is higher than the signal level by 71 dB at a shift of ≥900 kHz.

The power of the active intentional radio interference can leak into the DG output of each mixer and mix with the level of active intentional radio interference in the RF output of the mixer to obtain a constant current level proportional to the amplitude of the RF active intentional radio interference. Typically, active intentional radio interference is generated by the direct link of a competing radio base station. The power of active intentional radio interference may vary depending on the modulation or attenuation used. The worst active intentional radio interference can have amplitude modulation comparable to the desired signal bandwidth. In this case, the AM component falls to the top of any signal energy in the frequency band of the modulating signals after conversion with decreasing frequency, and it cannot be eliminated by filtering the frequency band of the modulating signals. This difficulty is worsened if the RF signal of the active intentional interference is amplified. If the RF signal of the active intentional radio interference increases by 10 dB, for example, then the distortion of the frequency band of the modulating signals increases by 20 dB. These distortions of the frequency band of the modulating signals can actually exceed the steepness of two-to-one, if both RF-in-GD isolation of RF mixers, which affects the self-mixing of active intentional radio interference, and TP2 of RF mixers, which characterizes the effects of second-order distortions, are insufficient.

The requirements for the leakage of active intentional RFI and DG for mixers in a receiver with direct conversion with decreasing frequency are very strict. Since there is no IF filtering in this type of receivers, therefore, the dynamic range of the elements of the frequency band of the modulating signals of the receiver may need to be increased by 30 dB or more, depending on the degree of analog filtering of the frequency band of the modulating signals, inter-detail, frequency and temperature fluctuations in the gain . Residual sideband specifications for various modulation standards must also be respected. Since a receiver of this type has a lower gain to its cascade of the baseband frequency, therefore, flicker noise in the baseband of the baseband has a greater effect on the receiver's ability to process FM modulated signals.

Therefore, there is a need to provide a direct conversion transceiver that can modulate RF signals in several ranges and several modes.

The disclosed embodiments of the invention illustrate novel and improved systems and methods for generating a local oscillator (GD) frequency in a direct frequency conversion radio communication device. According to one embodiment, the system comprises a voltage controlled oscillator (VCO), a divider, and a mixer. The divider has an input and an output, the signal of which is obtained by dividing the input signal. The input of the divider is operatively connected to the VCO. The mixer has a first mixer input, which is operatively connected to the VCO; the second input of the mixer, which is operatively connected to the output of the divider; and has a way out. The output signal of the mixer provides the main frequency for the phase shifter and the second divider in parallel.

According to other embodiments, the system comprises a VCO, a first divider, a second divider and a mixer. The first divider has an input and an output, the signal of which is obtained by dividing the input signal. The input of the first divider is operatively connected to the VCO. The second divider has an input and an output, the signal of which is obtained by dividing the input signal. The input of the second divider is operatively connected to the output of the first divider. The mixer has a first mixer input, operatively connected to the output of the first divider; the second input of the mixer, operatively connected to the output of the second divider, and has an output.

According to another embodiment, the system comprises a DG generator, a frequency band selection mechanism, and a configuration selection mechanism. The DG generator has one or more configurations and contains a mixer configured to mix the VCO frequency with a divided version of the VCO frequency. Each configuration refers to the frequency band of the RF signals and produces an output signal whose frequency refers to the frequency band of the RF signals. The frequency band selection mechanism is configured to select a frequency band of the RF signals. The configuration selection mechanism is configured to select a configuration related to the selected frequency band of the RF signals.

The signs, objectives and advantages of the disclosed embodiments are illustrated by the following detailed description in conjunction with the drawings, in which similar symbols indicate everywhere similar components, and in which:

FIG. 1 is a graph of saturation points and non-linear distortions and intersection points of the second and third orders.

FIG. 2 is a schematic block diagram of a conventional double conversion receiver.

FIG. 3 is a block diagram of a conventional double conversion receiver.

FIG. 4 is a schematic block diagram of a direct conversion receiver.

FIG. 5 is a block diagram of a direct conversion receiver.

FIG. 6 is a block diagram of a system for generating a local oscillator frequency according to an embodiment of the present invention.

FIG. 7 is a block diagram of a system for generating a local oscillator frequency according to an embodiment of the present invention.

FIG. 8 - implementation of the transmitter with zero IF.

FIG. 4 shows a schematic block diagram of a direct downconversion receiver 200 according to an embodiment of the present invention. The receiver 200 includes an RF signal path 210, a direct down-converter 220 and a baseband processor 230.

The RF signal path 210 receives the RF signals 201. The RF signals 201 may comprise signals modulated in several modes and transmitted in several frequency bands. The RF signal path 210 may include selection mechanisms for selecting from several different modes and different bands. In addition, the RF signal path 210 may include amplifiers or filters for preparing the RF signals 201 for further processing. These prepared signals are referred to as pre-processed RF signals 215: FIG. 4. The direct down-converter 220 receives the pre-processed RF signals 215 from the RF signal path 210 and converts them down to the baseband signals 225.

The processor 230 of the frequency band of the modulating signals can perform the subsequent processing of the modulating signals 225, for example: neutralization of the direct current, matched filtering and filtering of active intentional radio interference, decimation of the sample, automatic gain control, measurement of signal power (signal strength indicator of the received signal), signal convolution, reverse alternation error correction and decoding into digital data or audio streams. The processed information can then be sent to its intended purpose, for example, to the output mechanism in a radio device, for example, to a display, speaker or data output. It should be noted that the baseband processor 230 can also be used by a transmitter to supplement receiver 200.

FIG. 5 illustrates a receiver 200 in more detail. Antenna 301 performs the function of interconnecting receiver 200 with incoming RF signals. Antenna 301 may also broadcast RF signals from a transmitter connected to antenna 301. Several antennas can be used to separate the frequency bands and to separate simultaneously operating modes from each other. The interface 305 can extract the received RF signals from the transmitted RF signals, whereby the receiver 200 and the transmitter can share the antenna 301.

The interface 305 may include one or more antenna switches 312. The antenna switch 312 filters the signals in the incoming receive band. The antenna switch 312 also separates the signals in the incoming reception band from the signals in the outgoing transmission band. Multiple antenna switches 312 may be used if multiple operating bands are needed for a given receiver or transceiver application. According to FIG. 5, a single antenna switch 312 can process signals modulated in the MDCRC, FM, and MSES modes, assuming that all relevant operating bands are included in a given antenna switch band 312.

Interface 305 may also include one or more switches 314 and bandpass filters 316. Switch 314 selects between receive and transmit operations. For example, switch 314 may correspond to GSM or Bluetooth modes in which signals are not received and transmitted at the same time. A band-pass filter 316 filters the signals of the Global Satellite Radiodetermination System in the inbound reception band. Since the signals of the Global Satellite Radiodetermination System are received and not transmitted, the use of an antenna switch is therefore not necessary. Other bandpass filters 316 may be provided at receiver 200 for other analogue reception-only modes.

A low noise amplifier (LNA) 320 is coupled to an interface 305 and amplifies the received RF signals. The LNA 320 can be used to provide minimum noise characteristics in the reception band, but a sufficiently high gain to minimize noise from subsequent stages in the receiver 200. The gain of the LNA 320 can be adjusted by the LNA controller 324. Transmit power may seep into receiver 200 from interface 305. For example, antenna switch 312 may not completely filter transmit power. Therefore, the LNA 320 may require a high compression ratio and a third-order intersection point.

LNA 320 is connected to a band pass filter 330 (PF) reception. PF 330 also suppresses transmitter signals that go beyond the reception band. It should be noted that PF 330 may not be required in some embodiments of this invention. For example, as indicated above, signals modulated in GSM mode may not be received and transmitted at the same time if the maximum values of the data rate in the OSPRS (General Service Packet Radio Communication) are not provided.

FIG. 5 illustrates an RF signal path including one antenna switch 312, one LNA 320, and one PF 330. But the receiver 200 may also have several RF signal paths. Each signal path may correspond to one or more specific operating frequency bands of the receiver 200. For example, the receiver 200 may have corresponding signal paths of Cellular, MES, MSES and GSM. Each RF path may contain, if necessary, an antenna switch, a switch and / or a band-pass filter, an LNA, a PF, and I- and Q-mixers. In addition, for the simultaneous reception of the signals of the Global Satellite Radiodetermination System while working with other modes, you may need a separate HD generator, baseband amplifiers, analog low-pass filters, analog-to-digital converters, digital I / Q processing and demodulation.

The selection mechanism 310 switches between different paths of the RF signals, depending on the currently operating frequency bands. The selection mechanism 310 may include a band selection device connected, for example, to different antenna switches and PFs. The selector 310 can also be connected to mixers 340A, 340B of channels I and Q. For example, for received signals in the US Cellular band, selector 310 can switch to antenna switch 312, LNA 320 and PF 330 to filter and amplify received signals accordingly signals.

The output of the PF 330 is connected to the input of the mixers 340A 340V I- and Q-channels. According to an exemplary embodiment, PF 330 may have a differential output (not shown) connected to the differential inputs (not shown) of the mixers 340A, 340B. Accordingly, the positive and negative output terminals of the PF 330 can be connected to the positive and negative input terminals of the mixer 340A, and to the positive and negative input terminals of the mixer 340B. Such a differential arrangement of the signal path reduces the connection between the local oscillator and the transmitter with the RF signal path and increases the suppression of the common mode of the amplitude-modulated active intentional radio interference (higher, second order, level of intersection of the input signal at the inputs of the mixer). Thus, in the receiver 200, the isolation and suppression of active intentional radio interference is improved.

Or a transformer can be connected to the unbalanced output of PF 330. The transformer can convert an unbalanced signal into a differential signal, which can be routed to the differential inputs of the mixers 340A, 340V.

According to FIG. 5 local oscillator (DG) 350 is connected with buffer amplifiers 351A, 351B. Buffer amplifiers 351A, 351B are connected to the second input 342A of the amplifier 340A and to the second input 342B of the mixer 340B, respectively. Buffer amplifiers 351A, 351B can have differential outputs if I- and Q-mixers 340A, 340B have differential inputs. In some implementations, it is not necessary to include buffer amplifiers in the design of the receiver 200.

DG 350 may include a frequency generator that can generate output signals at various frequencies. For example, the DG 350 may output a first signal and a second signal that will have a phase shift of 90 ° compared with the first signal. DG 350 may include a phase locked loop (PLL), a voltage controlled oscillator (VCO); frequency mixing mechanism and phase shift mechanism. The DG 350 may include a band selector 354 that adjusts the DG 350 depending on the operating frequency of the received RF signals. In an exemplary embodiment, the DG 350 uses differential paths to reduce HD leakage and reduce noise coupling to signal paths in and out of the RF outputs of mixer I and Q.

Each of the mixers 340A, 340B mixes the received RF signal from the PF 330 with the signal received from the HD 350 at the second input 343A, 342B of the mixers 340A, 340B. The mixing process multiplies the signals. The mixers 340A, 340B therefore directly directly convert the received RF signals to modulating signals I and Q with decreasing frequency. In an exemplary embodiment: mixers 340A, 340B have a corresponding gain that can be adjusted using the mixer gain control 341A, 341B.

After down-conversion: the I and Q signals are processed in the respective signal paths 365A, 365B. Signal I path 365A characterizes both signal paths and may include an amplifier 360A, an anti-aliasing filter 370A, and channel I analog-to-digital converter 380A. Amplifier 360A is connected to the output of mixer 340A. After processing and analog-to-digital conversion in the respective signal paths: the digital I-channel data 382 and the Q-channel data 384 can be further processed. In some implementations, the signals I and Q can be processed in the working paths for certain modes. In accordance with other implementations, the signal paths I and Q can be used in different modes together.

Receiver 200 may include dedicated modules for Bluetooth mode. Depicted in FIG. 5, the direct down-converter 390 and the Bluetooth mode modulator 395 can functionally and structurally be similar to the structures described above. But since the Bluetooth mode can work simultaneously with other operating modes, such as MDCRC, therefore, the direct down-converter 390 and the processor 395 of the frequency band of the modulating signals of the Bluetooth mode can be performed as modules special for this mode. Similarly, the Global Satellite Radiodetermination System can act in conjunction with other modes, and it requires a separate modulating signal path and a generator circuit.

6 illustrates a system 400 for generating a local oscillator frequency according to an embodiment of the present invention. It should be noted that system 400 may be part of radios, transmitters, or transceivers. For example, system 400 may be included in receiver 200 as DG 350 — FIG. 5. System 400 comprises a phase locked loop (PLL) 410, a low-pass filter 401 of a synchronous detector, a mixer 450, a voltage controlled oscillator (VCO) 420, and a switch 440.

The switch 440 may be made in the form of a multi-position switch. 6, the switch 440 is a coordinate switch having three positions. In the first position (1-2), which is designated here as “forward”, the switch 440 connects the VCO 420 to the input of the divider 430. In the second position (2-3), “back”, the switch 440 connects the output of the mixer 450 to the input of the divider 430 In the third position (1-3), “bypass”, the switch 440 couples the VCO 420 to the output of the mixer 450 and the second input of the mixer 450 is taken out of action. Although it is shown that system 400 includes a switch, in other implementations of system 400, a switch is not needed.

For example, a VCO 420 can be directly connected to a divider 430. The position of the switch 440 can be controlled using a regulating mechanism (not shown), such as a band selector, depending on the frequency of the band of received RF signals.

VCO 420 may comprise a VCO with an unbalanced output that will be external to the microcircuit containing the corresponding receiver, transmitter, or transceiver. An external VCO may have more optimal characteristics in terms of phase noise than a VCO integrated into the SIS (specialized integrated circuit). But VCO as an integral part of the scheme may be sufficient - depending on the relevant requirements for suppressing active intentional radio interference in a given operating band. If an external VCO 420 is used: the PLL 410 can be directly connected to the VCO 420. In addition, the PLL 410 can be connected to the output of the mixer 450 if the PLL 410 is part of the 400 system. The PLL 410 receives a signal at a reference frequency of 405 to create discrete intervals channels in each working lane.

The implementation of FIG. 6 contains an input PLL switch 445. Switch 445 can couple the PLL 410 to the VCO 420, to the output of the mixer 450, or to the output of the divider 430. According to the well-known prior art PLL 410, the low-pass filter 401 of the synchronous detector and the VCO 420 interact in the sense of generating a signal having a VCO frequency. The frequency of the VCO can be higher or lower than the frequency of the received or transmitted signals. The divider 430 may include a frequency divider that outputs a signal whose frequency is a divided variant of the input signal. For example, the divider 430 may divide by an integer N, and the value of N can be set by the control signal.

The VCO 420 is connected to the first input of the mixer 450. In accordance with the position of the switch 440, as described above, the second input of the mixer 450 can be connected - through a divider 430 - to the VCO 420 ("forward"), the output of the mixer 450 ("back") or to an open circuit ("bypass"). The mixer 450 may include a single sideband frequency mixer (OBHF), a mirror frequency suppression mixer that provides only one primary component of the mixer. The OBFC mixer minimizes the undesired components of the mixer at the mixer output. In particular, the OBPCh mixer provides an output frequency signal that is either the sum of two input frequencies (upper side band, or PWB), or the difference of two input frequencies (lower side band, or NBP). The upper OBFC mixer retains the upper sideband and suppresses the lower OBPC. Conversely, the lower OBPCh mixer retains the lower sideband and suppresses the upper sideband. The mixer 450 can be configured to operate in either PFS or NBP modes, depending on the control signal input to the mixer 450.

System 400 may also include a second divider 470, which generates quadrature DG signals 490. The second divider 470 can divide the input frequency by an integer M and may consist of triggers. If the divider 470 consists of two triggers, then the first trigger can synchronize the front of the input signal, while the second trigger will synchronize the slice. The corresponding output signals of the triggers may not coincide in phase by 90 ^° . Moreover, each of the triggers can differentially actuate any of the I- and Q-mixers 340A, 340B. In accordance with other implementations: buffer amplifiers 351A, 351B can be installed between the second divider 470 and I - and Q-mixers 340A, 340B. If M = 2, i.e. the second divider 470 divides by 2, the second divider 470 functions as a broadband phase shifter, used when used in conjunction with the divider 430 for a wide range of frequencies. The second divider 470 can generate the DG signals of the I- and Q-mixer for the Cellular bands of the USA and Japan.

Phase shifter 460 may be included in system 400 in parallel with second divider 470. Or, system 400 may contain only phase shifter 460. Phase shifter 460, which may include an LCR network or active elements, may be connected to the output of mixer 450. Phaser 460 may receive an input signal and generate quadrature HD output signals 480. In the case of a receiver: each quadrature signal can be mixed with the received RF signals to convert the lower frequency RF signals into I- and Q-component frequency bands. signals. In an exemplary embodiment, the phase shifter 460 is valid for higher operating bands of the ESS (American or Korean) and MESC.

According to an embodiment of the present invention, the N value for the divider 430, the position of the switch 440, and the mode of the mixer 450 can be changed to generate a wide range of HD frequencies. In addition, you can change the value of M for the second divider 470. Although it is possible to generate a wide range of HD frequencies, it is necessary that the VCO 420 only work in a relatively narrow tuning range. Accordingly, system 400 may be implemented in a multi-band and multi-mode radio, transmitter, or transceiver.

In exemplary embodiments, system 400 has differential signal paths. For example, the output signal of the VCO 420 and the input and output signals of the mixer 450 and the divider 430 may be differential. Therefore, it is possible to minimize the radiated HD energy I and Q and the conducted communication into the RF signal path in a radio device containing system 400.

A microprocessor (not shown) in a radio device comprising system 400 is configured to determine an applicable frequency band for RF signals. Based on the selected band, a configuration selection mechanism such as the band selector 354 of FIG. 5, may select in the system 400 a configuration related to a selected frequency band. Therefore, the system 400 can generate appropriate control signals to set the N value for the divider 430, the position of the switch 440, the mixer mode 450, and the M value for the second divider 470.

Table 1 sets forth example configurations for a system 400 configured for a receiver. VCO 420 is tuned to operate in the approximate frequency range from 1600 to 1788 MHz. VCO 420 may be the primary source of radiated and conducted noise in a radio device. According to Table 1: VCO frequency ranges are different from the corresponding RF reception ranges. The configurations listed below minimize the effects of VCO noise in the radio.

Table 1
DG control configurations for multi-band receiver with direct downconversion RF band
frequencies RF range
reception frequencies
(MHz) Value
divider
N Mixer
OBHR Position
switch
la / value
second
divider, M Frequency
range
VCO in
receiver
(MHz) SES, USA 1930-1990 8 PFS forward 1716-1769 Cellular
USA 869-894 Not valid
howls Not valid
howls Bypass and de
2 years 1738-1788 Cellular
Japan 832-870 Not valid
howls Not valid
howls Bypass and de
2 years 1664-1740 MSEC 2110-
2170 0 PFS Forward 1688-1736 Scs
Korea 1840-
1875 8 PFS Forward 1635-1651

According to the present invention, to optimize the tuning range and the average carrier frequency of the VCO 420 for different reception frequency ranges and for different design methods, other configurations can be prepared, for example: an external DG or integral VCO. To provide these configurations, system 400 may include additional dividers.

Divider 430 and mixer 450 may generate unwanted spurious signals outside the desired reception band. But the output of mixer 450 will suppress these spurious signals. The leads 342A, 342B of the mixers 340A, 340B I and Q (see FIG. 5) may contain resonators that can also suppress these spurious signals. The paths of the RF signal can also have multiple RF characteristics of the PF, which can suppress the components of the active intentional radio interference at the same frequencies as the false signals of the main speaker.

As indicated above, the second divider 460 can create quadrature HD signals 480. Mixers 340A, 340B I and Q receive, as input signals, quadrature HD signals 480, which can be supplied by buffers 351A, 351B as input signals. Phase changes in the load resistance and the electrical capacitance of the mixers 340A, 340V I and Q can cause system errors. But due to the implementation of I- and Q-mixers 340A, 340V on the same chip, the requirements for phase matching can be observed. Therefore, it will be possible to comply with technical specifications regarding the residual sideband.

Amplitude matching may be necessary between I and Q channels. An example of amplitude matching methods may be to calibrate the gain of I- and Q-channels using analog or digital gain compensation. To perform analog gain compensation (not shown), an independent or switchable power detection mechanism can be connected to each of the I- and Q-channels to measure the level of the received signal of the channels and accordingly shift the gain. SIS remembers calibration values for I- and Q-channels. Using the digital bus interface between the SIS and the power detection mechanism, you can consult about the calibration values and compensate for the gain. To digitally compensate for the gain (not shown), the baseband path may include a digital multiplier after an analog-to-digital converter that multiplies the I- and Q-signals. Thus, the calibration values stored in the SIS can be used as reference values, and accordingly, the gain of the I- and Q-channels can be compensated.

According to another embodiment (not shown), the radio or transceiver may comprise a signal path specific to the Global Satellite Radio Determination System (GSSRO). GSSRO-modulated signals are received in only one frequency. Therefore, the receiver only needs to tune to one GSSRO frequency. In particular, a specialized GSSRO path may have a PLL and a VCO exclusively for GSSRO signals. A VCO, which can be located on or off the microcircuit, can operate at a frequency of 3150.84 MHz, or a dual GSSRO frequency. VCO GSSRO therefore can be connected to a divider (dividing by 2) and divide it by this divider to generate a GD frequency for direct conversion with decreasing frequency of RF signals of GSSRO. Although the receiver may have a separate GSSRO RF signal path, the GSSRO modulating signal band path may be separate or shared with signals modulated in accordance with other modulation standards. If this path is separate, then the processing of the frequency band of the modulating signals with respect to the GSSRO signals can occur simultaneously with the processing of the frequency band of the modulating signals with respect to other modulated signals. In the case of sharing, you can save current and area of the printed circuit board.

Since the Bluetooth mode can operate simultaneously with other operating modes, such as MSDCS, therefore, a separate VCO and a DG generator can be included in the receiver or transceiver to facilitate the generation of the DG frequency for direct conversion with lowering the frequency of the signals of the Bluetooth mode.

FIG. 7 illustrates an alternative system 500 for generating a local oscillator frequency. System 500 includes a PLL 570, a lowpass filter 560 of a synchronous detector, a multi-band VCO 501, a VCO divider 520, an OBPC mixer 540, an OBPC divider 530, and a reception divider 550. A multi-range VCO 501, PLL 570 and a low-pass filter 560 of a synchronous detector jointly derive a VCO frequency in variable frequency ranges. The band selector 510 determines the applicable frequency range for the multi-band VCO 501.

A VCO divider 520 is connected to a multi-band VCO 501. A VCO divider 520 can divide the VCO frequency by an integer P, for example, 2. A divided output signal from the VCO divider 520 is fed to the input of the OBCH divider 530. The VCH divider 530 can divide the output frequency of the VCO divider 530 by an integer, for example - 2. The output of the OBCH divider 530 and the output of the VCO divider 520 are sent to the corresponding inputs of the OBPC mixer 540. The OBPC mixer 540 mixes the signals together. Depending on the action of the OBPC mixer 540 as a PWM mixer or NBP mixer: the sum or difference of the input signals is output by the mixer 540. Therefore, the VCO divider 520, the OBPC divider 530 and the OBPC mixer 540 act together as a frequency multiplier with a fractional multiplication factor . The output of mixer 540 is routed to the input of the reception divider 550. A reception divider 550 divides the input signal by an integer, such as 1 or 2.

By changing the frequency band of the multi-band VCO 501 and also changing the mode of the OBCH divider 540, the division values of the VCO divider 520, the OBCH divider 530 and the reception divider 550, the system 500 can generate a wide range of frequencies of the HD. Table 2 shows exemplary configurations for the system 500, which make the system 500 suitable for a multi-band radio.

table 2
DG control configurations for multi-band receiver with direct downconversion RF band Range
RF reception frequencies
(MHz) VCO Divider Value The value of the OBL divider OBHH mixes
tel Receive Divider Value Receive VCO Frequency Range (MHz) SES, USA 1930-1990 2 2 PFS one 2573-2653 Cellular, USA 869-894 2 2 PFS 2 2317-2384 Cellular, Japan 832-870 2 2 PFS 2 2219-2320 MSEC 2110-2170 2 2 PFS one 2813-2893 SES, Korea 1840-1875 2 2 PFS one 2453-2500

In system 500, other configurations are possible. For example, system 500 may include a bypass multiplier switch 580 connected to a multi-band VCO 501 and a receive divider 550. With the switch closed, the multi-range VCO 501 can operate at 2- or 4-fold the operating frequency of the received signals. The reception divider 550 can therefore accordingly divide the output frequency of the VCO by 2 or 4 in order to generate the desired frequency of the HD. In particular, to generate the HD signals of the Cellular I and Q mixer: VCO 501 can operate at 4 times the receive frequency, and the reception divider 550 can divide by 4. But tuning can be difficult due to the wide operating range of the multi-range VCO 501. It should be noted that the system 500 can directly connect the multi-range VCO 501 to the reception divider 550 and that the bypass switch 580 of the multiplier, the OBPC divider 530, the OBPC mixer 540 and the VCO divider 520 can be excluded from the system 500.

In this case, the system 500 may include a switch (not shown) connected to the multi-band VCO 501 and input 545 of the OBCH mixer 540. With the switch closed: the OBPC mixer 540 can mix the VCO frequency with a divided version of the VCO frequency. Therefore, the system 500 can generate the HD signals of the I- and Q-mixer in an order similar to that carried out in the system 400.

FIG. 8 illustrates an embodiment of a direct conversion upconverter or zero IF transmitter 600. Transmitter 600 includes a system 602 that generates a DG frequency. The system 602 is similar to the system 400 described above, but is specially designed and operates in a radio transmitter with direct conversion with increasing frequency. The system 602 includes a PLL 610, a lowpass filter 601 of a synchronous detector, a first and second OBFC mixers 645, 650, VCO 620, an input PLL switch 641, DG switches 640A and 640B, and a second divider 670.

The transmitter phase noise requirements are less stringent than for a receiver that must comply with active intentional radio interference requirements. Therefore, the VCO 620 can be more easily integrated into the SIS transmitter or transceiver. But in other implementations, the VCO 620 can be performed outside the integrated circuit. A VCO 620, a lowpass filter 601 of a synchronous detector, a PLL 610 and a reference frequency generator 605 in conjunction generate a VCO output frequency. The PLL input switch 641 can selectively connect the PLL 610 to the VCO 620, to the output of the divider 630, or to the output of the first PLL mixer 645. Therefore, the input source for the PLL 610 can be switched from the VCO 620 either to the signal at the output of the divider 630, or to the signal on the output of the first OBFC mixer 645. Therefore, if the desired RF frequency is generated, frequency capture may occur.

Each of the switches 640A, 640B has two positions. In other implementations, additional provisions are possible. According to other implementations: provide switches 640A, 640B is not necessary. In the forward position of switch 640A, it connects the VCO 620 to the input of the first OBFH mixer 645. In the "back" position, switch 640A connects the output of the OBFH mixer 645 to the main output of the OBFH mixer 645. In the forward position of switch 640B, it connects VCO 620 to the input of the second OBPC mixer 650. In the “back” position, switch 640B connects the output of the OBPC mixer 650 to the main output of the OBPC mixer 650.

The VCO 620 is connected to the input of the divider 630. The divider 630 divides the output frequency of the VCO by an integer N. The divider 630 provides the first and second output signals. The first output of the divider 630 is connected to the first OBFC mixer 645. The second output of the divider 630 is connected to the second OBFC mixer 650. The signals in the first and second output signals of the divider are both divided variants of the input frequency, but with a phase difference of 90 °.

When the switch 640A is in the forward position, the first OBFC mixer 645 mixes the output frequency of the VCO with the first divided variant output by the divider 630. Similarly, the second OBPC mixer 650 mixes the output frequency of the VCO with the second divided variant output by the divider 630. Output signals the first and second OBPC mixers 645, 650 are the same in frequency, but with a phase difference of 90 °. The output signals of the first and second OBHCH mixers 645, 650 are the GD frequencies of the transmitter of the system 602.

The output of the second OBPC mixer 650 is connected to the second divider 670. The second divider 670 can divide the input frequency by an integer M. The second divider 670 gives the first and second output signals. The first and second output signals are quadrature. The output signals of the second divider 670 are the DG frequencies of the transmitter system 602.

By changing the N and M values, the OBFC mixers 645, 650, and the position of the switches 640A, 640B, the system 602 can generate a wide range of HF frequencies of the transmitter. Therefore, system 602 is useful for performing direct up-conversion transmitters such as transmitter 600. Table 3 provides examples of configurations related to transmitter operating bands. Further configurations may also be implemented in accordance with the concept of the present invention. As indicated above, the desired frequency band can be selected using the frequency band selection mechanism and the corresponding configuration can be selected using the configuration selection mechanism.

Table 3
DG control configurations for uplink multi-band direct conversion transmitter RF band
frequencies RF range
transmission frequencies (MHz) The value of the divisor N Mixer Switch position
la / value
second divider, M VCO frequency range in
broadcast (MHz) SES, USA 1850-1910 four PFS Forward 1480-1528 Cellular, USA 824-849 8 PFS forward and division by 2 1465-1509 Cellular, Japan 887-925 four PFS forward and division by 2 1419-1480 MSEC 1920-1980 four PFS Forward 1536-1584 SES, Korea 1750-1775 8 PFS Forward 1400-1424

In accordance with another implementation (not shown), the system 602 can generate a GD transmission frequency by mixing the GD reception frequency for the receiver with the GD fixed frequency shift. This method takes into account that the following modulation standards have a fixed frequency shift between transmit and receive channels, according to Table 4.

Table 4
Shift of the transmission frequency relative to the frequency of the reception channel Mode Transmission Shift (MHz) GSM
MSEC
Cellular, Japan
SES, Korea
SES, USA
Cellular, USA - 45
-190
+55
-90
-80
-45

In particular, the DG generation circuits in system 602 (PLL 610, synchronous detector lowpass filter 601, first and second OBPC mixers 645, 650, VCO 620 and switches 640A, 640B) can generate a DG reception frequency. The second generator, which is a fixed-shift DG, can be connected to the input of each mixer from the first and second OBPC mixers 645, 650. Accordingly, the first OBPC mixer 645 and the second OBPC mixer 650 can mix the reception GD frequency with a shift DG frequency to get the DG transmission frequency. But it should be noted that the receiving DG can generate false output signals. Off-chip filtering in the transmitter or transceiver may be required to fulfill the specifications regarding false leakage conducted for the receive band. This filtering can suppress the spurious component in the reception frequency.

Transmitter 600 may use the DG frequency generated by system 602 to transmit RF signals. The baseband processor 608 may be external to the transmitter 600 of FIG. 8, or form part of a transceiver comprising a transmitter 600. A baseband processor 608 provides a pair of output signals. Each output signal can be performed as a balanced or differential pair. Two output signals represent modulating analog signals I and Q for each mode and are provided as separate signal paths, and therefore quadrature modulation of the signals can be performed in subsequent stages of the transmitter 600.

In an exemplary embodiment, transmitter 600 comprises three RF outputs. Two outputs may correspond to the frequency bands of the MF or MSEC, and the others may correspond to the bands of Cellular. For RF output signals, the frequency response of the first RF mixer 651 is connected to the OBPC mixer 645 and to the first output of the frequency band of the modulating signals of the processor 608 of the frequency band of the modulating signals. The first RF mixer 651 upconverts the modulating signal directly to the desired RF frequency. A second RF mixer 653 is coupled to an OBFC mixer 650 and a second output of a baseband frequency band processor 608 of the baseband modulator signal. The second RF mixer 653 converts with increasing frequency the modulating signal directly into the same RF frequency as the output of the first RF mixer 651. The output signals of the first and second RF mixers 651, 653 are squared due to the relative phase difference signals used for upconversion of modulating signals.

The quadrature RF signals are then sent to a signal adder 660, which combines the two quadrature signals into a single signal. The input signals of the signal adder 660 can be balanced to match the balanced output signals from both the first and second RF mixers 651, 653. The output of the signal adder 660 can also be a balanced signal to minimize signal interference from common mode noise sources .

The output signal of the adder 620 signals can be simultaneously sent to two amplification circuits. Both amplifier circuits may be configured to operate in the frequency response band. According to FIG. 8, the first amplification circuit may comprise AGC amplifiers 662 and 664. The second amplification circuit may comprise AGC amplifiers 662 and 666.

For the RF output of the Cellular mode, a third RF mixer 652 is connected to the first output of the second divider 670 and to the first output of the baseband frequency of the baseband processor 608 of the baseband. The third RF mixer 652 upconverts the modulating signal directly to the desired RF frequency. The fourth RF mixer 654 is connected to the second output of the second divider 670 and to the second output of the baseband frequency of the baseband processor 608 of the baseband. The fourth RF mixer 654 upconverts the modulating signal directly to the same RF frequency as the output of the third RF mixer 652. The output signals of the third and fourth RF mixers 652 and 654 are quadrature due to the relative phase difference of the HD signals, used for upconverting baseband signals.

The quadrature RF signals are then sent to a signal adder 670, which combines the two quadrature signals into a single signal. The input signals of the signal adder 670 can be balanced to match the balanced output signals from both the third and fourth RF mixers 652, 654. The output of the adder can also be a balanced signal to minimize signal interference from common mode noise sources.

The output of the adder 670 signals can be connected to a third amplification circuit. A third amplifier circuit may be configured to operate in the Cellular mode transmission band. According to FIG. 8, the third amplification circuit may comprise AGC amplifiers 672 and 674.

Transmitter 600 may be configured such that only one amplifier circuit will act at any given time. Therefore, if the transmitter 600 is configured to transmit in a specific frequency band, then only one amplifier circuit can operate to provide this frequency band. An inactive amplifier circuit can be powered using control circuits (not shown) to save power. It should be noted that the three amplifier circuits shown in FIG. 8, and other similar amplification circuits may also include transmission filters, gates, or antenna switches according to prior art methods.

The foregoing detailed description relates to the accompanying drawings, which illustrate exemplary embodiments of the invention. Other embodiments are possible in which modifications can be made within the concept and scope of the invention. For example, many of the devices mentioned above can be directly connected to each other; as described, they are separated from each other by such intermediate devices as filters and amplifiers. In addition, the provisions of this invention can be applied to those modulation standards and operating bands that will be developed in the future. Therefore, it is understood that this detailed description does not limit the invention. The scope of the invention is defined by the attached claims.

Claims (50)

1. A method for generating a local oscillator (GD) frequency in a multi-band wireless communication device, the method comprises the steps of receiving a first signal from a voltage controlled oscillator (VCO) having a VCO frequency that is in the first frequency range, dividing the first signal by frequency programmable value N to obtain a second signal having a frequency divided by decreasing, wherein the programmable value N is changed based in part on the control signal, and the first signal is mixed with the second signal so that obtain an output signal having a frequency DG, which is in a second frequency range determined by a first frequency range and a programmable value N. 2. The method according to claim 1, in which the output signal is also divided by frequency by the value of M. 3. The method according to claim 1, in which the phase of the output signal is also shifted. 4. The method according to claim 1, in which the frequency of the first signal is divided by the number N when receiving a control signal. 5. The method according to claim 1, wherein said device comprises a receiver. 6. The method according to claim 1, wherein the output signal is also mixed with a third signal having a frequency shift representing a frequency shift between the transmitted and received signals of the wireless communication device to obtain a fourth signal for the transmitter. 7. The method according to claim 1, wherein said device comprises a transmitter. 8. The method according to claim 1, in which the frequency band of the RF signals is selected and the configuration of the DG generator related to the selected frequency band of the RF signals is selected, wherein the DG generator has one or more configurations, each configuration being interconnected with at least one frequency band of the RF signals and receive an output signal whose frequency is interconnected with at least one frequency band of the RF signals. 9. The method of claim 8, in which also perform the control of the generator of the DG, which is based on the choice of configuration. 10. A method for generating a local oscillator (GD) frequency in a multi-band wireless communication device, the method comprises the steps of: receiving a first signal from a voltage controlled oscillator (VCO) having a VCO frequency in the first frequency range, dividing the first signal by a value N, in order to obtain a second signal having a down-divided frequency, divide the second signal in frequency by a value of M to obtain a third signal having a further down-divided frequency and mix the second signal with a third signal scrap to obtain an output signal having a frequency of HD, which is in the second frequency range determined by the first frequency range. 11. The method according to claim 10, in which the output signal is also divided by frequency by the value of R. 12. The method of claim 10, wherein the VCO is a multi-range VCO. 13. A system for generating a local oscillator (GD) frequency in a multi-band wireless radio communication device, comprising a voltage controlled oscillator (VCO) operating in a first frequency range, a divider having an input and an output, the signal of which is obtained by dividing the input signal by a programmable value N, which changes based in part on the control signal, while the input of the divider is connected to the VCO, and the mixer having the first input of the mixer connected to the VCO, the second input of the mixer connected to the output of the divider, and the output Providing DG frequency in the second frequency range determined by the first frequency range. 14. The system according to item 13, in which the said VCO is made outside the microcircuit containing the system. 15. The system of claim 14, wherein the VCO has an unbalanced output. 16. The system according to item 13, in which the VCO is integrated into the microcircuit containing the system. 17. The system of claim 13, wherein the frequency at which the VCO operates is lower than the frequency of the RF signals. 18. The system of claim 13, wherein the frequency at which the VCO operates is higher than the frequency of the RF signals. 19. The system according to item 13, in which the operating frequency of the VCO is in the interval between 1600 and 1788 MHz. 20. The system according to item 13, in which the VCO is connected to a phase-locked loop (PLL), and also contains a second PLL and a second VCO for signals received when operating in the Global Satellite Radio Determination System (GSSRO), while the second VCO operates at a 2-fold frequency of received signals GSSRO. 21. The system according to claim 20, which also contains a third PLL and a third VCO for signals received when operating in Bluetooth mode. 22. The system according to item 13, in which the mixer is a mixer of one side frequency band (OBPC). 23. The system according to item 22, in which the OBPC mixer is an OBPC mixer of the upper side strip. 24. The system of claim 22, wherein the OBPC mixer is a top sideband OBPC mixer. 25. The system of item 13, in which the output of the mixer is connected to the PLL, while the PLL is internal to the microcircuit containing the system. 26. The system according to item 13, in which the input of the divider is selectively connected to the VCO. 27. The system of claim 26, wherein the switch selectively connects the divider input to the VCO. 28. The system of claim 27, wherein the switch is controlled by a switching regulator based on an RF signal band. 29. The system according to item 13, in which the input of the divider is selectively connected to the output of the mixer, using a switch controlled by a switching controller, based on the band of RF signals received by a multi-band wireless communication device. 30. The system according to item 13, in which the output of the mixer is selectively connected to the VCO. 31. The system according to item 13, which also contains a phase shifter, the input of which is connected to the output of the mixer, while the phase shifter has an output that generates quadrature signals. 32. The system of claim 31, wherein the phase shifter comprises an active phase shifter. 33. The system according to item 13, which also contains a second divider having an input connected to the output of the mixer, and an output whose output signal is obtained by dividing the input signal. 34. The system of claim 33, wherein the second divider divides by 2. 35. The system of claim 33, wherein the second divider outputs a first signal and a second signal; wherein the first signal has a phase shift of 90 ° with respect to the second signal. 36. The system of claim 35, further comprising a common-mode I-mixer and a quadrature Q-mixer, wherein the first signal drives one of the I- and Q-mixers in the device. 37. The system of claim 13, wherein the radio communications device comprises a receiver and the received RF signal band is a US Private Mode (RF) bandwidth and in which the VCO operating frequency range is between 1716 MHz and 1769 MHz, the divider divides by 8, and the mixer is an OBPC mixer of the upper side strip. 38. The system of claim 13, wherein the radio communication device comprises a receiver, and the band of received RF signals is the band of the International Telecommunication Union (MSEC), while the operating frequency range of the VCO is between 1688 MHz and 1736 MHz, the divider divides 4, and the mixer is an OBPC mixer of the upper side strip. 39. The system of claim 13, wherein said communication device is part of a wireless communication transceiver. 40. The system of claim 13, wherein said device comprises a transmitter. 41. The system of claim 40, in which the band of transmitted RF signals is the U.S. frequency band, and in which the operating frequency of the VCO is in the range between 1480 MHz and 1528 MHz, the divider divides by 4, and the mixer is an upper-frequency mixer side strip. 42. The system of claim 40, which also comprises an upconverter and an amplifier circuit configured to operate in a transmission frequency band, wherein the amplifier circuit is connected to an upconverter. 43. The system of claim 13, wherein said communication device comprises a receiver, and which also comprises a frequency shift HD representing a frequency shift between the transmitted and received signals of the wireless communication device connected to a third input of the mixer, wherein the output of the mixer provides a frequency of HD for the transmitter. 44. The system according to item 13, in which the first input of the mixer and the output of the mixer are differential. 45. The system of claim 13, wherein said radio communication device includes a receiver that comprises differential signal paths. 46. A system for generating a local oscillator frequency (LO) in a multi-band wireless communication device, comprising a voltage controlled oscillator (VCO) operating in a first frequency range, a first divider having an input signal and an output signal that is obtained by dividing the input signal of the first divider, wherein the input of the first divider is connected to the VCO, the second divider having an input signal and an output signal that is obtained by dividing the input signal of the second divider, while the input of the second divider is connected to the output of Vågå divider and a mixer, the mixer having a first input connected to the output of the first divider, a second input connected to the output of the second divider, and an output providing DG frequency which is in a second frequency range determined by the first frequency range. 47. The system according to item 46, which also contains a third divider connected to the output of the mixer. 48. The system of claim 46, wherein the VCO is a multi-range VCO. 49. The system of claim 13, comprising a DG generator having one or more configurations, each of which configurations relates to at least one frequency band of the RF signals, and generating an output signal whose frequency relates to at least one frequency band The RF signal, the DG generator contains a mixer and the first and second dividers, and a configuration selection mechanism configured to select a configuration related to the selected frequency band of the RF signals. 50. The system of claim 49, wherein the configuration generator controls the generator.

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