RU2296432C1 - Method for autocorrelation receipt of noise-like signals - Google Patents

Abstract

FIELD: radio engineering, possible use in digital communication systems, in particular, devices for synchronization and receipt of noise-like (phase-manipulated) signals and range-finding their emission source.

SUBSTANCE: device for realization of aforementioned method contains device for measuring signal range, frequency detector, impulse counter, arithmetic blocks, scaling multiplexers, delay lines, multiplexers, band filters, saw-shaped voltage generator, low frequencies filters, threshold block, key, registration block, receiving antenna, narrowband filters, phase shifter for 90 degrees, phase detectors, measuring devices, extreme controllers, adjustable delay blocks, correlators.

EFFECT: expanded functional capabilities of method due to range-finding of source of noise-like signals in two planes.

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Description

The proposed method relates to radio engineering and can be used in digital communication systems, in particular, in devices for synchronizing and receiving noise-like phase-shifted (PSK) signals and direction finding of a radiation source in two planes.

Known methods and devices for receiving noise-like signals (ed. Certificate) of the USSR No. 177771, 451166, 491187, 543194, 860276, 1417206; U.S. Patent Nos. 4,146,641, 4,811,363, 4,912,422; Germany patents No. 2646255, 3935911; Petrovich N.P. and other communication systems with noise-like signals. - M .: Owls. radio, 1969, p. 94, fig. 8.a; J. Spilker. Digital satellite communications. - M .: Communication, 1979, p. 281; Varakin L.E. communication systems with noise-like signals. - M .: Communication, 1985, p. 18, fig. 1.9, c and others).

, кратный тактовому периоду τ_э, выделяют суммарное напряжение, перемножают его с принимаемым сигналом, задержанным на время

, кратное тактовому периоду τ_э, выделяют напряжение тактовой частоты, перемножают его с принимаемым сигналом, задержанным на время τ, которое периодически изменяют по линейному закону, выделяют низкочастотное напряжение, пропорциональное автокорреляционной функции, сравнивают его с пороговым уровнем, при превышении порогового уровня измеряют циклический сдвиг, по которому определяют кодовую структуру принимаемого сигнала.Of the known methods, the closest to the intended method is the Autocorrelation method of "noise-like signals" (RF patent No. 2.248.102, H 04 L 27/22, 2003), which is selected as a prototype. This method provides the reception of noise-like signals with an a priori unknown code structure and consists in multiplying the received signal with a reference signal, measuring the duration of the received signal, thereby highlighting the moments of phase jump, determining the number and magnitude of clock pulses, while the reference signal is formed by delaying the received signal for a while

a multiple of the clock period τ _e , isolate the total voltage, multiply it with the received signal, delayed for a while

a multiple of the clock period τ _e , the clock frequency voltage is isolated, it is multiplied with the received signal delayed by the time τ, which is periodically changed according to the linear law, a low-frequency voltage is proportional to the autocorrelation function, it is compared with the threshold level, and when the threshold level is exceeded, the cyclic the shift by which the code structure of the received signal is determined.

However, this method does not fully realize its potential capabilities. Using correlation processing of received noise-like signals, it is possible to detect the source of their radiation.

An object of the invention is to expand the functionality of the method by direction finding a radiation source of noise-like signals in two planes.

, кратное тактовому периоду τ_э, выделяют суммарное напряжение, перемножают его с принимаемым сигналом, задержанным на время

, кратное тактовому периоду τ_э, выделяют напряжение тактовой частоты, перемножают его с принимаемым сигналом, задержанным на время τ, которое периодически изменяют по линейному закону, выделяют низкочастотное напряжение, пропорциональное автокорреляционной функции, сравнивают его с пороговым уровнем, при превышении порогового уровня измеряют циклический сдвиг, по которому определяют кодовую структуру принимаемого сигнала, шумоподобные сигналы принимают на приемные антенны, разнесенные на фиксированные расстояния d₁, d₂ и расположенные в виде геометрического прямого угла, в вершине которого помещают антенну опорного канала, общую для антенн двух пеленгационных каналов, расположенные в азимутальной и угломестной плоскостях, в каждом канале принимаемый шумоподобный сигнал перемножают самого на себя, выделяют гармоническое колебание, сдвигают по фазе на 90 градусов гармоническое колебание опорного канала, измеряют разности фаз между ним и гармоническими колебаниями пеленгационных каналов, формируя при этом фазовые шкалы отсчета азимута α и угла места β, источника излучения шумоподобных сигналов, точные но неоднозначные, перемножают шумоподобный сигнал опорного канала с задержанными по времени шумоподобными сигналами пеленгационных каналов, выделяют низкочастотные напряжения, пропорциональные взаимно корреляционным функциям, изменяют время задержки до получения максимального значения взаимно корреляционной функции, поддерживают эти значения, фиксируют временные задержки τ₁, τ₂, соответствующие максимальному значению взаимно корреляционных функций, и определяют азимут α и угол места β, источника излучения шумоподобных сигналовThe problem is solved in that according to the method of autocorrelation receiving noise-like signals, which consists in multiplying the received signal with a reference signal, measuring the duration of the received signal, performing frequency detection of the received signal, thereby highlighting the moments of the phase jump, determining the number and magnitude of the clock periods, while the reference signal is formed by delaying the received signal for a while

a multiple of the clock period τ _e , isolate the total voltage, multiply it with the received signal, delayed for a while

a multiple of the clock period τ _e , the clock frequency voltage is isolated, it is multiplied with the received signal delayed by the time τ, which is periodically changed according to the linear law, a low-frequency voltage is proportional to the autocorrelation function, it is compared with the threshold level, and when the threshold level is exceeded, the cyclic the shift by which the code structure of the received signal is determined, noise-like signals are received at the receiving antennas spaced at fixed distances d ₁ , d ₂ and is located In the form of a geometric right angle, at the apex of which a reference channel antenna is placed, common to the antennas of two direction-finding channels, located in the azimuth and elevation planes, in each channel the received noise-like signal is multiplied by itself, emit harmonic oscillation, phase shift by 90 degrees harmonic oscillation of the reference channel, measure the phase difference between it and the harmonic oscillations of direction finding channels, forming the phase scales of the azimuth α and elevation angle β, the source from radiation of noise-like signals, accurate but ambiguous, multiply the noise-like signal of the reference channel with time-delayed noise-like signals of direction finding channels, emit low-frequency voltages proportional to the cross-correlation functions, change the delay time to obtain the maximum value of the cross-correlation function, maintain these values, fix the time delays τ _1, τ _2, corresponding to the maximum value of cross-correlation functions, and determining the azimuth angle α and β place ISTO nick radiation of noise signals

,

,

,

,

where c is the speed of light propagation; thereby forming a time frame for reading the angular coordinates α and β, rough, but unambiguous.

The structural diagram of a device that implements the proposed method is presented in figure 1. timing diagrams explaining the essence of the proposed method are depicted in figure 2. the relative position of the receiving antennas is shown in Fig. 3. direction finding characteristic is shown in figure 4.

The device comprises a frequency lock 2, a pulse counter 3, a first arithmetic unit 4, connected in series to the output of the antenna 21.1, the second input of which is connected to the output of the antenna 21.1 through the signal duration meter 1, the first scaling multiplier 5. the first delay line 7, the second input of which is connected to the output of the antenna 21.1, the first multiplier 8, the second input of which is connected to the output of the antenna 21.1, the first bandpass filter 9, the second multiplier 11, the second input of which is connected through the delay line 10 to the output of the antenna 2 1.1 and the second scaling multiplier 6, the second band-pass filter 12, the third multiplier 15, the second input of which through the third delay line 14 is connected to the output of the antenna 21.1, the first low-pass filter 16, the threshold block 17, the key 18, the second input of which is connected to the output of the line 14 delays, the second arithmetic unit 19, the second input of which is connected to the output of the first arithmetic unit 4 and the registration unit 20, the second and third of which are connected to the outputs of the meter 1 of the signal duration and the arithmetic unit 4, the second input of the delay line ki 14 via a sawtooth voltage generator 13 is connected to the output of the threshold unit 17.

The device also contains one reference channel and two direction finding channels.

The reference channel contains a series-connected antenna 21.1, a multiplier 22.1, the second input of which is connected to the output of the antenna 21.1, a narrow-band filter 23.1 and a phase shifter 24 by 90 degrees.

The first (second) direction-finding channel contains series-connected antennas 21.2 (21.3), a multiplier 22.2 (22.3), the second input of which is connected to the output of the antenna 21.2 (21.3), a narrow-band filter 23.2 (23.3) and a phase lock 25.1 (25.2), the second input of which connected to the output of the phase shifter 24 by 90 degrees, and the output is connected to the fourth (fifth) input of the registration unit.

An adjustable delay unit 29.1 (29.2), a multiplier 22.4 (22.5), the second input of which is connected to the output of the antenna 21.1, a low-pass filter 26.1 (26.2) and an extreme controller 28.1 (28.2), the output of which is connected, are connected in series to the output of the antenna 21.2 (21.3) to the second input side 29.1 (29.2) adjustable delay. A filter 27.2 (27.2) is connected to the output of the filter 26.1 (26.2). the second input of the adjustable delay unit 29.1 (29.1) is connected to the sixth (seventh) input of the registration unit 20. These blocks form a correlator 30.1 (30.2).

The proposed method is implemented as follows. Suppose that a pseudo-random sequence (PSP) is used as the modulating function, the symbols of which are described by the recurrence relation

X ₁ = A ₁ X _I-1 ⊕A ₁ X _I-2 ⊕ ... ⊕A _m X _im ,

where a _i = {0, 1} are the coefficients of the correlating polynomial,

A (X) = X ⁰ ⊕a ₁ X ¹ ⊕a ₂ X ² ⊕ ... ⊕a _m X ^m ,

⊕ is the addition sign modulo two,

m is the bit depth of the pseudo-random sequence, the period of which is determined by the formula N = 2 ^m -1.

To transmit such a sequence M (t) through the communication channels (Fig.2.b), the high-frequency harmonic oscillation is manipulated in phase (Fig.2.a).

U _c (t) = V _c cos (ω _c t + φ _c ), 0≤t≤T _c ,

where V _c , ω _c , φ _c , T _c - amplitude, carrier frequency, initial phase and duration of high-frequency oscillations; as a result, a phase-shift (PSK) signal is formed (noise-like signal) (Fig.2.c).

, 0≤t≤Т_с,

, 0≤t≤T _s ,

where φ _k (t) = {0, π} is the manipulated component of the phase, which displays the law of phase manipulation in accordance with the modulating code M (t) (PSP) (Fig.2.b), and φ _k (t) = const for kτ _e <t <(k + 1) and can change abruptly at t = kτ _e , i.e. at the boundaries between elementary premises (k = 1, 2, ..., N-1);

τ _e , N is the duration and number of chips that make up a signal of duration T _s (T _s = Nτ _e ).

At the receiving location, FMN signals (noise-like signals):

U ₁ (t) = V _with cos [ω _with t + φ _k t + φ ₁ ],

U ₂ (t) = V _with cos [ω _s (t-τ ₁ ) + φ _k (t-τ ₁ ) + φ ₂ ],

U ₃ (t) = V _s cos [ω _s (t-τ ₂ ) + φ _k (t-τ ₂ ) + φ ₃ ], 0≤t≤T _s

where φ ₁ , φ ₂ , φ ₃ are the initial phases of the signals

- время запаздывания сигнала, приходящего на антенну 21.2 по отношению к сигналу, приходящему на антенну 21.1;

- the delay time of the signal arriving at the antenna 21.2 with respect to the signal arriving at the antenna 21.1;

- время запаздывания сигнала, приходящего на антенну 21.3 по отношению к сигналу, приходящему на антенну 21.1 (фиг.3);

- the delay time of the signal arriving at the antenna 21.3 with respect to the signal arriving at the antenna 21.1 (Fig. 3);

d ₁ , d ₂ - measuring base;

α, β — angles of arrival of radio waves in the azimuthal and elevation planes;

C is the speed of light propagation.

From the outputs of the antennas 21.1, 21.2 and 21.3, respectively, the signals are fed to the input of the multipliers 22.1, 22.2 and 22.3, at the output of which harmonic oscillations are formed:

U ₄ (t) = V ₄ cos [2ω _with t + 2φ ₁ ],

U ₅ (t) = V ₄ cos [2ω _s (t-τ ₁ ) + 2φ ₂ ],

U ₆ (t) = V ₄ cos [2ω _s (t-τ ₂ ) + 2φ ₃ ], 0≤t≤Ts,

;Where

;

K ₁ - transmission coefficient of the multipliers;

It should be noted that the width of the spectrum Δf _{from the} received FMN signals U ₁ (t), U ₂ (t), U ₃ (t) is determined by the duration of their elementary parcels τ _e (clock cycle)

,

,

while the width of the spectrum of the second harmonics U ₄ (t), U ₅ (t), U ₆ (t) is determined by the duration T _{s of the} signal

.

.

Therefore, when multiplying the FMN signals themselves, the vase manipulation is eliminated and their spectrum is “collapsed” N times

This circumstance makes it possible to isolate harmonic oscillations U ₄ (t), U ₅ (t), U ₆ (t) using narrow-band filters 23.1, 23.2, and 23.3, respectively, filtering out a significant part of noise and interference.

If harmonic oscillations U ₄ (t), U ₅ (t), U ₆ (t) from the outputs of narrow-band filters 23.1 and 23.2, 23.1 and 23.3 are directly fed to phase detectors 25.1 and 25.2, we obtain at the output of the latter:

Where

To ₂ - the transfer coefficient of the phase descriptors.

From the above relations it is seen that the voltage at the output of the phase descriptors 25.1 and 25.2 depends on the angles α and β. However, due to the fact that the cosine is an even function, the signs of U _o (α) and U _o (β) are independent of the side of the deviation. To eliminate this drawback, a phase shifter 24 of 90 degrees is included in the reference channel. In this case, the mismatch voltage at the output of the phase detectors 25.1 and 25.2 are determined by the expressions:

The above dependencies are usually called direction-finding characteristics (figure 4)

The steepness of the characteristics in the region of small angles α and β, where the characteristics are almost linear, is equal to:

и

Увеличение измеренных баз d₁, d₂ и уменьшение длины волны λ повышают крутизну K_α, K_β и увеличивают точность пеленгации источника излучения ФМН-сигналов.Thus, the steepness of the characteristics is determined by the ratio

and

An increase in the measured bases d ₁ , d ₂ and a decrease in the wavelength λ increase the steepness K _α , K _β and increase the accuracy of direction finding of the radiation source of the FMN signals.

However, the ambiguity of the reading of the angles α and β increases.

The steepness of the characteristic determines the dead zones, 2α _min and 2β _min when setting significant noise V _W (figure 4).

The number of ambiguity zones, i.e. areas where the phase differences:

change by an amount equal to 2π, are determined by the relations:

For a unique reference, it is necessary to choose n ₁ = 1 and n ₂ = 1, i.e. choose measuring bases based on the following conditions:

The phase differences Δφ ₁ and Δφ ₂ are fixed by the registration unit 20.

This is how the phase scales of reading the angular coordinates α and β are formed: accurate, but ambiguous.

Received FMN signals U ₁ (t) and U ₂ (t), U ₁ (t) and U ₃ (t) simultaneously come from the outputs of antennas 21.1 and 21.2, 21.1 and 21.3 to two inputs of the correlator 30.1 (30.2), consisting of block 29.1 (29.2) adjustable delay multiplier 22.4 (22.5) and a filter 26.1 (26.2) low pass. The cross-correlation functions R ₁ (τ) and R ₂ (τ) obtained at the output of correlators 30.1 (30.2), measured by measuring instruments 27.1 and 27.2, have a maximum for the value of the introduced controlled delay:

τ ₁ = t ₂ -t ₁ , τ ₂ = t ₃ -t ₁ ,

where t ₁ , t ₂ , t ₃ is the signal travel time for the distances R ₁ , R ₂ , R ₃ from the radiation source to the first 21.1, second 21.2 and third 21.3 receiving antennas:

ΔR ₁ = R ₂ -R ₁ , ΔR ₂ = R ₃ -R ₁ .

The maximum values of R ₁ (τ) and R ₂ (τ) are supported by the extreme controllers 28.1 and 28.2 acting on the second inputs of the adjustable delay units 29.1 and 29.2. The scales of the adjustable delay blocks 29.1 and 29.2 (angle indicators) are graded directly in the values of the angular coordinates α and β of the radiation source of the FMN signals:

where τ ₁ , τ ₂ are the introduced signal delays corresponding to the maximum of possibly correlation functions R ₁ (τ) and R ₂ (τ).

The values of the angular coordinates α and β are fixed by the registration unit 20. This is how the time scales for counting the angular coordinates α and β are formed: rough, but unambiguous.

Essentially, full phase differences are measured with measuring scales:

Δφ ₁ = m + Δφ ₁ , Δφ ₂ = n + Δφ ₂ ,

where m, n are the number of complete cycles of the measured phase differences, determined by time scales;

Δφ ₁ and Δφ ₂ are the phase differences measured by phase scales (0≤Δφ ₁ ≤2π, 0≤Δφ ₂ ≤2π).

It should be noted that the location of the receiving antennas 21.1, 21.2 and 21.3 in the form of a geometrically right angle, at the apex of which the first receiving antenna 21.1 of the reference channel is located, is dictated by the very ideology of direction finding of the radiation source of FMN signals in space.

The received FMN signal U ₁ (t) from the output of the receiving antenna 21.2 simultaneously receives sleep inputs of the meter 1 of the signal duration, frequency detector 2, multiplier 8, delay lines 7, 10, 14.

The output of the frequency detector 2 is formed of short bipolar pulses (Fig.2.g), the temporary position of which corresponds to the moments of the abrupt phase change of the received FMN signal U ₁ (t) (Fig.2.c).

These pulses are fed to the input of a 3-pulse counter, where the number ν of phase jumps is calculated. The following relationship exists between the number of phase jumps ν and the number N of chips:

ν = 0.5 (N-1).

The number of phase jumps ν calculated by the counter 3, arriving at the first input of the arithmetic unit 4, the second input of which is measured by the meter 1, the duration T of _the signal. In the arithmetic unit 4 of the signal is determined by the duration of the elementary parcels τ _e (clock period).

At the same time, the received FMN signal U ₁ (t) (Fig.2.c) is supplied to the first input of the multiplier. The value of τ _e through scaling multipliers 5 and 6 are supplied to the control inputs of the delay line 7 and 10, respectively, where the delays are set:

t _z1 = K ₁ τ _e , t _z2 = K ₂ τ _e ,

multiples of the clock period τ _e The second input of multiplier 8 is supplied the received PSK-signal delayed by an amount t _P1 (fig.2.d)

U ₇ (t) = U ₁ (t-τ _z1 ) = V _s cos [ω _s (t-τ _z1 ) + φ _k (t-τ _z1 ) + φ _s ], 0≤t≤T _s .

The output of the multiplier 8 forms the following oscillation:

U ₈ (t) = V ₈ cos [2ω _with t-ω _with τ _z1 + φ _k (t) -φ _k (t-τ _z1 ) + 2φ _s ] + V ₈ cos [ω _with τ _z1 + φ _k ( t) -φ _k (t-τ _z1 )], 0≤t≤T _s ,

Where

from which the band-pass filter 9, tuned to 2ω _s , the total voltage is allocated (Fig.2.e).

U _Σ (t) = V ₈ Cos [2ω _with t-ω _with τ _z1 + φ _k (t) -φ _k (t-τ _z1 ) + 2φ _s ], 0≤t≤T,

which is fed to the first input of the multiplier 11, to the second water of which the received FMN signal is applied, delayed by a value of τ _{s2 by} the delay line 10 (Fig.2.zh).

U ₉ (t) = U ₁ (t-τ _z2 ) = V _s cos [ω _s (t-τ _z2 ) + φ _k (t-τ _z2 ) + φ _s ], 0≤t≤T _s .

The output of the multiplier 11 forms the following oscillation:

U ₁₀ (t) = V ₁₀ cos [3ω _with t-ω _s (τ _z1 + τ _z2 ) + φ _k (t) + φ _k (t-τ _z1 ) + φ _k (t-τ _z1 ) + φ _k (t-τ _s2) + 3φ _c] + V ₁₀ cos [ω _c t-ω _c (τ _{s2 P1} -τ) + 3φ _c] + V ₁₀ cos [ω _c t + ω _c (τ _s2 -τ _P1) + φ _k (t) + φ _k (t-τ _z1 )] - φ _k (t-τ _z1 ) -φ _to (t-τ _z2 ),

0≤t≤T _s .

Where

from which the band-pass filter 12 tuned to ω _s , the voltage of the frequency difference is allocated (Fig.2.z)

U _p (t) = V ₁₀ cos [ω _c t + ω _c (τ _{s2 P1 -τ) + φ k (t) +} φ k (t-τ _P1) -φ _k (t-τ _s2)], 0 ≤t≤T _s ,

the manipulated phase of which has the form:

φ _cr (t) = φ _k (t) + φ _k (t-τ _z1 ) -φ _k (t-τ _z2 ) = φ _k (t-θτ _e ),

where θ is the cyclic shift expressed by the number of clock periods (elementary premises).

The voltage U _p (t) from the output of the band-pass filter 12 is supplied to the first input of the multiplier 15, to the second input of which the received PSK signal is delayed by a value of τ using the delay line 14, which is periodically tuned according to a linear law using a sawtooth voltage generator 13

U ₁₁ (t) = U ₁ (t-τ) = V _s cos [ω _s (t-τ) + φ _k (t-τ) + φ _s ], 0≤t≤T _s ,

where τ is a variable value of the delay value of the delay line 14.

The output of the multiplier 15 produces the following voltage:

₁₂ U (t) = V ₁₂ cos [2ω _c t + ω _c (τ _s2 -τ _P1 + τ) + φ _a (t) + φ _a (t-θτ _e) + 2φ _c] + V ₁₂ cos [ω _with (τ _s2 -τ _P1 + τ) + φ _a (t-θτ _e) -φ _to (t-τ)], 0≤t≤T _p.

Where

low-pass filter 12 emits a low-frequency voltage proportional to the autocorrelation function,

_H U (t) = V ₁₂ cos [2ω _c t + ω _c (τ _s2 -τ _P1 + τ) + φ _a (t-θτ _e) -φ _to (t-τ)],

which is compared with the threshold level in the threshold block 17. The threshold voltage V _{then is} exceeded only at the maximum voltage value U _N (t), which is obtained when the following condition is met:

τ ₀ = θτ _e, cos [w _c (τ _s2 + θτ _P1 -τ _e)] = 2πk, k = 1, 2, 3, ...

if the threshold level V _pores is exceeded, a constant voltage is generated in the threshold block 17, which is supplied to the control input of the sawtooth voltage generator 13, stopping its adjustment, and to the control input of the switch 18, opening it. In the initial state, the key 18 is always closed.

In this case, the value of the delay value τ ₀ = θτ _e corresponding to the maximum of the autocorrelation function R (τ), through the public key 18 enters the arithmetic unit 19, where the value of the duration τ _{e of} elementary messages from the output of the arithmetic unit 4 is also received. cyclic shift

,

,

which is fixed by the registration unit 20, where the measured values of the duration τ _{e of} elementary packets and the duration T _{from the} received FMN signal are also recorded. The specified shift establishes a unique correspondence between the code structure of the received FMN signal and the conversion function, which is specified by the parameters τ _z1 and τ _z2 :

θ↔θ [A (x), B (x)],

where A (x) is the generating polynomial that determines the code structure of the received FMN signal;

и

а коэффициент В₀=1.B (x) = B ₀ X ⁰ + B ₁ x ¹ + ... + B _n x ⁿ is a transformation function whose numbers of zero coefficients are defined as

and

and coefficient B ₀ = 1.

For example, for τ = 2τ _{P1 e} and τ _s2 = 3τ _e (θ = 8)

A (x) = x ⁰ ⊕x ² ⊕x ⁵ ;

B (x) = x ⁰ ⊕x ² ⊕x ³ .

By measuring the cyclic shift θ, the code structure (the law of phase manipulation) of the received FMN signal can be determined from the correspondence table. This provides the ability to receive noise-like signals with an a priori unknown code structure.

Thus, the proposed method in comparison with the prototype provides not only the reception of signals with a priori unknown code structure, but also accurate and unambiguous direction finding of their radiation source in two planes. Thus, the functionality of the method is expanded.

Claims (1)

The method of autocorrelation receiving noise-like signals, which consists in multiplying the received signal with a reference signal, measuring the duration of the received signal, performing frequency detection of the received signal, thereby highlighting the moments of phase jump, determining the number and magnitude of the clock periods, while the reference signal is formed by delaying the received signal _Z1 at time τ = K ₁ τ _e multiple of the clock period τ _e, allocate the total voltage, it is multiplied with the received signal m, delayed by time τ _s2 = K ₂ τ _e multiple of the clock period τ _e is isolated voltage difference frequency, multiplies it with the received signal delayed by time τ, which periodically varies in a linear fashion, separated low frequency voltage proportional to the autocorrelation function, compare it with a threshold level, when the threshold level is exceeded, a cyclic shift is measured, according to which the code structure of the determined signal is determined, characterized in that noise-like signals are received at receiving antennas Spaced a fixed distance d ₁ and d ₂ and arranged in a geometric right angle, the vertex of which is placed a reference channel antenna common to the antennas of the two direction-finding channels disposed in the azimuth and elevation planes, in each channel the received noise-like signal is multiplied with itself , emit harmonic oscillation, phase shift the harmonic oscillation of the reference channel by 90 degrees, measure the phase differences between it and harmonic oscillations of direction finding channels, thereby forming Accurate, but ambiguous, phase scales of the azimuth α and elevation angle β of the radiation source of the noise-generating signal multiply the noise-like signal of the reference channel with time-delayed noise-like signals of direction-finding channels, select low-frequency voltages proportional to the cross-correlation functions, change the delay time to obtain the maximum value of mutually correlation functions, support these values, fix the time delays τ ₁ and τ ₂ corresponding to the maximum value of relational functions, and determine the azimuth α and elevation angle β of the radiation source of noise-like signals

where c is the speed of light propagation, thereby forming a time frame for reading the angular coordinates α and β, rough, but unambiguous.

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