RU2270461C2 - Method for detecting targets by pulse radio-location station and radio-location station for realization of said method - Google Patents

Abstract

FIELD: radio-location equipment, possible use for engineering of radio-location stations, meant for navigation and detection of targets.

SUBSTANCE: in detection method, modulated probing signals are generator and emitted, reflected signals are received and process, length of proving pulses and digit capacity of phase manipulation code is selected in accordance to certain conditions, with closing in of target parameters of probing pulses are changed, let-through bands of receipt device, complex circling line of binary-quantized signals after phase detection and joined compressed signal. Radio-location station has synchronization device 22, frequency readjustment block 8, exciting device 5, phase manipulator 4, power amplifier 3, antenna switch 2, antenna 1, high frequency amplifier 9, mixer 10, intermediate frequency amplifier 11, phase detectors block 12, video amplifiers block 13, amplifier quantizing devices block 14, adjustable discontinuous filters 18₁, 18₂, block for joining quadratures 19, antenna drives block 23, device for calculating Doppler frequency 17, code-frequency transformer 16, compensator of Doppler frequency 15, block for threshold generation 21.

EFFECT: improved concealment of probing emission, improved resistance to interference and ecological cleanness of radio-location station.

2 cl, 11 dwg

Description

The invention relates to radar technology and can be used in radar stations (radars) intended for navigation and target detection (in particular surface targets), including followed by tracking the selected target in range and angular coordinates, mainly in radars operating in conditions of electronic countermeasures and installed on mobile carriers, as well as to improve the environmental cleanliness and electromagnetic compatibility of the radar with other radio equipment.

Currently, for radionavigation and detection of surface targets, pulsed radars are used, for example, of the “Furuno” type [1, 2], which use short pulsed signals with high duty cycle and relatively high pulsed power necessary for detecting echo signals from surface targets at the required range. The disadvantage of such radars is the use of powerful probe pulses with an incoherent principle of construction and operation at a constant carrier frequency, which is the reason for the low latency of the probe radiation, and therefore, low noise immunity, primarily in relation to response and impact interference, as well as low environmental cleanliness and electromagnetic compatibility.

A known method for target detection, implemented in a pulsed radar [3] (US patent No. 4338604, IPC G 01 S 13/24, publication 1982) and based on the use of complex coherent signals with low duty cycle and relatively low pulse power with intrapulse modulation and tuning the carrier frequency from pulse to pulse. At the same time, the required detection range and range resolution are provided and an increase in stealth and noise immunity, as well as environmental cleanliness and electromagnetic compatibility, is achieved.

The main disadvantage of the detection method and radar [3] is the relatively high level of residues (side lobes of the autocorrelation function) during compression of complex signals and, as a result, insufficient dynamic range and low probability of detecting small objects masked by echo residues from large objects. Another disadvantage is the non-optimal choice of parameters of the probe pulses in the absence of their regulation, as well as taking into account the own speed of the radar carrier.

Closest to the proposed invention is a method for detecting targets implemented in the radar [4] (RF patent No. 2039365, G 01 S 13/52 with priority from 09/27/93, publ. 09/07/95) and adopted as a prototype of the invention. The prototype method includes the generation of highly frequency-stable microwave oscillations, an abrupt change in their carrier frequency f _c from period to period of pulse repetition according to an arbitrary law, intra-pulse phase manipulation with a binary multi-bit code, reconstruction of the phase-shift code from period to period of pulse repetition, pulse amplitude modulation , radiation of probe pulses, reception of reflected signals, superheterodyne conversion of received signals using heterodyne microwave oscillations with high-stable frequency, tunable synchronously with the carrier frequency of the probe pulses, amplification of received signals at an intermediate frequency, quadrature phase detection of signals, amplification of quadrature components of signals by video frequency, binary quantization of quadrature components by zero level, digital matched filtering - time compression of quadrature components, combining compressed quadrature signals, binary quantization of combined signals with a threshold level selected from the condition allowable probability of exceeding its noise emission mezhperiodnoe accumulation quantized signal during the burst detection and burst pulse echo signal by comparing the accumulation result with a threshold level.

The radar [4], adopted as a prototype of the proposed device, contains a serially connected synchronizer, a frequency tuner, an exciter, a phase manipulator, a power amplifier, an antenna switch and an antenna, a serially connected high-frequency amplifier, a mixer, an intermediate frequency amplifier, a serially connected phase unit detectors, a block of video amplifiers and a block of amplitude quantizers, as well as two identical tunable discrete filters, the outputs of which are connected, respectively oh, with the first and second inputs of the quadrature combining unit, the output of which is connected to the first input of the information processing device, and the antenna drive unit, the output of which is kinematically connected to the antenna, the input of the high-frequency amplifier connected to the third arm of the antenna switch, the mixer heterodyne input, and the voltage reference of the reference frequency of the phase detector unit is connected to the corresponding outputs of the exciter, the first output of the anticipatory pulses of the synchronizer is also connected to the first input of the generator code, the second input of which, combined with the inputs for controlling the recording of tunable discrete filter codes, is connected to the output of bursts of clock pulses of the synchronizer, and the output is connected to the inputs of the phase manipulator code and tunable discrete filters, the synchronizer output of the synchronizer is connected via a pulse modulator to a power amplifier, and also - with the corresponding input of the information processing device, the output of the clock pulses of the synchronizer is connected to the clock inputs of the tunable discrete filters and information processing devices, and the information input of the latter is connected to the output of the azimuth codes of the antenna drive unit.

In the known invention provides a complete suppression of residuals after compression due to a combination of phase-shift code-shift (FM) tuning from pulse to pulse with digitally matched filtering of FM signals and subsequent inter-period accumulation of binary-quantized compressed signals during the burst, which allows reliable detection of echo signals from small objects.

, с - скорость света, Т_И - длительность зондирующих импульсов, используемых для обнаружения эхо-сигналов на максимальной требуемой дальности, а также - о недостаточной разрешающей способности по дальности и точности измерения дальности целей, находящихся на малых дальностях, что в результате может привести к ошибкам навигации.The disadvantage of the prototype is the insufficiently high secrecy of the probe radiation and, as a result, the radar noise immunity and environmental cleanliness both due to the non-optimal choice of parameters of the probe pulses in the absence of taking into account the own speed of the radar carrier, and due to the lack of regulation of signal parameters and processing as reduction in range to targets. In particular, we are talking about reducing the observability of echo signals from targets located at ranges R≪R ₀ , where R is the distance to the target,

, s is the speed of light, T _AND is the duration of the probe pulses used to detect echo signals at the maximum required range, and also about the insufficient resolution in range and accuracy of measuring the range of targets located at short ranges, which can result in navigation errors.

.An object of the invention is to increase the secrecy of the probe radiation and, as a consequence, the noise immunity and environmental cleanliness of the radar by optimally selecting the parameters of the probing radiation and signal processing parameters taking into account the own speed of the radar carrier, adjusting these parameters depending on the expected or measured range to detectable targets, and also optimal filtering of signals from targets in the near zone for ranges

.

The essence of the invention lies in the fact that in a method for detecting targets by a pulsed radar station, which includes generating highly stable microwave frequencies, the frequency-wise hopping of their carrier frequency f _c from period to pulse repetition period according to an arbitrary law, intra-pulse phase manipulation by binary multi-bit code, code restructuring phase manipulation from period to period of pulse repetition, pulse amplitude modulation, radiation of probe pulses, reception of reflected signals , superheterodyne conversion of received signals, amplification of received signals by intermediate frequency, quadrature phase detection of signals, amplification of quadrature components of signals by video frequency, binary quantization of quadrature components of signals by zero level, time compression of quadrature components, combining of compressed quadrature signals, binary quantization of combined signals with threshold level selected from the condition of acceptable probability of exceeding it by noise emissions, between Periodic accumulation of binary signals during the burst, detection of a burst of pulsed echo signals by comparing the accumulation result with a threshold level, the duration of T _AND probing pulses and the capacity of the N phase-shift keying code are selected from the conditions

where R _max - the maximum expected range to the target,

c is the speed of light

ΔR is the required range resolution,

и уменьшают импульсную мощность зондирующих импульсов пропорционально

, по значению V собственной скорости носителя радиолокационной станции, измеряемой бортовой системой навигации, вычисляют допплеровскую частоту F_Д в соответствии с соотношениемAs you approach the target and decrease R _max decrease T _AND while maintaining or decreasing N, at the same time adjust the passband of the receiving device in accordance with the ratio

and reduce the pulse power of the probe pulses proportionally

, according to the value of V own speed of the carrier of the radar station, measured by the on-board navigation system, calculate the Doppler frequency F _D in accordance with the ratio

, сдвигают по фазе в сторону отставания комплексную огибающую бинарно-квантованных сигналов после фазового детектирования на угол

, где 2^r - число уровней квантования угла 2π, r=2, 3,... - целые числа, и изменяют коэффициент сжатия N₁(R) к уровень квантования U(N₁) объединенных сжатых сигналов от целей на дальностях R<R₀ в соответствии с соотношениямиwhere ψ _A, ϑ _A are the angles that determine the direction of the antenna axis axis in azimuth and elevation relative to the direction of the vector V, then at discrete time instants with a period equal to

phase shift towards the lag the complex envelope of the binary-quantized signals after phase detection by an angle

, where 2 ^r is the number of quantization levels of the angle 2π, r = 2, 3, ... are integers, and the compression coefficient N ₁ (R) is changed to the quantization level U (N ₁ ) of the combined compressed signals from targets at ranges R < R ₀ in accordance with the ratios

- целая часть от

Where

- the whole part of

- монотонно возрастающая функция

- monotonically increasing function

N_min>1.

N _min > 1.

To a radar station that implements the indicated method, comprising a synchronizer connected in series, a frequency adjustment unit, an exciter, a phase manipulator, a power amplifier, an antenna switch and an antenna, a high-frequency amplifier, a mixer, an intermediate frequency amplifier, the output of which is connected to the signal (first) output the input of the phase detector block, the series-connected phase detector block, the video amplifier block and the amplitude quantizer block, as well as two identical tunable discrete filters, the outputs of which are connected, respectively, with the first and second inputs of the quadrature combining unit, the output of which is connected to the first input of the information processing device, and the antenna drive unit, the output of which is connected kinematically to the antenna, the input of the high-frequency amplifier connected to the third the shoulder of the antenna switch, the heterodyne input of the mixer and the voltage input of the reference frequency of the phase detector unit are connected to the corresponding outputs of the pathogen, the first output is of the giving pulses of the synchronizer is also connected to the first input of the code generator, the second input of which, combined with the inputs for controlling the recording of codes of tunable discrete filters, is connected to the output of the bursts of pulses of the clock frequency of the synchronizer, and the output is connected to the inputs of the code of the phase manipulator and tunable discrete filters, the output of clock pulses synchronizer is connected via a pulse modulator to a power amplifier, as well as to the corresponding input of the information processing device, the output of the clock pulses The synchronizer is connected to the clock inputs of the tunable discrete filters and the information processing device, and the information input of the latter is connected to the output of the azimuth codes of the antenna drive unit, additionally connected are a Doppler frequency calculator, a code-frequency converter and a Doppler frequency compensator, the first and second outputs which are connected respectively to the signal inputs of the first and second tunable discrete filters, as well as the threshold forming unit, the input to them whose clock frequency pulses and the corresponding input of the code-frequency converter are connected to the corresponding synchronizer output, the information outputs of the azimuth codes and elevation angle of the antenna drive unit and the information output of the frequency tuning block are connected to the corresponding inputs of the Doppler frequency calculator, the output of the synchronizer zeroing signals is connected to the corresponding the inputs of the threshold forming unit, the information processing device and tunable discrete filters, and the outputs of the command signals The information processing apparatus of the first to third coupled, respectively, to control inputs of video amplifier block synchronizer and a power amplifier, wherein the unit of phase detectors comprises 2 ^r-1 of the phase detector, the outputs of which through the corresponding 2 ^r-1 video amplifier unit video amplifier connected to the 2 ^{r -1} amplitude quantizers block amplitude quantizers, and the outputs of the latter are connected to the corresponding signal inputs of the Doppler frequency compensator.

According to the proposed method for detecting targets of a pulsed radar, it is proposed to choose a duration T _{AND of} probe pulses from the condition

where R _max - the maximum expected range to the target,

and the number of bits N of the FM code is from the relation

where ΔR is the required resolution in range,

reduce the duration of the probe pulses as they approach the targets and decrease R _max while maintaining (or decreasing) the number of bits of the FM code, at the same time adjust the passband of the receiving device in accordance with the ratio

, сдвигать по фазе в сторону отставания периодически - в дискретные моменты времени - комплексную огибающую бинарно-квантованных сигналов после фазового детектирования на угол

с периодом, равным

, где r=2, 3,... - целые числа,adjust the pulse power of the probing signals as R _max changes to the targets proportionally

phase shift towards the lag periodically - at discrete points in time - the complex envelope of the binary-quantized signals after phase detection by an angle

with a period equal to

where r = 2, 3, ... are integers,

F _D - Doppler frequency, which is determined from the ratio

where V is the own speed of the radar carrier;

ψ _A , ϑ _A - angles that determine the direction of the axis of the antenna pattern in azimuth and elevation relative to the direction of the vector V,

in particular, when r = 2, by circular switching of the direct and inverted quadrature components of these signals, and to change the compression coefficient N ₁ (R) with matched filtering, as well as the level U (N ₁ ) of quantization of the combined compressed signals from targets at ranges R < R ₀ where

according to the rules

- ′′целая часть от

N_min>1Where

- ′ ′ the integer part of

N _min > 1

- монотонно возрастающая функция

например

- monotonically increasing function

eg

We give the necessary explanations.

The fulfillment of condition (1) leads to the choice of the maximum possible duration T _{AND of the} probe pulse while providing a temporary isolation between transmission and reception, and therefore, at a given pulse energy, it minimizes its pulse power, i.e. utmost secrecy of probe radiation. Relations (2) and (3) determine the choice of the required resolution in range and bandwidth of the receiving device, consistent with the spectrum of the radar pulses of the signals.

The use of relatively long pulses in radar in coherent intra-pulse processing (in particular, when compressing FM signals) leads to a significant limitation of the channel bandwidth ΔF _D on the Doppler frequency, which is determined by the formula

following from the expression for the frequency characteristic of the Doppler frequency of the optimal intrapulse processing

valid for all pulsed signals with a rectangular amplitude envelope [5].

Therefore, if special measures are not taken, for radars mounted on board moving carriers, the Doppler frequency shift F _{D of the} signals reflected from the targets can exceed the half-width of the band 0.5ΔF _D and the corresponding signals will not be detected. This limits the choice of the duration of the T _AND probe pulses. In order to enable detection of the echo signals with longer duration _TI and provide thereby improving concealment, it is proposed to make compensation for the phase shift due to the Doppler frequency F _D of the reflected signals due to movement of the carrier.

For this purpose, it is proposed, firstly, to calculate this Doppler frequency, as indicated above, in accordance with expression (4), and secondly, to shift the phase (in the direction of the delay) of the received signals according to the quasilinear law, i.e.

при

at

j=1, 2, 3,...Where

j = 1, 2, 3, ...

через

часть периода

допплеровской частоты F_Д, т.к. при этом аппроксимируется требуемый линейный закон φ_Д(t)=φ₀-2πF_Дt, соответствующий вычитанию частоты Допплера F_Д из спектра комплексной огибающей принимаемых сигналов.Thus - at an angle

across

part of the period

Doppler frequency F _D , because while approximating the required linear law φ _D (t) = φ ₀ -2πF _D t corresponding to subtracting the Doppler frequency F _D from the spectrum of the complex envelope of the received signals.

Let us estimate the energy losses that the proposed stepwise approximation (9) of the linear phase shift φ _D (t) leads to.

The expression for the signal-to-noise ratio in voltage after compression and combining of the quadrature components has the form

Where

are the quadrature components of the signal before compression, and φ (t) in (11) is expressed as (9), t _i = t ₀ + iτ _И , i = 1,2, ..., N.

Further, at the time of matching, the weighting coefficients h _{i of the} squared filters and the code symbols q _i = ± 1 of the FM video signals are matched, so that h _Ni q _i = 1 for i = 1,2, ..., N.

In the ideal case, when φ _Д (t) = φ ₀ -2πF _Д t, substituting φ _Д (t _i ) instead of φ (t _i ) in (11), we obtain

In the case of the proposed stepwise approximation (9), we obtain from (10), (11), as is easy to see, the expression

where after simple transformations

, определяющие потери из-за дискретности предлагаемого закона (9) по сравнению с идеальным, приведены в таблицеCoefficient values

that determine the losses due to the discreteness of the proposed law (9) compared to the ideal are shown in the table

r K _r K _r , dB 2 0.90 0.9 3 0.974 0.2 four 0,999 0.02

Thus, even at r = 2, the energy loss due to the approximation of the linear law of phase change according to (9) does not exceed 1 dB.

Regulation of the pulsed power of the probing signals in the detection and tracking modes as the expected or measured range to the target decreases, respectively, ensures the utmost secrecy of the probing radiation. It should be borne in mind that, as we approach the target, the duration of T _AND probe pulses is proposed to be reduced in proportion to the decrease in range, therefore, to ensure the minimum necessary detection of the signal-to-noise ratio, it is proposed to adjust the pulse power of the probe signals in proportion to the cube of the expected value of the range to the targets. Finally, when reducing the range to targets to values R <R _{0, it is} proposed to ensure the matching of processing parameters — optimal filtering — with truncated received signals due to their partial overlap on the probing signals.

In this case, the compression ratio of the truncated received signals is determined by expression (6), and to achieve matching of these signals with the compression filter, it is advisable to change its length in the zone where R <R ₀ , in proportion to the current range. In this case, the noise dispersion after compression is reduced, and therefore it is advisable to also reduce the quantization level U of the combined compressed signals at ranges R <R ₀ so that the probability of exceeding it by noise emissions remains constant, while providing an additional increase in the radar noise immunity in this region.

In order to find the required law for changing the threshold level at R <R ₀ , we determine the probabilities of exceeding it in the presence and absence of a signal, i.e. P _N and R _W, respectively.

, получим для плотности распределения процесса после сжатия при объединении квадратур по правилу "корень квадратный из суммы квадратов"Assuming that the amplitude U _{0 of the} received signal is distributed according to the Rayleigh law, and the initial phase according to the uniform law in the interval

, we obtain for the distribution density of the process after compression when combining quadratures according to the rule "square root of the sum of squares"

, причем

- отношение сигнал/шум (по мощности) до сжатия.where x> 0,

, and

- signal-to-noise ratio (power) before compression.

, где y>0.When combining quadratures by the rule "sum of squares" we get

where y> 0.

Accordingly, for the probabilities of exceeding the levels x ₀ and y ₀ in the presence and absence of signals, we obtain

,

, так что

- в первом случае,

,

So that

- In the first case,

,

, причем

- во втором случае.

,

, and

- in the second case.

, а во втором случае - из соотношения

.Thus, the detection characteristics are the same in both cases, but the threshold level in the first case is selected from the relation

, and in the second case, from the relation

.

Therefore, depending on the rule of combining quadratures, the threshold level in the region R <R ₀ should be a linear or quadratic function of the square root of the compression coefficient N ₁ ≤N.

The invention is illustrated by a further description and drawings, which show:

- figure 1 is a structural diagram of a radar that implements the inventive method;

- figure 2 is a structural diagram of the pathogen (B) of the transmitter;

- figure 3 is a structural diagram of a frequency adjustment unit (BCH);

- figure 4 is a structural diagram of a compensator for Doppler frequency (CDF);

- figure 5 is a structural diagram of the Converter "code-frequency" (PCC);

- 6 is a structural diagram of tunable discrete filters (PDF);

- Fig.7 is a structural diagram of an information processing device (UOI);

- Fig. 8 is a structural diagram of a threshold formation unit (BFP);

- Fig.9 is a structural diagram of a primary processing unit (BPO);

- figure 10 - waveforms of the signals at the outputs of the synchronizer (C);

- 11 - structural diagram of the synchronizer.

In figure 1, the following notation:

1 - antenna (A);

2 - antenna switch (AP), which can be made in the form of a ferrite Y-circulator;

3 - power amplifier (UM), which is a pulse-modulated microwave amplifier, implemented depending on the required power and band of amplified frequencies on the basis of an electrovacuum device (amplitron, traveling wave lamp, multipath klystron, etc.) or a semiconductor device ( transistor) - see, for example, [6], pp. 19-52. The power control of the PA 3 can be carried out, for example, by introducing controlled attenuation into the excitation circuit or by switching in the output circuits;

4 - phase manipulator (FM), which can be performed according to the scheme given in the description of US patent No. 4338604 [3], and a segment of a strip waveguide switched by microwave diodes that are controlled by pulses from a generator can be used as a delay line code (see below);

5 - pathogen (B), a structural diagram of which is presented in figure 2;

6 - pulse modulator (IM), which, depending on the AM circuit, can be implemented according to one of the circuits given in [6], pp. 103-107, Figs. 43-45;

7 - tunable code generator (GK), which can be performed according to the scheme given in the description of patent No. 2039365 [4];

8 - block frequency adjustment (BFC), a structural diagram of which is shown in figure 3;

9 - high frequency amplifier (UHF), implemented in the form of a low-noise transistor microwave amplifier;

10 - mixer (SM), made in the form of a balanced mixer;

11 - intermediate frequency amplifier (UPCH);

на последующей относительно предыдущего, в частности, при r=2 БФД состоит из двух идентичных фазовых детекторов, на которые опорное напряжение подается со сдвигом фаз на

(на один относительно другого);12 is a block of phase detectors (BFD), consisting in the General case of 2 ^r-1 identical phase detectors, which are supplied with a reference voltage with phase shifts by

at the next relative to the previous one, in particular, at r = 2, the BFD consists of two identical phase detectors, to which the reference voltage is applied with a phase shift of

(one relative to the other);

13 is a block of 2 ^r-1 (in the general case) identical video amplifiers, the band of which can be changed according to the control signal, for example, by switching capacitors that determine the cutoff frequency of the frequency response (in particular, at r = 2, the BFD consists of two identical video amplifiers) ;

14 is a block of 2 ^r-1 (in the general case) amplitude quantizers (LHCs) that perform amplitude quantization of signals and noise arriving at their inputs at a zero threshold level in two gradations - for example, 1 or 0 (in particular, when r = 2 LHC consists of two identical amplitude quantizers);

15 - compensator for the Doppler frequency (CDF), the structural diagram of which is shown in figure 4 (for clarity, for the case r = 2);

16 - Converter "code-frequency" (PCC), a structural diagram of which is shown in Fig.5;

17 - Doppler frequency calculator (VDF), which performs the Doppler frequency calculations according to the formula

according to the known values of the speed V of the radar carrier received from the on-board navigation system, the sounding frequency f _C and the angles ψ _A and ϑ _A obtained from the frequency tuning unit 8 and the antenna drive unit 22 (see below), respectively;

18 ₁ , 18 ₂ - identical tunable discrete filters (PDF), the structural diagram of which is shown in Fig.6;

19 - block combining quadratures (BOK);

20 - information processing device (UOI), a structural diagram of which is shown in Fig.7;

21 - block formation of the threshold (BFP), a structural diagram of which is shown in Fig;

22 - synchronizer (C), the structural diagram of which is shown in Fig. 11, and Fig. 10 shows the waveforms of the signals at the outputs of the synchronizer;

23 is an antenna drive unit (BP), consisting of an azimuth drive, an elevation angle drive, and corresponding sensors.

In the diagram of FIG. 1, a synchronizer 22, a frequency tuner 8, a driver 5, a phase manipulator 4, a power amplifier 3, an antenna switch 2 and an antenna 1 are connected in series, a high-frequency amplifier 9, a mixer 10, an intermediate-frequency amplifier 11 are connected in series, a block of 12 phase detectors, a block of 13 video amplifiers, a block of 14 amplitude quantizers, and a Doppler frequency compensator 15 are connected, the control input of which is connected to the output of the Doppler calculator 17 through a code-frequency converter 16 frequency, and the first and second outputs with the first inputs of tunable discrete filters 18 ₁ , 18 _2, respectively, the outputs of which are connected to the corresponding inputs of the quadrature combining unit 19, and the output of the latter is connected to the first input of the information processing device 20.

The input of the high-frequency amplifier 9 is connected to the third arm of the antenna switch 2, the heterodyne input of the mixer 10 and the voltage input of the reference frequency of the phase detector unit 12 are connected to the corresponding (second and third) outputs of the exciter 5, the strip control input of the video amplifier block 13 is connected to the first output of the device 20 information processing, the second and third outputs of which are connected to the control inputs of the synchronizer 22 and the power amplifier 3, respectively.

The second input (modulation pulses) of the power amplifier 3 is connected to the output of the pulse modulator 6, the input of which and the fifth input of the information processing device 20 are connected to the fifth output of the synchronizer 22.

The first input of the code generator 7 is connected to the first output of the synchronizer 22, the output of which is connected to the control input of the phase manipulator 4 and the code (fifth) inputs of tunable discrete filters 18 ₁ and 18 ₂ . To the second output of the synchronizer 22 are connected the fourth inputs of tunable discrete filters 18 ₁ and 18 ₂ and the second input of the generator 7 codes.

The third output of the synchronizer 22, on which a sequence of clock pulses is generated, is connected to the second control input of the code-frequency converter 16, to the second inputs of the information processing device 20 and the threshold forming unit 21, as well as to the third inputs of the PDF 18 ₁ and 18 ₂ . The fourth output of the synchronizer 22 is connected to the third input of the information processing device 20 directly and, through the threshold generation unit 21, with its fourth input, as well as with the second inputs of the tunable discrete filters 18 ₁ and 18 ₂ .

The sixth input (azimuth codes) of the information processing device 20 and the first information input of the Doppler frequency calculator 17 are connected to the first information output of the antenna drive unit 23, the second information output (elevation codes) of which is connected to the second input of the Doppler frequency calculator 17, to the third input of which values of the carrier speed V from the on-board navigation system are provided. The fourth input of the Doppler frequency calculator 17 is connected to the information output of the frequency tuning unit 8.

The kinematic output of the drive unit 23 is connected to the control input of the antenna 1. The fourth output of the information processing device 20 represents the radar information output.

Figure 2 presents a diagram of the pathogen 5, where it is indicated:

24 ₁ , ..., 24 _m - crystal oscillators (KG ₁ , ..., KG _m );

25 ₁ , ..., 25 _m - gated amplifiers (US ₁ , ... US _m );

26 - frequency multiplier (Smart);

27 - mixer (cm);

28 - generator oscillations of the reference frequency (GOP);

29 - amplifier oscillations in the frequency of the signal (US _c );

30 - amplifier oscillations of the heterodyne frequency (US _g );

31 - amplifier oscillations of the reference frequency (US _op ).

In figure 2, the crystal oscillators 24 ₁ , ..., 24 _{m are} connected through the corresponding gated amplifiers 25 ₁ , ..., 25 _m , the control inputs of which form the input of the exciter 5, with the input of the frequency multiplier 26. The output of the frequency multiplier 26 through the amplifier 29 is connected to the first output (oscillation of the frequency f _C signal) of the pathogen 5, as well as directly to the input of the mixer 27, the second output of which is connected to the second output of the exciter 5 through the oscillator 30 of the local oscillator frequency. To the second input of the mixer 27 connected to the first output of the oscillator 28 of the reference frequency, the second output of the latter through the amplifier 31 of the oscillations of the reference frequency is connected to the third output of the pathogen 5.

Figure 3 presents the structural diagram of the block 8 frequency adjustment, where indicated:

32 - noise generator (GS);

33 - amplifier-limiter (UO);

34 - counter (MF);

35 - block of elements And;

36 - register (P);

37 - decoder (Dsh).

Figure 3, the control input of the block of elements 35 And is the input of the block 8 frequency adjustment, and its signal inputs are connected to the corresponding outputs of the counter 34, to the input of which through the amplifier-limiter 33 is connected the output of the generator 32 noise. The outputs of block 35 are bitwise connected to the register 36, the outputs of which are connected to the first output of frequency tuning block 8 through a decoder 37, and directly to the second output of block 8.

Figure 4 presents the structural diagram of the compensator 15 Doppler frequency for the case r = 2, i.e. when the angle 2π is quantized to 2 ^r = 4 levels, where it is indicated:

38 ₁ , 38 ₂ - inverters (INV ₁ , INV ₂ );

39 ₁ , 39 ₂ - switches (K ₁ , K ₂ ).

In Fig. 4, the first signal input of the Doppler frequency compensator 15 is connected directly to the first input of the first switch 39 ₁ and the fourth input of the second switch 39 ₂ , and through the first inverter 38 ₁ to the third input of the first switch 39 ₁ and the second input of the second switch 39 ₂ . A second signal input of the Doppler frequency of the compensator 15 is connected to a second input of the first switch 39 ₁ and the first input of the second switch 39 ₂ directly, and to a fourth input of the first switch 39 ₁ and the third input of the second switch 39 ₂ - via a second inverter 38 _2. The outputs of the first and second switches 39 ₁ , 39 ₂ form, respectively, the first and second outputs of the compensator 15 Doppler frequency. The control inputs of the switches 39 ₁ and 39 _{2 are} connected to the control input of the CDC 15.

In general, the Doppler frequency compensator 15 has

2 ^r-1 signal inputs, 2 ^r-1 inverters 38 ₁ , ..., 38 ₂ ^r-1 , and each of the switches 39 ₁ , 39 ₂ has 2 ^r inputs (1, 2, ..., 2 ^r ) . Below is the corresponding table of connections of the inputs of the compensator 15 with the inputs of the switches 39 ₁ , 39 ₂ - directly or through inverters (in the latter case this is indicated by the abbreviation "inv").

Connection table Compensator Input Number 15 Switch Input Number 39 ₁ Switch Input Number 39 ₂ one one 2 ^r-1 +2 ^r-2 +1 • • • • • • • • • 2 ^r-2 2 ^r-2 2 ^r 2 ^r-2 +1 2 ^r-2 +1 one • • • • • • • • • 2 ^r-1 2 ^r-1 2 ^r-2 1 inv 2 ^r-1 +1 2 ^r-2 +1 • • • • • • • • • 2 ^r-2 inv 2 ^r-1 +2 ^r-2 2 ^r-1 (2 ^r-2 +1) inv 2 ^r-1 +2 ^r-2 +1 2 ^r-1 +1 • • • • • • • • • 2 ^r-1 inv 2 ^r 2 ^r-1 +2 ^r-2

Figure 5 presents the structural diagram of the Converter 16 "code-frequency", where indicated:

40 - decoding coefficient of division;

41 - controlled divider.

In Fig. 5, the first input of the code-frequency converter 16 is connected to the control input of the controlled divider 41 through a decoder 40, the signal input of the controlled divider 41 forms the second input of the converter 16, and the output of the divider 41 is the output of the code-frequency converter 16.

Figure 6 presents the structural diagram of tunable discrete filters (PDF) 18 ₁ , 18 ₂ , where it is indicated:

42 - register shift signal (PC ₁ );

43 - register shift codes (RS ₂ );

44 - register shift signal logical "1" (RS ₃ )

45 ₁ , ..., 45 _N - adders modulo "2" (SM ₂ );

46 ₁ , ..., 46 _N - NAND elements;

47 - multi-input adder (MS);

48 - counter;

49 - block subtraction.

In the diagram of FIG. 6, the signal input of the signal shift register 42 forms the first input of a tunable discrete filter 18 ₁ (18 ₂ ). The input of the zeroing of the register 44 of the shift signal of the logical "1" is connected to the input of the zeroing of the counter 48 and forms the second input of the tunable discrete filter 18 ₁ (18 ₂ ). The third input of the PDF 18 ₁ (18 ₂ ) is connected to the clock inputs of the signal shift register 42, the logic shift signal register 44 "1" and the counter 48. The outputs of all N bits of the signal shift register 42 are connected to the first inputs of the corresponding adders 45 ₁ , ... , 45 _N modulo "2", their second inputs are connected to the corresponding outputs of the N bits of the code shift register 43, the signal input of which forms the fifth input of the PDF 18 ₁ (18 ₂ ), and the clock input of the signal shift register 43 is the fourth input of the PDF 18 ₁ (18 ₂ ). The outputs of all N adders 45 ₁ , ..., 45 _{N are} connected to the first inputs of the NAND elements 46 ₁ , ..., 46 _N , the second inputs of which are connected to the corresponding outputs of the register 44 of the signal shift logical "1", the signal input of which connected to the logical 1 bus.

The outputs of the AND-NOT elements 46 ₁ , ..., 46 _{N are} connected to the corresponding inputs of the multi-input adder 47, the output of which is connected to the input of the reduced subtraction unit 49. The input of the subtracted subtraction block 49 is connected to the output of the counter 48, and the output of the subtraction block 49 forms the output of the tunable discrete filter 18 ₁ (18 ₂ ).

Figure 7 presents the structural diagram of the device 20 information processing, where indicated:

50 - block comparison with the threshold (BSP);

51 - primary processing unit (BPO);

52 - key (C);

53 - block secondary processing (BVI).

In the diagram of Fig. 7, the first input of the UOI 20 through the threshold comparison unit 50 is connected to the first - signal - input of the primary processing unit 51, the second input of which is connected to the third input of the nulling signal of the information processing device 20, the fourth input of which is the control input of the block 50 comparison with the threshold. The second input of the information processing device 20 — clock pulses — is connected via a key 52 to the third input of the primary processing unit 51, the control input of the key 52 forms the fifth input of the device 20, and the fourth input of the primary processing unit 51 is the sixth input - azimuth values of the device 20. Output the primary processing unit 51 is connected to the signal input of the secondary processing unit 53, and the outputs of the latter from the first to the fourth form the corresponding outputs of the information processing device 20.

On Fig presents a structural diagram of the block 21 of the formation of the threshold, where indicated:

54 - counter (MF);

55 - decoder (Dsh);

56 - memory block (PSU);

57 - trigger (Tg);

58 - key (C).

In the diagram of Fig. 8, the first input of the threshold forming unit 21 is the zero setting of the counter 54, its outputs are bitwise connected to the corresponding inputs of the decoder 55 and the inputs of the least significant bits of the memory unit 56, the highest bit of which is connected to the output of the trigger 57, and the counting input of the latter is connected with the output of the decoder 55. The output of the trigger 57 is also connected to the control input of the key 58, the information input of which is the second input of the block 21, and the output is connected to the counting input of the counter 54. The first input of the block 21 is also connected to zero they trigger input 57 and the output 56 of the storage unit 21 is the output of block formation threshold.

Figure 9 presents the structural diagram of the block 51 of the primary processing, where indicated:

59 ₁ , ..., 59 _n-1 - shift registers (PC ₁ , ..., PC _n-1 );

60 - multi-input adder (MS);

61 - digital comparator (CC);

62 - polling unit (BO);

63 - range meter (ID);

64 - azimuth meter (IA);

65 - random access memory device goals (RAM).

In Fig. 9, n-1 registers 59 ₁ , ..., 59 of the _n-1 shift are connected in series, the input of the first register 59 _{1 of the} shift is connected to the first input of the multi-input adder 60 and at the same time is the first input of the primary processing unit 51, the inputs of the registers 59 ₂ , ..., 59 _{n-1 are} connected respectively to the inputs from the second to the n-1st, multi-input adder 60, the n-th input of which is connected to the output of the register 59 _n-1 .

The clock inputs of all registers 59 ₁ , ..., 59 _{n-1 of the} shift are interconnected and form the third input of the primary processing unit 51. The output of the multi-input adder 60 through a digital comparator 61 is connected to the input of the polling unit 62, the first and second outputs of which are connected through a range meter 63 and an azimuth meter 64, respectively, with the first and second inputs of the target memory 65, the output of which is the output of the primary processing unit 51. The second and third inputs of the range meter 63 are connected respectively to the inputs of the same name of the unit 51, and the second input of the azimuth meter 64 is the fourth input of the primary processing unit 51.

Figure 10 presents the waveforms of the signals at the outputs of the synchronizer 22, where indicated:

τ₀ и упреждением t₀ относительно начала периода повторения зондирующих импульсов, причем τ₀≪t₀, t₀≪T_П - на первом выходе синхронизатора 22;66 is a waveform of the control pulses of the frequency tuning unit 8 and the code generator 7 with a duration

τ ₀ and lead t ₀ relative to the beginning of the repetition period of the probe pulses, and τ ₀ ≪t ₀ , t ₀ ≪T _P - at the first output of the synchronizer 22;

67 - waveforms of bursts of pulses with a burst duration of T _AND , a pulse repetition period of τ _AND ≪T _AND , a pulse duration of τ ₀ ≪τ _And and a repetition period of bursts of T _P - to control the code generator 7 and tunable discrete filters 18 ₁ , 18 ₂ - on the second output of the synchronizer 22;

68 - waveforms of sequences of pulses with a repetition period τ _AND ≪T _И and pulse durations τ ₀ ≪τ _И - for controlling a code-frequency converter 16, tunable discrete filters 18 ₁ and 18 ₂ , an information processing device 20 and a threshold generating unit 21 - at the third output of the synchronizer 22;

69 - waveforms of periodic pulses with a period T _P, delay T _And relative to the beginning of the repetition period and duration τ ₀ ≪T _And - zeroing signals for controlling tunable discrete filters 18 ₁ , 18 ₂ , information processing device 20 and threshold forming unit 21 - on the fourth synchronizer output 22;

70 - waveforms of periodic pulses with a period T _P , the leading edge of which is the beginning of the repetition period, and the duration T _AND <T _P - for controlling the pulse modulator 6 and the information processing device 20 - at the fifth output of the synchronizer 22.

Figure 11 presents the structural diagram of the synchronizer 22, where indicated:

71 - generator of clock pulses (GTI);

72 - controlled divider (DU);

73 - key (C);

74 - counter (MF);

75 - decoder (Dsh);

76 - trigger block (BTg).

In the diagram of FIG. 11, a clock pulse generator 71, a controlled divider 72, and a counter 74 are connected in series. The outputs of the counter 74 through a decoder 75 are bitwise connected to the inputs of the trigger block 76. The input of the controlled divider 72 is the input of the synchronizer 22, the output of the controlled divider 72 through the key 73 is connected to the second output of the synchronizer 22 and directly to its third output. The first, fourth and fifth outputs of the synchronizer 22 are connected to the outputs of the trigger block 76, and the last output is also connected to the control input of the key 73.

In accordance with the structural diagram of figure 1, the radar that implements the inventive method, works as follows.

The transmitter exciter 5 generates continuous oscillations of the signal frequency f _{Ci, the} local oscillator f _Gi and the reference oscillations of the intermediate frequency f _pch , while the frequencies f _Ci , f _{Gi are} highly stable during one repetition period T _{P of the} probe pulses, but can vary stepwise from period to period randomly under the action of signals from the frequency hopping unit 8, assuming one of the values of m (r = 1, 2, ..., m), and so that always the relation | f _Ci -f _{plaster Gi} | = f _nq.

The causative agent 5 works as follows (figure 2).

Crystal oscillators 24 ₁ , ..., 24 _m generate continuous oscillations of stable frequencies f ₁ , f ₂ , ..., f _m, respectively, these oscillations are amplified by gated amplifiers 25 ₁ , ..., 25 _m, respectively, of which of each repetition period, only one is open, and the rest are closed - in accordance with the control code received at the control inputs of the amplifiers 25 ₁ , ..., 25 _m through the input of the pathogen 5 from the frequency tuning unit 8. Oscillations of frequency f _i through an open amplifier 25 _i are supplied to a frequency multiplier 26, where they are multiplied to frequency f _Ci , frequency fluctuations f _{Ci are} amplified in amplifier 29 and come to the first output of the exciter 5, and also go to the input of the mixer 27, to the other input of which vibrations of the intermediate frequency f _pch are generated, generated by the generator of the oscillations of the reference frequency 28. The oscillations of the heterodyne frequency f _Gi generated in the mixer 27 are amplified in the amplifier 30 and fed to the second output of the pathogen, and the oscillations of the reference frequency equal to the intermediate pass through the amplifier 31 to the third output of the pathogen 5.

The frequency _tuning f _Ci and f _{Gi is} performed using the block 9 frequency adjustment (BFCH), which operates as follows (figure 3).

, далее, эти колебания усиливаются и ограничиваются в усилителе-ограничителе 33 и поступают на счетчик 34, который осуществляет счет, например, положительных фронтов по модулю m и имеет, таким образом, m равновероятных состояний. В момент, определяемый синхроимпульсами, поступающими через период повторения Т_П с первого выхода синхронизатора 22 (66, фиг.11) с упреждением на время t₀ относительно начала следующего периода, показания счетчика 34 через элемент "И" БЭИ 35 записываются в регистр 36 и преобразуются в дешифраторе 37 в параллельный m-разрядный код с одним ненулевым разрядом, который сохраняется в течение всего периода повторения и определяет значения f_Ci и f_Гi в следующем периоде. Одновременно код частоты f_Ci с выхода регистра 36 поступает на второй выход блока 8 перестройки частоты.A noise generator 32, constructed, for example, based on a noise diode, generates a noise signal with a spectral width far exceeding the repetition frequency

further, these oscillations are amplified and limited in the amplifier-limiter 33 and fed to the counter 34, which counts, for example, positive edges modulo m and thus has m equiprobable states. At the moment determined by the clock pulses arriving through the repetition period T _P from the first output of the synchronizer 22 (66, Fig. 11) with a lead time of t ₀ relative to the beginning of the next period, the readings of the counter 34 through the element "And" BEI 35 are recorded in the register 36 and are transformed in the decoder 37 into a parallel m-bit code with one non-zero bit, which is stored throughout the repetition period and determines the values of f _Ci and f _Гi in the next period. At the same time, the frequency code f _Ci from the output of the register 36 is fed to the second output of the frequency tuning unit 8.

Fluctuations in the frequency of the signal f _Ci enter the phase manipulator 4, where they are phase-manipulated at the level 0π by a binary multi-digit code (the number of bits N) generated in the tunable code generator 7. The operation of the code generator 7, constructed in accordance with the code generator described in the prototype [4], occurs exactly the same as in [4]. Further, the oscillations are amplified by power in a power amplifier 3, which generates probe pulses with a duration T _AND with an intrapulse phase shift keying (FM) N bit binary code tunable from period to period of repetition under the action of a pulse modulator 6 controlled by a synchronizer 22 and an information processing device 20 T _n The probe pulses generated in the transmitter power amplifier 3 pass through the antenna switch 2 into the antenna 1 and are emitted into space.

In this case, the duration T _{AND of the} probe pulses and the number of bits of the code N FM are selected from relations (1) and (2), respectively, and the pulse power is set to maximum to detect targets at maximum range. The antenna drive unit 23 operates autonomously, providing an overview of the space.

The received signals passing through the antenna switch 2 fall into the high-frequency amplifier 9 and, after amplification, into the mixer 10, to the heterodyne input of which the oscillations of the local oscillation frequency from the pathogen 5 are received. The intermediate-frequency signals generated in the mixer 10 are amplified, further, in the amplifier 11 of the intermediate frequency and come to the phase detectors of block 12, to which the oscillations of the intermediate frequency from the pathogen 5 are received as oscillations of the reference frequency.

- на последующий относительно предыдущего (в частности, при r=2 БФД состоит из двух фазовых детекторов со сдвигом фаз по опорному напряжению на

), поэтому на выходах БФД 12 образуютсяBlock 12 phase detectors consists of 2 ^r-1 identical phase detectors, which are supplied with a reference voltage with a shift

- the next relative to the previous one (in particular, at r = 2, the BFD consists of two phase detectors with a phase shift in the reference voltage by

), therefore, at the outputs of the BFD 12 are formed

2 ^r-1 signals S _l (t), l = 1, 2, ..., 2 ^r-1 , of the form

a (tt _R ) cos [2πF _Д t + ψ (tt _R ) + φ],

where a (t), ψ (t) are the functions of the amplitude and phase modulation (manipulation) of the signals,

- задержка, соответствующая дальности R до цели,

- the delay corresponding to the range R to the target,

φ is the initial phase shift between the oscillations of the received signals and the reference oscillations, and

here Nτ _И = Т _И , υ _i = 1 or 0 is the FM code.

, т.е. составляющим

периода допплеровской частоты, соответствующей радиальной составляющей скорости носителя РЛС в направлении на цель, наблюдаемую в данный момент.These signals pass through the corresponding series-connected video amplifiers of block 13 and amplitude quantizers of block 14, where they are quantized by 2 gradations (0 or 1) at a zero threshold level, the quantized signals are transmitted to the corresponding signal input 2 ^{r-1 of the} channel compensator 15 of the Doppler frequency, in which (Fig. 4) directly and through inverters 38 ₁ , ..., 38 ₂ ^r-1 (in Fig. 4 r = 2) are fed to 2 ^r - input switches 39 ₁ , 39 ₂ , to which as control (commuting ) signals come from the output of the converter 16 "code-hour tota "pulses with a period equal to

, i.e. constituting

period of the Doppler frequency corresponding to the radial component of the speed of the radar carrier in the direction of the currently observed target.

, поступающих на второй вход управляемого делителя 41 с третьего выхода (68 на фиг.11) синхронизатора 22.To generate these pulse values of the Doppler frequency F _D, calculated from the formula (4) in the calculator 17, the Doppler frequency from the values of the azimuth and elevation angle of the antenna coming from the information output 23 of the antenna drive unit, the velocity V medium supplied from the onboard navigation system, and the values frequency signals f _Ci currently coming to the calculator 17 from the frequency hopping unit 8 is transmitted to the first input of the converter 16 "frequency-code", wherein they pass through the decoder 40 (see FIG. 5) and further convert the Xia in divider 41 with a controlled pulse repetition frequency

entering the second input of the controlled divider 41 from the third output (68 in FIG. 11) of the synchronizer 22.

As a result, pulses with the required frequency 2 ^r F _Д are generated at the output of the code-frequency converter 16, controlling the switches 39 ₁ , 39 ₂ in the compensator 15 of the Doppler frequency. Under the influence of these pulses, the switches 39 ₁ , 39 ₂ synchronously connect their respective input contacts 1, 2, ..., 2 ^r to their outputs.

в ПДФ 18)At the multi-channel signal input of the compensator 15, random processes are formed after amplitude quantization in block 14 according to the rule (taking into account the subsequent subtraction

in PDF 18)

where ε _l (t) = n _l (t) + s _l (t), and n _l (t) are normal random processes with zero mathematical expectations and the same standard deviations σ _Ш (i.e., noise).

(т.е. при малых отношениях сигнал/шум до сжатия)Then for mathematical expectations and standard deviations of the processes η _l (t) we obtain for

(i.e., at low signal to noise ratios before compression)

от сигналов на одноименных входах коммутатора 39₁.Further, as can be seen from the connection table and figure 4, the signals at the inputs of the switch 39 ₂ lag in phase by

from signals at the inputs of the same name switch 39 ₁ .

,As a result of switching at the outputs of the compensator 15 of the Doppler frequency, processes η _C (t), η _S (t) are formed with mathematical expectations

,

,

,

Where

j=1, 2, 3,...at

j = 1, 2, 3, ...

those. the phase incursions will be compensated due to the Doppler frequency F _D with an accuracy of distortion, the smaller the greater r. As shown above, these distortions lead to energy losses, which even for r = 2 do not exceed 1 dB, and for r≥4 they are negligible.

Thus, at the first and second outputs of the Doppler frequency compensator 15, due to the proposed connection scheme, quadrature signal components are formed, which are fed to the first inputs of tunable discrete filters 18 ₁ and 18 _2, respectively.

Tunable discrete filters 18 ₁ and 18 ₂ work as follows (Fig.6).

Both PDFs (18 ₁ and 18 ₂ ) in each repetition period are consistent with the probe signal according to the phase manipulation code (FM). To ensure this coordination in each repetition period, the PDF 18 ₁ , 18 _{2 are} rebuilt in accordance with the change in the FM code in the code generator 7.

The discrete filters 18 ₁ and 18 _{2 are} tuned to the required FM code during the emission of probe pulses by writing the code generated in this period into the shift registers 43 of each PDF 18 ₁ , 18 ₂ . The FM code is fed to the fifth inputs of the discrete filters 18 ₁ , 18 ₂ . Writing this code to the registers 43 is carried out by a packet of N clock pulses arriving at the fourth inputs of discrete filters 18 ₁ , 18 ₂ from the second output (67, Fig. 10) of the synchronizer 22. At the end of the packet of N clock pulses, discrete filters 18 ₁ , 18 ₂ are tuned to the expected reflected FM signal.

At the end of the probe pulse, which is the moment the range begins, from the fourth output of the synchronizer 22, a zeroing signal (69, Fig. 10) comes to the second inputs of the PDF 18 ₁ and 18 ₂ , under which the shift registers 44 and counters 48 are reset.

The received signal at clock pulses coming from the third output of the synchronizer 22 to the third inputs of the PDF 18 ₁ , 18 ₂ , is moved into the shift registers 42. At the same time, the clock pulses provide the recording of “units” in the shift register 44 and the current range count by the counter 48. The binary numbers recorded in the registers 42 and 43 are summed bitwise modulo 2 in the adders 45 ₁ , ..., 45 _N. Single-digit numbers are fed to the first inputs of elements 46 ₁ , ..., 46 _N AND NOT, the second inputs of which receive enable ("1") or disable ("0") signals from shift register 44. At short ranges R <R _0, with each clock pulse, the enable signal is sent to the next AND-NOT element, because the next "unit" is pushed into the register 44, so the number of open items "NAND" is proportional to the current range. When R≥R _0, all bits of the register PC ₃ 44 are filled with "units" and all the elements "AND NOT" are open. Single-digit numbers from the outputs of the adders 45 ₁ , ..., 45 _N pass through the open AND-NOT elements 46 ₁ , ..., 46 _N and are added to the multi-input adder 47, after which a number proportional to the result is subtracted from it current range (the number of clock pulses counted from zero range by the counter 48). Due to the alternate opening of the outputs of the bits of the signal registers 42 with an increase in the current range in the zone R <R ₀ , the "effective" length of the PDF 18 ₁ (18 ₂ ) changes in accordance with (6), and the filter is always aligned in length with the received signal.

At the moment when the received signal is completely pushed into the shift registers 42, i.e. when the moment of matching comes, the main peaks of the quadrature components of the compressed signal are formed at the outputs of the discrete filters 18 ₁ and 18 ₂ . The quadrature components of the compressed signals are fed to the first and second inputs of the quadrature combining unit 19, where they are squared in the corresponding quadratures and summed. These operations ensure the elimination of the unknown initial phase of the reflected signals.

, начинают поступать на вход приемника, когда еще не закончено излучение зондирующего импульса и приемник, соответственно, закрыт, так что первые

разрядов ФМ-сигналов не принимаются, а принимаемый сигнал состоит из оставшихся

разрядов, которые в момент согласования занимают N₁ первых разрядов сигнальных регистров 42 сдвига в ПДФ 18₁, 18₂. Однако, в отличие от прототипа [4], обеспечивается полное согласование ПДФ с принимаемыми сигналами независимо от дальностей до целей, их отражающих.Signals reflected from targets at ranges

, begin to arrive at the input of the receiver, when the radiation of the probe pulse is not yet completed and the receiver, respectively, is closed, so the first

bits of FM signals are not received, and the received signal consists of the remaining

bits that at the time of coordination occupy N _{1 the} first bits of the signal registers 42 of the shift in the PDF 18 ₁ , 18 ₂ . However, unlike the prototype [4], full matching of the PDF with the received signals is provided, regardless of the ranges to the targets that reflect them.

From the output of the quadrature combining unit 19, the received signal, represented by a multi-bit positive number, is sent to the comparison unit 50 with the threshold of the information processing device 20 (Fig. 7), where it is compared with the numerical threshold generated by the threshold forming unit 21 (Fig. 8), which operates in the following way.

. В этот момент по сигналу с выхода дешифратора 55 срабатывает триггер 57, который подает сигнал на старший разряд БП 56 и одновременно блокирует вход счетчика 54. Таким образом, устанавливается постоянный (максимальный) порог на выходе блока 21 формирования порога на дальностях R≥R₀. Благодаря этому в блок 50 сравнения с порогом через четвертый вход устройства 20 поступают пороговые значения, которые при R<R₀ растут пропорционально N₁, где N₁ - число действующих разрядов ПДФ 18₁, 18₂, что обеспечивает постоянство ложных выбросов из-за шумов в области R<R₀. Если значение сигнала, поступающее на сигнальный вход блока 50 сравнения с порогом, превышает пороговое значение, приходящее на его второй вход, вырабатывается единичный сигнал, в противном случае - нулевой. Пачка нормированных импульсов сигналов от цели поступает на первый вход блока 51 первичной обработки, проходит цепь из n-1 последовательно соединенных регистров 59₁,... 59_n-1 сдвига, на которые в качестве сдвигающих импульсов через третий вход блока 51 поступают импульсы с третьего выхода синхронизатора 22 (68, фиг.10), стробируемые на время T_И излучения зондирующих импульсов в ключе 52, на который в качестве управляющих приходят импульсы с длительностью Т_И с пятого выхода (70, фиг.10) синхронизатора 22. Число разрядов в регистрах 59₁,..., 59_n-1 следует выбирать равным

, для того чтобы обеспечить накопление n импульсов пачки со всех элементов дальности. В процессе накопления благодаря перестройке кодов ФМ от импульса к импульсу при бинарном квантовании перед сжатием происходит, как в прототипе [4], подавление боковых лепестков. Накопленные за n периодов повторения сигналы образуются в сумматоре 60, с выхода которого они поступают в цифровой компаратор 61, в котором они сравниваются с заранее выбранным пороговым числом

, реализуя обнаружение по правилу

. При условии превышения порога в компараторе 61 образуются нормированные импульсы, которые поступают в блок 62 опроса, осуществляющий объединение первого и последнего импульсов, превысивших порог, и считывание значений дальности и азимута обнаруженных целей. Это достигается в измерителе 63 дальности и измерителе 64 азимута, на первый из которых с этой целью приходят обнуляющие импульсы с четвертого выхода синхронизатора 22, фиксирующие начало отсчета дальности, и импульсы с частотой

с третьего входа блока 51 первичной обработки, осуществляющие счет дальности в измерителе 63 дальности, а на второй - коды значений азимута с датчика углов антенны, поступающие из блока 23 приводов антенны через шестой вход устройства 20 обработки информации на четвертый вход блока 51 первичной обработки.At the first input of the threshold forming unit 21, a nulling signal from the fourth output (69, Fig. 10) of the synchronizer 22 is received, which resets the counter 54 and the trigger 57 at the beginning of the range reference. Next, the clock pulses from the third output of the synchronizer 22 (68, Fig. 10) are supplied through the second input of the block 21 and the public key 58 to the counting input of the counter 54, in which, thus, the current range is counted in units of _And . Codes of the current range from the outputs of the counter 54 are transmitted to the lower bits of the address of the memory block 56, in which the threshold values are written as functions of the range, passing to the output of the block 21. At the same time, the signals from the outputs of the counter 54 are transmitted to the decoder 55, which is activated when the current range R reaches a value

. At this moment, a trigger 57 is triggered by the signal from the output of the decoder 55, which sends a signal to the senior bit of the PSU 56 and simultaneously blocks the input of the counter 54. Thus, a constant (maximum) threshold is set at the output of the threshold forming unit 21 at ranges R≥R ₀ . Because of this, threshold values are received in the comparison unit 50 with the threshold through the fourth input of the device 20, which at R <R ₀ increase in proportion to N ₁ , where N ₁ is the number of active PDF discharges 18 ₁ , 18 ₂ , which ensures the constancy of spurious emissions due to noise in the region R <R ₀ . If the signal value arriving at the signal input of the comparison unit 50 with a threshold exceeds a threshold value arriving at its second input, a single signal is generated, otherwise it is zero. A pack of normalized pulses of signals from the target goes to the first input of the primary processing unit 51, a chain of n-1 series-connected shift registers 59 ₁ , ... 59 _n-1 passes through which pulses from the third input of block 51 are received from the third output of the synchronizer 22 (68, FIG. 10), gated for the time T _{And the} radiation of the probe pulses in the key 52, to which pulses with a duration T _And come from the fifth output (70, FIG. 10) of the synchronizer 22 as the control pulses. Number of discharges in registers 59 ₁ , ..., 59 _n-1 should be selected equal

, in order to ensure the accumulation of n burst pulses from all range elements. In the process of accumulation due to the restructuring of the FM codes from pulse to pulse during binary quantization, before compression, as in the prototype [4], side lobes are suppressed. The signals accumulated over n repetition periods are generated in the adder 60, from the output of which they enter the digital comparator 61, in which they are compared with a pre-selected threshold number

realizing discovery by rule

. If the threshold is exceeded, normalized pulses are generated in the comparator 61, which are fed to the polling unit 62, combining the first and last pulses that have exceeded the threshold, and reading the range and bearing values of the detected targets. This is achieved in the range meter 63 and the azimuth meter 64, the first of which for this purpose are resetting pulses from the fourth output of the synchronizer 22, fixing the origin of the range, and pulses with a frequency

from the third input of the primary processing unit 51, performing range calculation in the range meter 63, and on the second, the azimuth codes from the antenna angle sensor coming from the antenna drive unit 23 through the sixth input of the information processing device 20 to the fourth input of the primary processing unit 51.

The coordinates of the detected targets, grouped in pairs, are recorded in the random access memory 65 of the targets, from where they are sent to the output of block 51 and, further, to the input of secondary processing unit 53.

The secondary processing unit 53 may be implemented as a programmable computing device. Its work consists in inter-review processing and the formation of teams associated with changes in the parameters of the probing (and therefore received) signals as the ranges to the targets change. Inter-review processing consists in accumulating primary detection decisions made in primary processing block 51 for a cycle of M reviews and making final decisions of the form “K from M” for each of the targets found in block 51, and also in measuring the coordinates of these targets - range and azimuth by "smoothing", i.e. averaging them over M reviews, taking into account the movement of targets relative to the radar during this processing.

, и, когда условие

перестает выполняться, принимается решение об уменьшении длительности Т_И зондирующих сигналов, например, при сохранении разрядности N кода ФМ, одновременно регулируется полоса пропускания приемного устройства в соответствии с соотношением

и уменьшается мощность зондирующих сигналов по мере уменьшения максимальной дальности пропорционально

. Соответствующие команды подаются через первый, второй и третий выходы устройства 20 обработки информации на входы управления блока 13 видеоусилителей, синхронизатора 22 и усилителя 3 мощности, соответственно, одновременно значения координат целей поступают на четвертый - информационный выход устройства 20 обработки информации.The value of the current maximum range to the target from the number of detected R _max (t) is compared with the value

, and when the condition

ceases to be executed, a decision is made to reduce the duration of T _AND probing signals, for example, while maintaining the bit depth N of the FM code, the passband of the receiving device is simultaneously adjusted in accordance with the ratio

and the power of the probing signals decreases as the maximum range decreases proportionally

. Corresponding commands are sent through the first, second, and third outputs of the information processing device 20 to the control inputs of the video amplifier block 13, the synchronizer 22, and the power amplifier 3, respectively, simultaneously, the coordinates of the targets are sent to the fourth — information output of the information processing device 20.

The operation of the synchronizer 22 is to generate control signals (11), while the signal 68 (third output) is formed by dividing the frequency of the pulses generated by the clock generator 71 (11), the required number of times using a controlled divider 72, the input of which receives a command signal from the second output of the information processing device 20. Signals 66, 69, and 70 — at the first, fourth, and fifth outputs — are generated using a counter 74, a decoder 75, and an RS-flip-flop unit 76, which generates signals of the required duration and delay (lead) relative to the beginning of the repetition period, and signal 67 at the second output - by logically multiplying the signals 69 and the inverted signal 70, performed using the key 73.

The technical advantage of the proposed detection method compared with the methods implemented in the prototype is that it allows to increase the secrecy of the probe radiation and noise immunity, primarily in relation to the response interference induced by the intelligence stations, when detecting targets in a wide range of ranges (when changing the maximum range to the target by 10 or more times during the observation process) including radars mounted on mobile carriers, and also provides an increase in the resolution of lities of range and accuracy of range measurement with decreasing distance to targets.

), в принимаемых сигналах перед сжатием, как это и предусмотрено в заявляемом способе.Increasing secrecy and interference resistance with respect to the mating interference determined by the possibility of significant increase of the duration of the complex FM signal over the prior art while maintaining the energy signal through the optimal choice of pulse duration T _and that at long range detection can be increased by at least an order of magnitude ( i.e., 10 times or more) in comparison with the prototype, pulse power can be reduced by the same amount, which provides a corresponding increase in stealth. For radars mounted on mobile carriers, a prerequisite for the use of relatively long FM signals is the compensation of the Doppler frequency shift corresponding to the carrier’s own radial speed (with accuracy

), in the received signals before compression, as provided for in the claimed method.

, одновременно с соответствующей регулировкой порога обнаружения импульсов сжатых сигналов приводит к повышению наблюдаемости в

раз по мощности (например, в 2-3 раза - благодаря уменьшению дисперсии шумов при сжатии) по сравнению с прототипом, не предусматривающим этих мер, что эквивалентно соответствующему повышению помехозащищенности по отношению к шумовым помехам, в частности, на 3-5 дБ.The proposed - as the maximum range decreases - a proportional decrease in the duration of the probe pulses simultaneously with the expansion of the spectrum of FM signals and the passband of the receiver while maintaining (or decreasing) the number of bits of the FM code and decreasing the pulse power of the probing signals in proportion to the cube of the maximum range value allows maintaining high stealth in these conditions, as well as to increase the resolution in range and the accuracy of its measurement several times (for example, 5-10 times). Finally, automatic matching of the compression filter with the duration of the FM signals from targets located at ranges R less than

, simultaneously with the corresponding adjustment of the threshold for detecting pulses of compressed signals, increases the observability

times in power (for example, 2-3 times - due to a decrease in the dispersion of noise during compression) compared with the prototype that does not provide for these measures, which is equivalent to a corresponding increase in noise immunity with respect to noise interference, in particular by 3-5 dB.

Thus, the implementation of the proposed measures will lead to a significant increase in stealth, noise immunity, resolution in range and accuracy of range measurement, including for radars mounted on mobile carriers.

Using the information presented in the application materials, the proposed method can be implemented in a radar made according to the above description and drawings using known materials, elements, units and technologies, and used to detect signals from targets and measure their coordinates, which proves the industrial applicability of the object of the invention .

INFORMATION SOURCES

1. Digitized Marine Radar. Furuno Electric Co., LTD, Catalog ¹ . R-089f, Radio Holland Group, Electr. Systems, Marine.

2. Reference radar. Ed. M. Skolnik. Per. from English. Volume 4 (Ch. 3 "Civil Ship Radar, p. 96), M .: Sov. Radio, 1978.

3. Pulse coherent Doppler radar with frequency tuning. US patent No. 4338604, CL. G 01 S 13/24, publ. 07/06/82.

4. Radar station. RF patent No. 2039365, cl. G 01 S 13/52, with priority from 09/27/93, publ. 07/09/95 (prototype).

5. Cook C. and Bernfeld M. Radar systems, M .: Sov. Radio, 1971 (p. 103, table 41).

6. Reference radar. Ed. M. Skolnik. Translation from English. Volume 3, M. Sov. Radio, 1979.

Claims (2)

1. A method for detecting targets by a pulsed radar station, which includes emitting pulsed phase-shift keyed signals with step-wise tuning of the carrier frequency f _C , tuning of the phase manipulation code and pulse amplitude modulation from period to period of the probe pulses, receiving the reflected signals and processing them, including superheterodyne conversion and amplification at an intermediate frequency, quadrature phase detection and amplification of the quadrature components of the signals at the video frequency, bi explicit quantization of the quadrature component signals at zero level, time compression of the quantized quadrature component signals, combining the compressed quadrature component signals, binary quantization of the combined signals with a threshold level selected from the condition of an acceptable probability of exceeding it by noise emissions, inter-period accumulation of binary signals during the burst time and detection bursts of pulsed echo signals by comparing the accumulation result with a threshold level, characterized in that the duration l T _{And the} probe pulses and the capacity N of the phase-shift keying code are selected from the conditions

where R _max - the maximum expected range to the target, c is the speed of light ΔR is the required range resolution, as they approach the target, they reduce the duration of the probe pulses while maintaining or decreasing the bit depth of the phase manipulation code, at the same time adjust the passband of the receiving device in accordance with the ratio

and reduce the pulse power of the probe pulses proportionally

at discrete time instants with a period equal to

quantized quadrature components of the signals are shifted in phase to the backward direction by an angle

where 2 ^r is the number of quantization levels of the angle 2π, r = 2, 3, ... are integers, F _D is the Doppler frequency determined by the relation

Where V is the own speed of the carrier of the radar station, ψ _A ,

- angles that determine the direction of the axis of the antenna pattern in azimuth and elevation relative to the direction of the vector V, and change the compression ratio N ₁ (R) and the quantization level U (N ₁ ) of the combined compressed signals from targets at ranges R <R ₀ , in accordance with the relations

,

Where

- the whole part of

- monotonically increasing function

2. A radar station containing a serially connected synchronizer, a frequency tuner, an exciter, a phase manipulator, a power amplifier, an antenna switch and an antenna, a serially connected high-frequency amplifier, a mixer, an intermediate frequency amplifier, a phase detector unit, a video amplifier unit and an amplitude quantizer unit, as well as two identical tunable discrete filters, the outputs of which are connected respectively to the first and second inputs of the quadrature combining unit, the output otorogo connected to the first input of the information processing device, and the drive unit of the antenna, the output of which is connected by kinematic connection with the antenna, and the input of the high-frequency amplifier is connected to the third arm of the antenna switch, the heterodyne input of the mixer and the voltage input of the reference frequency of the phase detector unit are connected to the corresponding outputs of the pathogen , the first output of anticipatory pulses of the synchronizer is also connected to the first input of the code generator, the second input of which, combined with the control inputs using tunable discrete filter codes, it is connected to the output of bursts of clock pulses of the synchronizer frequency, and the output is connected to the inputs of the phase manipulator code and tunable discrete filters, the synchronization clock output is connected via a pulse modulator to a power amplifier, and also to the corresponding input of the information processing device, the output pulses of the clock frequency of the synchronizer is connected to the clock inputs of tunable discrete filters and information processing devices, and information input q the latter is connected to the output of the azimuth codes of the antenna drive unit, characterized in that a Doppler frequency calculator, a code-frequency converter and a Doppler frequency compensator are introduced into it, the signal inputs of which are connected to the corresponding outputs of the amplitude quantizer unit, and the first and second the outputs are connected respectively to the signal inputs of the first and second tunable discrete filters, as well as a threshold generating unit, the input of which clock pulses and the corresponding input of the code-frequency converter is connected to the corresponding output of the synchronizer, the information outputs of the azimuth codes and elevation angle of the antenna drive unit and the information output of the frequency tuning unit are connected to the corresponding inputs of the Doppler frequency calculator, the output of the synchronizer zeroing signals is connected to the corresponding inputs of the information processing device, tunable discrete filters and threshold forming unit, the output of which is connected to the fourth input of the processing device and information and outputs command signals to the information processing apparatus from the first to the third coupled, respectively, to control inputs of video amplifier block synchronizer and a power amplifier, wherein the unit of phase detectors comprises 2 ^r-1 of the phase detector, the outputs of which through the corresponding 2 ^rl block video amplifiers video amplifiers are connected to 2 ^r-1 amplitude quantizers of the amplitude quantizer block, and the outputs of the latter are connected to the corresponding signal inputs of the Doppler frequency compensator.

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