JPS5989002A - Comb-line type band-pass filter - Google Patents
Comb-line type band-pass filterInfo
- Publication number
- JPS5989002A JPS5989002A JP19364482A JP19364482A JPS5989002A JP S5989002 A JPS5989002 A JP S5989002A JP 19364482 A JP19364482 A JP 19364482A JP 19364482 A JP19364482 A JP 19364482A JP S5989002 A JPS5989002 A JP S5989002A
- Authority
- JP
- Japan
- Prior art keywords
- conductor
- dielectric
- internal
- internal conductors
- conductors
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
Classifications
-
- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
- H01P1/00—Auxiliary devices
- H01P1/20—Frequency-selective devices, e.g. filters
- H01P1/201—Filters for transverse electromagnetic waves
- H01P1/205—Comb or interdigital filters; Cascaded coaxial cavities
- H01P1/2056—Comb filters or interdigital filters with metallised resonator holes in a dielectric block
Landscapes
- Physics & Mathematics (AREA)
- Electromagnetism (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
Description
本発明は、マイクロ波用コムライン型帯域通過ろ波器に
関するものである。(以下、帯域通過ろ波器をBPF
と略記する。)
従来のコムライン型BPFは、共振器の内部導体の軸長
を管内波長の偽に形成し、電界結合にょって共振器間を
結合するように構成しでいるため、共振器として誘電体
共振器を用いた場合には誘電体の誘電率に応じて段間結
合係数が変るため、その設計製作は容易ではない。
又、従来のコムライン型BPFは、共振器の内部導体を
一列に配設しているため全体の形状が横長となり、共振
器として誘電体共振器を用いて小型化を図ってもその目
的を十分に達することが出来ず、更に有極型BPFを構
成する場合には比較的長い同軸ケーブル又はストリップ
ライン等より成る間接結合回路を必要とするため有極化
には不適である等の欠点を有する。
本発明は、段間結合を磁界結合によって行うように構成
することにより共振器として誘電体共振器を用いた場合
でも誘電率の変化によって段間結合係数に変化を生ずる
ことなく、シたがって設計製作が容易で、又、共振器を
フの字型に配設することにより全体を極めて小型化し、
有極型BPFを構成するに当って間接結合回路を極めて
簡潔小型ならしめることが出来、この点からも組立調整
が容易で、電気的特性の良好なコムライン型BPFを実
現することを目的とする。
第1図は本発明の一実施例を示す断面図(第2図のC−
C断面図)、第2図は第1図のA−A断面図、第3図は
第1図のB−B断面図で、各図tこおいて、1は共振器
の外部導体を兼ねる筐体、2は導体より成る隔壁、3は
筐体1内に充てんした誘電体で、倒木ばチタン酸バリウ
ム磁器又はチタン酸マグネシウム磁器等より成る。4I
ないし4ル(nはろ波器の次数)は誘電体3に穿った
孔隙、51ないし51は半同軸型共振器を形成する内部
導体で、電気長で管内波長λ、の4の軸長を有し、銀又
は銅等を蒸着等の手段によって孔隙4I な(\し4
ユの各内壁面に付着せしめた金属被膜より成る。金属被
膜の代りに棒状導体を孔隙4I ないし4n、内に嵌入
して内部導体を形成してもよい。6は半同軸型共振器の
外部導体で、誘電体3の開放面(空間部7を介して筐体
1の土壁と対向する誘電体3の上面)を除いた表面に銀
又はtl!1等の金属被膜を蒸着等の手段により付着せ
しめて形成する。
金属被膜の代りに導体板を付着上しめてもよく、又、内
部導体5I ないし51の各下端部(短絡部ンと筐体
l との電気的接続を確保し得る場合には金属被膜又は
導体板を付着することなく、筐体1及び隔壁2を外部導
体としてもよい。更に筐体1を省いて誘電体3の表面に
付着せしめられて外部導体6を形成する金属被膜又は導
体板が最外側に露出するように構成しでもよい。この場
合にも隔壁2は、これを省くことなく残置せしめる。8
o及び8?I+1 は入出力同軸端子、9o及び971
4 (は入出力結合素子で、内部導体51 ないし5n
と同様、誘電体3に穿った孔隙4o及び41→fの各内
壁面に付着せしめた金属被膜又は孔隙4o及び471や
、内に嵌入した棒状導体より成り、何れのi4台もそめ
軸長は内部導体51 ないし5rLと同様である。
第4図は上記本発明BPF’の等価口路図で、T。
及びTnう1 は入出力端子、R+ ないしR,は内部
導体5I ないし57L及び外部導体6によって形成
される共4辰回S各、Me、r %M+、a % 1争
・・・M(n−+)、a % My、1yi*+)は磁
界結合度である。
本発明BPF’は内部導体5I ないし51の各軸長を
電気長で管内波長の4に形成し、磁界結合によって段間
結合を行うように構成しであるため共振器の内部導体部
分においてはTEMモード電磁波の結合伝ばん理論を適
用し、内部4体間の誘電体部分においてはH11モード
電磁波の結合伝ばん理論を当て嵌めることが出来るから
設計を極めて容易ならしめ得る。
即ち、筐体1内に充てんしである誘電体の材質に関係な
くその透磁率μは空気の場合と全く同じ1であるから段
間結合回路の設計理論は内部導体の周りが空気である場
合と全く同じこととなる。
但し、内部導体の軸長はε−*(εは誘電体の誘電率)
に比例し、Huモード電磁波の管内遮断波長入。は6丁
に比例する。つまり本発明BPFにおいでは内部導体の
周りに誘電体を設けたことによる影響は、内部導体の軸
長とHuモード電磁波の管内遮断波長に及ぶのみである
から設計製作は従来の誘電体共振器を用いたコムライン
型BPFに比し遥かに容易である。
まず本発明BPFにおいては入出力結合素子9σ及び9
□1、半同軸型共振器を形成する内部41本51ないし
5礼の各軸長ノdを、
J、=’x ・・山・・・(1)とすること
によりTEMモード電磁波の共振を可能ならしめるもの
であるが、上式において島はTEMモード電磁波信号の
自由空間における波長(mm)で、共振周波数fa と
の間に、
入。 = 300/ f、(CIHz) = 30
0000/ f、(MHz)なる関係を有すること周知
の通りである。
半同軸型共振器の内部導体に共振電流が流れると、内部
導体の軸方向と直角方向に磁界を生じ、この磁界の中、
筐体1の横幅方向に平行な磁界成分を廟すると共に内部
導体の軸方向に平行な電界成分を有する旧モード電磁波
の磁界成分によって隣接する内部導体相互の結合が行わ
れ、このHnモード電磁波の管内遮断波長入、は、第3
図示のように筐体1の横幅をH(mm)とすると、^o
=2Hu ・・・・・・・・(3)とな
る。
又、入出力結合素子9o及び内部導体51 開における
H++モード電磁波の磁界減衰量L+84+ % 5+
及び5、間における磁界減衰量LH++! ・・・・
・・5TL−1及び5n開における磁界減衰量LHt*
−1J、% 、内部導体51及び入出力結合素子9□0
間におけるH++モード電磁波の磁界減衰量ら)(y
+、(u++1をまとめてL l”l K、(Kや+1
(Kは0ないしn)と表わし、入出力結合素子9.及び
内部導体51開の中心軸開隔C6,l内部導体5I 及
び5コの中心軸開隔C+、2 % 52及び5」の中心
軸部Il#IC2,J1・・・・・・5社−1及び5n
の中心軸開隔C(n:、1.n、内部導体5It及び入
出力結合素子9ルl の中心軸開隔C7L、(71+I
)をまとめてCに、〔に十〇と表わすと、各段の磁界減
衰量””+(K+Qと入出力結合素子及び内部導体の中
心軸開隔Cに、(ハ中すの関係は、で表わされる。
上式は理論的に求めた計算式であるが、本発明者が基礎
実験を重ねた結果によれば内部導体5Iないし51の外
径に応じで等偏磁界面に誤差を生じ減
で居り、次の実験式から磁界≠衰量を求める方が正確で
ある。
d:内部導体の外径
したがって、
次に許容通過帯域幅Bwrの上限周波数をf、い下限周
波数をfThe present invention relates to a combline bandpass filter for microwaves. (Hereinafter, the bandpass filter is referred to as BPF
It is abbreviated as ) Conventional combline type BPFs are configured so that the axial length of the internal conductor of the resonator is equal to the wavelength in the tube, and the resonators are coupled by electric field coupling. When a resonator is used, the interstage coupling coefficient changes depending on the dielectric constant of the dielectric material, so it is not easy to design and manufacture the resonator. In addition, in conventional comb-line BPFs, the internal conductors of the resonators are arranged in a row, making the overall shape elongated, and even if a dielectric resonator is used as the resonator for miniaturization, the purpose of the BPF cannot be achieved. Furthermore, when configuring a polarized BPF, an indirect coupling circuit consisting of a relatively long coaxial cable or strip line is required, making it unsuitable for polarization. have The present invention has a structure in which interstage coupling is performed by magnetic field coupling, so that even when a dielectric resonator is used as a resonator, the interstage coupling coefficient does not change due to a change in dielectric constant. It is easy to manufacture, and by arranging the resonator in a fold-back shape, the overall size is extremely small.
When constructing a polarized BPF, the indirect coupling circuit can be made extremely simple and compact, and from this point of view, the aim is to realize a combline type BPF that is easy to assemble and adjust and has good electrical characteristics. do. Figure 1 is a sectional view showing an embodiment of the present invention (C--C in Figure 2).
Figure 2 is a cross-sectional view taken along line A-A in Figure 1, and Figure 3 is a cross-sectional view taken along line B-B in Figure 1. In each figure, 1 also serves as the external conductor of the resonator. The casing, 2, is a partition made of a conductor, and 3 is a dielectric filled in the casing 1, which is made of barium titanate porcelain, magnesium titanate porcelain, or the like. 4I
4 through 4 (n is the order of the filter) are holes drilled in the dielectric 3, and 51 through 51 are internal conductors forming a semi-coaxial resonator, each having an axial length of 4, where the electrical length is the tube wavelength λ. Then, the pores 4I (\shi4
It consists of a metal coating attached to each inner wall surface of the unit. Instead of the metal coating, a rod-shaped conductor may be inserted into the holes 4I to 4n to form the inner conductor. Reference numeral 6 denotes an outer conductor of the semi-coaxial resonator, which is coated with silver or tl! It is formed by depositing a first grade metal coating by means such as vapor deposition. A conductive plate may be attached instead of the metal coating, or a metal coating or a conductive plate may be attached to each lower end of the internal conductors 5I to 51 (if electrical connection between the short-circuit part and the housing l can be ensured). The casing 1 and the partition wall 2 may be used as external conductors without adhering to the outer conductor 6.Furthermore, the casing 1 is omitted and the metal coating or conductive plate that is attached to the surface of the dielectric 3 to form the external conductor 6 is the outermost conductor. The partition wall 2 may also be configured so as to be exposed to the outside.In this case as well, the partition wall 2 is left without being omitted.8
o and 8? I+1 is input/output coaxial terminal, 9o and 971
4 (input/output coupling element, internal conductor 51 to 5n
Similarly, the holes 4o and 41→f formed in the dielectric 3 are made of metal coatings attached to the inner walls of the holes 4o and 471, and the rod-shaped conductors fitted inside the holes 4o and 471. This is similar to the internal conductors 51 to 5rL. FIG. 4 is an equivalent path diagram of the BPF' of the present invention. and Tn1 are input/output terminals, R+ to R, are four circular turns S each formed by the inner conductors 5I to 57L and the outer conductor 6, Me, r %M+, a % 1 contest...M(n −+), a % My, 1yi**) is the degree of magnetic field coupling. The BPF' of the present invention is configured such that the axial length of each of the internal conductors 5I to 51 is electrically lengthed to 4, which is the tube wavelength, and the interstage coupling is performed by magnetic field coupling. The coupled propagation theory of mode electromagnetic waves can be applied, and the coupled propagation theory of H11 mode electromagnetic waves can be applied to the dielectric portion between the four internal bodies, making the design extremely easy. In other words, regardless of the material of the dielectric filled in the housing 1, its magnetic permeability μ is 1, which is exactly the same as that of air, so the design theory of the interstage coupling circuit is based on the case where the internal conductor is surrounded by air. It's exactly the same thing. However, the axial length of the internal conductor is ε-* (ε is the dielectric constant of the dielectric)
The cutoff wavelength of Hu mode electromagnetic waves in the tube is proportional to . is proportional to 6 guns. In other words, in the BPF of the present invention, the effect of providing a dielectric material around the internal conductor is only on the axial length of the internal conductor and the cut-off wavelength of the Hu mode electromagnetic wave in the tube, so the design and manufacture can be performed using a conventional dielectric resonator. This is much easier than the comline type BPF used. First, in the BPF of the present invention, the input/output coupling elements 9σ and 9
□1. The resonance of the TEM mode electromagnetic waves can be achieved by setting the axial length no. This is possible, but in the above equation, the island is the wavelength (mm) in free space of the TEM mode electromagnetic wave signal, and it is between the resonant frequency fa. = 300/f, (CIHz) = 30
As is well known, the relationship is 0000/f, (MHz). When a resonant current flows through the internal conductor of a semi-coaxial resonator, a magnetic field is generated in a direction perpendicular to the axial direction of the internal conductor, and within this magnetic field,
The adjacent internal conductors are coupled to each other by the magnetic field component of the old mode electromagnetic wave, which has a magnetic field component parallel to the width direction of the housing 1 and an electric field component parallel to the axial direction of the internal conductor. The cut-off wavelength in the tube is the third
As shown in the figure, if the width of the housing 1 is H (mm), then ^o
=2Hu (3). In addition, the magnetic field attenuation amount of H++ mode electromagnetic waves when input/output coupling element 9o and internal conductor 51 are open L+84+ % 5+
and 5, magnetic field attenuation amount LH++!・・・・・・
...Magnetic field attenuation LHt* at 5TL-1 and 5n open
-1J, %, internal conductor 51 and input/output coupling element 9□0
magnetic field attenuation of H++ mode electromagnetic waves between
+, (u++1 together L l”l K, (K and +1
(K is 0 to n), and the input/output coupling element 9. and center axis gap C6 of inner conductor 51, l inner conductor 5I and center axis gap C+ of 5 pieces, 2% 52 and 5'' center axis part Il #IC2, J1...5 companies - 1 and 5n
Center axis spacing C(n:, 1.n, center axis spacing C7L of internal conductor 5It and input/output coupling element 9L, (71+I
) are collectively represented as C and [ as 10], then the magnetic field attenuation amount of each stage "" + (K + Q and the central axis gap C of the input/output coupling element and the internal conductor, (C) is expressed as The above formula is a calculation formula obtained theoretically, but according to the results of repeated basic experiments by the present inventor, an error occurs in the uniform polarized magnetic interface depending on the outer diameter of the internal conductors 5I to 51. Therefore, it is more accurate to find magnetic field ≠ attenuation from the following experimental formula. d: Outer diameter of the internal conductor Therefore, Next, the upper limit frequency of the allowable passband width Bwr is f, and the lower limit frequency is f.
【 で表わすと、共振周波数(中心周波数)f
、及びBwrは、
t−= (ft −fn )’
・・ ・” ・ (7)swr =
r、−fL−・−・−(s)で表わされる。
又、内部導体5I ないし賄の各々に関連する回路素
子値をgl ないしgrLとすると、入出力結合素子9
゜と内部導体51 間の磁界結合係数M01.及び内部
導体51と入出力結合素子9?t41間の磁界結合係数
MTI、(7!+りは−
、22BwrZ。
Ma、+ =Mn<n+1r−= (−・) −一・・
・・(9)g+ gh f=Zat
za:ろ波器の特性インピーダンス
Zal :誘電体共振器の特性インピーダンスで表わ
され、内部導体5I と5−間の磁界結合係数M+、
コ%内gl導体5コと5J 間の磁界結合係数M)、J
・・・・・・内部導体5?iイ と5n間の磁界結合係
数M(7(−r)、tLをまとめてMX、[3+1)と
表すと、
で表わされる。
(4)式の磁界減衰量LH%、(N++l t (10
) 式’)hji界結合係数MK、(へやりで表わすと
、
Lk’s、(Her) ” 20ρOgMH,tK◆
l)となる。
本発明BPFの伝送特性を所要の特性とするためには従
来一般に用いられている手法と同様に、所化
要の伝送特性と同様の特性を有する基準A低域通過ろ波
器の回路素子値を求め、この回路素子値と本発明BPF
’の各対応する回路素子値とが一致するように、例えば
内部導体の各中心軸間隔を調整することにより目的を達
することが出来る。
例えば本発明BPFの伝送特性を通過帯域がチェビシェ
フ特性で、減衰域がワグナ特性を呈するような伝送特性
ならしめようとする場合には、まず設計上許容される電
圧定在波比(VSWR)をSとおいて次式から通過帯域
内の許容リップルLarを求める。
次に第5図に等価回路図を、第6図(横軸:@過周波数
f、縦軸:減衰量ATT 、 ft :管内遮断周波数
)に伝送特性曲線図を示すようなチェビシェフ型基準化
低域通過ろ波器における通過帯域内の許容リップルを
(12)式から求めると共に、その次数を本発明BPF
の次数nと一致せしめて回路素子値g1 ないしgt
tを(13)及び(14)式から求メる。
上式において、
(11)式の許容通過帯域幅Bwr及び中心周波数f、
に所要値を代入すると共に、(13)及び(14)式か
ら求めた回路素子値g+ ないしglを(11)式に
代入して各段間の磁界減衰量LH×、C式今−)を求め
、この”)’++11++1の値を(6)式に代入する
と共に、筐体1の横幅H1内部導体の外径d、誘電体3
の誘電率ε及び波長洩の各設定値を(6)式に代入して
各隣接内部導体毎の中心軸間隔Cに、(K+、l)を求
め、内部導体51 ないし5nの各中心軸間隔を(6
)式から得られたC〆、(K411に一致せしめること
により本発明BPF’の伝送特性を前記所要の伝送特性
とすることが出来、他の伝送特性を呈せしめようとする
場合にも同様の手法によって目的を達することが出来る
。
前記のようにして通過帯域をチェとシェフ特性、減衰域
をワクナ特性となした本発明BPFの伝送特性は次式で
表わされる。
上式におしゴしは減衰量、7.<x>はチェビシェフの
多項式で、x(I の場合、
7;(x) =cos (n cos−’x )x>I
の場合、
7、、(x) =cosh (n aoeh−’
x’)fo f fI+
X = −(−−−)
Bw r fa f
第7図は前記特性を有する本発明BPFの伝送特性を示
す曲線図で、横軸は伝送周波数f (MHz)、縦軸は
減衰量L (dB)である。
以上は入出力結合素子として内部導体5.ないし5□と
同様構造の結合導体9a 及び9n+Iを用い、導体九
と5+ 間及び導体5nと97t+、開の各変成器結
合によって入出力結合を行うようC二構成した場合を例
示したが、第8図及び第9図に要部断面図を示すように
入出力結合素子として内部導体5f及び5nの各頂部と
各別に対向する電柵板9’a 及び9階、を用い、内部
導体51 及び5TLとの開に形成される容量を介して
入出力結合を行うように構成しでも本発明を実施するこ
とが出来る。この場合には入出力結合容量Cctr戸及
びCcn、tn+りが初段及び終段の共振回路に付加さ
れるため内部導体51X、らt : Cco、+の容量
リアクタンスXCyr(w++l ’ CerL(n+
+1の容量リアクタンスとなり、他の内部導体の軸長に
比し短縮す仔るから内gII導体5I 及び5…の周り
の誘電体の高がも他の内部導体の周りにおける誘電体の
高谷より低く形成し得るから、この低い部分に入出力結
合素子9λ及び9’l+Iを配設することが出来る。し
たがって前実施例のように素子9.及び髄+、を設ける
ために筐体1の寸法を大ならしめる必要なく、全体を小
型に構成することが出来る。
尚、入出力結合素子9シ及び9石lの容量リアクタンス
XcQ、1 y Xca、(n・鴫・喀) 飄容量Cc
o、+ NCcn、(3++l、回路素子値g+ %
gn %許容通過帯域幅Bwr 、中心周波数f。、中
心角周波数ω2、ろ波器の特性インピーダンスZa−誘
電体共振器の特性インピーダンスzaε の開には次式
の関係が成立する。
入出力結合素子外及び9’ro+を電極板を以て形成す
る代りにコンデンサ2以て形成してもよい。
第10図は、第8図及び第9図に示した実施例の等価回
路図で、入出力結合容MCco、r 及びCcn、tn
+r+以外の符号は第4図と同様である。
この実施例においでは入出力結合を容量結合を以て行う
ように構成した点が前実施例と異なるのみで、他の構成
及び作動は前実施例と全く同様である。
以上側れの実施例においても誘電体3を除いて空気と置
換えても本発明を実施することが出来、この場合にはε
=1とおくことにより前記各計算式を適用することが出
来る。
型
次に、第11図は本発明BPFを有極牛に構成した一例
を示す断面図で、次数nを6に選ぶと共に入出力結合を
結合容量を介して行うように構成した場合を示しである
。入出力結合素子は示しでいない。、同図において、I
O+、& は間接結合素子で、例えばループより成り、
隔壁2に穿った貫通孔11の内壁との間を絶縁を保って
貫通孔■に挿通支持され、内部導体51 と5&間を
磁界結合により間接結合する。l Oa、tもまた間接
結合素子で、例えば一端が内部導体5コとの間に結合容
量を形成し、他端が内部導体島との間に結合容量を形成
する容量素子より成り、隔壁2に穿った貫通孔12に挿
通し、隔壁2との開を絶縁を保って取付けである。他の
符号及び構成は前記各実施例と同様である。
第12図はその等価回路図で、’M1.6は素子10+
、cによる磁界結合度、C】、ダは素子10p、すによ
る結合容量、T、及びTり は入力端子、Cce、r
及びCcA、り は入出力結合容量%RI ないし
R6は共振回路、MbaないしMり、6は磁界結合度で
ある。
第11図における内部導体5I ないし56及び外部
導体6より成る半同軸型共振回路R1ないしR6(第1
2図)の中、縦続接続関係において隣接する共振回路間
には第13図に要部の等価口路を示すような変成器結合
回路が形成される。第13図にオイテ、LIRは共振回
路におけるインダクタンス分、ORは共振回路における
容量分、MRは相互インダクタンスである。第13図に
宗した回路は第14図に示すように共振作用を行う回路
部分JT−と、位相回路を構成する回路部分jpに分け
ることが出来る。第14図においてMMは磁気相互イン
ダクタンス%’CMは等価位相回路の容量である。
位相回路を構成する回路部分ノφを[F]マトリクスを
用いて理論計算すると、
jωsMMミjω・CM ・・・・・・(19)
・・・・(20)
但し、
1−ω。MMOM=0
とする。
位相回路を構成する回路部分すの特性インピーダンスを
2゜、特性アドミタンスをYoaとすると、Zaa
−ω、MM=1/ω。C’M ++++・・(2
1)Yoa =ωIC□・・・・・・(22)位相回
路を構成する回路部分1.と等価な特性インピーダンス
Zaa を有し、電気長で入a/4の仮想ケーブルの
等測的な長さを1とし、m=2π/入を及びZea =
Yoa :I とすると、となり、位相回路を構成
する回路部分1pは+90’の位相回路となり、第12
図の回路は第15図に示す等価回路となる。同図におい
てP、ないしP、7は+90゛の位相回路で、他の符号
は第12図と同様である。
第15図の回路を伝送する信号の位相関係、即ち、共振
周波数の信号と減衰極を生ずべき周波数の信号(以下、
減衰極信号と略記する)の位相関係を検討すると、第1
5図のA点及びA′点点間主回路を伝送する間に共振周
波数の信号に対して減衰極信号は、共振回i?8Ra
ないしR&において士90°×4=±3600 の位
相差を生ずると共に位相回路pJ ないしp、においで
+9σX3=+270” の位相差を生じ、結局+2
700 の位相差を生ずる。
一方、A及びに点から分岐する間接結合回路10a、L
を伝送する信号は一90″″の位相差を生ずるから、主
回路を伝送してA′点に達した減衰極信号と間接結合回
路を介してA′点に達した減衰極信号との間にはl80
1の位相差を生ずる。したがって減衰極信号がA−A’
間の主回路を伝送する間に生ずる減衰量と、A及びに点
から分岐する間接結合回路の結合量とが等しくなるよう
に結合容量c2.&の大きさを定めることにより減衰極
信号はに点において理論的には完全に打消し合って減衰
極を生ずる。
同様に第15図におけるB点及びB′点の間の主回路を
伝送する間に共振周波数の信号に対して減衰極信号は、
共振回路R1ないしRhにおいて+96×6=±540
1 の位相差【生ずると共に位相回路P1ないしR6
において+qd×s = + aso’ の位相差を
生じ、総合的には一90°の位相差を生ずる。
一方、BELUB′点から分岐されたループより成る間
接結合回路101.6を伝送する信号は+90’の位相
差を生ずるから、キロi!8を伝送してB′点に到った
減衰極信号と間接結合回路を介してB′点に達した減衰
極信号との間には180°の位相差を生ずる。
したがって、減衰極信号がB−8’開の主回路を伝送す
る間に生ずる減衰量と、B及びB′点から分岐する間接
結合回路の結合量とが等しくなるように結合度M +4
を足りことにより減衰極信号の周波数に減衰極を生
ずる。
上記有極型本発明BPPの伝送特性は次式から理論的に
求めることが出来る。
上記実施例のように次数nが6、即ち偶数の場nが奇数
の場合は、
・・・・(27)
娼=(1−司) ・・・・・・(2C)
(26)式におけるRrは実数部をとるの意、(27)
式におけるエヨは虚数部をとるの意、(28)式におけ
るf−i は中心周波数f、と減衰極を生ずる周波数の
差で、f−1′、ω とfすることにより一般の通過帯
域チェビシェフ特性、減衰域ワグナ特性のBPFとする
ことが出来る。
以上は共振回i!8RaとR5間を容量結合素子を以て
間接結合した場合を例示したが、この他R+とRe間又
はR」とR6間或はRrとR6間のように2個又はその
整数倍の個数の共振回路を隔てた共振回路相互を容量結
合素子を以て間接結合することによって目的を達するこ
とが出来る。又、磁界結合素子を以て間接結合を行う場
合には4個又はその整数倍の個数の共振回路を隔てた共
振回路相互を開4妾結合することにより減衰極を生せし
めることが出来る。
第11図はる波器の次数が6で、容量結合素子及び磁界
結合素子より成る間接結合回路を各1組設けた場合を例
示したが、ろ波器の次数を任意に選び得ると同様、間接
結合回路の数も任意にl!!ぶことが出来、又、複数個
の間接結合回路を設けた場合にはすべての間接結合回路
を容量結合素子によって形成するか、すべての間接結合
口路を磁界結合素子を以て形成するか、又は間接結合回
路の一部適宜数を容量結合素子を以て形成し、他の間接
結合回路を磁界結合素子を以て形成するようにしても本
発明を実施することが出来る。
以上の説明から明らかなように、本発明コムライン型B
PPは段間結合を磁界結合を以て行うように構成するこ
とにより設計製作を容易ならしめ、又、共振器をコの字
型に配設することにより全体を極めて小型化し得ると共
に、非縦続接続関係において隔壁を隔てて隣接する共振
器相互を間接結合することにより有極型伝送特性となし
得るので、間接結合素子を極めて簡潔小型化することが
出来、この点からも組立調整が容易となり良好な電気的
特性を得ることが出来る等の特長を有するものでその効
果甚だ大である。[ Resonant frequency (center frequency) f
, and Bwr are t-= (ft-fn)'
・・” ・ (7) swr =
r, −fL−·−·−(s). Further, if the circuit element values related to each of the internal conductors 5I and 5I are gl to grL, then the input/output coupling element 9
゜ and the internal conductor 51 magnetic field coupling coefficient M01. and the internal conductor 51 and the input/output coupling element 9? Magnetic field coupling coefficient MTI during t41, (7! + Riha-, 22BwrZ. Ma, + = Mn<n+1r-= (-・) -1...
...(9) g+gh f=Zat za: Characteristic impedance of the filter Zal: Represented by the characteristic impedance of the dielectric resonator, magnetic field coupling coefficient M+ between the internal conductors 5I and 5-,
Coefficient of magnetic field coupling between 5 gl conductors and 5J in % M), J
...Inner conductor 5? When the magnetic field coupling coefficient M(7(-r), tL between i and 5n is collectively expressed as MX, [3+1), it is expressed as follows. Magnetic field attenuation LH% of equation (4), (N++l t (10
) Equation') hji field coupling coefficient MK, (expressed in Heyari, Lk's, (Her) ” 20ρOgMH,tK◆
l). In order to set the transmission characteristics of the BPF of the present invention to the required characteristics, the circuit element values of the standard A low-pass filter having the same characteristics as the required transmission characteristics are used, as well as the conventionally generally used method. This circuit element value and the BPF of the present invention
The purpose can be achieved by, for example, adjusting the distance between the center axes of the internal conductors so that the values of the corresponding circuit elements of ' coincide with each other. For example, when trying to make the transmission characteristics of the BPF of the present invention such that the pass band is a Chebyshev characteristic and the attenuation region is a Wagner characteristic, first, the voltage standing wave ratio (VSWR) allowed in the design is determined. Let S be the allowable ripple Lar within the passband from the following equation. Next, the Chebyshev-type normalized low voltage is shown as an equivalent circuit diagram in Fig. 5 and a transmission characteristic curve diagram in Fig. 6 (horizontal axis: @ overfrequency f, vertical axis: attenuation amount ATT, ft: pipe cutoff frequency). The allowable ripple in the passband of a bandpass filter is
(12), and its order is determined by the BPF of the present invention.
The circuit element value g1 to gt is made to match the order n of
Calculate t from equations (13) and (14). In the above equation, the allowable passband width Bwr and center frequency f of equation (11),
Substituting the required value into , and substituting the circuit element value g+ or gl obtained from equations (13) and (14) into equation (11), the magnetic field attenuation between each stage LH x, C equation now -) is obtained. Substitute the value of ")'++11++1 into equation (6), and also calculate the width H1 of the housing 1, the outer diameter d of the internal conductor, and the dielectric material 3.
By substituting the set values of permittivity ε and wavelength leakage into equation (6), (K+, l) is determined for the center axis spacing C of each adjacent internal conductor, and each center axis spacing of internal conductors 51 to 5n is calculated. (6
By making C〆 obtained from the equation ( The purpose can be achieved by this method. The transmission characteristics of the BPF of the present invention, in which the pass band is made to have Che and Chef characteristics and the attenuation region is made to be Wakuna characteristics, as described above, are expressed by the following equation. is the attenuation, 7.<x> is the Chebyshev polynomial, and if x(I, 7; (x) = cos (n cos-'x )x>I
If 7,, (x) = cosh (naoeh-'
x') fo f fI+ is the attenuation amount L (dB). The above is an internal conductor 5. as an input/output coupling element. The case where the C2 configuration is illustrated is that the coupled conductors 9a and 9n+I having the same structure as 5□ to 5□ are used, and the input/output coupling is performed between the conductors 9 and 5+, and between the conductors 5n and 97t+, and the open transformer coupling. As shown in the cross-sectional views of main parts in FIGS. 8 and 9, electric fence plates 9'a and 9th floor, which respectively face the tops of the internal conductors 5f and 5n, are used as input/output coupling elements, and the internal conductors 51 and The present invention can also be implemented by configuring input/output coupling to be performed via a capacitance formed open to the 5TL. In this case, since the input/output coupling capacitance Cctr and Ccn, tn+ are added to the first and final stage resonant circuits, the capacitive reactance XCyr(w++l' CerL(n+
Since the capacitive reactance becomes +1 and is shortened compared to the axial length of the other internal conductors, the height of the dielectric material around the inner gII conductors 5I and 5... is also lower than the height of the dielectric material around the other internal conductors. Therefore, the input/output coupling elements 9λ and 9'l+I can be arranged in this low portion. Therefore, as in the previous embodiment, element 9. There is no need to increase the size of the casing 1 to provide the casing 1 and the lining +, and the entire structure can be made compact. In addition, the capacitance reactance XcQ, 1 y
o, + NCcn, (3++l, circuit element value g+%
gn % allowable passband width Bwr, center frequency f. , the center angular frequency ω2, the characteristic impedance Za of the filter and the characteristic impedance zaε of the dielectric resonator. The outside of the input/output coupling element and 9'ro+ may be formed using the capacitor 2 instead of using the electrode plate. FIG. 10 is an equivalent circuit diagram of the embodiment shown in FIGS. 8 and 9, in which the input/output coupling capacitors MCco,r and Ccn,tn
The symbols other than +r+ are the same as in FIG. 4. This embodiment differs from the previous embodiment only in that the input/output coupling is performed by capacitive coupling, and the other configurations and operations are completely the same as the previous embodiment. Even in the above-mentioned embodiments on the side, the present invention can be carried out even if the dielectric 3 is removed and replaced with air, and in this case, ε
By setting =1, each of the above calculation formulas can be applied. Next, FIG. 11 is a cross-sectional view showing an example of the BPF of the present invention configured in a polarized form, in which the order n is chosen to be 6 and input/output coupling is performed via a coupling capacitance. be. Input/output coupling elements are not shown. , in the same figure, I
O+, & are indirect coupling elements, for example, consisting of a loop,
It is inserted and supported in the through hole 2 while maintaining insulation between the inner wall of the through hole 11 bored in the partition wall 2, and the internal conductors 51 and 5& are indirectly coupled by magnetic field coupling. l Oa, t are also indirect coupling elements, for example, one end is a capacitive element that forms a coupling capacitance with the internal conductor 5, the other end is a capacitive element that forms a coupling capacitance with the internal conductor island, and the partition wall 2 It is inserted into a through hole 12 drilled in the hole 12, and installed while maintaining insulation from the partition wall 2. Other symbols and configurations are the same as in each of the above embodiments. Figure 12 is its equivalent circuit diagram, where 'M1.6 is element 10+
, the degree of magnetic coupling due to c, C], da is the coupling capacitance due to the element 10p, T, and Tri are the input terminals, Cce, r
and CcA, ri is the input/output coupling capacity %RI to R6 is the resonance circuit, Mba to M, and 6 is the magnetic field coupling degree. Semi-coaxial resonant circuits R1 to R6 (first
In FIG. 2), a transformer coupling circuit is formed between adjacent resonant circuits in a cascade connection, as shown in FIG. 13, the equivalent circuit of which is the main part. In FIG. 13, LIR is the inductance in the resonant circuit, OR is the capacitance in the resonant circuit, and MR is the mutual inductance. A circuit similar to that shown in FIG. 13 can be divided into a circuit portion JT- which performs a resonance action and a circuit portion JP which constitutes a phase circuit, as shown in FIG. In FIG. 14, MM is the magnetic mutual inductance %'CM is the capacitance of the equivalent phase circuit. Theoretically calculating the circuit portion φ that constitutes the phase circuit using the [F] matrix, jωsMMmijω・CM (19)
...(20) However, 1-ω. Let MMOM=0. If the characteristic impedance of the circuit part constituting the phase circuit is 2°, and the characteristic admittance is Yoa, then Zaa
−ω, MM=1/ω. C'M +++++...(2
1) Yoa = ωIC□ (22) Circuit portion configuring the phase circuit 1. Let us assume that the isometric length of a hypothetical cable with characteristic impedance Zaa equivalent to , and electrical length of input a/4 is 1, m = 2π/in and Zea =
If Yoa :I, then the circuit portion 1p constituting the phase circuit becomes a +90' phase circuit, and the 12th
The circuit shown in the figure becomes an equivalent circuit shown in FIG. In the figure, P, to P, 7 are +90° phase circuits, and other symbols are the same as in FIG. 12. The phase relationship of the signals transmitted through the circuit of FIG.
When considering the phase relationship of the attenuated pole signal (abbreviated as attenuation pole signal), the first
During transmission through the main circuit between points A and A' in Figure 5, the attenuation pole signal for the signal at the resonant frequency is at the resonant turn i? 8Ra
A phase difference of +90° x 4 = ±3600 is produced in R&, and a phase difference of +9σ
This results in a phase difference of 700°. On the other hand, indirect coupling circuits 10a and L branching from points A and
Since the signal transmitted through the main circuit produces a phase difference of -90'', there is For l80
This produces a phase difference of 1. Therefore, the attenuation pole signal is AA'
The coupling capacitance c2. By determining the magnitude of &, the attenuation pole signals theoretically cancel each other out completely at the point to produce an attenuation pole. Similarly, the attenuation pole signal for the signal at the resonant frequency during transmission through the main circuit between points B and B' in FIG.
+96×6=±540 in resonant circuit R1 to Rh
1 phase difference [occurs and the phase circuits P1 to R6
A phase difference of +qd×s=+aso' is generated, and a total phase difference of -90° is generated. On the other hand, since the signal transmitted through the indirect coupling circuit 101.6 consisting of a loop branched from the BELUB' point produces a phase difference of +90', the kilo i! A phase difference of 180° is generated between the attenuated pole signal that has reached point B' after transmitting 8 and the attenuated pole signal that has reached point B' via the indirect coupling circuit. Therefore, the degree of coupling M +4 is set so that the amount of attenuation that occurs while the attenuated pole signal is transmitted through the B-8' open main circuit is equal to the amount of coupling of the indirect coupling circuit branching from points B and B'.
By adding , an attenuation pole is generated at the frequency of the attenuation pole signal. The transmission characteristics of the polarized BPP of the present invention can be theoretically determined from the following equation. As in the above example, if the order n is 6, that is, if the order is an even number, then n is an odd number, ...(27) Pro = (1 - Tsukasa) ...... (2C)
Rr in equation (26) means taking the real part, (27)
Eyo in the equation means taking the imaginary part, f-i in equation (28) is the difference between the center frequency f and the frequency that produces the attenuation pole, and by setting f-1', ω and f, the general passband Chebyshev It can be a BPF with a characteristic and an attenuation region Wagner characteristic. The above is the resonance time i! 8Ra and R5 are indirectly coupled using a capacitive coupling element, but in addition, two or an integral multiple of two resonant circuits, such as between R+ and Re, between R' and R6, or between Rr and R6, are used. This objective can be achieved by indirectly coupling the resonant circuits separated by a capacitive coupling element. Further, when indirect coupling is performed using a magnetic field coupling element, an attenuation pole can be generated by open-coupling four or an integer multiple of four resonance circuits to each other. Fig. 11 shows an example in which the order of the filter is 6 and one set of indirect coupling circuits each consisting of a capacitive coupling element and a magnetic coupling element is provided, but the order of the filter can be arbitrarily selected. The number of indirect coupling circuits can also be arbitrarily l! ! In addition, when multiple indirect coupling circuits are provided, all indirect coupling circuits are formed by capacitive coupling elements, all indirect coupling ports are formed by magnetic coupling elements, or indirect coupling circuits are The present invention can also be practiced by forming an appropriate number of some of the coupling circuits using capacitive coupling elements, and forming other indirect coupling circuits using magnetic coupling elements. As is clear from the above explanation, the comline type B of the present invention
PP allows for easy design and manufacture by configuring interstage coupling using magnetic field coupling, and by arranging the resonators in a U-shape, the overall size can be made extremely small. By indirectly coupling adjacent resonators across a partition wall, a polarized transmission characteristic can be achieved, making it possible to make the indirect coupling element extremely simple and compact, and from this point of view as well, assembly and adjustment are easy and good results are achieved. It has features such as being able to obtain electrical characteristics, and its effects are tremendous.
第1図ないし第3図は本発明の一実施例を示す断面図、
第4図はその等価回路図、第5図は基準化低域通過ろ波
器の等価回路の一例を示す図、第6図はその特性曲線図
、第7図は本発明ろ波器の伝送特性の一例を示す図、第
8図及び第9図は本発明の他の実施例を示す断面図、第
10図はその等価回路図、第11図は本発明の他の実施
例を示す図、第12図ないし第15図はその作動説明の
ための等価回路図で、ド筐体、2:隔壁、3:誘電体、
46ないし4m+I :孔隙、5Iないし5rL=内部
4体、6:外部導体、7:空間部、86及び871+1
:入出力同軸端子、9o % 97L++、9;及び
9’?L*(:入出力結合素子、IQ+、4及びl0a
j ’間接結合素子、11及び12:貫通孔、T#及び
T?Iやl二人出力端子、R+ないしR?L:共振回路
、P、ないしPり:位相回路である。
第1図
第2図
第、3図
第4図
第5図
f−1(Ml−IZJ−
第8図
第9図
第10図
第11図1 to 3 are sectional views showing one embodiment of the present invention,
Fig. 4 is its equivalent circuit diagram, Fig. 5 is a diagram showing an example of the equivalent circuit of the normalized low-pass filter, Fig. 6 is its characteristic curve diagram, and Fig. 7 is the transmission of the filter of the present invention. FIG. 8 and FIG. 9 are cross-sectional views showing other embodiments of the present invention, FIG. 10 is an equivalent circuit diagram thereof, and FIG. 11 is a diagram showing another embodiment of the present invention. , Fig. 12 to Fig. 15 are equivalent circuit diagrams for explaining the operation thereof, which include a housing, 2: partition, 3: dielectric,
46 to 4m+I: pore space, 5I to 5rL=4 internal bodies, 6: external conductor, 7: space, 86 and 871+1
: Input/output coaxial terminal, 9o% 97L++, 9; and 9'? L*(: input/output coupling element, IQ+, 4 and l0a
j 'Indirect coupling elements, 11 and 12: through holes, T# and T? I or l two output terminals, R+ or R? L: resonant circuit, P or P: phase circuit. Figure 1 Figure 2 Figure 3 Figure 4 Figure 5 f-1 (Ml-IZJ- Figure 8 Figure 9 Figure 10 Figure 11
Claims (1)
配設された複数個の内部導体と、この複数個の内部導体
の周りに一体に設けたコの字型の誘電体と、この誘電体
の開放面を除く表面に設けた外部導体と、前記複数個の
内部導体の中、初段及び終段の内部導体に各対向せしめ
た入出力結合素子と、前記複数個の内部導体の中、伝送
信号の往路を形成する内部導体と復路を形成する内部導
体の開に介在せしめた導体隔壁と、前記各部品を内装す
る筐体とを備えたことを特徴とするコムライン型帯域通
過ろ波器。 (2)電気長で管内波長の4の軸長を有し、コの字型に
配設された複数個の内部導体と、この複数個の内部導体
の周りに一体に設けたコの字型の誘電体と、この誘電体
の開放面を除く表面に設けた外部導体と、前記複数個の
内部導体の中、初段及び終段の内部導体に各対向せしめ
た入出力結合素子と、前記複数個の内部導体の中、伝送
信号の往路を形成する内部導体と復路を形成する内部導
体の間に介在せしめた導体隔壁と、前記複数個の内部導
体及び外部導体より成る複数個の半同軸型共振器の中、
縦続接続関係にある2個又はその整数倍の個数の半同軸
型共振器を隔てた半同軸型共振器相互を間接結合する容
量結合素子と、前記各部品を内装する筐体とを備えたこ
とを特徴とするコムライン型帯域通過ろ波器。 (3)電気長で管内波長の4の軸長を有し、コの字型に
配設された複数個の内部導体と、この複数個の内部導体
の周りに一体に設けたコの字形の誘電体と、この誘電体
の開放面を除く表面に設けた外部導体と、前記複数個の
内部導体の中、初段及び終段の内部導体に各対向せしめ
た入出力結合素子と、前記複数個の内部導体の中、伝送
信号の往路を形成する内部導体と復路を形成する内部導
体の間に介在せしめた導体隔壁と、前記複数個のつ部導
体及び外部導体より成る複数個の半同軸型共振器の中、
縦続接続関係にある4個又はその整数倍の個数の半同軸
型共振器を隔てた半同軸型共振器相互を間接結合する磁
界結合素子と、前記各部品を内装する筐体とを備えたこ
とを特徴とするコムライン型帯域通過ろ波器。 (4)電気長で管内波長の4の軸長を有し、コの字型に
配設された複数個の内部導体と、この複数個の内部導体
の周りに一体に設けたコの字型の誘電体と、Cの誘電体
の開放面を除(表面に設けた外部導体と、前記複数個の
内部導体の中、初段及び終段の内部導体に各対向せしめ
た入出力結合素子と、前記複数個の内部導体の中、伝送
信号の往路を形成する内部導体と復路を形成する内部導
体の間に介在せしめた導体隔壁と、前記複数個の内部導
体及び外部導体より成る複数個の半同軸型共振器の中、
縦続接続関係にある2個又はその整数倍の個数の半同軸
型共振器を隔てた半同軸型共振器相互を間接結合する容
量結合素子と、前記複数個の半同軸型共振器の中、縦続
接続関係にある4個又はその整数倍の個数の半同軸型共
振器を隔てた半同軸型共振器相互を間接結合する磁界結
合素子と、前記各部品を内装する筐体とを備えたことを
特徴とするコムライン型帯域通過ろ波器。 (5)複数個の内部導体の中、縦続接続関係において隣
接対向する内部導体の各中心軸間隔CK、い。、)(k
は1ないしn、nはろ波器の次数)が、cK、(K++
) =0.36 + ””K、(x+B54.6η d:内部導体の外径 H:外部導体の横幅 t:誘電体の誘電率 入。:伝送信号の自由空間波長 gK及びgH++ ’基準化低域通過ろ波器の回i18
素子値 BWr :許容通過帯域幅 f。:中を周波数 で与えられる特許請求の範囲第1項なL\し第4項の何
れかに記載のコムライン型帯域通過ろ波器。 (6)内部導体が誘電体に穿った孔隙σン内壁面に付着
せしめた金属被膜より成る特許請求の範囲第1項ないし
第4項の何れかに記載のコムライン型帯域通過ろ波器。 (7)内部導体が誘電体に穿った孔隙内(こ嵌入した棒
状導体より成る特許請求の範囲第1項なし\し第4項の
何れかに記載のコムライン型帯域通過−ろ;皮脂。 (8)外部導体が誘電体の表面(こ付着せしめた金属被
膜より成る特許請求の範囲第1項なし1し第4項の何れ
かに記載のコムライン型帯域通過ろ波器0 (9)外部導体が誘電体の表面に付着せしめた導体板よ
り成る・特許請求の範囲第1項なし1し第4項の何れか
に記載のコムライン型帯域通過ろ、皮Ha(lO)”外
部導体が筐体及び導体隔壁より成る特許請求の範囲第1
項ならXシ第4項の何九カ・(こ乙己軌のコムライン型
帯域鴨過ろ波器。 (11)入出力結合素子が容量結合素子より成る特許請
求の範囲第1項なし\し第4項の何れ力・fこg己載の
コムライン型帯域通過ろ波器。 成 (12)入出力結合素子が変成器形−子より成る特許請
求の範囲第1項なし1し第4項σノ何れ力\(こ言己載
のコムライン型帯域通過ろ波器。 (13)容量結合素子より成る入出力結合素子fこより
形成される入出力結合容量の付加分に応じて初段及び終
段の内部導体の軸長及びその周りの区秀電体の高さを短
縮して形成された誘電体の段部に前記入出力結合素子を
配設した特許請求の範囲第1項ないし第4項の何れかに
記載のコムライン型帯域通過ろ波器。 (14)誘電体を空気を以て置換し、内部導体を棒状導
体を以て形成すると共に外部導体を筐体及び導体隔壁を
以て形成した特許請求の範囲第1項ないし第4項の何れ
かに記載のコムライン型帯域通過ろ波器。 (15)筐体を省いた特許請求の範囲第1項ないし第4
項の何れかに記載のコムライン型帯域通過ろ波器。[Scope of Claims] (1) A plurality of internal conductors having an electrical length of 4 times the pipe wavelength and arranged in a U-shape; A U-shaped dielectric provided, an external conductor provided on the surface of this dielectric except for the open surface, and an input/output device each facing the first and final internal conductors of the plurality of internal conductors. A coupling element, a conductor partition interposed between an internal conductor forming an outgoing path and an internal conductor forming an incoming path of a transmission signal among the plurality of internal conductors, and a casing in which each of the components is housed. A combline type bandpass filter characterized by: (2) A plurality of internal conductors having an axial length equal to 4 of the pipe wavelength in electrical length and arranged in a U-shape, and a U-shape integrally provided around the plurality of internal conductors. a dielectric material, an external conductor provided on the surface of the dielectric material except for the open surface, an input/output coupling element each facing the first stage and final stage internal conductors among the plurality of internal conductors; a conductor partition interposed between an internal conductor forming an outgoing path of a transmission signal and an internal conductor forming an incoming path among the internal conductors, and a plurality of semi-coaxial type inside the resonator,
A capacitive coupling element that indirectly couples two semi-coaxial resonators separated by two or an integral multiple of the number of semi-coaxial resonators in a cascade connection relationship, and a casing in which each of the above-mentioned components is housed. A combline type bandpass filter featuring: (3) A plurality of internal conductors having an electrical length equal to 4 of the pipe wavelength and arranged in a U-shape, and a U-shaped conductor integrally provided around the plurality of internal conductors. a dielectric, an external conductor provided on the surface of the dielectric other than the open surface, an input/output coupling element facing each of the first-stage and final-stage internal conductors among the plurality of internal conductors; a plurality of semi-coaxial type conductor partitions interposed between the inner conductor forming the outgoing path of the transmission signal and the inner conductor forming the return path, and the plurality of corner conductors and the outer conductor; inside the resonator,
A magnetic field coupling element that indirectly couples the semi-coaxial resonators separated by four or an integral multiple of the semi-coaxial resonators in a cascade connection relationship, and a casing in which each of the above-mentioned components is housed. A combline type bandpass filter featuring: (4) A plurality of internal conductors having an axial length equal to 4 of the pipe wavelength in terms of electrical length and arranged in a U-shape, and a U-shape integrally provided around the plurality of internal conductors. a dielectric material C, an external conductor provided on the surface of the dielectric material C (excluding the open surface), and input/output coupling elements each facing the first and final stage internal conductors among the plurality of internal conductors; Among the plurality of internal conductors, a conductor partition is interposed between an internal conductor forming an outgoing path of a transmission signal and an internal conductor forming a return path, and a plurality of semicircular conductors comprising the plurality of internal conductors and the external conductor. Inside the coaxial resonator,
a capacitive coupling element that indirectly couples two semi-coaxial resonators separated by two or an integral multiple thereof in a cascade connection; A magnetic field coupling element that indirectly couples the semi-coaxial resonators separated by four or an integral multiple of the number of semi-coaxial resonators in a connection relationship, and a casing in which each of the above-mentioned components is housed. Features a combline type bandpass filter. (5) Among the plurality of internal conductors, the center axis distance CK between adjacent internal conductors facing each other in a cascade connection relationship. ,)(k
is 1 to n, n is the order of the filter), cK, (K++
) = 0.36 + ""K, (x + B54.6η d: Outer diameter of the inner conductor H: Width of the outer conductor t: Dielectric constant of the dielectric.: Free space wavelength of the transmission signal gK and gH++ 'Referenced low Pass filter time i18
Element value BWr: Allowable passband width f. : A combline type bandpass filter according to any one of claims 1 and 4, in which the middle is expressed as a frequency. (6) A combline type bandpass filter according to any one of claims 1 to 4, wherein the internal conductor is made of a metal coating adhered to the inner wall surface of a hole σ formed in a dielectric material. (7) A combline type band-pass filter according to any one of Claims 1 to 4, in which the internal conductor is a rod-shaped conductor inserted into a hole drilled in a dielectric; (8) The combline type bandpass filter 0 according to claim 1 or any one of claims 1 to 4, in which the outer conductor is made of a metal coating attached to the surface of a dielectric (9) The outer conductor consists of a conductive plate attached to the surface of a dielectric material.The combline type bandpass filter according to any one of claims 1 to 4, with a skin Ha(lO)" outer conductor Claim 1 consists of a casing and a conductor partition wall.
If the term is X, how many times is the fourth term? (This is a comline type bandpass filter. A self-mounted combline type bandpass filter according to claim 4.Constitution (12) Claims 1 to 4 in which the input/output coupling element is a transformer. (13) An input/output coupling element f consisting of a capacitive coupling element. Claims 1 to 5, wherein the input/output coupling element is disposed in a stepped portion of a dielectric material formed by shortening the axial length of the final stage internal conductor and the height of the electrical conductor around it. The combline type bandpass filter according to any one of Clause 4. (14) A patent claim in which the dielectric is replaced with air, the inner conductor is formed by a rod-shaped conductor, and the outer conductor is formed by a casing and a conductor partition. The combline type bandpass filter according to any one of claims 1 to 4. (15) Claims 1 to 4 in which the casing is omitted.
2. The combline bandpass filter according to any one of Items 1 to 1.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP19364482A JPS5989002A (en) | 1982-11-04 | 1982-11-04 | Comb-line type band-pass filter |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP19364482A JPS5989002A (en) | 1982-11-04 | 1982-11-04 | Comb-line type band-pass filter |
Publications (1)
Publication Number | Publication Date |
---|---|
JPS5989002A true JPS5989002A (en) | 1984-05-23 |
Family
ID=16311366
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP19364482A Pending JPS5989002A (en) | 1982-11-04 | 1982-11-04 | Comb-line type band-pass filter |
Country Status (1)
Country | Link |
---|---|
JP (1) | JPS5989002A (en) |
Cited By (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS6098902U (en) * | 1983-12-09 | 1985-07-05 | 富士電気化学株式会社 | dielectric filter |
JPS6390202A (en) * | 1986-10-02 | 1988-04-21 | Mitsubishi Electric Corp | High-frequency filter |
JPS63169802A (en) * | 1987-01-08 | 1988-07-13 | Yuniden Kk | Dielectric resonator |
JPH0211001A (en) * | 1988-06-29 | 1990-01-16 | Matsushita Electric Ind Co Ltd | Dielectric filter |
JP2012199623A (en) * | 2011-03-18 | 2012-10-18 | Ube Ind Ltd | Dielectric resonator component |
-
1982
- 1982-11-04 JP JP19364482A patent/JPS5989002A/en active Pending
Cited By (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS6098902U (en) * | 1983-12-09 | 1985-07-05 | 富士電気化学株式会社 | dielectric filter |
JPS6390202A (en) * | 1986-10-02 | 1988-04-21 | Mitsubishi Electric Corp | High-frequency filter |
JPS63169802A (en) * | 1987-01-08 | 1988-07-13 | Yuniden Kk | Dielectric resonator |
JPH0420521B2 (en) * | 1987-01-08 | 1992-04-03 | Uniden Kk | |
JPH0211001A (en) * | 1988-06-29 | 1990-01-16 | Matsushita Electric Ind Co Ltd | Dielectric filter |
JP2012199623A (en) * | 2011-03-18 | 2012-10-18 | Ube Ind Ltd | Dielectric resonator component |
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