JPS5931316B2 - Synchronous motor control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a synchronous motor control device configured to maintain a power factor of approximately 1 and to control terminal voltage approximately constant.

Conventionally, DC motors have been used to drive steel rolling mills and cement gearless drives.

Its drive characteristic is proportional to speed ω from the time of startup to base speed ωb, and constant terminal voltage Vt above base speed ωb.
characteristics are required. When a DC motor is used in this way, there is a drawback that maintenance and inspection of the commutator, brushes, etc. are always required. Therefore, it has been considered to use a synchronous motor having almost the same characteristics as a DC motor instead of a DC motor. However, if a synchronous motor is simply driven using the conventional speed control method, the terminal voltage Vt will be proportional to the speed ω, and the current I flowing through the armature will be in phase with the voltage induced in the armature winding. , the disadvantage is that the power factor deteriorates. The present invention was made in view of these circumstances, and its purpose is to have a drive characteristic proportional to the speed at the base speed, and to control the terminal voltage to be almost constant regardless of the speed when the base speed is exceeded. A synchronous motor control device is provided.

Hereinafter, details of the present invention will be explained with reference to the drawings.

FIG. 1 is a block diagram showing a circuit for driving and controlling a three-phase synchronous motor as an embodiment of the present invention.

FIG. 2 is a block diagram showing a specific configuration of the main part in FIG. 1. FIG. 3 is a vector diagram showing the relationship between voltage, current, and magnetic flux when a synchronous motor is driven with a power factor of 1.
A indicates power running, and b indicates regeneration.

However, the vector diagram shows the case where the resistance component is ignored and the field of the synchronous motor is cylindrical. FIG. 4 is a characteristic diagram showing the operating characteristics of the function generator FG used for power factor control.

The basic principle of the present invention will be explained with reference to these drawings.

The relationship between the voltage, magnetic flux, and speed ω of each part of the synchronous motor in FIG. 3 is generally expressed by the following equation. Terminal voltage Vt-
Kφω ...1) Leakage reactance drop Xt=K(1)tω ...2) Synchronous reactance drop 1XS=K(1)aω ...3) Horizontal axis magnetic flux φ5
=? +φt ...4) Induced voltage EO by field = K (!) fω ・゜・5) Internal induced voltage V, = Kφiω ...6) Main magnetic flux φ=φ
f+φ2...7) Leakage reactance Xt=ωLt...8) Synchronous reactance Xs-ωL
s...9) However, speed = ω, field magnetic flux = φf1 current =, internal composite magnetic flux = φi (=φf + φa), magnetic flux generated based on armature reaction = φa1 leakage magnetic flux = φt1 proportional coefficient = K1 terminal voltage t (It is expressed as phase difference between ψ and induced voltage EO (φf) = θ.

First, a theoretical explanation will be given for controlling the terminal voltage Vt to be constant.

The terminal voltage Vt is determined by the magnetic flux φ as shown in equation (1) above. In order to make the terminal voltage Vt proportional to ω up to the base speed ωb and constant above the base speed ωb as shown in Fig. 4, the magnetic flux φ as the output is changed to the base using a function generator using the speed ω as an input. It may be configured such that the magnetic flux φ is constant up to the speed ωb and is inversely proportional to the speed ω, with the magnetic flux φ=Vt/Kω above the base speed ωb. Next, a theoretical explanation will be given for controlling the power factor of the electric motor to 1. As mentioned above, the magnetic flux φ is obtained by a function generator. On the other hand, magnetic flux φ
2 is φ5 from the above equations (2), (3)) and (4).
It is determined as =I(LS+Lt)/K. If we calculate field magnetic flux φf vectorially from magnetic flux φ5 and magnetic flux φ, φf=t~
〒27, and SlNθ and CO which are functions of phase difference θ
Sθ is SlNθ=φ7φF, COSθ=φ/φf, respectively.
It can be found as Further, the field current command 1fR for actually generating the field magnetic flux by the signal of the field magnetic flux φf becomes IfR-K·φf if magnetic saturation is ignored. Next, each voltage E induced in the U-phase, V-phase, and W-phase armature windings of the synchronous motor. From the reference sine wave SiNωt which is synchronized with the reference sine wave SiNωt and the reference cosine wave COSωt1 which is 90 degrees ahead of this in phase, and from the COSθ and SiNθ obtained as described above, SiNθ
・COs(!)t+COsθ・SiNωt=SiN(ω
t+θ) is obtained, and a current reference signal SiN(ωt+θ) which is in phase with the terminal voltage Vt is obtained. Next, the first
An embodiment of the present invention shown in the figures will be described. Figure 1 shows an example of driving a 3-phase, 2-pole synchronous motor. In the figure, 1 indicates the 3-phase synchronous motor, 2 indicates the field of its rotor, and 3U, 3V, and 3W indicate each phase. The armature winding is shown.

Each of the armature windings 3U, 3V, and 3W is connected to an output end of a cyclone converter main body 4, which is a power conversion device.

The cyclone converter main body 4 includes a U that can independently supply input to each of the armature windings 3U, 3V, and 3W.
It is composed of a phase converter 4U, a V-phase converter 4V, and a W-phase converter 4W. The above converters 4U, 4V, and 4W are each formed equally, and use a thyristor as an active element, and as is well known, three-phase bridge circuits are connected in antiparallel.
It is formed so that phase-to-single phase conversion is possible. The input terminals of each of the converters 4U, 4V, and 4W are 3
It is connected to a phase alternating current power supply EP, and its conduction is controlled by a signal from a gate control device 5. The control device 5 includes each of the converters 4U and 4V.
, 4W independently controlled U-phase gate control circuit 5
It consists of a U and V phase gate control circuit 5V and a W phase gate control circuit 5W. Above each phase gate control circuit 5U
, 5V, and 5W are configured equally, and the current flowing through each armature winding 3U, 3, and 3W is connected to the current transformer 3.
1U, 31V, 31W and current detector 32U, 32V,
The signal converted by 32W and the arithmetic circuit 6 described later
The output signal of each of the converters 4U, 4 is input.
It is configured to control V and 4W continuously and in a current controlled manner.

Therefore, the rotor of the synchronous motor 1 has a position detection plate P.
A position detector 90 having three position detecting means is disposed at a stationary position facing the position detecting plate PS.

The position detector 90 is connected to the armature windings 3U, 3V, 3W.
For example, a magnetic plate with an electrical angle width of 1800 (in this case, a mechanical angle width of 1800) is connected directly to the rotor as a position detection plate PS.
It consists of a proximity switch, etc. which sends out a signal of 01'' when this magnetic plate approaches, and sends out a signal of 60'' at other times. , 3V, 3W, the output signal is switched each time the polarity of the speed electromotive force induced by the voltages 3V, 3W changes.The output signal of the position detector 90 is output from the U-phase pattern generating circuit 13U, 3V, 3W. It is applied to a V-phase pattern generation circuit 13V and a W-phase pattern generation circuit 13W, respectively.

Each of the circuits 13U, 13V, and 13W has the same configuration, and receives the output signal of the position detector 90 and the output signal of the tachogenerator TG that detects the rotational speed of the rotor of the synchronous motor, and This generates a sine wave (SinO)t and a cosine wave (COSωt) synchronized with the output signal of 90. These pattern generation circuits 13U, 13
The output signal of V, 13W is the arithmetic circuit 6U, 6V mentioned above.
, 6W. On the other hand, the output signal ωR of the speed setter 22 for setting the speed of the synchronous motor 1 is transmitted to the comparator 2.
It is introduced at one input end of 3.

The output signal ω of the tachogenerator TG is introduced into the other input terminal of the comparator 23, and the deviation between the signals ωR and ω is sent out from the comparator 23, and the difference between the signals ωR and ω is sent to the speed control circuit 2.
5. Therefore, the amplitude 1R of the current to be passed through the armature windings 3U, 3V, and 3W is set according to the deviation between the speed reference ωR and the speed ω. The output signal of the speed control circuit 25 is sent to the arithmetic circuit 26, where the magnitude of the field current IfR is determined to keep the terminal voltage Vt constant and the power factor to 1.
and the phase difference θ are set. The specific circuit configuration of the arithmetic circuit 26 and the arithmetic circuits 6U, 6V, and 6W is as shown in FIG.

That is, the speed signal ω from the tacho generator TG and the current amplitude command signal R from the speed control circuit 25 are introduced into the arithmetic circuit 26. The inner velocity signal ω of this introduced signal
is input to the function generator FG forming the arithmetic circuit 26. This function generator FG receives the speed signal ω and generates a signal for setting the magnetic flux φ. 0A1 is an amplifier which receives the current amplitude command signal R and generates a signal for setting the magnetic flux φ5.

The signals corresponding to the magnetic flux φ and magnetic flux φ5 generated in this way are
They become inputs to multipliers MLPl, Ml, P2, respectively, and output signals from multipliers MLPl, MLP2 are φ2,
φ′2 is obtained. 0A2 is an adder, and multipliers Ml, P
1 and the output signals of MLP2 are added.

The output signal of the adder 0A1 becomes an input to the square root circuit SQR, which generates a signal corresponding to the field magnetic flux φf to be set.
This signal is further amplified by an amplifier 0A3 to generate a field current command signal 1fR for flowing a field current If that generates the necessary actual magnetic field/magnetic flux φf. This signal is sent to the field control circuit 30 to control the firing phase of the converter 31 that supplies current to the field 2. still,
The field current is detected by the current detector 32 and the field current is detected by the field control circuit 30.
Feedback will be sent to you. On the other hand, multiplier MLPl, ML
Magnetic flux φ and magnetic flux φ obtained from P2 and square root circuit SQR
2 and the field magnetic flux φf are transmitted through dividers DIVl and DIV
2, where signals corresponding to COS θ and SiN θ are generated.

These signals are multiplied by the current amplitude command signal 1R by multipliers MLP3 and MLP4, and IR-COSθ and R-S
It becomes a signal of INθ (phase shift setting signal). The arithmetic circuit 26 is configured as described above. Multiplier MLP3, MLP4
Output signal and pattern generation circuit 13U, 13V, 13
The output signal of W is sent to arithmetic circuits 6U, 6V, and 6W corresponding to the U-phase, V-phase, and W-phase, respectively, for calculation, but the following explanation will be omitted and only the U-phase will be described. In the arithmetic circuit 6U, the reference cosine wave signal COsO)t and the signal IR-S
Multiplier MLP5 which receives INθ as input, and multiplier M which receives reference sine wave signal SINωt and signal 1R-COSθ as input.
Each output signal of LP6 is added by adder 0A5.
Then, the adder 0A5 outputs a current command signal 1RU=IR-S for flowing the actual current U necessary to generate the speed ω.
A signal corresponding to IN(ωt+θ) is generated. This signal is sent to the gate control circuit 5U. Similarly, output signals 1R and IRW of other arithmetic circuits 6V and 6W are sent to gate control circuits 5V and 5W. The output signals of these gate control circuits 5U, 5V, 5W control the firing phases of the cycloconverters 4U, 4V, 4W, and the currents supplied to the armature windings 3U, 3, 3W. Ru. Note that the output currents of the cycloconverters 4U, 4V, and 4W are detected by current detectors 34U, 31, and 31W, respectively, and further converted into predetermined voltage signals by signal converters 32U, 32V, and 32W, respectively, and then sent to the gate control circuit 5U. ,5V,
Feedback is given to 5W. With the above configuration, each phase current and terminal voltage are approximately in phase, and the power factor can be made close to 1.

Furthermore, the terminal voltage can also be kept constant at approximately the base speed ωb or higher. Furthermore, since the vector diagram in Figure 3 mentioned above ignores the resistance component of the armature windings, etc., errors caused by this will appear in the control characteristics of the synchronous motor 1, but in general, the resistance component is very small. Since it is small, it hardly affects the control characteristics.

FIG. 5 shows another embodiment of the present invention, in which the same parts as in FIG. 1 are designated by the same reference numerals, and detailed explanation will be omitted here.

This embodiment differs from the embodiment shown in FIG. The signal ω is sent to the voltage converter 100 to obtain a signal ω proportional to the speed, which is used as an input signal for the comparator 23 and pattern generation circuits 13U, 13V, and 13W. Here, the frequency-to-voltage converter 100 generates a voltage proportional to the reciprocal of the input pulse interval (pulse period). Furthermore, in this embodiment, the frequency-to-voltage converter 10
The pulse signal obtained by differentiating the output pulse of the position detector 90 obtained from 0
The difference from the embodiment shown in FIG. 1 is that the signal is supplied to the function generator FGA instead of G. The specific structure of this function generator FGA is as shown in FIG. 7, and is different in structure from the function generator FG shown in FIG. That is, the function generator FGA is supplied with a pulse signal obtained by differentiating the output pulse of the position detector 90 from the frequency-voltage converter 100, and the interval Tps between the pulses is TPS=2πP/6.
It is expressed as ω (where P is the number of synchronous motor poles and ω is the speed). Therefore, in order to obtain the speed ω-one-terminal voltage Vt characteristic shown in FIG. 4, it is sufficient to generate a function of the pulse period Tps main magnetic flux φ characteristic shown in FIG. Here, Tb22πP/ωb. This Tb indicates the pulse interval when the basis flux is ωb. To generate this function, the function generator FGA comprises a first and a second monostable circuit 1 which are supplied with a differential pulse signal.
01, 102 and integrator 103, sampling circuit 10
It consists of 4. The integrator 103 is a switch SW2.
, operational amplifier 0A1 capacitor C and constant voltage diode D are connected in parallel with each other, and a resistor R is connected to one end (input end) of the parallel circuit so that the operational amplifier 0A does not become saturated even when integrated over a period of Tb. A constant voltage -Vx is applied thereto, and the input terminal of the sampling circuit 104 is connected to the other end (output end). The switch SW2 is closed by the output pulse signal of the second monostable circuit 102. The sampling circuit 104 is composed of a switch SWl and a hold circuit H in series. The switch SWl is
It is designed to be closed by the output pulse signal of the monostable circuit 101. Here, the differential pulse signal Q1 taken out from the frequency-voltage converter 100, the output pulse signals MSl, MS of the first and second monostable circuits 101, 102
2. The temporal changes in the output signal 01 of the integrator 103 and the output signal φ of the sampling circuit 104 are shown in FIG.
Shown in e. However, in FIG. 8, the horizontal axis is time and the vertical axis is voltage, showing the state in which the synchronous motor is accelerating. To explain the operation, the output signal 01 of the integrator 103 is sampled by the output pulse signal MS1 of the first monostable circuit 101 and held in the hold circuit H. Then, the fall of the output pulse of the first monostable circuit 101 activates the second monostable circuit 102 to generate the pulse signal MS2.
is occurring. This pulse signal MS42 is used as a reset pulse to close the switch SW2, reset the integrator 103, and start a new integration.
In addition, in Fig. 2, amplifier 0A3 is used to make the current FR proportional to magnetic flux φf, but if controllability is taken into account, a function generator with saturation characteristics as shown in Fig. 9 can be used instead of amplifier 0A3. desirable.

The relationship between the field magnetic flux φf and the field current 1f is as follows, since the field magnetic flux φf is not proportional to the increase in the field current 1f due to the saturation of iron.If the current FR and the magnetic flux φf are proportionally approximated, Since this amount of error occurs, in order to perform accurate control, it is better to provide a relational generator that simulates this saturation characteristic. In the embodiment described above, the mechanical positional relationship between the field pole and the armature winding is determined by the position detector 90, and the relative position of the two is determined. It may also be used to find the relative positions of the two.

Furthermore, the present invention can also be applied to a linear motor. According to the present invention described in detail above, the power factor can be made approximately 1, and the terminal voltage can be made almost constant at a predetermined speed or higher, making it possible to obtain a device suitable for various uses.

[Brief explanation of drawings]

Fig. 1 is a block diagram showing an embodiment of the synchronous motor control device according to the present invention, Fig. 2 is a block diagram showing the specific configuration of the main parts of Fig. 1, and Fig. 3 is a block diagram showing the voltage and current of the synchronous motor. , a characteristic diagram showing the relationship between the magnetic fluxes in vector form, FIG. 4 is a characteristic diagram showing the speed-terminal voltage characteristic of the synchronous motor of the present invention and the speed-main magnetic flux characteristic to obtain it, and FIG. 5 is a characteristic diagram showing the speed-main magnetic flux characteristic of the synchronous motor of the present invention A block diagram showing another embodiment of the synchronous motor control device,
Fig. 6 is a characteristic diagram showing the pulse period vs. main magnetic flux characteristics, Fig. 7 is a circuit diagram showing the specific configuration of the function generator used in the embodiment shown in Fig. 5, and Fig. 8 is a signal diagram of each part in Fig. 7. FIG. 9 is a characteristic diagram showing the field magnetic flux-current characteristic. 1...Synchronous motor main body, 2...Field,
4...Cycloconverter, 5...Gate control circuit, 25...Speed control circuit, 6,2
6... Arithmetic circuit, 13U, 13V, 13W...
...Pattern generation circuit, 90...Position detector, TG...Tachogenerator, 100...
...Frequency-to-voltage converter.

Claims (1)

[Claims] 1. A power conversion device that converts the AC frequency of an AC power supply that receives power and applies it to the armature winding of a synchronous motor;
A gate control circuit that controls the output current of the power converter, a field current control circuit that controls the field current of the synchronous motor, and a position that detects the relative position of the armature winding of the synchronous motor and the field. a detection means; a speed detection means for detecting a relative speed between an armature winding of the synchronous motor and a field; and setting an amplitude of a current flowing through the armature winding in response to the speed detected by the speed detection means. a current amplitude command signal generating means for generating a current amplitude command signal for controlling the synchronous motor; a current amplitude command signal obtained by the current amplitude command signal generating means and a signal obtained by the speed detecting means; A field current command signal is generated to be input to the field current control circuit by calculating a field current such that the main magnetic flux is constant up to a base speed and is inversely proportional to the speed above the base speed, and the amplitude command signal is input to the field current control circuit. a first arithmetic circuit that generates a phase shift setting signal based on the current amplitude command obtained by the generating means and the sine and cosine values of the phase difference between the main magnetic flux and the field magnetic flux; and a first arithmetic circuit that is synchronized with the output signal of the position detecting means. a pattern generating means for generating sine wave and cosine wave signals; and a pattern generating means for generating sine wave and cosine wave signals; A synchronous motor control device comprising: a second arithmetic circuit that obtains a current command signal for causing current to flow through the armature winding and sends this current command signal to the gate control circuit.