JPH08265056A - Tuning amplifier - Google Patents

Abstract

PURPOSE: To obtain the tuning amplifier in which a tuning frequency and a maximum attenuation are optionally adjusted without mutual interference. CONSTITUTION: The tuning amplifier is made up of two phase shift circuits each comprising an operational amplifier 12 receiving an input signal at its inverting input terminal via a resistor 18, a series circuit comprising a capacitor 14 between both terminals of which the input signal voltage is applied and a variable resistor 16, a phase inverting circuit 80 inverting an output of the poststage phase shift circuit 10, and an adder circuit adding a signal outputted from the phase inversion circuit 80 via each of a feedback resistor 70 and an input resistor 74 and the input signal received by an input terminal 90 at a prescribed ratio. The tuning frequency is adjusted by changing a time constant of the series circuit to change a resistance ratio of the input resistor 74 and the feedback resistor 70 thereby adjusting a maximum attenuation.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a tuning amplifier which can be easily integrated, and more particularly to a tuning amplifier which can arbitrarily adjust the tuning frequency and the maximum attenuation without interfering with each other.

[0002]

2. Description of the Related Art Conventionally, various types of amplifier circuits using active elements and reactance elements have been proposed and put into practical use as tuning amplifiers.

[0003]

In the conventional tuning amplifier, when the tuning frequency is adjusted, the Q and gain depending on the LC circuit are changed, and when the maximum attenuation is adjusted, the tuning frequency is changed. As shown by the characteristic curves A and B of No. 21, since the gain at the tuning frequency changes when the maximum attenuation amount is adjusted, the tuning frequency, the gain at the tuning frequency, and the maximum attenuation amounts C1 and C2 are adjusted without interfering with each other. It was extremely difficult.

Further, it has been difficult to form a tuning amplifier capable of adjusting the tuning frequency and the maximum attenuation amount by an integrated circuit.

Therefore, the present invention was devised to solve such a problem.

[0006]

In order to solve the above-mentioned problems, the tuning amplifier of the present invention has an input side impedance element to which an AC signal input to an input terminal is input at one end and a feedback signal to one side. An adder circuit including a feedback-side impedance element that is input to an end, and an addition circuit that adds the AC signal input to the input terminal and the feedback signal, and one end of the first resistor is connected to the inverting input terminal. Differential input amplifier, a second resistor connected between the inverting input terminal and the output terminal of the differential input amplifier, and a third resistor and a capacitor connected to the other end of the first resistor. And a series circuit formed by connecting the third resistor and the capacitor to a non-inverting input terminal of the differential input amplifier, and a phase shift circuit for inverting an input AC signal. Output Comprising a phase inverting circuit, and each of the two phase shifting circuits and a phase inverting circuit connected in cascade,
When the signal added by the adding circuit is input to the first-stage circuit of the plurality of circuits connected in cascade, the signal output from the last-stage circuit is used as the feedback signal and the feedback-side impedance is used. It is characterized in that it is inputted to one end of the element and the output of any one of these plural circuits is taken out as a tuning signal.

Further, the tuning amplifier of the present invention includes a phase inverting circuit that inverts the phase of an input AC signal and outputs the inverted signal.
An operational amplifier, a time constant circuit composed of a resistor and a capacitor to which the input AC signal is applied, a circuit for inputting a signal generated in the time constant circuit to a non-inverting input terminal of the operational amplifier, and an inversion of the operational amplifier. Connected to the input terminal,
A two-stage phase shift circuit having an input resistance to which an input signal is applied and a feedback resistance connected between the output terminal and the inverting input terminal of the operational amplifier, and for shifting an AC signal in the same direction; A cascade connection including the phase inversion circuit and the two-stage phase shift circuit, a feedback impedance element connected between an output and an input of the cascade connection, and an input impedance element for transmitting an AC signal to the cascade connection. And an input circuit for inputting via the input circuit.

Further, the tuning amplifier of the present invention includes a phase inverting circuit that inverts the phase of an input AC signal and outputs the inverted signal.
An operational amplifier, a time constant circuit composed of a resistor and a capacitor to which the input AC signal is applied, a circuit for inputting a signal generated in the time constant circuit to a non-inverting input terminal of the operational amplifier, and 2 has an input resistance connected to the inverting input terminal and to which an input signal is applied and a feedback resistance connected between the output terminal of the operational amplifier and the inverting input terminal, and shifts the AC signal in the same direction. -Stage phase shift circuit, a cascade connection including the phase inversion circuit and the two-stage phase shift circuit, a feedback-side impedance element connected between an output and an input of the cascade connection, and an alternating current to the cascade connection An input circuit for inputting a signal via an input-side impedance element, and a constant conversion circuit for converting the capacitance of the capacitor are provided.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A tuning amplifier according to an embodiment of the present invention will be specifically described below with reference to the drawings.

The features of the tuning amplifier of each of the following embodiments are that the phase shift circuit at the previous stage shifts the phase of the AC signal input via the impedance element on the input side (for example, the input resistance), and the phase shift circuit at the previous stage. Is the output of the phase inversion circuit that shifts the phase of the AC signal so that the phase relationship between the input and output voltages is the same, the phase inversion circuit that inverts the phase of the output of the phase shift circuit of the latter stage, And a feedback-side impedance element (for example, a feedback resistor) that feeds back to the input of the phase shift circuit of the previous stage, sets the gain of the entire system to approximately 1, and tunes at a frequency at which the total phase difference of the closed circuit becomes 0 °. There is an amplification operation.

(First Embodiment) FIG. 1 is a circuit diagram showing a configuration of a tuning amplifier of a first embodiment to which the present invention is applied.
The tuning amplifier 1 shown in the figure has two phase shift circuits 10 each of which performs a total phase shift of 180 ° at a predetermined frequency by shifting a phase of an input signal by a predetermined amount.
, A phase inversion circuit 80 that inverts the phase of the output signal of the subsequent phase shift circuit 10, a feedback resistor 70 and an input resistor 74 (input resistor
74 has a resistance value that is n times the resistance value of the feedback resistor 70).
It is configured to include an adder circuit that adds a signal (feedback signal) output from 80 and a signal (input signal) input to the input terminal 90 at a predetermined ratio.

FIG. 2 shows the extracted structure of the phase shift circuit 10 at the front stage and the rear stage shown in FIG. The phase shift circuit 10 in the preceding stage shown in the figure is an operational amplifier (operational amplifier) 12 which is a kind of differential input amplifier, and a non-inverting input of the operational amplifier 12 by shifting a phase of a signal input to an input end 22 by a predetermined amount. Capacitor 14 and variable resistor 16 input to the terminal,
The resistor 18 is inserted between the input end 22 and the inverting input terminal of the operational amplifier 12, and the resistor 20 is inserted between the output end 24 of the operational amplifier 12 and the inverting input terminal.

In this specification, it is assumed that the operational amplifier 12 and the like operate ideally, and when deviation from the ideal is a problem in actually designing a circuit, description will be added each time.

In the phase shift circuit 10 having such a configuration, when a predetermined AC signal is input to the input end 22, the voltage VR1 appearing across the variable resistor 16 is applied to the non-inverting input terminal of the operational amplifier 12. To be done.

Since there is no potential difference between the two inputs (the inverting input terminal and the non-inverting input terminal) of the operational amplifier 12 shown in FIG. 2, the potential of the inverting input terminal of the operational amplifier 12, the capacitor 14 and the variable resistor 16 are different. Is equal to the potential at the connection point.
Therefore, the same voltage VC1 that appears across the capacitor 14 appears across the resistor 18.

Here, when the resistance values of the resistor 18 and the resistor 20 are equal, the same current flows through these two resistors 18 and 20, so that the voltage VC1 also appears across the resistor 20. Moreover,
The voltage VC1 appearing across each of these two resistors 18 and 20 has the same vector direction, and considering the inverting input terminal (voltage VR1) of the operational amplifier 12 as a reference, the voltage VC1 across the resistor 18 is Input voltage Ei is the vector addition
Then, the voltage VC1 of the resistor 20 is vector-subtracted to obtain the output voltage Eo.

FIG. 3 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 10 and the voltage appearing at the capacitor or the like.

As shown in the figure, the voltage VR1 appearing across the variable resistor 16 and the voltage VC1 appearing across the capacitor 14 are 90 ° out of phase with each other, and the vector addition of these is the input voltage. It becomes Ei. Therefore, when the amplitude of the input signal is constant and only the frequency changes,
Along the circumference of the semi-circle shown in FIG.
R1 and the voltage VC1 across the capacitor 14 change.

The output voltage Eo is obtained by subtracting the voltage VC1 from the voltage VR1 in a vector manner. Considering the voltage VR1 applied to the non-inverting input terminal as a reference, the input voltage Ei
The output voltage Eo differs from the output voltage Eo only in the direction in which the voltage VC1 is combined, and their absolute values are equal. Therefore, the relationship between the magnitude and the phase of the input / output voltage can be expressed by an isosceles triangle having the input voltage Ei and the output voltage Eo as hypotenuses and the base being twice the voltage VC1, and the amplitude of the output signal is related to the frequency. It can be seen that the amplitude is the same as that of the input signal and the phase shift amount is represented by φ1 shown in FIG.

As is clear from FIG. 3, the voltage VR1
And the voltage VC1 intersect at a right angle on the circumference, theoretically, the phase difference between the input voltage Ei and the voltage VR1 changes from 90 ° to 0 ° as the frequency ω changes from 0 to ∞. The phase shift amount φ1 of the entire phase shift circuit 10 is twice that, and changes from 180 ° to 0 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be verified quantitatively.

When the input voltage Ei is applied to the input end 22, the current flowing through the resistors 18 and 20 from the input end 22 to the output end 24 is I, and the resistance values of the resistors 18 and 20 are equal to each other. Is r, the voltage across the resistors 18 and 20 is −
I · r.

By the way, as described above, since no potential difference should occur between the two inputs of the operational amplifier 12 shown in FIG. 2, the voltage VR1 across the variable resistor 16 applied to the non-inverting input terminal of the operational amplifier 12 and the output. Between the voltage Eo,

[Equation 1] There is a relationship.

In order to prevent a potential difference between the two inputs of the operational amplifier 12, the voltage VC1 across the capacitor 14 and the resistance
Since the value obtained by adding the voltage across both ends of 18 −I · r must be 0,

[Equation 2] Becomes From equation (1) and equation (2),

(Equation 3) Becomes

Since the sum of the voltages VR1 and VC1 across the variable resistor 16 and the capacitor 14 is the voltage Ei applied to the input terminal 22, the voltage between these voltages is

[Equation 4] There is a relationship. From equation (3) and equation (4),

(Equation 5) Becomes Here, C is the capacitance of the capacitor 14, R is the resistance value of the variable resistor 16, and the time constant of the CR circuit is T (= C
R).

Substituting s = jω in the equation (5) and transforming it,

(Equation 6) Becomes When the absolute value of the output voltage Eo is calculated from the equation (6),

(Equation 7) Becomes That is, the equation (7) is the phase shift circuit 10 of this embodiment.
Indicates that no matter how the phase between input and output rotates, the amplitude of the output signal is equal to the amplitude of the input signal and is constant.

From the equation (6), the input voltage E of the output voltage Eo is
When the phase shift amount φ1 for i is calculated,

(Equation 8) Becomes From this equation (8), for example, ω is 1 / T (= 1 /
(CR)) Phase shift amount φ at the frequency
1 becomes 90 °, and only the phase can be shifted by 90 ° without attenuating the amplitude of the input signal. Therefore, considering the case where the time constants T of the two phase shift circuits 10 are equal to each other for simplification of explanation, it is possible to shift the phase by 90 ° in each case when ω is 1 / T, that is, 180 ° in total. Moreover, by changing the resistance value R of the variable resistor 16, the frequency ω at which the total phase shift amount of the two phase shift circuits 10 becomes 180 ° can be changed.

The phase inverting circuit 80 shown in FIG. 1 includes an operational amplifier 82 in which an input signal is input to an inverting input terminal via a resistor 84 and a non-inverting input terminal is grounded, and an inverting input of the operational amplifier 82. The resistor 86 is connected between the terminal and the output terminal. When an AC signal is input to the inverting input terminal of the operational amplifier 82 via the resistor 84, a reverse-phase signal whose phase is inverted is output from the output terminal of the operational amplifier 82, and the reverse-phase signal is shown in FIG. It is adapted to be taken out from the output terminal 92 of the tuning amplifier 1.

The output of the phase inverting circuit 80 is fed back to the input side of the preceding phase shift circuit 10 via the feedback resistor 70, and this fed back signal and the input resistor 74 are input. The signals are added, and the added voltage is applied to the input end (input end 22 shown in FIG. 3) of the phase shift circuit 10 at the previous stage.

By forming such a feedback loop, the phase is shifted by 180 ° by the two phase shift circuits 10 at a certain frequency, and the phase is inverted by the phase inverting circuit 80. The phase shift amount of is 0 °. At this time, the phase inversion circuit
The tuning operation is performed by setting the amplification factor of 80 to a predetermined value and setting the loop gain of the entire tuning amplifier 1 to approximately 1.

FIG. 4 is a system diagram in which the entire two phase shift circuits 10 and 30 having the above-mentioned configuration are replaced with a circuit having a transfer function K1, and a resistor R0 is provided in parallel with the circuit having the transfer function K1. Feedback resistor 70 is connected in series with feedback resistor 70
An input resistor 74 having a resistance value (nR0) which is n times the resistance value of n is connected. FIG. 5 is a system diagram obtained by converting the system shown in FIG. 4 by the Miller's theorem, and the transfer function A of the entire system after conversion is

[Equation 9] Can be represented by

By the way, as is clear from the equation (5), the transfer function K2 of the phase shift circuit 10 is

[Equation 10] And the overall transfer function K3 when the phase shift circuit 10 is cascaded in two stages is

[Equation 11] Becomes Therefore, the overall transfer function K1 when the phase inversion circuit 80 is further connected after the phase shift circuit 10 is connected in two stages is

(Equation 12) Becomes Substituting this equation (12) into the above equation (9),

(Equation 13) Becomes

According to this equation (13), it can be seen that when ω = 0 (DC region), A = −1 / (2n + 1) and the maximum attenuation is given. It can also be seen that the maximum attenuation is given when ω = ∞. Furthermore, for example, ω = 1
Tune point of / T (when the time constants of the two phase shift circuits 10 are different and they are T ₁ and T ₂ , respectively, ω = 1
It can be seen that A = 1 at / √ (T ₁ · T ₂ ), which is irrelevant to the resistance ratio n between the feedback resistor 70 and the input resistor 74. In other words, as shown in FIG. 6, even if the value of n is changed, the tuning point does not shift and the amount of attenuation at the tuning point does not change.

As described above, according to the tuning amplifier 1 of this embodiment, the tuning frequency and the gain at the time of tuning are constant even if the resistance ratio n of the feedback resistor 70 and the input resistor 74 is changed, and only the maximum attenuation amount is obtained. Can be changed. On the contrary, since the maximum attenuation amount is determined by the resistance ratio n described above, even when the tuning frequency is changed by changing the resistance value of the variable resistor 16 in each phase shift circuit 10, the maximum attenuation amount is The tuning frequency, the gain at the tuning frequency, and the maximum amount of attenuation can be adjusted without interfering with each other without interfering with each other.

Further, the tuning amplifier 1 of the first embodiment is configured by combining an operational amplifier, a capacitor or a resistor, and since any constituent element can be formed on a semiconductor substrate, the tuning frequency and the maximum attenuation amount can be reduced. It is also easy to form the entire adjustable tuning amplifier 1 on a semiconductor substrate to form an integrated circuit.

(Second Embodiment) FIG. 7 is a circuit diagram showing a configuration of a tuning amplifier according to a second embodiment of the present invention.
The tuning amplifier 1a shown in the figure has two phase shift circuits 30 each performing a total phase shift of 180 ° at a predetermined frequency by shifting the phase of the input signal by a predetermined amount.
, A phase inversion circuit 80 that inverts the phase of the output signal of the subsequent phase shift circuit 10, a feedback resistor 70 and an input resistor 74 (input resistor
74 has a resistance value that is n times the resistance value of the feedback resistor 70).
It is configured to include an adder circuit that adds a signal (feedback signal) output from 80 and a signal (input signal) input to the input terminal 90 at a predetermined ratio. Each of the two phase shift circuits 30 described above has a relative phase relationship of input and output voltages opposite to that of the phase shift circuit 10 included in the tuning amplifier 1 of the first embodiment. Have the same configuration.

FIG. 8 shows the extracted structure of the phase shift circuit 30 at the front stage and the rear stage shown in FIG. The phase shift circuit 30 shown in the figure includes an operational amplifier 32, which is a type of differential input amplifier, and a variable resistor for shifting the phase of the signal input to the input terminal 42 by a predetermined amount and inputting it to the non-inverting input terminal of the operational amplifier 32. 36 and capacitor 34, input 42 and operational amplifier
The resistor 38 is inserted between the inverting input terminal 32 and the inverting input terminal, and the resistor 40 is inserted between the output terminal 44 of the operational amplifier 32 and the inverting input terminal.

In the phase shift circuit 30 having such a configuration, when a predetermined AC signal is input to the input end 42, the voltage VC2 appearing across the capacitor 34 is applied to the non-inverting input terminal of the operational amplifier 32. It

Further, since there is no potential difference between the two inputs (the inverting input terminal and the non-inverting input terminal) of the operational amplifier 32 shown in FIG. 8, the potential of the inverting input terminal of the operational amplifier 32, the variable resistor 36 and the capacitor 34. Is equal to the potential at the connection point.
Therefore, the same voltage VR2 that appears across the variable resistor 36 appears across the resistor 38.

Here, when the resistance values of the resistor 38 and the resistor 40 are the same, the same current flows through these two resistors 38 and 40, so that the voltage VR2 also appears at both ends of the resistor 40. Moreover,
The voltages VR2 appearing at both ends of these two resistors 38 and 40 are oriented in the same vector direction, and considering the inverting input terminal (voltage VC2) of the operational amplifier 32 as a reference, the voltage VR2 at both ends of the resistor 38 becomes a vector. Input voltage Ei
Then, the output voltage Eo is obtained by vector-wise subtracting the voltage R2 across the resistor 40.

FIG. 9 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 30 and the voltage appearing at the capacitor or the like.

As shown in the figure, the voltage VC2 appearing across the capacitor 34 and the voltage VR2 appearing across the variable resistor 36 are 90 ° out of phase with each other, and the vector addition of these is the input voltage. It becomes Ei. Therefore, when the amplitude of the input signal is constant and only the frequency changes,
The voltage VC2 across the capacitor 34 and the voltage VR2 across the variable resistor 36 change along the circumference of the semicircle shown in FIG.

As described above, the voltage VC2 to the voltage V
The output voltage Eo is obtained by subtracting R2 in vector.
Considering the voltage VC2 applied to the non-inverting input terminal as a reference, only the direction in which the voltage VR2 is combined with the input voltage Ei output voltage Eo is different and the absolute values thereof are equal. Therefore, the relationship between the magnitude of input / output voltage and the phase is as follows.
It can be represented by an isosceles triangle having i and the output voltage Eo as hypotenuses and the base being twice the voltage VR2. The amplitude of the output signal is the same as the amplitude of the input signal regardless of the frequency, and the amount of phase shift is It can be seen that it is represented by φ2 shown in FIG.

As is clear from FIG. 9, the voltage VC2
And the voltage VR2 intersect at a right angle on the circumference, theoretically, the phase difference between the input voltage Ei and the voltage VC2 changes from 0 ° to 90 ° as the frequency ω changes from 0 to ∞. The shift amount φ2 of the entire phase shift circuit 30 is twice that, and changes from 0 ° to 180 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be quantitatively verified.

Similar to the case of the phase shift circuit 10 shown in FIG.
When the voltage Ei is applied to the input terminal 42, the current flowing from the input terminal 42 to the output terminal 44 through the resistors 38 and 40 is I, and the resistance values of the resistor 38 and the resistor 40 are equal, and the value is r. ,
The voltage across the resistors 38 and 40 is -I.r.

Since there must be no potential difference between the two inputs of the operational amplifier 32 shown in FIG. 8, the voltage VC2 across the capacitor 34 applied to the non-inverting input terminal of the operational amplifier 32 and the output voltage Eo are not present. ,

[Equation 14] There is a relationship.

Further, in order to prevent a potential difference between the two inputs of the operational amplifier 32, the voltage VR2 across the variable resistor 36 and the resistor 38 are applied.
Since the value obtained by adding the voltage −I · r across both ends of must be 0,

(Equation 15) Becomes From equation (14) and equation (15),

[Equation 16] Becomes

Since the sum of the voltages VC2 and VR2 across the capacitor 34 and the variable resistor 36 is the voltage Ei applied to the input terminal 42, the voltage Ei applied between these voltages is

[Equation 17] There is a relationship. From equation (16) and equation (17),

(Equation 18) Becomes Here, C represents the electrostatic capacitance of the capacitor 34, and R represents the resistance value of the variable resistor 36, which is the same as in the case of the phase shift circuit 10.
The time constant of the R circuit is T (= CR). Substituting s = jω in Eq. (18) and transforming

[Formula 19] Becomes

The equations (18) and (19) described above are phase shift circuits.
Only the signs are different from the equations (5) and (6) shown for 10. Therefore, as for the absolute value of the output voltage Eo, the equation (7) can be applied as it is, and the amplitude of the output signal of the phase shift circuit 30 is the same as that of the input signal, no matter how the phase between the input and the output rotates. It can be seen that it is equal to and constant.

Further, when the phase shift amount φ2 of the output voltage Eo with respect to the input voltage Ei is obtained from the equation (19),

(Equation 20) Becomes From this equation (20), for example, ω is 1 / T (= 1 /
(CR)) Phase shift amount φ at the frequency
2 becomes 90 °, and only the phase can be shifted by 90 ° without attenuating the amplitude of the input signal. Therefore, as in the first embodiment, in order to simplify the description,
Considering the case where the time constants T of the two phase shift circuits 30 are equal,
When ω is 1 / T, the phase is 90 ° in each case, 2
The total of the one phase shift circuits 30 can be shifted by 180 °, and furthermore, by changing the resistance value R of the variable resistor 36, the frequency ω at which the total amount of phase shift is 180 ° can be changed.

The phase inversion circuit 80 shown in FIG. 7 is the same as that shown in FIG. 1 in the first embodiment, and the phase of the signal output from the phase shift circuit 30 in the subsequent stage is further inverted to obtain the tuning amplifier. It is output from the output terminal 92 of 1a. Further, the output of the phase inversion circuit 80 is fed back to the input side of the phase shift circuit 10 of the previous stage via the feedback resistor 70, and the fed-back signal and the signal input via the input resistor 74 are The added voltage is added and applied to the input end (input end 42 shown in FIG. 8) of the phase shift circuit 30 at the previous stage.

By forming such a feedback loop, the phase is shifted by 180 ° by the two phase shift circuits 30 at a certain frequency, and the phase is inverted by the phase inversion circuit 80. The phase shift amount of is 0 °. At this time, the phase inversion circuit
The tuning operation is performed by setting the amplification factor of 80 to a predetermined value and setting the loop gain of the entire tuning amplifier 1a to approximately 1.

By the way, the tuning amplifier 1a of the second embodiment including the two phase shift circuits 30 and the phase inverting circuit 80 described above.
Can be represented by the system diagram shown in FIG. 4 by replacing it with a circuit having a transfer function K1 as in the case of the first embodiment. Therefore, it can be expressed by the system diagram shown in FIG. 5 by conversion by the Miller's theorem, and the transfer function A of the entire system after conversion can be expressed by the equation (9).

As is clear from the equation (18), the transfer function K21 of each of the two phase shift circuits 30 is

[Equation 21] Since K21 is different from K2 in the above equation (10) only in sign, the overall transfer function when the phase shift circuit 30 is cascaded in two stages is the same as K3 shown in equation (11). Can be applied. Further, the overall transfer function when the phase inversion circuit 80 is further connected after connecting the two stages of the phase shift circuit 30, K1 shown in the equation (12) is further converted by the Miller's theorem. A shown in equation (13)
Can be applied as is.

Therefore, the tuning amplifier 1a of the second embodiment.
Has a characteristic similar to that of the tuning amplifier 1 of the first embodiment, and A = -1 / (2n +) when ω = 0 (DC region).
It can be seen that the maximum attenuation is given in 1). It can also be seen that the maximum attenuation is given when ω = ∞. Furthermore, for example, ω = ₁ / T T 1, respectively in a case where the time constant different tuning points (two phase shifting circuits 30,
When T ₂ is set, ω = 1 / √ (T ₁ · T ₂ ) tuning point)
, A is 1 and is irrelevant to the resistance ratio n of the feedback resistor 70 and the input resistor 74, and the tuning point does not shift even if the value of n is changed as shown in FIG. The amount of attenuation does not change either.

As described above, the tuning amplifier 1a of this embodiment is
According to this, even if the resistance ratio n of the feedback resistor 70 and the input resistor 74 is changed, the tuning frequency and the gain at the time of tuning are constant, and only the maximum attenuation amount can be changed. On the contrary, since the maximum attenuation amount is determined by the resistance ratio n described above, even if the tuning frequency is changed by changing the resistance value of the variable resistor 36 in each phase shift circuit 30, the maximum attenuation amount is The tuning frequency, the gain at the tuning frequency, and the maximum amount of attenuation can be adjusted without interfering with each other without interfering with each other.

Further, the tuning amplifier 1a of the second embodiment is constructed by combining an operational amplifier, a capacitor or a resistor, and since any constituent element can be formed on a semiconductor substrate, the tuning frequency and the maximum attenuation amount can be reduced. It is also easy to form the entire adjustable tuning amplifier 1 on a semiconductor substrate to form an integrated circuit.

(Other Embodiments) The tuning amplifiers 1 and 1a of the above-described embodiments are provided with two phase shift circuits 10 or 2.
It is composed of one phase shift circuit 30 and a phase inversion circuit 80, and a predetermined tuning operation is performed by setting the total phase shift amount to 0 ° by the whole of these three circuits. Therefore, focusing only on the amount of phase shift, there is some degree of freedom in the order in which the three circuits are connected, and the connection order can be determined as necessary.

10 and 11 show two phase shift circuits 10.
Alternatively, it is a diagram showing a connection state of 30 and a phase inversion circuit 80. In these figures, the feedback impedance element 70a and the input impedance element 74a are for adding the output signal of each tuning amplifier and the input signal at a predetermined ratio, and most commonly in FIG. As shown in the figure, the feedback resistor 70 is used as the feedback impedance element 70a.
Is used as the input-side impedance element 74a.

However, the feedback-side impedance element 70a and the input-side impedance element 74a need only be added without changing the phase relationship of the signals input to the respective elements, so the feedback-side impedance element 70a and the input-side impedance element 74a. May be formed by capacitors, or both the feedback impedance element 70a and the input impedance element 74a may be formed by inductors. Alternatively, each impedance element may be formed by combining resistors, capacitors, or inductors so that the ratio of the real number and the imaginary number of impedance can be adjusted at the same time.

FIG. 10A shows a configuration in which the phase inversion circuit 80 is arranged at the subsequent stage of the two phase shift circuits 10 and corresponds to the tuning amplifier 1 shown in FIG. 2 in FIG. 10 (B)
A configuration is shown in which a phase inversion circuit 80 is arranged at the subsequent stage of one phase shift circuit 30, and corresponds to the tuning amplifier 1a shown in FIG. As described above, when the phase inverting circuit 80 is arranged in the subsequent stage, a large output current can be taken out by giving the phase inverting circuit 80 a function as an output buffer.

In FIG. 10C, the phase inversion circuit 80 is arranged between the two phase shift circuits 10, but in FIG. 10D, the phase inversion circuit 80 is arranged between the two phase shift circuits 30. Each of the arranged configurations is shown. In this way, the phase inversion circuit 80
With the above arrangement, mutual interference between the two phase shift circuits can be completely prevented.

FIG. 11A shows a configuration in which the phase inversion circuit 80 is arranged in the preceding stage of the two phase shift circuits 10, and FIG. 11B shows the phase inversion circuit 80 in the preceding stage of the two phase shift circuits 30. Each of the arranged configurations is shown. As described above, when the phase inversion circuit 80 is arranged in the preceding stage, the phase shift circuit 10 or 30 in the preceding stage is arranged.
It is possible to minimize the influence of the feedback impedance element 70a and the input impedance element 74a on the.

Further, the phase shift circuits 10 and 30 shown in the above-mentioned embodiments include the variable resistors 16 and 36.
These variable resistors 16 and 36 can be specifically realized by using a junction type or MOS type FET.

FIG. 12 is a diagram showing the structure of the phase shift circuit in the case where the variable resistor 16 or 36 in the two phase shift circuits shown in each embodiment is replaced with an FET.

FIG. 9A shows a configuration in which the variable resistor 16 is replaced with an FET in the two phase shift circuits 10 shown in FIG. FIG. 7B shows a configuration in which the variable resistor 36 in the two phase shift circuits 30 shown in FIG. 7 is replaced with an FET.

As described above, the channel formed between the source and drain of the FET is used as a resistor to change the variable resistance.
When used in place of 16 or 36, the gate voltage can be variably controlled to arbitrarily change the channel resistance within a certain range to change the amount of phase shift in each phase shift circuit. Therefore, in each tuning amplifier, the frequency at which the amount of phase shift of the signal that makes one round becomes 0 ° can be changed, and the tuning frequency of the tuning amplifier of each embodiment can be arbitrarily changed.

In each phase shift circuit shown in FIG. 12, the variable resistance is composed of one FET, that is, a p-channel or n-channel FET.
Alternatively, T and n-channel FETs may be connected in parallel to form one variable resistor, and gate voltages of equal magnitude but different polarities may be applied between the gate and substrate of each FET. When changing the resistance value, the magnitude of the gate voltage may be changed. As described above, by combining the two FETs to form the variable resistance, the non-linear region of the FET can be improved, and thus the distortion of the tuning signal can be reduced.

Further, the phase shift circuit 10 or 30 shown in each of the above-mentioned embodiments changes the resistance value of the variable resistor 16 or 36 connected in series with the capacitors 14 and 34 to change the amount of phase shift. Although the whole tuning frequency is changed by means of, the capacitors 14 and 34 may be formed by variable capacitance elements, and the whole tuning frequency may be changed by changing the capacitance thereof.

FIG. 13 is a diagram showing the structure of the phase shift circuit in the case where the capacitors 14 or 34 in the two types of phase shift circuits 10 or 30 shown in each embodiment are replaced by variable capacitance diodes.

FIG. 1A shows a configuration in which the variable resistor 16 is replaced with a fixed resistor and the capacitor 14 is replaced with a variable capacitance diode in the two phase shift circuits 10 shown in FIG. FIG. 2B shows a configuration in which, in the two phase shift circuits 30 shown in FIG. 7, the variable resistor 36 is replaced with a fixed resistor and the capacitor 34 is replaced with a variable capacitance diode.

In FIGS. 13 (A) and 13 (B), the capacitor connected in series to the variable capacitance diode changes its DC current when a reverse bias voltage is applied between the anode and the cathode of the variable capacitance diode. The impedance is extremely small at the operating frequency, that is, it has a large capacitance. In addition, since the potentials at both ends of the capacitor shown in FIGS. 13A and 13B are constant when the DC component is seen, by applying a reverse bias voltage larger than the amplitude of the AC component between the anode and the cathode, Each variable capacitance diode can function as a variable capacitance capacitor.

As described above, the capacitor 14 or 34 is composed of a variable capacitance diode, and the magnitude of the reverse bias voltage applied between the anode and the cathode is variably controlled so that the capacitance of the variable capacitance diode falls within a certain range. The amount of phase shift in each phase shift circuit can be changed by changing the value arbitrarily. Therefore, in each tuning amplifier, the frequency at which the amount of phase shift of the signal that goes around once becomes 0 ° can be changed, and the tuning frequency of the tuning amplifier can be arbitrarily changed.

By the way, although the variable capacitance diode is used as the variable capacitance element in FIGS. 13A and 13B described above, the FET in which the source and the drain are connected to a fixed potential in a direct current and a variable voltage is applied to the gate is used. May be used. As described above, since the potentials across the variable capacitance diodes shown in FIGS. 13A and 13B are fixed in terms of direct current, these variable capacitance diodes are
It is only necessary to replace it with T, and the gate capacitance, that is, the electrostatic capacitance of the FET can be changed by changing the voltage applied to the gate.

Further, although only the electrostatic capacitance of the variable capacitance diode is changed in FIGS. 13A and 13B described above, the resistance value of the variable resistor 16 or 36 may be changed at the same time. FIG. 13C shows a configuration in which the variable resistor 16 is used and the capacitor 14 is replaced with a variable capacitance diode in the two phase shift circuits 10 shown in FIG. In the same figure (D), in the two phase shift circuits 30 shown in FIG.
A configuration is shown in which the variable resistor 36 is used and the capacitor 34 is replaced with a variable capacitance diode. It goes without saying that the variable capacitance diode may be replaced with an FET having a variable gate capacitance.

Needless to say, the variable resistance shown in FIGS. 13C and 13D can be formed by utilizing the channel resistance of the FET as shown in FIG. In particular,
When a p-channel FET and an n-channel FET are connected in parallel to form one variable resistor and gate voltages of equal magnitude and different polarities are applied between the base and substrate of each FET, Since the nonlinear region can be improved, the distortion of the tuning signal can be reduced.

As described above, even when the variable resistance and the variable capacitance element are combined to form the phase shift circuit, the resistance value of the variable resistance and the capacitance of the variable capacitance element can be arbitrarily changed within a certain range. The amount of phase shift in each phase shift circuit can be changed. Therefore, in each tuning amplifier, the frequency at which the amount of phase shift of the signal that goes around once becomes 0 ° can be changed, and the tuning frequency of the tuning amplifier can be arbitrarily changed.

In addition to the case where the variable resistance or the variable capacitance element is used as described above, a plurality of resistors or capacitors having different element constants are prepared and the switch is switched to select from among the plurality of elements. One or more may be selected. In this case, the element constants can be discontinuously switched by the number of elements to be connected and the connection method (serial connection, parallel connection, or a combination thereof) by switching the switches. For example, instead of a variable resistor, a plurality of 2 n-th series resistors whose resistance values are R, 2R, 4R, ...
By selecting one or an arbitrary plurality and connecting them in series, it is possible to easily realize switching of resistance values at equal intervals with a smaller number of elements. Similarly, instead of the capacitors, prepare a plurality of 2 n-th series capacitors having capacitances of C, 2C, 4C, ... And select one or an arbitrary plurality of capacitors for parallel connection. Thus, it is possible to easily realize the switching of the electrostatic capacitances at equal intervals with a smaller number of elements. Therefore, the tuning amplifier of this embodiment is applied to a circuit having a plurality of tuning frequencies, for example, an AM radio, and is suitable for the purpose of selecting and receiving one station from a plurality of broadcasting stations.

When the tuning amplifier 1 of each of the above-mentioned embodiments is formed on a semiconductor substrate, a capacitor is practically used.
It is not possible to set a large capacitance as 14 or 34. Therefore, if the small capacitance of the capacitor actually formed on the semiconductor substrate can be apparently increased by devising the circuit, the time constant T can be set to a large value to reduce the tuning frequency. It is convenient for.

FIG. 14 shows the phase shift circuits 10 and 30 shown in FIG.
FIG. 11 is a diagram showing a modified example in which the capacitor 14 or 34 used for is composed of a circuit instead of a single element, and functions as a capacitance conversion circuit that makes the capacitance of the capacitor actually formed on the semiconductor substrate look large. To do. In addition,
The entire circuit shown in FIG. 14 corresponds to the capacitor 14 or 34 included in the phase shift circuit 10 or 30.

The capacitance conversion circuit 14a shown in FIG. 14 includes a capacitor 210 having a predetermined capacitance C0, two operational amplifiers 212 and 214, and four resistors 216, 218, 220 and 222. Has been done.

In the first-stage operational amplifier 212, a resistor 218 (whose resistance value is R18) is connected between the output terminal and the inverting input terminal.
It is grounded through 216 (this resistance is R16).

Between the voltage E1 applied to the non-inverting input terminal of the first stage operational amplifier 212 and the voltage E2 appearing at the output terminal,

[Equation 22] There is a relationship. The first-stage operational amplifier 212 mainly functions as a buffer that performs impedance conversion, and the gain may be 1. R18 when the gain is 1
When / R16 = 0, that is, R16 is set to infinity (removing the resistor 216), or R18 is set to 0Ω (direct connection).

Further, the operational amplifier 214 of the second stage has a resistor 222 between the output terminal and the inverting input terminal.
And a resistor 220 (whose resistance value is R20) is connected between the inverting input terminal and the output terminal of the operational amplifier 212 described above, and the non-inverting input terminal is grounded. There is.

When the voltage appearing at the output terminal of the second stage operational amplifier 214 is E3, the voltage E3 between this voltage E3 and the voltage E2 appearing at the output terminal of the first stage operational amplifier 212 is:

(Equation 23) There is a relationship. In this way, the second-stage operational amplifier 214 functions as an inverting amplifier, and the first-stage operational amplifier 212 is set in order to set its input side to high impedance.
Is used.

Further, the non-inverting input terminal of the first-stage operational amplifier 212 and the second-stage operational amplifier 2 thus connected are connected.
As described above, the capacitor 210 having a predetermined electrostatic capacity is connected between the 14 output terminals.

In the capacitance conversion circuit 14a shown in FIG. 14, assuming that the transfer function of the entire circuit excluding the capacitor 210 is K4, the capacitance conversion circuit 14a can be represented by the system diagram shown in FIG. FIG. 16 is a system diagram in which this is converted by the Miller's theorem.

When the impedance Z0 shown in FIG. 15 is used to represent the impedance Z1 shown in FIG.

[Equation 24] Becomes Here, the capacitance conversion circuit 14a shown in FIG.
In the case of, the impedance Z0 = 1 / (jωC0), and by substituting this into the equation (24),

(Equation 25)

(Equation 26) Becomes In this equation (26), the capacitance C0 of the capacitor 210 in the capacitance conversion circuit 14a is apparently (1-
K4) shows that it has doubled.

Therefore, when the gain of the amplifier is negative, K4 is always larger than 1, so that the capacitance C0 can be changed to the larger one.

By the way, the gain of the amplifier in the capacitance conversion circuit 14a shown in FIG.
The gain K4 of the amplifier constituted by 214 as a whole is given by the following equations (22) and (23):

[Equation 27] Becomes Substituting equation (27) into equation (26),

[Equation 28] Becomes Therefore, by setting the resistance value of the four resistors 216, 218, 220, 222 to a predetermined value, the two terminals 22
The apparent capacitance C between 4 and 226 can be increased.

When the gain of the amplifier by the first-stage operational amplifier 212 is 1, that is, when R16 is set to infinity (resistor 216 is removed) or R18 is set to 0Ω as described above, R18 / R16 If = 0, then (2
Equation 8 is simplified to

[Equation 29] Becomes

FIG. 17 is a diagram showing the configuration of the capacitance conversion circuit 14b in which the resistor 216 connected to the inverting input terminal of the first operational amplifier 212 shown in FIG. 14 is removed. In this case, since the electrostatic capacitance C appearing between the terminals 224 and 226 is expressed by the equation (28), it is possible to change it from C0 to the larger one simply by changing the ratio of R22 and R20.

As described above, the capacitance conversion circuit 14 described above is used.
a or 14b is the resistance ratio R22 / R between the resistance 220 and the resistance 222.
20 or by changing the resistance ratio R18 / R16 of the resistor 216 and the resistor 218, the electrostatic capacitance C0 of the capacitor 210 actually formed on the semiconductor substrate can be converted to an apparently larger one. Therefore, when the entire tuning amplifier 1 shown in FIG. 1 or the like is formed on a semiconductor substrate, a capacitor 210 having a small electrostatic capacitance C0 is formed on the semiconductor substrate, and the capacitor 210 shown in FIG. The circuit shown in FIG. 17 allows conversion into a large capacitance C, which is convenient for integration. In particular, if a large capacitance can be secured in this way, the overall mounting area of the tuning amplifier 1 shown in FIG. 1 can be downsized, and the material cost can be reduced.

Also, at least one of the resistors 216, 218, 220, 222 (at least one of the resistors 220, 222 in the case of the capacitance conversion circuit 14b shown in FIG. 17) is formed by a variable resistor. Therefore, specifically, the junction type or MOS type F
By connecting the ET or p-channel FET and the n-channel FET in parallel to form a variable resistance, it is possible to easily form a capacitor having a variable capacitance.
Therefore, by using this capacitor instead of the variable capacitance diode shown in FIG. 13, the phase shift amount can be arbitrarily changed within a certain range. Therefore, it is possible to change the frequency at which the phase shift amount of the signal that makes one round in the tuning amplifier becomes 0 °, and it is possible to arbitrarily change the tuning frequency of the tuning amplifier of each embodiment.

Since the first-stage operational amplifier 212 is used as a buffer for increasing the input impedance as described above, the operational amplifier 212 may be replaced with an emitter follower circuit or a source follower circuit.

FIG. 18 is a diagram showing the configuration of a capacitance conversion circuit 14c using an emitter follower circuit in the first stage. The capacitance conversion circuit 14c shown in the figure includes an emitter follower circuit 22 including a first-stage operational amplifier 212 and two resistors 216 and 218 shown in FIG.
It has a configuration replaced with 8.

FIG. 19 is a diagram showing the structure of a capacitance conversion circuit 14d using a source follower circuit in the first stage. The capacitance conversion circuit 14d shown in the figure has a configuration in which the first-stage operational amplifier 212 and the two resistors 216 and 218 shown in FIG. 14 are replaced with a source follower circuit 230 including an FET and a resistor.

The capacitance conversion circuits 14c and 14d described above are also included.
Each of the resistors 22 connected to the operational amplifier 214
The point that the apparent capacitance C between the terminals 224 and 226 can be arbitrarily changed by changing the resistance ratio of 0 and 222 is the same as the capacitance conversion circuit 14a and the like shown in FIG. .. Therefore, at least one of the resistors 220 and 222 is connected to a junction-type or MOS-type FET or p-channel FET and n-type.
A capacitance variable capacitor can be constructed by replacing the channel FET with a variable resistor connected in parallel. By using this capacitor instead of the variable capacitance diode shown in FIG. 13, the phase shift amount can be increased. Can be arbitrarily changed within a certain range. Therefore, in each tuning amplifier, the frequency at which the amount of phase shift of the signal that goes around once becomes 0 ° can be changed, and the tuning frequency of the tuning amplifier of each embodiment can be arbitrarily changed.

The present invention is not limited to the above embodiment, and various modifications can be made within the scope of the gist of the present invention.

For example, in each tuning amplifier shown in FIG. 1 or the like, the feedback resistor 70 having a fixed resistance value is used as the feedback impedance element, and the input resistor 74 having a fixed resistance value is used as the input impedance element. However, at least one of the resistors may be a variable resistor so that the maximum attenuation amount can be changed arbitrarily. In this case, it goes without saying that the variable resistor can be formed by utilizing the channel resistance of the FET as shown in FIG. In particular, when a p-channel FET and an n-channel FET are connected in parallel to form one variable resistor and gate voltages of equal magnitude and different polarities are applied between the base and substrate of each FET, Since the non-linear region of the FET can be improved, the distortion of the tuning signal can be reduced.

Similarly, when the feedback impedance element and the input impedance element are capacitors, at least one of them is composed of a variable capacitance diode or a gate capacitance variable FET so that the maximum attenuation can be changed arbitrarily. Good.

Further, although the tuning amplifier 1 or 1a of each of the above-described embodiments includes two phase shift circuits, when varying the tuning frequency, the CR circuits included in both phase shift circuits are used. In addition to the case where the element constant of at least one of the resistance and the capacitor included is changed, the case where the element constant of at least one of the resistance and the capacitor included in the CR circuit included in the one phase shift circuit is changed can be considered. In this case, it is easier to change the element constant of the element whose one end is grounded in either one of the phase shift circuits. Also, by fixing each element constant of all resistors and capacitors,
It is also possible to construct a tuning amplifier with a fixed tuning frequency.

When the tuning amplifier of each embodiment is integrated on a semiconductor substrate, electrodes are formed by sandwiching an insulating film such as a silicon oxide film or the gate capacitance of the FET is used as described above. To form a capacitor in the phase shift circuit.

Further, in each of the above-mentioned embodiments, a circuit with high stability can be constructed by constructing the phase shift circuits 10 and 30 by using the operational amplifier, but it is used as in this embodiment. In this case, since the offset voltage and the voltage gain are not required to have high performance, a differential input amplifier having a predetermined amplification degree may be used instead of the operational amplifier in each phase shift circuit.

FIG. 20 is a circuit diagram in which a portion necessary for the operation of the phase shift circuit of each embodiment is extracted from the configuration of the operational amplifier, and the whole operates as a differential input amplifier having a predetermined amplification degree. The differential input amplifier shown in the figure has a differential input stage 100 composed of FETs, a constant current circuit 102 for supplying a constant current to the differential input stage 100, and a predetermined bias voltage for the constant current circuit 102. Bias circuit 104 and differential input stage 100
And an output amplifier 106 connected to. As shown in the figure, the offset adjusting circuit and the like included in the actual operational amplifier can be omitted to simplify the configuration of the differential input amplifier. Since the upper limit of the operating frequency can be increased by simplifying the circuit in this way, the upper limit of the operating frequency of the tuning amplifier 1 and the like configured by using this differential input amplifier is correspondingly increased. You can

[0107]

As is clear from the description based on the above embodiments, each element constituting the tuning amplifier of the present invention can be formed by an integrated circuit manufacturing method.
The tuning amplifier can be miniaturized as an integrated circuit on a semiconductor wafer, and can be manufactured inexpensively by mass production.

Particularly, the channel between the source and drain of the FET is used as the variable resistance of the CR circuit in each phase shift circuit, and the resistance of the channel is changed by changing the control voltage applied to the gate of this FET. Then, it is possible to avoid the influence of the inductance and the capacitance of the wiring to which the control voltage is applied, and it is possible to obtain a tuned amplifier having ideal characteristics almost as designed.

In the tuning amplifier of the present invention, the maximum amount of attenuation is determined by the resistance ratio of the input side impedance element and the feedback side impedance element, and the tuning frequency is determined by the time constant of the CR circuit in each phase shift circuit. The attenuation amount, the tuning frequency, and the gain at the tuning frequency can be set without interfering with each other.

In the conventional tuning amplifier utilizing LC resonance, the tuning frequency ω is 1 / √LC. Therefore, when the capacitance C or the inductance L is changed to adjust the tuning frequency, the tuning frequency is changed. Although it changes in proportion to the square root of the amount of change, in the tuning amplifier of the present invention, since the tuning frequency ω is 1 / (CR), the tuning frequency should be changed in proportion to the resistance value R or the capacitance C. Therefore, it is possible to make large changes and adjustments.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration of a tuning amplifier of a first embodiment to which the present invention is applied.

FIG. 2 is a diagram showing an extracted configuration of a phase shift circuit shown in FIG.

FIG. 3 is a vector diagram showing a relationship between an input / output voltage of a phase shift circuit and a voltage appearing in a capacitor or the like.

FIG. 4 is a system diagram in which the entire two phase shift circuits are replaced with a circuit having a transfer function K1.

FIG. 5 is a system diagram obtained by converting the system shown in FIG. 4 according to Miller's theorem.

FIG. 6 is a diagram showing a tuning characteristic of the tuning amplifier of the first embodiment.

FIG. 7 is a diagram showing a configuration of a tuning amplifier of a second embodiment.

FIG. 8 is a diagram showing an extracted configuration of the phase shift circuit shown in FIG. 7.

FIG. 9 is a vector diagram showing a relationship between an input / output voltage of a phase shift circuit and a voltage appearing in a capacitor or the like.

FIG. 10 is a diagram showing a connection form of a phase shift circuit and a phase inversion circuit.

FIG. 11 is a diagram showing a connection form of a phase shift circuit and a phase inversion circuit.

FIG. 12 is a diagram showing a configuration of a phase shift circuit in which a variable resistance of the phase shift circuit is replaced with an FET.

FIG. 13 is a diagram showing a configuration of a phase shift circuit in which a capacitor of the phase shift circuit is replaced with a variable capacitance diode.

FIG. 14 is a diagram showing a configuration of a capacitance conversion circuit that apparently increases the capacitance that a capacitor actually has.

FIG. 15 is a diagram showing the circuit shown in FIG. 14 using a transfer function.

16 is a diagram in which the configuration shown in FIG. 15 is converted by the mirror theorem.

17 is a diagram showing a configuration of a capacitance conversion circuit which is a simplified version of the circuit of FIG.

FIG. 18 is a diagram showing a configuration of a capacitance conversion circuit using an emitter follower circuit in the first stage.

FIG. 19 is a diagram showing a configuration of a capacitance conversion circuit using a source follower circuit in the first stage.

FIG. 20 is a circuit diagram in which a portion necessary for the operation of the phase shift circuit of the present invention is extracted from the configuration of the operational amplifier.

FIG. 21 is a characteristic curve diagram showing an example of the relationship between the tuning frequency, the gain at the tuning frequency, and the maximum attenuation amount in the conventional tuning amplifier.

[Explanation of symbols] 1 Tuning amplifier 10, 30 Phase shift circuit 12, 32 Operational amplifier 14, 34 Capacitor 16, 36 Variable resistance 18, 20, 38, 40 Resistance 70 Feedback resistance 74 Input resistance 80 Phase inversion circuit 90 Input terminal 92 Output Terminal

Claims (25)

[Claims] 1. An input-side impedance element having an input terminal to which an AC signal input to an input terminal is input, and a feedback-side impedance element having a feedback signal input to one end, the input-side impedance element being input to the input terminal. An addition circuit for adding the AC signal and the feedback signal, a differential input amplifier having one end of the first resistor connected to the inverting input terminal, an inverting input terminal and an output terminal of the differential input amplifier, A second resistor connected between the first resistor and a series circuit including a third resistor and a capacitor connected to the other end of the first resistor, and a connecting portion of the third resistor and the capacitor. Two phase shift circuits connected to the non-inverting input terminals of the differential input amplifier, and a phase inverting circuit that inverts and outputs the phase of the input AC signal are provided. Inversion circuit Each of them is connected in cascade, and the signal added by the adder circuit is input to the first-stage circuit of the plurality of circuits connected in cascade, and the signal output from the last-stage circuit is input. A tuning amplifier, wherein the feedback signal is input to one end of the impedance element on the feedback side, and an output of any one of the plurality of circuits is taken out as a tuning signal. 2. The tuning amplifier according to claim 1, wherein the third resistor and the capacitor forming the series circuit are connected in the same manner in the two phase shift circuits. 3. The tuning amplifier according to claim 1, wherein the differential input amplifier is an operational amplifier. 4. The tuning amplifier according to claim 1, wherein each of the input impedance element and the feedback impedance element is a resistor. 5. The device according to claim 4, wherein at least one of the input impedance element and the feedback impedance element is formed by a variable resistor, and the resistance ratio of the input impedance element and the feedback impedance element is changed. A tuned amplifier characterized by changing maximum attenuation. 6. The tuning amplifier according to claim 1, wherein each of the input impedance element and the feedback impedance element is a capacitor. 7. The capacitance element according to claim 6, wherein at least one of the input impedance element and the feedback impedance element is formed of a variable capacitance element, and a capacitance ratio of the input impedance element and the feedback impedance element is changed. A tuned amplifier characterized in that the maximum attenuation is changed by changing the maximum attenuation. 8. The tuning frequency according to claim 1, wherein the third resistor included in at least one of the two phase shift circuits is formed by a variable resistor, and the resistance value is changed. A tuning amplifier characterized by being changed. 9. The tuning amplifier according to claim 5, wherein the variable resistance is formed by a channel of an FET, and a channel voltage is changed by changing a gate voltage. 10. The variable resistance according to claim 5, wherein the variable resistor is formed by connecting a p-channel type FET and an n-channel type FET in parallel, and changing the magnitude of the gate voltage of each FET having a different polarity. A tunable amplifier characterized by changing the channel resistance. 11. The tuning frequency is changed according to claim 1, wherein the capacitor included in at least one of the two phase shift circuits is formed of a variable capacitance element, and the capacitance is changed. A tuned amplifier characterized by: 12. The tuning amplifier according to claim 7, wherein the variable capacitance element is formed of a variable capacitance diode whose reverse bias voltage can be changed or an FET whose gate capacitance can be changed by changing the gate voltage. . 13. The method according to claim 1, wherein the third resistor included in at least one of the two phase shift circuits has a plurality of resistors having a fixed resistance value, and is switched by a switch. A tuning amplifier characterized in that the tuning frequency is changed by selectively connecting. 14. The capacitor according to claim 1, wherein the capacitors included in at least one of the two phase shift circuits have a plurality of capacitors having a fixed electrostatic capacitance, and are selectively switched by switching. A tuning amplifier characterized in that the tuning frequency is changed by connecting to the. 15. The capacitor according to claim 1, wherein the capacitor included in at least one of the two phase shift circuits is connected in parallel between an amplifier having a negative gain and an input / output of the amplifier. The tuned amplifier is characterized in that the capacitance viewed from the input side of the amplifier is made larger than the capacitance actually possessed by the capacitor element by replacing the capacitor element. 16. The tuning amplifier according to claim 15, wherein the tuning frequency is changed by changing the gain of the amplifier to change the capacitance viewed from the input side of the amplifier. 17. A phase inverting circuit for inverting and outputting the phase of an input AC signal, an operational amplifier, a time constant circuit including a resistor and a capacitor to which the input AC signal is applied, and the time constant circuit. A circuit for inputting a signal generated at the non-inverting input terminal of the operational amplifier, an input resistance connected to the inverting input terminal of the operational amplifier, to which an input signal is applied, and an output terminal and an inverting input terminal of the operational amplifier. A two-stage phase shift circuit for shifting an AC signal in the same direction, a cascade connection including the phase inversion circuit and the two-stage phase shift circuit, and the cascade connection. A feedback-side impedance element connected between an output and an input of the connection, and an input circuit for inputting an AC signal to the cascade connection via the input-side impedance element, the tuning amplifier. Vessel. 18. The tuning according to claim 17, wherein the tuning frequency is changed by changing the resistance of the time constant circuit of the preceding phase shift circuit and / or the resistance of the time constant circuit of the subsequent phase shift circuit. amplifier. 19. The tuned amplifier according to claim 17, wherein the maximum attenuation amount is adjusted by changing the ratio of the input impedance and the feedback impedance. 20. The tuning amplifier according to claim 17, wherein the resistance of each time constant circuit is formed by a channel of the FET. 21. A phase inverting circuit for inverting and outputting the phase of an input AC signal, an operational amplifier, a time constant circuit including a resistor and a capacitor to which the input AC signal is applied, and the time constant circuit. A circuit for inputting a signal generated at the non-inverting input terminal of the operational amplifier, an input resistance connected to the inverting input terminal of the operational amplifier, to which an input signal is applied, and an output terminal and an inverting input terminal of the operational amplifier. A two-stage phase shift circuit for shifting an AC signal in the same direction, a cascade connection including the phase inversion circuit and the two-stage phase shift circuit, and the cascade connection. A feedback side impedance element connected between the output and the input of the connection, an input circuit for inputting an AC signal to the cascade connection via the input side impedance element, and converting the capacitance of the capacitor That constant conversion circuit and, tuned amplifier, characterized in that it comprises a. 22. The tuning amplifier according to claim 21, wherein the tuning frequency is changed by changing the capacitance in the constant conversion circuit of the phase shift circuit of the front stage and / or the rear stage. 23. The tuned amplifier according to claim 21, wherein the maximum attenuation amount is changed by changing the ratio of the input impedance and the feedback impedance. 24. The tuning amplifier according to claim 21, wherein a resistor for changing the electrostatic capacitance is formed in the channel of the FET in the constant conversion circuit. 25. The tuning amplifier according to claim 1, which is formed as a semiconductor integrated circuit.