JPH0678552A - Control circuit for voltage inverter - Google Patents

Abstract

PURPOSE:To obtain an excellent protective function against an overcurrent condition which occurs on a load side, by providing a current regulating means which outputs a control signal to null the deviation of an inverter current value detected on the output side of the inverter from a control signal outputted by a voltage regulating means. CONSTITUTION:Concerning to a voltage inverter 3 for inverting DC power from a DC power source 2 into AC power, and supplying this AC power to a load 7 through the medium of an LC filter 4, a current regulating loop is provided on the inside of a voltage control loop containing a voltage regulator 18. Namely, an inverter current detector 21 is provided on the output side of the inverter 3, and the deviation of the inverter current iI detected by this detector from an inverter current command value iIS outputted by the voltage regulator 18 is inputted to the inverter current detector 21, and a signal for performing control to null the deviation of the inputs is outputted from the inverter current detector 21. In this way, overcurrent protection of the inverter 3 is performed by controlling the inverter 3 with the output signal of the inverter current detector 21.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control circuit for a voltage source inverter which has a filter composed of a reactor and a capacitor on the output side and which converts direct current into alternating current by instantaneous voltage control.

[0002]

2. Description of the Related Art FIG. 10 is a circuit diagram showing a conventional example of a control circuit of a voltage source inverter which is provided with a filter composed of a reactor and a capacitor on the output side to perform instantaneous voltage control. The operation of the conventional circuit shown in FIG. That is, the DC power from the DC power supply 2 is converted into AC power by the inverter 3, and this AC power is loaded through the LC filter 4 (composed of the filter reactor 5 and the filter capacitor 6) provided on the output side of the inverter 3 into the load 7 Is supplied to. Here detects the output voltage v _O of the LC filter 4 by the output voltage detector 11, via a rectifier 12 for converting the average value voltage v _M. By inputting the deviation between the average value voltage command value v _MS set by the voltage setter 13 and the average value voltage v _M to the average value voltage adjuster 14, the average value voltage adjuster 14 receives the input deviation. Outputs a signal that controls to zero. On the other hand, a sine wave signal having a desired frequency is obtained by the V / F converter 15 and the memory 16 which convert the voltage signal into a frequency signal corresponding to the voltage signal. The multiplier 17 calculates the product of this sine wave signal and the output signal of the average value voltage regulator 14 described above, and outputs the output voltage command value v _OS whose amplitude and frequency are desired values with a sine waveform.

The voltage regulator 18 uses this output voltage command value v _OS.
And an output voltage v _O detected by the output voltage detector 11 are input, and a control signal for making the input deviation zero by instantaneous value control is output. The inverter 3 uses this voltage regulator 18
Is supplied to the load 7 under the control of the output signal of AC.

[0004]

However, in the conventional circuit for controlling the inverter 3 shown in FIG. 10, protection is provided when an inrush current becomes excessive when the load 7 starts up, and when a load short circuit accident occurs. There is the inconvenience that it is not easy to add a protection function such as protection against short-circuit current. Further, when a load having a non-linear characteristic such as a rectifier is connected, there is a disadvantage that the distortion rate of the voltage waveform output via the LC filter 4 becomes large.

Therefore, an object of the present invention is to provide a voltage-type inverter of instantaneous value control which outputs a sinusoidal waveform AC with a protection function capable of coping with an overcurrent on the load side, and to distort the waveform even when the load has a non-linear characteristic. It is to be able to suppress the rate to a small value.

[0006]

In order to achieve the above object, a control circuit for a voltage source inverter according to the present invention is provided with a filter composed of a reactor and a capacitor on an AC output side of an inverter for converting DC into AC. The voltage adjusting means is provided which inputs a deviation between an output voltage detection value detected on the filter output side and an output voltage command value which is separately set, and outputs a control signal for making the input deviation zero. In the control circuit of the voltage source inverter which controls the inverter by the output signal of, the deviation between the control signal output by the voltage adjusting means and the inverter current detection value detected at the inverter output side is input, and this input deviation is calculated. A current adjusting means for outputting a control signal to make it zero is provided, and the inverter is controlled by the output signal of the current adjusting means. The command value, the output voltage detection value, and the inverter current detection value detected on the inverter output side are converted into two components of the synchronous rotation coordinate system based on the d axis and the q axis, respectively, and the output voltage command value and the output voltage for each axis are converted. Voltage adjusting means for separately outputting a control signal for making the deviation from the detected value zero, control signals for each axis output by the voltage adjusting means, and inverter current detection for each axis detected at the inverter output side Current control means for separately outputting a control signal for making the deviation from the value zero, and the d-axis output signals of these current control means and q
A control signal conversion means for inputting a shaft output signal and converting the control rate and the phase angle for controlling the inverter,
A non-interference network is separately provided on the output side of the voltage adjusting means and the current adjusting means. Alternatively, the voltage adjusting means calculates a quantity proportional to a deviation value between the output voltage detection value and an output voltage command value set separately, and a differential of the output voltage detection value detected on the filter output side. Voltage differential value calculation means for calculating a value, second gain for calculating an amount proportional to the voltage differential value, addition means for adding the output of the first gain and the output of the second gain, and the addition calculation The result is proportionally integrated with a proportional integral calculating means.

[0007]

A current controller is provided to control the inverter, but the voltage commander outputs the current command value to this current controller, so limiting the current command value causes the inverter output current to become excessive. By suppressing the current signal and the voltage signal at this time into a d-axis component and a q-axis component, which are rotational coordinate systems, respectively, and installing a non-interference circuit in the current adjusting circuit, these d-axis components The test adjustment is facilitated by controlling the and q-axis components by independent scalar quantities. Further, since the current command value to the current regulator is given by the voltage regulator of the voltage control loop provided outside the current control loop, a non-interference circuit is also installed in this voltage regulation circuit to facilitate the test adjustment. Further, the voltage adjusting means is of a variable structure type so as to perform so-called sliding state control, so that the target value can be tracked at high speed.

[0008]

1 is a circuit diagram showing a first embodiment of the present invention. The DC power supply 2, inverter 3, LC filter 4, load 7, output voltage detector shown in the first embodiment circuit are shown. 1
1, rectifier 12, voltage setting device 13, average value voltage regulator 1
4, the names / applications / functions of the V / F converter 15, the memory 16, the multiplier 17, and the voltage regulator 18 are the same as those in the conventional circuit described above with reference to FIG. To do.

In the circuit of the first embodiment, the voltage regulator 18
The current regulation loop is provided inside the voltage control loop whose component is. That is, an inverter current detector 21 is provided on the output side of the inverter 3, and the deviation between the inverter current i _I detected by the inverter current detector 21 and the inverter current command value i _IS output by the voltage regulator 18 is calculated by the inverter. By inputting to the current detector 21, this inverter current detector 21 outputs a signal for controlling the input deviation to zero. The inverter 3 is controlled by the output signal of the inverter current detector 21. Therefore, by limiting the above-described inverter current command value i _IS , the overcurrent protection of the inverter 3 can be easily performed.

FIG. 2 is a circuit diagram showing a second embodiment of the present invention. The DC power supply 2, the inverter 3, the LC filter 4, the load 7, the output voltage detector 11 shown in the circuit of the second embodiment. , Rectifier 12, voltage setting device 13, average value voltage regulator 14, V / F converter 15, memory 16, and multiplier 1
The name, application, and function of 7 are the same as those in the conventional circuit described above in FIG. 10, and the names, applications, and functions of the inverter current detector 21 and the current regulator 22 are the same as those of the first embodiment described above with reference to FIG. The description is omitted because it is the same as the case of the circuit.

In the circuit of the second embodiment, an output current detector 23 for detecting the output current i _O is provided on the output side of the LC filter 4, and the voltage regulator 30 having a variable structure is used instead of the voltage regulator 18. Equipped with. This voltage regulator 30 constrains the control response to the switching straight line by the sum of the amount proportional to the differential value of the output voltage v _O of the output voltage v _O Toko, by a so-called slip state control, as non-linear loads Since a large change in the load parameter is suppressed, a great effect can be obtained in improving the waveform distortion of the load voltage. Details of the variable structure type voltage regulator 30 will be described later.

FIG. 3 shows the LC shown in the second embodiment circuit of FIG.
It is the circuit diagram which described the voltage / current of each part of a filter, and is showing the filter reactor 5, the filter capacitor 6, and the load 7 by the three-phase circuit called U, V, and W. When the inductance of the filter reactor 5 is L and the capacitance of the filter capacitor 6 is C, the following equation 1
And the number 2 holds.

[0013]

[Equation 1]

[0014]

[Equation 2]

However, the inductance L and the capacitance C described in the formulas 1 and 2 are the formula 3, and the inverter voltage (phase voltage) v _I , the output voltage (phase voltage) v _O, and the potential v _{N with} respect to the inverter neutral point voltage. Is expressed by Equation 4, and the inverter current i _I , output current i _O, and capacitor current i _C are expressed by Equation 5, respectively.

[0016]

[Equation 3]

[0017]

[Equation 4]

[0018]

[Equation 5]

Note that v _IU = v _UN + v _N, v _IV = v _VN + v
_N, v _IW = v _WN + v _N. If the three phases are balanced, v _UN + v _VN + v _WN = 0, so the potential v _{N with} respect to the inverter neutral point voltage shown in Formula ₂ is as shown below. v _N = (v _IU + v _IV + v _WN ) / 3 Therefore, the above-mentioned equation 2 becomes the following equation 6. However, G described in Expression 6 becomes as shown in Expression 7.

[0020]

[Equation 6]

[0021]

[Equation 7]

The three-phase / two-phase conversion matrix C ₀ is shown in the following _equation 8. By using this conversion matrix C ₀ , the voltage V and the current I are converted into the dq axis coordinate system as shown in the following Expressions 9 and 10.

[0023]

[Equation 8]

[0024]

[Equation 9]

[0025]

[Equation 10]

[Mathematical formula-see original document] Here, from the equations 9 and 10, the aforementioned equation 6 becomes the following equation 11. Therefore, Equation 12 is obtained.

[0027]

[Equation 11]

[0028]

[Equation 12]

However, from the following equations 13 and 14, equation 15 is established. As a result, the above formula 12 becomes the following formula 16.

[0030]

[Equation 13]

[0031]

[Equation 14]

[0032]

[Equation 15]

[0033]

[Equation 16]

Similarly, since the above-mentioned equation 1 becomes equation 17, the result is equation 18.

[0035]

[Equation 17]

[0036]

[Equation 18]

However, since it is the following equation 19, the equation 17
Is the number 20.

[0038]

[Formula 19]

[0039]

[Equation 20]

From these equations 16 and 20, the following equation 21,
Equation 22 is obtained.

[0041]

[Equation 21]

[0042]

[Equation 22]

Two-axis conversion of the output voltage is performed by the following equations 23 and 2
It is 4.

[0044]

[Equation 23]

[0045]

[Equation 24]

Since V _C = V _O , the following equation 2
5 is obtained. Therefore, I _C becomes as shown in Expression 26.

[0047]

[Equation 25]

[0048]

[Equation 26]

Therefore, the voltage setting value v in the dq coordinate system
_S becomes as shown below. v _S = [v _Sd , v _Sq ] = [√3 / 2E, 0] From the above Expression 21, the following Expression 27, and therefore Expression 28, are obtained.

[0050]

[Equation 27]

[0051]

[Equation 28]

FIG. 11 is an explanatory diagram for explaining a conventional example of two-axis conversion of an LC filter.
Is represented in this illustration. These equations 27 and 28 are interference systems, and v _C = [v _Cd , v _Cq ] ^T is a disturbance.
Here, in order to make the biaxial components non-interfering, it is sufficient to insert a circuit network having a transfer characteristic F shown in Expression 29 into FIG. 11, and as a result, Expression 30 is obtained.

[0053]

[Equation 29]

[0054]

[Equation 30]

As a result, the biaxial components are independent. This output signal of the current regulator 22 constituting the current control loop u = [u _d, u _q] means a performing an operation shown in Formula 31 with respect to ^T, the number 31 is the number 32 It can be transformed into

[0056]

[Equation 31]

[0057]

[Equation 32]

FIG. 4 is an explanatory view of the biaxial conversion of the LC filter showing the third embodiment of the present invention. Here, _d (u d)
If / dt = u _dA and d (u _q ) / dt = u _qA , then the second non-interference circuit having the transfer characteristic F ₁ (see Eq. 32) at the output of the current regulator 22 as shown in FIG. by installing the network 50, the output of the current regulator 22 u _dA, the u _qA.
However, in FIG. 4, f = 1 and g = ω. Current regulator 2
Reference numeral 2 is a proportional-integral type, and currents I _d and I _q converted by two axes are
Is controlled so as to match the current command values I _Sd and I _Sq .
The output voltage command values v _Sd and v _Sq on the dq axes are output to the output u of the current regulator 22 via the second non-interference network 50 shown in FIG.
It can be _calculated from _dA and u _qA .

FIG. 6 is an explanatory view for explaining the conversion of the signal for controlling the pulse width modulation of the inverter, in which the pulse width of the inverter is changed from the output voltage command values v _Sd and v _{Sq on the} dq axes obtained as described above. The control signal conversion circuit 70 performs the calculation for obtaining the control rate λ and the phase angle θ for the modulation control. That is, the pulse width modulation control signal of the inverter is obtained by independently controlling the biaxial components of the output voltage.

Next, output current command values i _Sd , i on the dq axes
_Sq is created by the following. That is, the following expression 33 and thus expression 34 are obtained from the above expression 20.

[0061]

[Expression 33]

[0062]

[Equation 34]

FIG. 12 is an explanatory diagram for explaining a conventional example of 2-axis conversion of an LC filter, which is the above-mentioned formula 33 or formula 34.
Is represented in this illustration. These equations 33 and 34 are an interference system, and the output current i _o = [i _od , i _oq ] ^T becomes a disturbance. Here, in order to make the two axis components non-interfering, it suffices to insert a circuit network having a transfer characteristic F shown in Formula 35 into FIG. 12, as in the above-described current adjusting system. 36 is obtained.

[0064]

[Equation 35]

[0065]

[Equation 36]

As a result, the biaxial components are independent. This is meant to applying operation shown in Formula 37 with respect to the output signal w = [w _d, w _q] ^T of the voltage regulator 30 constituting the voltage control loop, this number 37 number 38 It can be transformed into

[0067]

[Equation 37]

[0068]

[Equation 38]

Similar to FIG. 4, FIG. 5 is an explanatory view of the biaxial conversion of the LC filter showing the third embodiment of the present invention. F Here, the differential value of w _d and w _dA, the differential value of w _q When w _qA, the output of the voltage regulator 30, as shown in FIG. 5
_If the first non-interfering network 40 having the transfer characteristic of ₂ (see Eq. 38) is installed, the output of the voltage regulator 30 becomes w _dA and w _qA , respectively. However, in FIG. 5, m = 1 and n = ω.

FIG. 7 is a circuit diagram showing a fourth embodiment of the present invention and shows the configuration of the voltage regulator 30. d-q A first gain 31 (gain C _V ) for amplifying the deviation between the voltage setting value v _OS of each axis and the component voltage actual value v _{o of} each axis is provided, and C is the calculation result of the deviation and the gain. Obtain _V (v _S −v). On the other hand, the capacitor current i _{C is set} to the second gain 32.
Since K _C i _C is obtained through (gain K _C ), if the sum of these two is S _L , this S _L is
It is represented by.

[0071]

[Formula 39]

[0072] The S _L a by proportional integral calculation in the proportional-plus-integral calculator 33, the inverter current command value i _IS represented by the number 40 is obtained.

[0073]

[Formula 40]

The inverter current command value i _IS calculated by the equation 40 is used as the current command value for each axis. Further, the capacitor current i _C is obtained from i _Cd = i _Id -i _Od and i _Cq = i _Iq -i _Oq . Here, the capacitor current i _C becomes as shown in Formula 41.

[0075]

[Formula 41]

Therefore, if v _S −v _O = Δv, then
The value S _L shown in Expression 39 can be expressed by Expression 42 described below.

[0077]

[Equation 42]

That is, S _L is proportional-integral controlled as shown in the equation 40, and S _L → 0 is always maintained. That is, the constraint control is performed at S _L = 0. The three-phase / two-phase conversion is converted into the α-β coordinate system by the conversion matrix C ₁ shown in the following Expression 43.
The voltage v _(ab) and the current i _(ab) in the α-β coordinate system are as shown in Formula 44.

[0079]

[Equation 43]

[0080]

[Equation 44]

Since these v _(ab) and i _(ab) converted into the α-β coordinate system are vectors that rotate at the angular frequency ω, they are converted into d- by using the conversion matrix C _{2 shown} in the equation 45. By converting to the q-axis rotating coordinate system, the voltage v
_(dq) and the current i _(dq) are as shown in Formula 46.

[0082]

[Equation 45]

[0083]

[Equation 46]

At this time, since C ₀ = C ₁ C ₂ , this conversion matrix C ₀ is expressed by the above-mentioned _equation 8.
As a result of the above-mentioned conversion, the three-phase set voltage v _S is expressed by Equations 47 and 48 in the α-β coordinate system.

[0085]

[Equation 47]

[0086]

[Equation 48]

In the dq coordinate system, the following equation 49 is obtained.

[0088]

[Equation 49]

FIG. 8 is a block diagram for executing the conversion from three phases to two phases, and expresses the operations of the equations 43 and 45 described above. The 3-phase / 2-phase conversion circuit shown in FIG.
Three amplifiers 1, 62, 63 and 64, 65, 6
It is composed of four multipliers 6, 67, which convert the three-phase voltage / current to the dq axes.
Cos ωt and sin ωt described in the equation 45 are external signals,
A two-phase sine wave oscillator 60 controls the frequency. When the external signal is fixed, the internal synchronization is achieved. Therefore, if the two-phase sine wave oscillator 60 is made to follow the frequency of the commercial power source by using, for example, a phase locked loop circuit, the commercial synchronization is achieved.

Here, if there is a phase difference of θ between the three-phase voltages E _R , E _S , and E _T and the output of the two-phase sine wave oscillator 60, the coordinate axes rotating at the commercial power supply frequency, that is, dq The components of these three-phase voltages E _R , E _S , and E _{T on} the axis are E _d ,
The value of E _q is E _d = E ₁ · E ₂ cos ωt E _q = E ₁ · E ₂ sin ωt, which is a signal on the dq coordinate axes.

[0091]

According to the present invention, in controlling the voltage source inverter, a current control loop is provided inside the voltage control loop to limit the current command value given from the voltage control loop to the current control loop. Therefore, an effect of facilitating protection against an overcurrent state generated on the load side, for example, a short-circuit accident or a large inrush current when the load starts, that is, current protection, can be obtained.

These voltage / current signals are decomposed into so-called rotating coordinate system dq axis components, and a current controller forming a current control loop with the output side of the voltage controller forming a voltage control loop. Since the non-interference network is separately provided on the output side of each, the d-axis component and the q-axis component can be independently controlled by the scalar amount, so that the test and adjustment of these control loops can be easily performed. The effect can be obtained.

Further, the voltage adjusting means has a variable structure so as to perform a so-called sliding state control. FIG. 9 is a graph showing the response of the variable structure type voltage regulator. As shown in the graph of FIG. 9, even if the operating point is at the point P due to the slip control, the operation is switched to the straight line S _L = 0. It is constrained on top of it and is controlled toward the origin. On the straight line of S _L = 0, the differential equation shown in the above equation 42 is followed, so that it has nothing to do with the parameter of the load. Therefore, even when a non-linear characteristic load such as a rectifier is connected, the output voltage is always instantaneously controlled toward the target value, and the distortion factor of the output voltage waveform can be suppressed to a small value. Then, when the control is performed toward the origin on the straight line S _L = 0, the voltage deviation Δv converges on the origin without causing an excessive control. That is, the output voltage quickly follows the target value without causing a time delay.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a first embodiment of the present invention.

FIG. 2 is a circuit diagram showing a second embodiment of the present invention.

FIG. 3 is a circuit diagram showing voltages and currents of respective parts of the LC filter shown in the second embodiment circuit of FIG.

FIG. 4 is an LC filter 2 showing a third embodiment of the present invention.
Illustration of axis conversion

FIG. 5 is a second LC filter showing a third embodiment of the present invention.
Illustration of axis conversion

FIG. 6 is an explanatory diagram illustrating conversion of a pulse width modulation control signal of an inverter.

FIG. 7 is a circuit diagram showing a fourth embodiment of the present invention.

FIG. 8 is a block diagram for performing conversion from three phases to two phases.

FIG. 9 is a graph showing the response of the variable structure type voltage regulator.

FIG. 10 is a circuit diagram showing a conventional example of a control circuit of a voltage source inverter that is provided with a filter composed of a reactor and a capacitor on the output side and performs instantaneous voltage control.

FIG. 11 is an explanatory diagram illustrating a conventional example of two-axis conversion of an LC filter.

FIG. 12 is an explanatory diagram illustrating a conventional example of two-axis conversion of an LC filter.

[Explanation of symbols]

2 DC power supply 3 Inverter 4 LC filter 5 Filter reactor 6 Filter capacitor 7 Load 11 Output voltage detector 12 Rectifier 13 Voltage setting device 14 Average value voltage regulator 15 V / F converter 16 Memory 17 Multiplier 18 Voltage regulator 21 Inverter Current detector 22 Current regulator 23 Output current detector 30 Voltage regulator 31 First gain 32 Second gain 33 Proportional integral calculator 40 First non-interference network 50 Second non-interference network 60 Two-phase oscillator 70 Control signal Conversion circuit i _C Capacitor current i _I Inverter current i _IS Inverter current command value i _O Output current v _C Capacitor voltage v _I Inverter voltage v _M Average value voltage v _MS Average value voltage command value v _O Output voltage v _OS Output voltage command value

Claims (5)

[Claims] 1. A filter comprising a reactor and a capacitor is provided on an AC output side of an inverter for converting DC to AC, and a deviation between an output voltage detection value detected at the filter output side and an output voltage command value set separately. In the control circuit of the voltage source inverter for controlling the inverter with the output signal of the voltage adjusting means, the voltage adjusting means outputs Inputting the deviation between the control signal and the detected value of the inverter current detected on the inverter output side, the current adjusting means for outputting a control signal for making the input deviation zero is provided. A control circuit for a voltage source inverter, which controls the inverter. 2. A deviation between an output voltage detection value detected at the filter output side and an output voltage command value set separately is provided on the AC output side of an inverter for converting DC to AC, and a filter including a reactor and a capacitor is provided. In the control circuit of the voltage source inverter for controlling the inverter by the output signal of the voltage adjusting means, the voltage adjusting means for outputting a control signal for inputting The output voltage detection value and the inverter current detection value detected at the inverter output side are respectively d
Convert to two components of the synchronous rotation coordinate system by the axis and q axis,
Voltage adjusting means for separately outputting a control signal for making the deviation between the output voltage command value for each axis and the output voltage detection value zero, a control signal for each axis output by the voltage adjusting means, and the inverter output side The current adjusting means for separately outputting a control signal for making the deviation from the detected value of the inverter current for each axis detected by 1. and the d-axis output signal and the q-axis output signal of these current adjusting means are inputted, A control circuit for a voltage source inverter, comprising: a control signal converting means for converting a control rate for controlling the inverter and a phase angle. 3. A filter comprising a reactor and a capacitor is provided on the AC output side of an inverter for converting DC to AC, and a deviation between an output voltage detection value detected on the filter output side and an output voltage command value set separately. In the control circuit of the voltage source inverter for controlling the inverter by the output signal of the voltage adjusting means, the voltage adjusting means for outputting a control signal for inputting The output voltage detection value and the inverter current detection value detected at the inverter output side are respectively d
Convert to two components of the synchronous rotation coordinate system by the axis and q axis,
Voltage adjusting means for separately outputting a control signal for making the deviation between the output voltage command value and the output voltage detected value for each axis zero, and a first non-interference circuit network provided on the output side of these voltage adjusting means,
Current adjusting means for separately outputting a control signal for making the deviation between the control signal for each axis output from the first non-interference circuit network and the inverter current detection value for each axis detected at the inverter output side zero. A second non-interference network provided on the output side of the current adjusting means, and a control rate and a phase angle for controlling the inverter by inputting the d-axis output signal and the q-axis output signal of the second non-interference network. A control circuit for a voltage source inverter, comprising: 4. The control circuit for a voltage source inverter according to claim 1, 2, or 3, wherein the voltage adjusting means sets an output voltage command value that is set separately from the output voltage detection value. The first to calculate the amount proportional to the deviation value from
A gain, a voltage differential value calculation means for calculating a differential value of an output voltage detection value detected on the filter output side, a second gain for calculating an amount proportional to the voltage differential value, and outputs of these first gains. A control circuit for a voltage source inverter, comprising: an addition means for adding the output of the second gain and a proportional-plus-integral calculation means for proportionally integrating the result of the addition operation. 5. The control circuit for a voltage source inverter according to claim 4, wherein the voltage differential value calculation means for calculating a differential value of the output voltage detection value detected at the filter output side is an inverter output input to the filter. A control circuit for a voltage source inverter, which is obtained from a difference between a current and an output current of the filter.