JP7329971B2 - converter - Google Patents

Description

The present invention relates to converters.

In recent years, various electronic devices equipped with batteries such as electric motorcycles and EVs have become popular. There is a demand for converters (power supplies) that are compatible with widely different electronic devices.

However, a converter (converter section) that converts an input voltage into a predetermined output voltage has a limited range of output voltage. Therefore, if an attempt is made to change the magnitude of the output voltage from the original input voltage to a voltage that is greatly changed, it is necessary to take measures such as mounting an additional converter section after the converter section. As such a converter, for example, the following converter (conventional converter 900) is conceivable.

FIG. 11 is a diagram showing a conventional converter 900. As shown in FIG. In FIG. 11, the symbol Vin indicates an input terminal, Vout indicates an output terminal, C3 indicates an input capacitor, and C4 indicates an output capacitor.
As shown in FIG. 11, a conventional converter 900 is connected to a transformer T1 having a primary winding T1-1 and a secondary winding T1-2, the primary winding T1-1 side of the transformer T1, and a switching element. A converter section 910 having a primary side switch section SW1 in which a bridge circuit is formed by Q1 and Q2, and a resonant capacitor C1 and a resonant inductor L1 connected in series with the primary winding T1-1, and a converter section. A rectifying section 930 connected to the secondary winding T1-2 side of the transformer T1 of 910 and having a bridge circuit composed of four diodes D1, D2, D3 and D4, and a switching element provided after the rectifying section 930. An additional converter section 960 that forms a boost chopper circuit with Q7, diode D5 and inductor L3 is provided.

According to the conventional converter 900, since the additional converter section 960 is provided, the output voltage can be greatly changed from the input voltage by duty-controlling the additional converter section 960. FIG.

Note that Patent Document 1 discloses a converter that includes a high-voltage battery type filter unit composed of a coil and a capacitor instead of a boost chopper circuit as an additional converter unit, and boosts voltage with the high-voltage battery type filter unit. .

Japanese Patent Application Laid-Open No. 2009-213202

However, in the conventional converter 900, since it is necessary to prepare an additional converter section 960, the area occupied by the components increases, and there is a problem that it is difficult to reduce the size of the converter (power supply).

SUMMARY OF THE INVENTION Accordingly, it is an object of the present invention to provide a compact converter capable of handling a wide range of output voltages while solving the above-described problems.

[1] A converter of the present invention is an isolated resonant converter that outputs a desired output voltage by switching between a low voltage output mode, a medium voltage output mode, and a high voltage output mode, comprising: A transformer having a primary winding and a secondary winding, a primary side switch unit connected to the primary winding side of the transformer and configured to form a bridge circuit with primary side switching elements, and the primary winding two converter sections connected in parallel, each having a resonant capacitor connected in series with the , a bridge circuit connected to the secondary winding side of each transformer of the first and second converter sections and comprising two diodes connected to the high side and two secondary side switching elements connected to the low side; is configured, and the two secondary windings connected in series are connected in parallel with the two diodes and the two secondary side switching elements respectively; In the high voltage output mode, on/off timings of the primary side switching element of the first converter section and the primary side switching element of the second converter section are synchronized, and in the low voltage output mode, is such that a phase difference occurs between the phase of the control voltage of the primary side switching element of the first converter section and the phase of the control voltage of the primary side switching element of the second converter section. a primary-side control unit for controlling on/off timing of the primary-side switching elements of the first and second converter units; side switching elements are turned off, and in the high voltage output mode, the two secondary side switching elements of the rectifying section are turned on in synchronization with the timing when the primary side switching elements of the first and second converter sections are turned on. a secondary side control section that controls the two secondary side switching elements of the rectifying section so that at least one of them is turned on for a predetermined time.

[2] In the converter of the present invention, in the low voltage output mode, the primary side control section controls the phase of the control voltage of the primary side switching element of the first converter section and the phase of the control voltage of the primary side switching element of the second converter section. Preferably, the output voltage is controlled by adjusting the phase difference between the phase of the control voltage of the primary side switching element and the phase of the control voltage of the primary side switching element.

[3] In the converter of the present invention, in the high voltage output mode, the secondary side control section adjusts the time to turn on at least one of the two secondary side switching elements of the rectifying section. Preferably, the output voltage is controlled by:

[4] In the converter of the present invention, in the high voltage output mode, the secondary side control section controls the rectification section so that the two secondary side switching elements of the rectification section are synchronously turned on for a predetermined period of time. Preferably, the two secondary side switching elements of are controlled.

[5] In the converter of the present invention, in the high-voltage output mode, the secondary-side control section controls the above-described One of the two secondary side switching elements of the rectifying section is turned on and kept on for a predetermined time, and the rectifying section is synchronized with the primary side switching elements of the first and second converter sections. It is preferable to turn on the other of the two secondary side switching elements of the rectifying section and control the two secondary side switching elements of the rectifying section so as to alternately repeat this.

[6] In the converter of the present invention, it is preferable that the primary side switching elements form a half bridge circuit in the primary side switch section.

[7] In the converter of the present invention, it is preferable that the primary side switching elements form a full bridge circuit in the primary side switch section.

[8] In the converter of the present invention, the first and second converter sections further include an inductor connected in series with the primary winding, and the first and second converter sections each include: It is preferable that the inductor, the resonant capacitor and the primary winding form a resonant circuit.

[9] In the converter of the present invention, it is preferable that the parasitic inductor, the resonant capacitor and the primary winding form a resonant circuit in the first and second converter sections.

According to the converter of the present invention, in the medium voltage output mode, the primary side control section synchronizes the on/off timings of the input side switching elements of the first and second converter sections, and the secondary side control section synchronizes the rectification section. In order to turn off the switching element on the output side, a desired output voltage is output within a predetermined output range by performing frequency control within the range of voltage that can be output, as in the conventional current resonance type converter 900. can be done.

Further, according to the converter of the present invention, in the low voltage output mode, the primary side control section controls the phase of the control voltage of the input side switching element of the first converter section and the phase of the control voltage of the input side switching element of the second converter section. The on/off timing of the input-side switching elements of the first and second converter units is controlled so that a phase difference occurs between the phase of the control voltage of the secondary-side control unit and the output-side switching element of the rectifier unit. For turning off, a reverse voltage section can be generated in the secondary winding of each transformer of the first and second converter sections to lower the output voltage. As a result, a relatively low output voltage can be output.

Further, according to the converter of the present invention, in the high voltage output mode, the primary side control section synchronizes the on/off timings of the input side switching elements of the first and second converter sections, and the secondary side control section: The two output sides of the rectifying section so that at least one of the two output side switching elements of the rectifying section is turned on for a predetermined time in synchronization with the timing when the input side switching elements of the first and second converter sections are turned on. Since the switching element is controlled, the rectifying section can perform the boost chopper operation only for the predetermined period, and the output voltage can be increased. As a result, a desired output voltage can be output at a relatively high voltage.

Therefore, according to the converter of the present invention, by adopting the configuration described above, it is possible to switch between the low voltage output mode, the medium voltage output mode, and the high voltage output mode. A compatible converter.

Further, according to the converter of the present invention, with the configuration described above, the low-voltage output mode, the medium-voltage output mode, and the high-voltage output mode can be achieved without mounting a non-isolated chopper circuit in the subsequent stage of the isolated converter. and can be implemented by switching between them, the converter can be downsized while being compatible with a wide range of output voltages.

1 is a diagram showing a circuit diagram of a converter 1 according to Embodiment 1; FIG. 4 is a waveform diagram of a medium voltage output mode (common mode) in Embodiment 1. FIG. 4 is a circuit diagram shown for explaining circuit operation in a middle voltage output mode (common mode) according to the first embodiment; FIG. 4 is a waveform diagram when the phase difference is 90° in the low voltage output mode (phase shift mode) in Embodiment 1. FIG. 4 is a circuit diagram for explaining circuit operation when the phase difference is 90° in the low voltage output mode (phase shift mode) according to the first embodiment; FIG. 4 is a waveform diagram when the phase difference is 180° (opposite phase) in the low voltage output mode (phase shift mode) in Embodiment 1. FIG. 4 is a waveform diagram in a high voltage output mode (boost mode) in Embodiment 1. FIG. 4 is a circuit diagram for explaining circuit operation in a high voltage output mode (boost mode) according to the first embodiment; FIG. 4 is a circuit diagram for explaining circuit operation in a high voltage output mode (boost mode) according to the first embodiment; FIG. FIG. 10 is a waveform diagram of a high voltage output mode (boost mode) in the converter according to Embodiment 2; 1 shows a conventional converter 900; FIG.

Hereinafter, the converter of the present invention will be described based on the embodiments shown in the drawings. Each embodiment described below does not limit the invention according to the scope of claims. Also, not all of the elements and their combinations described in each embodiment are essential to the solution of the present invention. In each embodiment, the same reference numerals may be used across the embodiments for configurations and elements having the same basic configuration, features, functions, etc., and repetitive descriptions may be omitted.

[Embodiment 1]
1. Configuration of Converter 1 According to Embodiment 1
FIG. 1 is a diagram showing a circuit diagram of a converter 1 according to Embodiment 1. FIG.
The converter 1 according to the first embodiment is an isolation resonant converter that outputs a desired output voltage by switching between a low voltage output mode, a medium voltage output mode, and a high voltage output mode.
As shown in FIG. 1, the converter 1 according to the first embodiment includes a first converter section 10, a second converter section 20, a rectifying section 30, a primary side control section 40, and a secondary side control section. 50.

The first converter section 10 is connected to a transformer T1 having a primary winding T1-1 and a secondary winding T1-2, the primary winding T1-1 side of the transformer T1, and a switching element (primary side switching Element) A current resonance type current resonance type circuit having a primary side switch section SW1 in which a half bridge circuit is formed by Q1 and Q2, and a resonance capacitor C1 and a resonance inductor L1 connected in series with the primary winding T1-1. This is the converter section. The primary winding T1-1, the resonant capacitor C1 and the resonant inductor L1 constitute an LLC type resonant circuit. The order of the primary winding T1-1, resonance capacitor C1 and resonance inductor L1 is arbitrary.

The second converter section 20 is connected to a transformer T2 having a primary winding T2-1 and a secondary winding T2-2, the primary winding T2-1 side of the transformer T2, and a switching element (primary side switching Element) A current resonance type current resonance type circuit having a primary side switch section SW2 in which a half bridge circuit is formed by Q3 and Q4, and a resonance capacitor C2 and a resonance inductor L2 connected in series with the primary winding T2-1. This is the converter section. The primary winding T2-1, the resonant capacitor C2 and the resonant inductor L2 constitute an LLC type resonant circuit. The order of the primary winding T2-1, resonance capacitor C2 and resonance inductor L2 is arbitrary.

The first converter section 10 and the second converter section 20 are connected in parallel to the input terminal Vin, and the primary side switch sections SW1 and SW2 are connected to the input terminal Vin. An input capacitor C3 is connected to the input terminal Vin side. Also, the respective secondary windings T1-2 and T2-2 are connected in series with each other.
In the first embodiment, the switching elements Q1, Q2, Q3, Q4, Q5 and Q6 use MOSFETs, but other suitable switching elements such as IGBTs may be used.

The rectifying section 30 is connected to the secondary windings T1-2 and T2-2 of the transformers T1 and T2 of the first converter section 10 and the second converter section 20, and two diodes connected to the high side. A bridge circuit is composed of D1, D2 and two switching elements (secondary side switching elements) Q5, Q6 connected to the low side, and two secondary windings T1-2, T2-2 connected in series are connected. , two diodes D1 and D2, and two switching elements Q5 and Q6, respectively. The cathode electrodes of the two diodes D1 and D2 are connected to the output terminal Vout (hereinafter the output voltage may be referred to as Vout), and the two switching elements Q5 and Q6 are grounded (connected to the ground terminal). It is An output capacitor C4 is connected between the output terminal Vout and the other terminal (for example, ground terminal) on the output side.

The rectifying section 30 functions as a full-bridge rectifying circuit when the two switching elements Q5 and Q6 are off, and functions as a boost chopper circuit and a rectifying circuit when the two switching elements Q5 and Q6 are on.

The primary-side control section 40 controls on/off of the switching elements Q1 and Q2 of the first converter section 10 and the switching elements Q3 and Q4 of the second converter section 20 . Specifically, in the medium voltage output mode and the high voltage output mode, the switching element Q1 (Q2) and the switching element Q3 (Q4) are synchronized (turned on and off at the same time with a phase difference of 0), and in the low voltage output mode , ON/OFF of the switching elements Q1, Q2, Q3 and Q4 are controlled so that a phase difference occurs between the phase of the switching element Q1 (Q2) and the phase of the switching element Q3 (Q4).

The secondary side control section 50 controls on/off of the switching elements Q5 and Q6 of the rectifying section 30 . Specifically, in the low voltage output mode and the medium voltage output mode, the switching elements Q5 and Q6 are turned off, and in the high voltage output mode, the switching elements Q1, Q2, Q3, and Q4 are turned on in synchronization with the timing. Both of the two switching elements Q5 and Q6 are controlled to turn on for a predetermined time.

2. Operation of Converter 1 According to First Embodiment Next, the operation of the converter 1 according to the first embodiment will be described. The converter 1 according to the first embodiment switches between a low-voltage output mode, a medium-voltage output mode, and a high-voltage output mode depending on the object connected to the output terminal (for example, each output is switched depending on the voltage required by the battery). change the mode).

(1) Medium voltage output mode (common mode)
First, the middle voltage output mode (common mode) will be described.
FIG. 2 is a waveform diagram of a medium voltage output mode (common mode) in the first embodiment. In FIG. 2, Q1, Q2, Q3, Q4, Q5, and Q6 represent gate voltage waveforms of switching elements Q1, Q2, Q3, Q4, Q5, and Q6, and _VT1 represents the voltage of secondary winding T1-2 of transformer T1. V _T2 denotes the voltage of the secondary winding T2-2 of the transformer T2, and V _T1 +V _T2 is the sum of the voltage of the secondary winding T1-2 and the voltage of the secondary winding T2-2. , and Vout indicates the output voltage (the same applies to FIGS. 4, 6, 7, and 10 below).
Further, times t1 and t9 indicate timings at which the switching element Q2 is turned off, times t2 and t10 indicate timings at which the switching element Q1 is turned on, times t5 and t13 indicate timings at which the switching element Q1 is turned off, Times t6 and t14 indicate timings at which the switching element Q2 is turned on (the same applies hereinafter to FIGS. 4, 6, 7, and 10).
FIG. 3 is a circuit diagram for explaining the middle voltage output mode (common mode) in the first embodiment. FIG. 3(a) is a circuit diagram for explaining the load current and excitation current when switching elements Q1 and Q3 are on and Q2 and Q4 are off, and FIG. 3(b) is a circuit diagram showing switching elements Q1 and Q3 FIG. 4 is a circuit diagram for explaining load current and exciting current when Q2 and Q4 are off and on;

The medium voltage output mode is a mode in which switching element Q1 (Q2) of first converter section 10 and switching element Q3 (Q4) of second converter section 20 are synchronized to perform frequency control.

(1-1) Q1, Q3 ON, Q2, Q4 OFF First, as shown in FIG. 2, switching elements Q2, Q4 are turned off at time t1, and switching elements Q1, Q3 are turned on at time t2 3A, in the first converter section 10 on the primary side, the switching element Q1, the resonance inductor L1, the resonance capacitor C1, the primary winding T1− A load current flows through the path 1 and returns to the ground terminal of the input terminal. Correspondingly, on the secondary side, a load current flows as indicated by the solid line in FIG. 3(a). At this time, an exciting current is generated from the input capacitor C3 that returns to the input capacitor C3 via the switching element Q1, the resonant inductor L1, the resonant capacitor C1, and the primary winding T1-1 (see the dashed line in FIG. 3(a)). .
Similarly, in the second converter section 20, the path from the input terminal Vin to the ground side terminal of the input terminal via the switching element Q2, the resonant inductor L2, the resonant capacitor C2, and the primary winding T2-1. A load current flows (see the solid line in FIG. 3(a)). Correspondingly, on the secondary side, the load current flows in the same direction through the secondary windings T1-2 and T2-2 of the first converter section (see the solid line in FIG. 3(a)). At this time, an exciting current is generated from the input capacitor C3 that returns to the input capacitor C3 via the switching element Q3, the resonant inductor L2, the resonant capacitor C2, and the primary winding T2-1 (see the dashed line in FIG. 3(a)). .

(1-2) When Q1 and Q3 are off and Q2 and Q4 are on Eventually, when the LLC resonance ends, only the excitation current (see the dashed line in FIG. 3(a)) remains, turning off the switching elements Q1 and Q3 ( At time t5), in the first converter section 10, an exciting current is generated that returns to the resonant capacitor C1 via the resonant capacitor C1, the primary winding T1-1, the parasitic diode of the switching element Q2, and the resonant inductor L1 ( See the one-dot chain line in FIG. 3(b).). In the second converter section 20 as well, an exciting current is generated that returns to the resonant capacitor C2 through the resonant capacitor C2, the primary winding T2-1, the parasitic diode of the switching element Q4, and the resonant inductor L2 (Fig. 3 (b ). See dash-dotted line.).

Next, at time t6, switching elements Q2 and Q4 are turned on. Then, a load current flows through the resonant capacitor C1 through the resonant capacitor C1, the resonant inductor L1, the switching element Q2, and the primary winding T1-1 (see the solid line in FIG. 3(b)). Also in the second converter section 20, a load current flows through resonance capacitor C2-resonance inductor L2-switching element Q4-primary winding T2-1-resonance capacitor C2 (see solid line in FIG. 3B). .

Accordingly, on the secondary side, the load current flows in the same direction (downward direction in FIG. 2B) in the secondary windings T1-2 and T2-2.

That is, as can be seen from (1-1) and (1-2) above, both the two converter units 10 and 20 generate a voltage that causes a current to flow in the same direction on the secondary side. The applied voltage is the sum of the voltage _VT1 of the secondary winding T1-2 of the first converter section 10 and the voltage _VT2 of the secondary winding T2-2 of the second converter section 20. becomes.

In the middle voltage output mode, the operation is similar to that of the conventional current resonance converter except that the first converter section 10 and the second converter section 20 are connected in parallel. Therefore, by performing frequency control, a desired output voltage (for example, a predetermined voltage V ₀ ) can be output within a predetermined range.

(2) Low voltage output mode (phase shift mode)
In the low power output mode, a phase difference is provided between the first converter section 10 and the second converter section 20, and a reverse voltage period is provided on the secondary winding side of the transformer, thereby outputting a low output voltage. mode. First, for ease of explanation, the case where the phase difference between the first converter section 10 and the second converter section 20 is 90° will be explained.

FIG. 4 is a waveform diagram when the phase difference between the switching elements Q1 and Q3 (Q2 and Q4) is 90° in the low voltage output mode (phase shift mode) in the first embodiment. In FIG. 4, times t3 and t12 indicate timings at which the switching element Q4 is turned off, times t4 and t13 indicate timings at which the switching element Q3 is turned on, and times t7 and t15 indicate timings at which the switching element Q3 is turned off. , and times t8 and t16 indicate the timings at which the switching element Q4 is turned on.
FIG. 5 is a circuit diagram for explaining a case where the phase difference is 90° in the low voltage output mode (phase shift mode) according to the first embodiment. FIG. 5(a) is a circuit diagram showing the load current and excitation current when switching elements Q1 and Q4 are on and Q2 and Q3 are off, and FIG. 5(b) is a circuit diagram showing switching elements Q1 and Q4 FIG. 4 is a circuit diagram for explaining load current and exciting current when Q2 and Q3 are off and Q3 is on;

(2-1) Time t2-t3
At time t1, the switching element Q2 is turned off, and at time t2, when the switching element Q1 is turned on, as shown by the solid line in FIG. A load current flows through the switching element Q1, the resonance inductor L1, the resonance capacitor C1, the primary winding T1-1, and back to the ground terminal of the input terminal. Correspondingly, on the secondary side, a load current flows as indicated by the solid line in FIG. 5(a). At this time, an exciting current is generated from the input capacitor C3 that returns to the input capacitor C3 via the switching element Q1, the resonant inductor L1, the resonant capacitor C1, and the primary winding T1-1 (see the dashed line in FIG. 5(a)). .

On the other hand, in the second converter section 20, since there is a phase difference with the first converter section 10, the switching element Q3 is turned off, the switching element Q4 is turned on, and the resonance capacitor C2-primary winding T2-1 A load current flows through the switching element Q4, the resonant inductor L2, and the resonant capacitor C2 (see the solid line in FIG. 5(a)). Along with this, on the secondary side of the second converter section 20, the secondary winding T2-2 is directed in the opposite direction to the secondary winding T1-2 in the first converter section 10 (see FIG. 5A). The load current flows downward (see the dashed arrow).

Therefore, since the load currents flow in the secondary winding T1-2 and the secondary winding T2-2 in opposite directions, a reverse voltage section is generated, and the output voltage can be lowered by canceling each other. (See Vout in FIG. 4). It should be noted that the output voltage Vout is rectified, DC-converted and output, and a voltage lower than the predetermined voltage _V0 is output.
Then, at time t3, the switching element Q4 is turned off.

(2-2) Time t4-t5
Next, at time t4, switching elements Q1 and Q2 are left unchanged (Q1: ON, Q2: OFF), and switching element Q3 is turned ON (Q4 is OFF). (See FIG. 3(a).). Then, at time t5, the switching element Q1 is turned off.

(2-3) Time t6-t7
Next, at time t6, when the switching element Q2 is turned on, in the first converter section 10, the load current flows through the path of capacitor C1-primary winding T1-1-switching element Q2-resonant inductor L1-capacitor C1. (See FIG. 5(b).). Along with this, on the secondary side of the first converter section 10, the secondary winding T1-2 is provided with a current flowing in the direction opposite to that of T2-2 (the upward direction in FIG. 5(b), see the dashed arrow). Load current flows.
Therefore, since the direction of load current flow is opposite between the secondary winding T1-2 and the secondary winding T2-2, a reverse voltage section can be generated to lower the output voltage (Vout in FIG. 4). See the graph in ). Then, at time t7, the switching element Q3 is turned off.

(2-4) Time t8-t9
Next, at time t8, when the switching element Q4 is turned on, the same operation as in the case of (1-2) of the middle voltage output mode is performed (see FIG. 3(b)).
Then, at time t9, the switching element Q2 is turned off.

After that, this process is repeated.

In the low voltage output mode, the phase of the control voltage of the switching element Q1 (Q2) of the first converter section 10 and the switching element Q3 (Q4) of the second converter section 20 are controlled by the primary side control section 40. Control is performed to output a desired output voltage by adjusting the phase difference between the phases of the voltages.
FIG. 6 is a waveform diagram when the phase difference between the on/off timings of the switching elements Q1 and Q3 (Q2 and Q4) is 180° (opposite phase) in the low voltage output mode (phase shift mode) according to the first embodiment.
For example, when the phase difference is 180° (opposite phase), the period in which the reverse voltage section occurs continues for a long time, so the output voltage can be considerably lowered (see FIG. 6). The output voltage can also be adjusted by changing the winding ratio of the transformers T1 and T2.

(3) High voltage output mode (boost mode)
The high voltage output mode is a mode in which the rectifying section 30 performs a boost chopper operation to increase the output voltage only for a predetermined period.
FIG. 7 is a waveform diagram in the high voltage output mode (boost mode) according to the first embodiment. In FIG. 7, Q'Id indicates the drain current of one of the switching elements Q1 to Q4, times t3, t7, t11, and t15 indicate timings when the switching elements Q5 and Q6 are turned off, and times t4, t8 and t12 indicate timings when Q'Id becomes 0. The solid line of Q'Id indicates the drain currents of the switching elements Q1 and Q3, and the broken line of Q'Id indicates the drain currents of the switching elements Q2 and Q4.
FIG. 8 is a circuit diagram for explaining the high voltage output mode (boost mode) in the first embodiment. FIG. 8(a) is a circuit diagram for explaining the load current and the excitation current when the switching elements Q1 and Q3 are on, Q2 and Q4 are off, and Q5 and Q6 are on (time t2 to t3). FIG. 8(b) is a circuit diagram for explaining the load current and excitation current when switching elements Q1 and Q3 are on, Q2 and Q4 are off, and Q5 and Q6 are off (time t3-t4).
FIG. 9 is a circuit diagram for explaining the high voltage output mode (boost mode) in the first embodiment. FIG. 9(a) is a circuit diagram for explaining load current and excitation current when switching elements Q1 and Q3 are off, Q2 and Q4 are on, and Q5 and Q6 are on (time t6 to t7). FIG. 8(b) is a circuit diagram for explaining the load current and excitation current when switching elements Q1 and Q3 are off, Q2 and Q4 are on, and Q5 and Q6 are off (time t7-t8).

In the high voltage output mode, the on/off of the switching elements is controlled so that the switching elements Q1 and Q3, Q2 and Q4 are synchronized. Therefore, the waveforms of the switching elements Q1, Q2, Q3 and Q4 are the same as in the medium voltage output mode (common mode) (see FIG. 2). The two switching elements Q5 and Q6 are controlled so that both of the two switching elements Q5 and Q6 of the rectifying section 30 are turned on for a predetermined time in synchronization with the timing when Q4 is turned on. In other words, among the engines that turn on the switching elements Q1 and Q3 or Q2 and Q4, the switching elements Q5 and Q6 are turned on only for a predetermined period of time after turning on.

(3-1) Time t2-t3
As shown in FIG. 7, at time t2, switching elements Q1 and Q3 are turned on and switching elements Q2 and Q4 are turned off. A load current flows through the path returning to the ground terminal of the input terminal via the element Q1, the resonant inductor L1, the resonant capacitor C1, and the primary winding T1-1 (see the solid line in FIG. 8(a)). Correspondingly, on the secondary side, a load current flows as indicated by the solid line in FIG. 8(a). At this time, an exciting current is generated from the input capacitor C3 that returns to the input capacitor C3 via the switching element Q1, the resonant inductor L1, the resonant capacitor C1, and the primary winding T1-1 (see the dashed line in FIG. 8A). .
Similarly, in the second converter section 20, the path from the input terminal Vin to the ground side terminal of the input terminal via the switching element Q2, the resonant inductor L2, the resonant capacitor C2, and the primary winding T2-1. A load current flows (see the solid line in FIG. 8(a)). Correspondingly, on the secondary side, the load current flows in the same direction through the secondary windings T1-2 and T2-2 of the first converter section (see broken line in FIG. 8(a)). At this time, an exciting current is generated from the input capacitor C3 to return to the input capacitor C3 via the switching element Q2, the resonant inductor L2, the resonant capacitor C2, and the primary winding T2-1 (see the dashed line in FIG. 8(a)). .

Between times t2 and t3, the two switching elements Q5 and Q6 of the rectifying section 30 are turned on. As a result, the secondary windings T1-2 and T2-2 of the transformers T1 and T2 are short-circuited (see the solid line in FIG. 8(a)). At this time, boosting energy is accumulated in the resonant inductors L1 and L2. A current component induced corresponding to the load current corresponding to the boosted voltage flows through the switching elements Q1 and Q3 (see Q'Id solid line in FIG. 7).

(3-2) Time t3-t4
Next, at time t3, the switching elements Q5 and Q6 are turned off (opened), so that a current flows as indicated by the solid line in FIG. 8(b).
In this way, between times t2 and t4, the secondary windings T1-2 and T2-2 of the transformers T1 and T2 are partially short-circuited and then opened, so that the boost chopper operation is performed in the rectifying section 30. As a result, the output voltage Vout can be boosted.
Then, at time t4, the current components Q'Id of the switching elements Q1 and Q3 become zero.

Eventually, when the LLC resonance ends, only the excitation current (see one-dot chain line in FIG. 8B) remains, and when switching elements Q1 and Q3 are turned off (time t5), resonance capacitor C1 - Primary winding T1-1 - Parasitic diode of switching element Q2 - An exciting current is generated that returns to resonant capacitor C1 through resonant inductor L1 (see dashed line in FIG. 9(a)). Also in the second converter section 20, an exciting current is generated that returns to the resonant capacitor C2 through the resonant capacitor C2, the primary winding T2-1, the parasitic diode of the switching element Q4, and the resonant inductor L2 (Fig. 9 (a ). See dash-dotted line.).

(3-3) Time t6-t7
Next, at time t6, switching elements Q2 and Q4 are turned on. Then, a load current flows through resonance capacitor C1-resonance inductor L1-switching element Q2-primary winding T1-1-resonance capacitor C1 (see solid line in FIG. 9A). Also in the second converter section 20, a load current flows through resonance capacitor C2-resonance inductor L2-switching element Q4-primary winding T2-1-resonance capacitor C2 (see solid line in FIG. 9A). .

Between times t6 and t7, the two switching elements Q5 and Q6 of the rectifying section 30 are turned on. As a result, the secondary windings T1-2 and T2-2 of the transformers T1 and T2 are partially short-circuited (see solid line in FIG. 9(a)). At this time, boosting energy is accumulated in the resonant inductors L1 and L2. Further, a current component induced corresponding to the boosted load current flows through the switching elements Q2 and Q4 (see Q'Id dashed line in FIG. 7).

(3-4) Time t7-t8
Next, at time t7, the switching elements Q5 and Q6 are turned off (opened) to allow current to flow as indicated by the solid line in FIG. 9(b).
In this manner, the boost chopper operation is performed in the rectifying section 30 between times t6 and t8, and the output voltage Vout can be boosted.
Then, at time t8, the current components Q'Id of the switching elements Q2 and Q4 become zero.

Eventually, when the LLC resonance ends, only the excitation current (see one-dot chain line in FIG. 9B) remains, and switching elements Q2 and Q4 are turned off (time t9). An exciting current is generated from C3 that returns to the input capacitor C3 via the switching element Q1, the resonant inductor L1, the resonant capacitor C1, and the primary winding T1-1 (see one-dot chain line in FIG. 8(a)). Also in the second converter section 20, an exciting current is generated from the input capacitor C3, which returns to the input capacitor C3 via the switching element Q2, the resonant inductor L2, the resonant capacitor C2, and the primary winding T2-1 (see FIG. 8). (a) see dashed line).

Next, at time t10, the switching elements Q1 and Q3 are turned on. The same is repeated below.

3. Effects of the converter 1 according to the first embodiment According to the converter 1 according to the first embodiment, the primary side control unit 40 controls the switching element Q1 of the first and second converter units 10 and 20 in the intermediate voltage output mode. The on/off timings of Q3, Q2 and Q4 are synchronized, and the secondary side control unit 50 turns off the switching elements Q5 and Q6 of the rectifying unit 30. A desired output voltage can be output within a predetermined output range by performing frequency control within the voltage range.

Further, according to the converter 1 according to the first embodiment, in the low voltage output mode, the primary side control unit 40 is positioned between the phase of the switching element Q1 (Q2) and the phase of the switching element Q3 (Q4). On/off of the switching elements Q1, Q2, Q3, and Q4 is controlled so as to generate a phase difference (phase difference between on/off timings), and the secondary side control unit 50 turns off the switching elements Q5 and Q6. A reverse voltage section can be generated at T1-2 and T2-2 to lower the output voltage. As a result, a relatively low output voltage can be output.

Further, according to the converter 1 according to the first embodiment, in the high voltage output mode, the primary side control unit 40 synchronizes the switching elements Q1 and Q3, Q2 and Q4, and the secondary side control unit 50 synchronizes the switching elements Q1 and Q3, Q2 and Q4. In order to control the switching elements Q5 and Q6 so as to turn on the two switching elements Q5 and Q6 of the rectifying section 30 for a predetermined time in synchronization with the timing when the elements Q1 and Q3 or Q2 and Q4 are turned on, the rectification is performed only during the predetermined period. The section 30 can perform the boost chopper operation, and the output voltage can be increased. As a result, a desired output voltage can be output at a relatively high voltage.

Therefore, according to the converter 1 according to the first embodiment, with the configuration described above, it is possible to switch between the low-voltage output mode, the medium-voltage output mode, and the high-voltage output mode. It becomes a converter that can handle the output voltage.

Further, according to the converter 1 according to the first embodiment, by adopting the configuration described above, a low voltage output mode, a medium voltage output mode, and a high Since the voltage output mode can be switched and implemented, the converter can handle a wide range of output voltages and can be downsized.

Further, according to the converter 1 according to the first embodiment, in the low voltage output mode, the primary side control unit 40 controls the phases of the switching elements Q1 and Q2 of the first converter unit 10 and the switching of the second converter unit 20. Since the output voltage is controlled by adjusting the phase difference between the elements Q3 and Q4 (the phase difference between the switching elements Q1 and Q3 or Q2 and Q4), the desired output voltage can be reliably obtained at relatively low voltages. can be output.

Further, according to the converter 1 according to the first embodiment, in the high voltage output mode, the secondary side control unit 50 adjusts the time during which the two switching elements Q5 and Q6 of the rectifying unit 30 are turned on, thereby increasing the output voltage. Because of the control, it is possible to reliably output the desired output voltage at relatively high voltages.

Further, according to the converter 1 according to the first embodiment, in the primary side switch units SW1 and SW2, the switching elements Q1 and Q2 and the switching elements Q3 and Q4 form half bridge circuits, respectively. It is possible to eliminate the need to use a switching element with a high breakdown voltage, and to obtain an efficient converter.

Further, according to the converter 1 according to the first embodiment, the first and second converter sections 10, 20 are connected in series with the primary windings T1-1, T2-1, respectively, and the resonant inductors L1, L2 , and in the first and second converter sections 10 (second converter section 20), resonant inductor L1 (L2), resonant capacitor C1 (C2), and primary winding T1-1 (T2-1) resonate Since the circuit is configured, soft switching can be realized with a relatively simple configuration.

[Embodiment 2]
FIG. 10 is a waveform diagram of the high voltage output mode (boost mode) in the converter according to the second embodiment. In FIG. 10, Q'Id indicates the drain current of the switching element, times t3 and t11 indicate timings at which the switching element Q6 is turned off, and times t7 and t15 indicate timings at which the switching element Q5 is turned off. , t4, t8, and t12 indicate timings when Q'Id becomes zero.

The hardware configuration circuit of the converter according to the second embodiment is the same as that of the converter 1 according to the first embodiment shown in FIG. It is different from the case of 1. That is, in the high-voltage output mode of the converter according to the second embodiment, the secondary-side control unit 50 operates the rectifying unit in synchronization with the timing when the switching elements Q1 and Q3 are turned on (see times t2 and t10 in FIG. 10). The switching element Q6 is turned on for a predetermined period of time (time t2-t3, t10-t11) (booster period), and in synchronization with the switching elements Q1 and Q3 of the first and second converter sections 10 and 20. The switching element Q5 of the rectifier is turned on (synchronous rectification period). In addition, the switching element Q5 of the rectifying section is turned on in synchronization with the timing when the switching elements Q2 and Q4 are turned on (see times t6 and t14 in FIG. 10), and is turned on for a predetermined time (time t6 to t7, t14 to t15). state (boosting period), and the switching element Q6 of the rectifying section 30 is turned on in synchronization with the on period of the switching elements Q2 and Q4 (synchronous rectification period). The two switching elements Q5 and Q6 of the rectifying section 30 are controlled so as to repeat this alternately (see FIG. 10).

Thus, the converter according to the second embodiment is similar to the converter 1 according to the first embodiment, although the on/off timings of the switching elements Q5 and Q6 in the voltage output mode are different from those of the converter 1 according to the first embodiment. In the middle voltage output mode and the high voltage output mode, the switching element of the first converter section and the switching element of the second converter section are synchronized, and in the low voltage output mode, the switching element of the first converter section is synchronized. A primary side that controls on/off of the switching elements of the first and second converter sections such that a phase difference occurs between the phase of the control voltage of the element and the phase of the control voltage of the switching element of the second converter section. In the low voltage output mode and the medium voltage output mode, the control unit turns off the switching elements of the rectifier unit, and in the high voltage output mode, synchronizes with the timing of turning on the switching elements of the first and second converter units. and a secondary side control unit that controls the two switching elements of the rectifying unit so that at least one of the two switching elements of the rectifying unit is turned on for a predetermined time. Since the mode and the high voltage output mode can be switched for implementation, the converter can handle a wide range of output voltages.

Further, according to the converter according to the second embodiment, in the high voltage output mode, the secondary side control unit performs two switching operations of the rectifying unit in synchronization with the timing when the switching elements of the first and second converter units are turned on. One of the elements is turned on for a predetermined period of time, and the other of the two switching elements of the rectifying section is turned on in synchronization with the switching elements of the first and second converter sections. Since the two switching elements in the section are controlled, synchronous rectification is performed, and power loss during output rectification can be reduced. As a result, it contributes to the improvement of power conversion efficiency and the suppression of heat generation.

Note that the converter according to the second embodiment has the same configuration as the converter 1 according to the first embodiment except for the on/off timings of the switching elements Q5 and Q6 in the voltage output mode. It has the corresponding effect among the effects it has.

Although the present invention has been described based on the above embodiments, the present invention is not limited to the above embodiments. Various aspects can be implemented without departing from the spirit of the invention, and for example, the following modifications are also possible.

(1) The numbers, positions, and the like of the components described in each of the above embodiments are examples, and can be changed within a range that does not impair the effects of the present invention.

(2) In each of the above-described embodiments, the input-side switching elements constitute a half-bridge circuit in the primary-side switch units SW1 and SW2, but the present invention is not limited to this. In the primary side switch section, a full bridge circuit may be configured with input side switching elements.

(3) In each of the above embodiments, the first and second converter sections 10 and 20 have the inductors L1 and L2 connected in series with the primary windings T1-1 and T2-1, and the inductor L1 , L2, resonance capacitors C1 and C2, and primary windings T1-1 and T2-1, the present invention is not limited to this. In the first and second converter sections 10 and 20, a resonance circuit may be configured with a parasitic inductor, a resonance capacitor, and a primary winding.

1, 900... converter 10, 910... first converter section 20... second converter section 30, 930... rectifier section 960... additional converter section Q1, Q2, Q3, Q4, Q5, Q6... Switching element SW1, SW2... Primary side switch part, C1, C2, C3, C4... Capacitor, L1, L2... Inductor, T1, T2... Transformer, T1-1, T2-1... Primary winding, T1- 2, T2-2... Secondary winding, D1, D2, D3, D4... Diodes (D3 and D4 are parasitic diodes)

Claims (9)

An isolated resonant converter that outputs a desired output voltage of the same polarity by switching between a low voltage output mode, a medium voltage output mode, and a high voltage output mode,
A transformer having a primary winding and a secondary winding, a primary side switch section connected to the primary winding side of the transformer and configured to form a bridge circuit with primary side switching elements, and the primary winding. Two converter sections connected in parallel each having a resonant capacitor connected in series with the line, the respective secondary windings being connected in series with each other and,
A bridge circuit is formed by two diodes connected to the high side and two secondary side switching elements connected to the low side, which are connected to the secondary winding side of each transformer of the first and second converter sections. a rectifying unit in which the two secondary windings configured and connected in series are connected in parallel with the two diodes and the two secondary switching elements, respectively;
in the medium voltage output mode and the high voltage output mode, synchronizing the on/off timings of the primary side switching element of the first converter section and the primary side switching element of the second converter section; In the low voltage output mode, between the phase of the control voltage of the primary side switching element of the first converter section and the phase of the control voltage of the primary side switching element of the second converter section. a primary side control section that controls on/off timing of the primary side switching elements of the first and second converter sections so as to generate a phase difference;
In the low-voltage output mode and the medium-voltage output mode, the secondary-side switching element of the rectifying unit is turned off, and in the high-voltage output mode, primary-side switching of the first and second converter units. (2) controlling the two secondary side switching elements of the rectifying section so that at least one of the two secondary side switching elements of the rectifying section is turned on for a predetermined period of time in synchronization with the timing at which the element is turned on; A converter, comprising: a secondary side control unit. In the low voltage output mode, the primary side control section controls the phase of the control voltage of the primary side switching element of the first converter section and the primary side switching element of the second converter section. 2. A converter as claimed in claim 1, wherein the output voltage is controlled by adjusting the phase difference between the phases of the voltages. In the high-voltage output mode, the secondary-side control unit controls the output voltage by adjusting the ON time of at least one of the two secondary-side switching elements of the rectifying unit. 3. A converter according to claim 1 or 2, characterized by: In the high-voltage output mode, the secondary-side control section controls the two secondary-side switching elements of the rectifying section so that the two secondary-side switching elements of the rectifying section are synchronously turned on for a predetermined period of time. 4. The converter according to any one of claims 1 to 3, characterized in that it controls In the high-voltage output mode, the secondary-side control unit switches the two secondary-side switching elements of the rectifying unit in synchronization with the timing at which the primary-side switching elements of the first and second converter units turn on. One of the elements is turned on and kept on for a predetermined time, and the two secondary side switching elements of the rectifying section are switched in synchronization with the primary side switching elements of the first and second converter sections. 4. The converter according to claim 1, wherein said two secondary side switching elements of said rectifying section are controlled such that the other of them is turned on and this is alternately repeated. 6. The converter according to claim 1, wherein in said primary side switch section, said primary side switching elements constitute a half bridge circuit. 6. The converter according to claim 1, wherein in said primary side switch section, said primary side switching elements constitute a full bridge circuit. The first and second converter sections further have an inductor connected in series with the primary winding,
8. The converter according to claim 1, wherein in said first and second converter sections, said inductor, said resonant capacitor and said primary winding constitute a resonant circuit. 8. The converter according to claim 1, wherein the parasitic inductor, the resonant capacitor, and the primary winding constitute a resonant circuit in the first and second converter sections.

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