JP7275897B2 - radar system - Google Patents

Description

The present invention relates to radar systems.

In recent years, the use of MIMO (Multiple-Input and Multiple-Output) technology in radar systems has been studied. Since a radar system using MIMO technology can virtually increase the number of antennas that form a phased array, it is possible to improve the resolution of the azimuth angle in which a target exists.

Transmission channel separation systems for radar systems are classified into a TDM-MIMO system in which a transmitter is turned on and off for each channel, and a DDM-MIMO system in which the phase of a transmission signal is rotated for each channel. Since the TDM-MIMO system operates only one transmission channel during a certain period of time, the channel separation performance can be improved. However, since the TDM-MIMO system requires a longer time interval for sampling the received signal of each channel, the frequency of the observable received signal becomes lower, lowering the upper limit of the measurable speed.

On the other hand, in the DDM-MIMO system, since the transmitters of each channel operate simultaneously, the channel separation performance is low, but the time interval for sampling the received signal of each channel can be shortened, and the speed detection range can be widened. . For this reason, it is desirable to use the DDM-MIMO system, especially for a vehicle-mounted radar system.

The MIMO radar discriminates the radar waves of each transmission branch when receiving the signals reflected by the target from the radar waves transmitted by a plurality of transmission branches. In the technique described in Patent Document 1, P (=M×N) pulse waves that have arrived are subjected to demodulation processing corresponding to each of N antennas, P demodulated signals are generated, and known discrete values are obtained. The phase differences of the P demodulated signals are aligned based on the sequence of , and the phases are adjusted by generating the P in-phase signals. This is a method of separating signals by performing code modulation corresponding to each transmission channel and acquiring the corresponding code and correlation peak value on the receiving side. Moreover, the circuit size increases due to the increase in memory.

Also, when a radar system uses a DDM-MIMO system using multi-level phase shift keying such as QPSK, the phase and amplitude of the transmitter tend to vary due to changes in the manufacturing process and environmental temperature. This variation tends to be larger than when BPSK modulation is applied. For this reason, the performance of separating received signals corresponding to transmission signals from a plurality of transmitters is likely to deteriorate.

JP 2016-50778 A

An object of the present disclosure is to provide a radar system that can be configured without deteriorating the separation performance of received signals as much as possible even if a change in manufacturing process or environmental temperature occurs.

The invention according to claim 1 is a radar system ( 1). A plurality of transmitters (31 . . . 3N) are provided for at least three or more N channels, and are configured to irradiate the target 2 with radar waves using transmission signals of the transmission channels for the N channels. A plurality of receivers (41 . . . 4M) are provided for a plurality of M channels and receive a plurality of reception signals corresponding to a plurality of transmission signals. A signal processing unit (9) processes a plurality of received signals.

Each of the plurality of transmitters includes a binary phase shifter (11) that shifts the phase to a phase shift value (φ1 ... φN) of 0 or π in chirp period units of the FCM modulation method, and at least one The above transmitters (33; 33, 34) are provided with a transmission power adjuster (10) capable of adjusting transmission power in chirp cycle units. The plurality of transmitters is configured to irradiate the target with radar waves with transmission signals generated at phase steps of higher resolution than the phase step π of the binary phase shifter.

The signal processing unit obtains a plurality of received signals corresponding to a plurality of transmitters from the reflected waves reflected by the target, samples the plurality of received signals at a repetition interval (T2) of the chirp period (T1), and performs a first FFT. Received signals corresponding to a plurality of transmitted signals are separated from each other based on the result of fast Fourier transform and first FFT. A plurality of transmitters are provided for three channels including a first transmitter (31), a second transmitter (32) and a third transmitter (33). The first transmitter irradiates the target with a radar wave using a transmission signal generated with a predetermined transmission power while setting the phase shift value (φ1) of the binary phase shifter to 0. The second transmitter repeatedly changes the phase shift value (φ2) of the binary phase shifter with two cycles of 0 and π in units of chirp cycles, and transmits a radar signal generated with a predetermined transmission power. Shoot a wave at a target. The third transmitter repeats the change with four cycles of 0, π/2, π, and 3π/2 while shifting the reference phase value (θ3) from 0 by phase steps of π/2 in chirp cycle units. , the phase shift value (φ3) of the binary phase shifter is set to 0, 0 or π, π, 0 or π, respectively, and the transmission power is set to zero when the reference phase value is π/2 or 3π/2. When the reference phase values are 0 and π, the target is irradiated with a radar wave by a transmission signal having a predetermined transmission power. The signal processing unit performs sampling processing on a plurality of received signals according to the resolution of the repetition interval (T2) of the chirp period (T1) and fast Fourier transforms the first transmitter, the second transmitter, and the third transmitter. 0, .pi., .pi./2, or 3.pi./2 corresponding to each phase difference in units of chirp cycles are detected as angular velocities, and a plurality of received signals are separated from each other according to the detection results of the angular velocities.

According to the first aspect of the present invention, a plurality of transmitters adjust transmission power while changing the phase shift keying method in units of chirp periods for each transmission channel, thereby transmitting radar waves with high-resolution phase steps. can be output. For this reason, compared to the case of applying the DDM-MIMO system using QPSK modulation, which requires control of the phase of the transmission signal with relatively high resolution, the phase and amplitude of the transmission signal vary due to the manufacturing process and temperature changes. can be suppressed. On the receiving section side, the separation performance of the received signals of all the transmission channels is not degraded as much as possible.

Electrical configuration diagram of a radar system according to one embodiment Explanatory diagram of control method for transmission power adjustment unit and binary phase shifter Electrical block diagram of a radar system with 3 transmitter channels Explanatory diagram showing the relationship between the reference phase value, the number of repetitions, the phase shift value of the phase shifter, and the voltage amplitude value, and the frequency change of the transmission signal Flowchart showing outline of processing contents Frequency transition diagram with respect to time of transmitted signal and received signal Schematic representation of power distribution with respect to angular velocity Results of plotting the results of peak detection using FFT in a matrix Explanatory diagram showing the phase change with respect to the reference phase of the received signal Electrical block diagram of a radar system with 4 transmitter channels Explanatory diagram showing the relationship between the reference phase value, the number of repetitions, the phase shift value of the phase shifter, and the voltage amplitude value, and the frequency change of the transmission signal Schematic representation of power distribution with respect to angular velocity

Hereinafter, embodiments of radar systems 1, 201, and 301 will be described with reference to the drawings.
The radar system 1 shown in FIG. 1 irradiates a target 2 with a radar wave frequency-modulated by the FCM (Fast Charp Modulation) modulation method, and measures the measurement target based on the received signal of the reflected wave reflected by the target 2. System. The objects to be measured include the distance from the radar system 1 to the target 2, the moving speed of the target 2, the azimuth angle at which the target 2 exists, and the like. In the FCM modulation method, the frequency is gradually increased or decreased from a predetermined initial frequency to the final frequency, and then instantaneously returned to the initial frequency. (see FIG. 6).

The radar system 1 includes a transmitter 3 including a plurality of transmitters 31 . . . 3N for at least three or more N channels, a receiver 4 including a plurality of receivers 41 . An integrated circuit 8 having a local oscillation signal generator 5 , a control register 6 and a logic circuit 7 , and a signal processing section 9 are connected to each other.
The signal processing unit 9 causes the control register 6 built in the integrated circuit 8 to store the control command, so that the logic circuit 7 of the integrated circuit 8 executes the control command. Although the integrated circuit 8 and the signal processing unit 9 are shown separately configured, the present invention is not limited to this and may be configured integrally, and mounted on one circuit (for example, the integrated circuit 8). The functions provided may be incorporated in the other circuit (for example, the signal processing section 9).

3N, and one of the transmitters 31 . . . 3N is hereinafter referred to as the "nth transmitter 3n" when necessary. Also, the channel variable of the receivers 41 . . . 4M is set to m, and one of the receivers 41 . The transmitters 31 . . . 3N irradiate the targets 2 with radar waves using the millimeter-wave band transmission signals TX generated respectively. The receivers 41 . . . 4M receive a plurality of reception signals RX corresponding to the plurality of transmission signals TX, respectively, and the signal processing unit 9 processes the reception signals RX by the receivers 41 .

The local oscillation signal generator 5 is composed of a so-called PLL circuit, generates a local oscillation signal based on a control signal input from the logic circuit 7, thereby generating a modulated signal by the FCM modulation method, Output to the unit 4 respectively. A frequency doubler or frequency tripler may be provided between the local oscillation signal generator 5 and the transmitter 3 and receiver 4 as required.

Transmitters 31 . . . 3N are configured by connecting a transmission power adjustment section 10, a binary phase shifter 11, a power amplifier 12, and an antenna element 13, respectively. The transmission power adjustment unit 10 enables adjustment of transmission power for each chirp period T1. The transmission power adjustment unit 10 may be provided in one or more of the transmitters 31 . . . 3N. good. Also, as will be described later, when the number of transmission channels N is four, the transmission power adjustment section 10 may be provided in the third transmitter 33 and the fourth transmitter 34 .

The binary phase shifter 11 shifts the phase with a phase shift value φ1 . . . φN of 0 or .pi. A signal is output to the power amplifier 12 . The power amplifier 12 power-amplifies the signal phase-shifted by the binary phase shifter 11 and outputs the amplified signal to the antenna element 13 .

As schematically shown in FIG. 2, the transmission power adjustment section 10 includes an input stage, a power output stage, etc. An N-channel MOSFET 20 is configured in the input stage. The transmission power adjusting unit 10 adjusts the transmission power of the transmission signal TX at the input stage and outputs it. The input stage of the binary phase shifter 11 is also configured with an N-channel MOSFET 21, and the phase can be shifted by adjusting the phase as it is or by inverting the phase to the opposite phase.

On the other hand, the logic circuit 7 includes a variable current source 22 , a MOSFET 23 whose drain-gate is connected, and an analog switch 24 . The logic circuit 7 outputs a bias control signal to the variable current source 22 so that the variable current source 22 can change the output constant current.
The analog switch 24 is connected between the gates of the MOSFETs 20 and 23, and can switch between conduction and interruption between the gates when the logic circuit 7 inputs an enable signal. When the logic circuit 7 validates (enables) the enable signal, the analog switch 24 conducts and the gates of the MOSFETs 20 and 23 conduct. Then, a current mirror circuit is configured at the input of the transmission power adjustment unit 10 . The input bias of the MOSFET 20 can be changed by the logic circuit 7 outputting the bias control signal to the variable current source 22 . Thereby, the logic circuit 7 can adjust the power amplification degree of the transmission power adjusting section 10 .

When the logic circuit 7 disables the enable signal, the analog switch 24 cuts off the gates of the MOSFETs 20 and 23, and grounds the input bias of the MOSFET 20 in the input stage of the transmission power adjustment unit 10 (not shown). Hold. As a result, the transmission power adjustment unit 10 can set the power amplification degree to zero. Since the input circuit of the transmission power adjustment section 10 of each transmitter 31...3N is configured in this manner, the logic circuit 7 can individually stop or operate the transmission signal output of each transmitter 31...3N.

The logic circuit 7 is provided with a voltage dividing circuit 25 using, for example, ladder resistors. The gate bias of MOSFET 21 in the input stage of device 11 can be adjusted. For this reason, the logic circuit 7 can switch between positive phase and reverse phase of the input signal amplitude, and the binary phase shifter 11 can shift the phase of each input signal by 0 or π, thereby performing phase modulation. realizable.

Returning the reference drawing to FIG. 1, the description continues. The antenna elements 13 of the transmitters 31 . . . 3N are arranged in a predetermined arrangement, and the transmission area of the radar wave can be switched by the phased array antenna system. The logic circuit 7 adjusts and outputs the bias control signal, the enable signal, and the phase control signal while causing the local oscillation signal generator 5 to output the modulated signal by controlling the timing. At this time, each n-th transmitter 3n changes the voltage amplitude value An [θn] based on the following equation (1) while changing the reference phase value θn for each chirp cycle T1 of the FCM modulation method. Radar waves are output so that

The reference phase value θn in equation (1) is a phase value that changes each time the chirp period T1 of FCM modulation is repeated, and the phase changes by phase steps 2π/{2̂(n−1)} from the initial value 0. indicate a value. This reference phase value θn changes at each phase step 2π/{2̂(n−1)}, and returns to 0 when the number of repetitions Kn=2̂(n−1) reaches 2π. is. Also, PT in equation (1) indicates the average output power of the transmission signal TX. By each n-th transmitter 3n outputting a radar wave based on this rule, the transmission unit 3 outputs a transmission signal TX whose resolution is higher than the phase step π of each binary phase shifter 11. It is possible to irradiate the target 2 with radar waves.

4M, on the other hand, comprise antenna elements 26, LNAs 27, mixers 28, and intermediate frequency amplifiers 29. In FIG. The antenna elements 26 of the receivers 41 . . . 4M are arranged in a predetermined arrangement, and the reception scanning area of the radar wave can be switched by the phased array antenna system. An antenna element 26 of each receiver 41 . . . 4M receives the radar wave reflected by the target 2 . The LNA 27 performs low-noise amplification on the received signal RX from the antenna element 26 and outputs the amplified signal to the mixer 28 . Mixer 28 mixes the amplified signal of LNA 27 and the modulated signal output from local oscillation signal generator 5 and outputs the result to intermediate frequency amplifier 29 . The intermediate frequency amplifier 29 amplifies the mixed signal input from the mixer 28 and outputs the amplified signal to the signal processing section 9 .

The signal processing unit 9 is composed of a microcomputer having functions such as A/D converters 51 . . . 5M, a data storage unit 60, and FFT processing units 61 and 62. The signal processing unit 9 A/D-converts the signals amplified by the intermediate frequency amplifiers 29 of the receivers 41 . . . 4M using the A/D converters 51 . The FFT processing units 61 and 62 of the signal processing unit 9 have a function of sampling the A/D conversion data stored in the data storage unit 60 at predetermined low resolution and high resolution and performing fast Fourier transform. The signal processing unit 9 uses the A/D conversion data stored in the data storage unit 60 to calculate the measurement object (the distance to the target 2, the moving speed of the target 2, the azimuth angle where the target 2 exists, etc.). do. Details of the processing executed by the signal processing unit 9 will be described in specific first and second embodiments below.

(Example 1)
Up to this point, a generalized configuration in which the number of transmission channels is N and the number of reception channels is M has been described as an example. Hereinafter, an embodiment of a radar system 201 provided with transmitters 31 . . . 33 for N=3 channels will be described. , 3 . . . , 9 . In FIG. 3, the same reference numerals as those of the configuration of FIG. 1 are used for explanation. As shown in FIG. 3, the transmitter 3 has the antenna element 13 of the first transmitter 31 installed in the center thereof, and the antenna elements 13 of the second transmitter 32 and the third transmitter 33 installed on both sides thereof. are doing. It is desirable that the antenna elements 13 are arranged on a straight line at regular intervals.

The n-th transmitter 3n generates the transmission signal TX having the voltage amplitude value Av based on the equation (1) as described above. FIG. 4 shows the reference phase value θn, the number of repetitions kn, the phase shift value φn of the phase shifter, and the voltage amplitude value Av shows the correspondence between
In the above equation (1), the number of iterations K1 corresponding to n=1 is 1, and k1={0} and θ1={0}. Assume that the phase shift value φ1 of the phase shifter 11 is {0}. Further, the logic circuit 7 sets the voltage amplitude value A1 [θ1] to a predetermined reference value SQRT (2PT) (described as a relative value 0 [dB] in FIG. 4). SQRT indicates square root. By setting the logic circuit 7 in this way, the first transmitter 31 irradiates the target 2 with radar waves having a predetermined transmission power.

In the above equation (1), the number of repetitions K2 corresponding to n=2 is 2, k2={0, 1} and .theta.2={0, .pi.}. The logic circuit 7 repeatedly changes the phase shift value φ2 of the binary phase shifter 11 of the second transmitter 32 with two cycles of {0, π}. As a result, the second transmitter 32 repeats changing the phase shift value φ2 of the binary phase shifter 11 twice with 0 and π in units of the chirp period T1.
At this time as well, the logic circuit 7 sets the voltage amplitude value A2 (θ2) of each chirp wave to the predetermined reference value SQRT (2PT) (relative value 0 [dB]). By setting the logic circuit 7 in this manner, the second transmitter 32 irradiates the target 2 with a radar wave having a predetermined transmission power while performing BPSK modulation in units of chirp period T1.

In the above equation (1), the number of repetitions K3 corresponding to n=3 is 4, k3={0, 1, 2, 3}, θ3={0, π/2, π, 3π/2}. . At this time, the third transmitter 33 shifts the reference phase value θ3 from 0 by phase steps of π/2 in units of chirp cycle T1, and changes four times of 0, π/2, π, and 3π/2 as one cycle. repeat.
When the reference phase value θ3=π/2 and 3π/2, the voltage amplitude value A3(θ3) in the equation (1) is zero. Therefore, when the logic circuit 7 sets the voltage amplitude value A3 (θ3) of the third transmitter 33 to 0 and does not output the transmission signal TX, the phase shift value φn of the binary phase shifter 11 is set to 0. or π. At this time, the logic circuit 7 repeatedly changes the phase shift value φ3 of the binary phase shifter 11 four times {0, 0 or π, π, 0 or π} as one cycle. By setting the logic circuit 7 in this way, the third transmitter 33 performs pseudo QPSK modulation in units of chirp period T1, and sets the transmission power to zero when the reference phase value θ3 is π/2 and 3π/2. , the target 2 is irradiated with a radar wave having a predetermined transmission power when the reference phase value .theta.3 is 0 or .pi.

Each of the transmitters 31 . . . 33 outputs radar waves to the target 2 while changing the reference phase value θn, the phase shift value φn of the phase shifter, and the number of repetitions kn based on such rules.

When the radar wave from the transmitter 3 is reflected by the target 2 , it is input to the receiver 4 of the radar system 1 . 4M receive the radar wave reflected by the target 2, the receivers 41 . . .

As shown in FIG. 5, when the signal processing unit 9 receives the reception signal RX of the receiving unit 4, it performs A/D conversion processing by the A/D converters 51 . . . 5M. get. After acquiring the AD conversion data of the reception signal RX corresponding to the transmission signal TX of the transmitters 31 . By executing the speed FFT (corresponding to the first FFT) by , the received signal RX is sampled at the low resolution of the repetition interval T2 of the adjacent chirp period T1 and fast Fourier transformed. Since the phase rotation speeds of the reference phase values θ1 . Angular velocities corresponding to the phase rotation velocities of the reference phase values θ1 . . . θ3 can be detected by performing FFT. As a result, 0, π, π/2, or 3π/2 corresponding to each phase difference of the first transmitter 31, second transmitter 32, and third transmitter 33 in units of chirp period T1 can be detected as angular velocities.

In S3, the signal processing unit 9 uses the received signal RX in one chirp period T1 shown in FIG. By executing the distance FFT by the FFT processing unit 62 of the signal processing unit 9, the delay time from when the radar wave is output to when it is reflected by the target 2 and the signal is received can be detected. You can find the distance up to 2. The signal processing unit 9 can also obtain the distance based on detecting the difference between the frequency of the transmission signal TX and the frequency of the reception signal RX as the beat frequency.
Then, the signal processing unit 9 can obtain the power distribution on the angular velocity axis as schematically shown in FIG. 7 by performing peak detection by CFAR (Constant False Alarm Rate) in S4.

For example, when an m-th receiver 4m receives radar waves reflected by two targets 2 at different distances, peak detection results using distance FFT and velocity FFT are plotted in a matrix, as shown in FIG. A power spectrum distribution is obtained as shown. In FIG. 8, transmission signals Tx1 . . . Tx3 of transmission channels CH1 . . . In S5, the signal processing unit 9 pairs-matches the power spectra of a group of signals Z1 that are considered to have the same distance to the target 2, thereby selecting the group of signals Z1 having the pair-matched power spectra.

Next, the signal processing unit 9 detects the reception signal RX of the m-th receiver 4m that has received the transmission signals Tx1 . . . , Tx3 of all the transmitters 31 . Get the phase. At this time, when the phase obtained by the signal processing unit 9 exceeds the number of transmission channels N, it is assumed that the image component is detected by the mixer 28 . In this case, the signal processor 9 removes the image component in S6. At this time, the signal processing unit 9 extracts and separates N signals whose phases are considered to be at equal intervals in S6.

For example, when the signal processing unit 9 detects phases as shown in FIG. 9, it is possible to extract and separate a group of N signals Zx that can be regarded as having equal intervals. The other signal Tx3 (Image) can be regarded as removed or discarded. The phase of the signal received by the receiver 4 also changes based on the relationship between the installation positions of the antenna elements 13 of the transmission channels CH1 . . . CH3 and the azimuth angle of the target 2 . For example, as schematically shown in FIG. 3, the antenna element 13 of the second transmitter 32 is located relatively close to the target 2 and the antenna element 13 of the third transmitter 33 is located relatively far from the target 2. In this case, the phase of the reception signal RX corresponding to the transmission signals Tx2, Tx1, and Tx3 of the second transmitter 32, the first transmitter 31, and the third transmitter 33 increases in order with respect to the reference phase. Therefore, the signal processing unit 9 can estimate the azimuth angle at which the target 2 exists in S7 based on the phase detection values detected at approximately equal intervals.

The relationship with the azimuth angle at which the target 2 exists is determined when the arrangement of the antenna elements 13 of the transmitters 31 . . . Therefore, the phase of the image component is also determined corresponding to the azimuth angle of the target 2 . For example, when a manufacturing company mass-produces and inspects a large number of radar systems 1, a corner reflector is regarded as a pseudo target 2 and installed at a predetermined angle with respect to the radar system 1 to detect image components. Then, the correspondence between the azimuth angle at which the target 2 exists and the phase of the image component can be obtained for each manufactured radar system 1 .

The manufacturing company stores this correspondence relationship in a non-volatile memory (not shown) mounted on the signal processing unit 9 before shipment of the signal processing unit 9, so that the signal processing unit 9 can detect the peak by the above-mentioned CFAR. After detection, the image component can be easily removed by selecting the received signal RX based on the correspondence stored in the non-volatile memory. As a result, necessary signals can be easily separated and obtained.

(Example 2)
A specific example of the radar system 301 provided with the transmitters 31 to 34 for four channels will be described below with reference to FIGS. 10 to 12. FIG. As shown in FIG. 10, the transmitter 3 has a first transmitter 31 and a second transmitter 32 installed in the center thereof, and a third transmitter 33 and a fourth transmitter 34 installed on both sides thereof. . It is desirable that the antenna elements 13 of these transmitters 31 . . . 34 are arranged on a straight line at regular intervals. Since other configurations are the same as those of the first embodiment, description thereof is omitted.

As shown in the first embodiment, the n-th transmitter 3n generates the transmission signal TX based on the voltage amplitude value An(θn) shown in equation (1). FIG. 11 shows the reference phase value θn, the number of repetitions kn, and the binary phase shifter 11 that change over time in the radar wave based on the transmission signal TX output by the transmitters 31 . . . 34 of the transmission channels CH1 . and the voltage amplitude value Av. The reference phase value θn, the number of repetitions kn, the phase shift value φn of the phase shifter, and the voltage amplitude value Av of the first transmitter 31, the second transmitter 32, and the third transmitter 33 are the same as those in the first embodiment. omitted.

In equation (1), the number of repetitions K4 corresponding to n=4 is 8, k4={0, 1, 2, 3, 4, 5, 6, 7}, θ4={0, π/4, π /2, 3π/4, π, 5π/4, 3π/2, 7π/4}. The fourth transmitter 34 shifts the reference phase value θ4 from 0 by phase steps of π/4 in units of chirp period T1, and shifts the reference phase value θ4 to 0, π/4, π/2, 3π/4, π, 5π/4, 3π. /2 and 7π/4 are repeated eight times as one cycle.
When the reference phase value θ4=π/2 and 3π/2, the voltage amplitude value A4(θ4) in the equation (1) becomes 0. Therefore, when the logic circuit 7 does not output the transmission signal TX by setting the voltage amplitude value A4 (θ4) of the fourth transmitter 34 to 0, the phase shift value φn of the binary phase shifter 11 is set to It may be set to 0 or π. When the reference phase value θ4 = π/4, 3π/4, 5π/4, 7π/4, the voltage amplitude value A4 (θ4) is SQRT(PT) based on the equation (1), and the relative value is -3 [dB ].

The logic circuit 7 repeatedly changes the phase shift value φ4 of the binary phase shifter 11 of the fourth transmitter 34 with {0, 0, 0 or π, π, π, π, 0 or π, 0} eight times as one cycle. . By setting the logic circuit 7 in this way, the fourth transmitter 34 outputs a predetermined transmission power when the reference phase value θ4 is 0 or π, and a predetermined transmission power when the reference phase value θ4 is π/2 or 3π/2. When the transmission power is zero and the reference phase value θ4 is π/4, 3π/4, 5π/4, or 7π/4, the transmission signal TX is generated so as to set the transmission power to half of the predetermined value. to irradiate.

Radar waves based on the transmission signals of the transmitters 31 . 4M receive the radar wave reflected by the target 2, the receivers 41 . . .

As shown in FIG. 5, the signal processing unit 9 receives the received signal RX from the receivers 41 . Get to 60. After acquiring the AD converted data of the received signal RX, the signal processing unit 9 uses the AD converted data in S2 to execute a speed FFT (corresponding to the first FFT) by the FFT processing unit 61, thereby reducing the adjacent chirp period T1. The received signal RX is sampled at a low resolution of repetition interval T2 and fast Fourier transformed. At this time, since the phase rotation speed at each repetition interval T2 of the reference phase values θ1 . . . . , the angular velocity corresponding to the phase rotation velocity can be detected by sampling and velocity FFT as described above. As a result, the signal processing unit 9 calculates π, 0, π corresponding to each phase difference in units of chirp period T1 in the first transmitter 31, the second transmitter 32, the third transmitter 33, and the fourth transmitter 34. /2 or 3π/2, or π/4 or 7π/4 can be detected as angular velocities, respectively.

In S3, the signal processing unit 9 uses the received signal RX during one chirp period T1 to perform a distance FFT (equivalent to the second FFT) at a high resolution using the FFT processing unit 62, thereby obtaining a target 2 as described above. can be calculated.
The signal processing unit 9 can obtain a power distribution with respect to the angular velocity corresponding to the moving velocity of the target 2 as shown in FIG.

After that, the signal processing unit 9 can separate the received signal RX by performing the same signal processing as in the first embodiment, and can estimate the moving speed of the target 2 and the azimuth angle where the target 2 exists. Although an example of N=3 channels was shown in the first embodiment and an example of N=4 channels was shown in the second embodiment, the present invention is not limited to this, and can be generalized to N=5 or more channels.

Hereinafter, the configuration of this embodiment will be conceptually summarized and its effects will be described.
According to this embodiment, the transmitters 31 . A transmission power adjustment unit 10 is provided which enables adjustment of the transmission power in units of period T1. The transmission unit 3 is configured to irradiate the target 2 with a radar wave using a transmission signal TX generated at a phase step having a higher resolution than the single phase step π of the binary phase shifter 11 .

The signal processing unit 9 obtains a plurality of received signals RX corresponding to the plurality of transmitters 31 . A fast Fourier transform (speed FFT) is performed, and the received signals RX corresponding to a plurality of transmitted signals TX are separated from each other based on the result of this transform.

In this embodiment, attention is focused on the BPSK modulation method, which has relatively little variation due to manufacturing processes and temperature changes, and the high separation performance of the received signal RX by the TDM-MIMO method, and each transmitter 31 . . . By changing the amplitude of the transmission signal TX stepwise while changing the phase shift keying method in units of chirp period T1 for each CHN, a radar wave is output with a high-resolution phase step. For this reason, compared to the case of applying the DDM-MIMO system using QPSK modulation, which requires the phase of the transmission signal TX to be controlled with a relatively high resolution, the phase and amplitude of the transmission signal TX are affected by changes in the manufacturing process and temperature. variation can be suppressed. As a result, on the receiving section 4 side, the separation performance of the received signals RX of all the transmission channels CH1 .

In addition, the signal processing unit 9 performs sampling processing at high resolution in one chirp period T1, performs fast Fourier transform (distance FFT) by the FFT processing unit 62, and calculates up to the target 2 calculated using the FFT processing unit 62. A group of signals Z1 and Z2 whose distances are considered to be the same are pair-matched, and after selecting the pair-matched group of signals Z1 and Z2, the FFT processing unit 61 is used to convert by velocity FFT. separates the signals based on Therefore, after separating the signals according to the distance to the target 2, the signal processing unit 9 can separate and discriminate the signals based on the angular velocities of the signals received from all the transmission channels CH1 . . . distinguishable. This allows easy discrimination.

Further, the signal processing unit 9 acquires the phase of the separated received signal RX with respect to the reference phase, and when the acquired phase exceeds the number of transmission channels N, extracts N signals whose phases are considered to be at equal intervals. Then, the azimuth angle at which the target 2 exists is estimated based on the extracted N received signals RX. Therefore, the image component generated in the mixer 28 can be removed due to the hardware configuration, and the azimuth angle at which the target 2 exists can be obtained with high accuracy.

(Other embodiments)
The present invention is not limited to the above-described embodiments, and for example, the following modifications or extensions are possible.
In the above-described embodiment, an example in which the number of transmission channels is 3 or 4 has been described, but it can be generalized to N channels. At this time, the n-th transmitter 3n changes the reference phase value θn by a phase step of 2π/{2̂(n−1)} in units of the chirp cycle T1, and repeats the number of repetitions Kn=2̂(n−1) for one cycle. , the voltage amplitude value Av of the transmission signal TX of the n-th transmitter 3n is used as the transmission signal TX based on the equation (1), and the target 2 is irradiated with the radar wave.

Then, the signal processing unit 9 samples a plurality of reception signals RX received by being reflected by the target 2 at a low resolution of the repetition interval T2 of the chirp period T1, and fast Fourier transforms the plurality of reception signals RX. As a result, 2π/{2̂(n−1)} corresponding to each phase difference in units of chirp period T1 of the n-th transmitter 3n can be detected as angular velocities, and a plurality of received signals are obtained based on the angular velocity detection results. RX can be isolated from each other. Therefore, even if it is generalized to N channels, a plurality of received signals RX can be separated from each other in the same way, and the same effects as those of the above-described embodiment can be obtained.

In the above-described embodiment, the configuration in which the transmitters 31 . . . 3N of all the transmission channels CH1 . . . At least one or more transmitters (for example, 33; 33 and 34) that require transmission power adjustment may be provided with the transmission power adjustment unit 10 . As a method of modulating the phase in units of chirp period T1, a form using BPSK or pseudo QPSK was used. method may be applied.

Although the present disclosure has been described in accordance with the embodiments described above, it is understood that the present disclosure is not limited to such embodiments or structures. The present disclosure also includes various modifications and modifications within the equivalent range. In addition, various combinations and configurations, as well as other combinations and configurations including one, more, or less elements thereof, are within the scope and spirit of this disclosure.

In the drawings, 1 is a radar system, 2 is a target, 3 is a transmitter, 31 . . . 3N is a transmitter, 4 is a receiver, 41 . indicates a binary phase shifter, and 61 and 62 indicate FFT processing units (first FFT and second FFT).

Claims (5)

A radar system (1) that irradiates a target (2) with a radar wave frequency-modulated by an FCM (Fast Charp Modulation) modulation method and measures a measurement target based on a signal that receives a reflected wave reflected by the target, ,
a plurality of transmitters (31...3N) provided for at least three or more N channels and configured to irradiate the target with the radar wave according to the transmission signal of each transmission channel;
a plurality of receivers (41...4M) provided for a plurality of M channels and respectively receiving a plurality of reception signals corresponding to the transmission signals for the N channels;
A signal processing unit (9) for signal processing a plurality of received signals,
Each of the plurality of transmitters includes a binary phase shifter (11) that shifts the phase to a phase shift value (φ1 . . . φN) of 0 or π in chirp period units of the FCM modulation method,
At least one or more of the plurality of transmitters (33; 43, 44) includes a transmission power adjustment unit (10) capable of adjusting transmission power in units of the chirp period, configured to irradiate the target with the radar wave by a transmission signal generated at a phase step having a higher resolution than the phase step π of the phase shifter,
The signal processing unit is
When the reflected waves reflected by the target are acquired as a plurality of received signals corresponding to the plurality of transmitters, the plurality of received signals are sampled according to the resolution of the repetition interval (T2) of the chirp period (T1), and fast Fourier processing is performed. A first FFT (61) for conversion is provided, and the plurality of received signals corresponding to the N channels of transmission signals are separated from each other based on the conversion result converted by the first FFT,
The plurality of transmitters are provided for three channels including a first transmitter (31), a second transmitter (32), and a third transmitter (33),
The first transmitter irradiates the target with a radar wave using a transmission signal generated at a predetermined transmission power while setting a phase shift value (φ1) of the binary phase shifter to 0;
The second transmitter repeatedly changes the phase shift value (φ2) of the binary phase shifter with two cycles of 0 and π in units of the chirp cycle, and is generated at the predetermined transmission power. irradiating the target with a radar wave by a transmission signal;
The third transmitter shifts the reference phase value (θ3) from 0 by phase steps of π/2 in units of the chirp cycle, and changes four times of 0, π/2, π, and 3π/2 as one cycle. is repeated, the phase shift value (φ3) of the binary phase shifter is respectively set to 0, 0 or π, π, 0 or π, and the transmission power when the reference phase value is π/2, 3π/2 is set to zero and the target is irradiated with a radar wave by a transmission signal having the predetermined transmission power when the reference phase value is 0 or π;
The signal processing unit performs sampling processing on the plurality of received signals with a resolution of a repetition interval (T2) of the chirp period (T1) and fast Fourier transforms them, so that the first transmitter, the second transmitter, and 0, π, π/2, or 3π/2 corresponding to each phase difference in units of the chirp period of the third transmitter are detected as angular velocities, and the plurality of received signals are separated from each other according to the detection results of the angular velocities. radar system. the plurality of transmitters are provided for four channels including the first transmitter, the second transmitter, the third transmitter, and a fourth transmitter (34);
The fourth transmitter shifts the reference phase value (θ4) from 0 by phase steps of π/4 in units of the chirp period to 0, π/4, π/2, 3π/4, π, 5π/4. , 3π/2, and 7π/4 are repeated eight times as one cycle, the phase shift value of the binary phase shifter is changed to 0, 0, 0 or π, π, π, π, 0 or π, 0, the predetermined transmission power when the reference phase value is 0 or π, the transmission power is zero when the reference phase value is π/2 or 3π/2, the reference phase value is π/4, irradiating the target with the radar wave by the transmission signal generated so that the transmission power is half the predetermined transmission power when 3π/4, 5π/4, or 7π/4;
The signal processing unit performs sampling processing on the plurality of received signals according to the resolution of the repetition interval of the chirp period and fast Fourier transforms them, thereby obtaining the first transmitter, the second transmitter, the third transmitter, and π, 0, π/2 or 3π/2, or π/4 or 7π/4 corresponding to each phase difference in units of the chirp period in the fourth transmitter are detected as angular velocities, respectively, and based on the detection results of the angular velocities 2. The radar system of claim 1, wherein said plurality of received signals are separated from each other. A radar system (1) that irradiates a target (2) with a radar wave frequency-modulated by an FCM (Fast Charp Modulation) modulation method and measures a measurement target based on a signal that receives a reflected wave reflected by the target, ,
a plurality of transmitters (31...3N) provided for at least three or more N channels and configured to irradiate the target with the radar wave according to the transmission signal of each transmission channel;
a plurality of receivers (41...4M) provided for a plurality of M channels and respectively receiving a plurality of reception signals corresponding to the transmission signals for the N channels;
A signal processing unit (9) for signal processing a plurality of received signals,
Each of the plurality of transmitters includes a binary phase shifter (11) that shifts the phase to a phase shift value (φ1 . . . φN) of 0 or π in chirp period units of the FCM modulation method,
At least one or more of the plurality of transmitters (33; 43, 44) includes a transmission power adjustment unit (10) capable of adjusting transmission power in units of the chirp period, configured to irradiate the target with the radar wave by a transmission signal generated at a phase step having a higher resolution than the phase step π of the phase shifter,
The signal processing unit is
When the reflected waves reflected by the target are acquired as a plurality of received signals corresponding to the plurality of transmitters, the plurality of received signals are sampled according to the resolution of the repetition interval (T2) of the chirp period (T1), and fast Fourier processing is performed. A first FFT (61) for conversion is provided, and the plurality of received signals corresponding to the N channels of transmission signals are separated from each other based on the conversion result converted by the first FFT,
The plurality of transmitters are composed of first to N-th transmitters (31 to 3N) where n is a channel variable of 1 to N of the N channels,
The n-th transmitter (3n) repeats Kn=2^(n-1) while changing the reference phase value (θn) by phase steps 2π/{2^(n-1)} in units of the chirp period. When it is repeated as one cycle, the average output power PT of the transmission power of the transmission signal of the n-th transmitter and the voltage amplitude value An [θn] are defined as follows. (1) irradiating the target with a radar wave as a transmission signal based on the formula;
The signal processing unit performs a fast Fourier transform on a plurality of received signals received as the radar waves are reflected from the target by sampling the plurality of received signals according to the resolution of the repetition interval of the chirp cycle, 2π/{2̂(n−1)} corresponding to each phase difference in units of the chirp period of the n-th transmitter is detected as an angular velocity, and the plurality of received signals are separated from each other based on the detection result of the angular velocity. radar system.

further comprising a second FFT (62) for sampling and fast Fourier transforming in one said chirp period;
The signal processing unit pairs-matches a group of signals that are considered to have the same distance to the target calculated using the second FFT, selects the pair-matched group of signals, and performs the first FFT. 4. A radar system according to any one of claims 1 to 3, wherein said received signals are separated based on said transformation result transformed by . After separating the received signal, the signal processing unit obtains a phase of the separated received signal with respect to a reference phase, and when the obtained phase exceeds the N, the phases of the received signal are equal intervals. 5. A radar system according to any one of claims 1 to 4, wherein the N signals to be considered are sampled and based on the sampled N signals the azimuth angle in which the target lies is estimated.

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