JP6870135B2 - element - Google Patents

Description

The present invention relates to an element that oscillates or detects a terahertz wave.

As a current injection type light source that generates electromagnetic waves (hereinafter referred to as "terahertz waves") in the frequency range from the millimeter wave band to the terahertz band (30 GHz or more and 30 THz or less), an oscillator in which a resonator is integrated in a negative resistance element is used. Well known. Patent Document 1 describes a terahertz wave oscillator in which a double barrier type resonance tunnel diode (hereinafter, sometimes simply referred to as “RTD”), which is a negative resistance element, and a microstrip resonator are integrated on the same substrate. Are listed.

It is known that an oscillator using a negative resistance element causes parasitic oscillation caused by a bias circuit including a power supply and wiring for adjusting the bias voltage of the negative resistance element. Parasitic oscillation refers to parasitic oscillation in a frequency band on the low frequency side different from the desired frequency, and reduces the oscillation output at the desired frequency.

In order to suppress parasitic oscillation, if the wavelength of the terahertz wave oscillated from the oscillator is λ _THz and the oscillation frequency is f _THz , it is necessary to lower the impedance on the bias circuit side in the frequency region of DC or more and _{less than f THz.} As a countermeasure, a method of arranging a circuit having a resistor and a capacitance at a position within _{λ THz} / 4 toward the power supply side when viewed from the RTD has been proposed. As a means for realizing this, Non-Patent Document 1 describes an oscillator using a slot type resonator. In Non-Patent Document 1, as shown in FIG. 11, the rectifier diode 15 is integrated at a position within _{λ THz} / 4 toward the power supply 16 side when viewed from the S-RTD 11. The resistance 17 is the sum of the internal resistance of the power supply 16 and the resistance of the connecting wire.

Japanese Unexamined Patent Publication No. 2006-101495

IEEE Electronics Letters, Vol 18, 218 (1997)

Since the method described in Non-Patent Document 1 only describes the configuration for a slot type resonator, it cannot be applied to an oscillator using a microstrip type resonator such as a patch antenna disclosed in Patent Document 1. It wasn't easy. _{For example, if the resonator is a patch antenna, the region within λ THz} / 4 toward the bias circuit side when viewed from the negative resistance element is the patch antenna itself or a region with peripheral members close to the patch antenna. Is. Therefore, in the microstrip type resonator, it is not easy to arrange the circuit without interfering with the resonator. Further, since the wiring structure required for adjusting the bias voltage of the negative resistance element is close to the resonator, it is necessary to reduce the parasitic oscillation at a relatively high frequency due to the structure.

In view of the above problems, it is an object of the present invention to provide an element capable _{of reducing parasitic oscillation caused by a wiring structure and oscillating a terahertz wave having a desired oscillation frequency of f THz more stably.}

The element as one aspect of the present invention is an element that oscillates or detects a terahertz wave, and has a resonant portion having a differential negative resistance element, a first conductor, a second conductor, and a dielectric, and the differential negative. It has a bias circuit that supplies a bias voltage to the sex resistance element and a line that connects the resonance portion and the bias circuit, and the differential negative resistance element and the dielectric are the first conductor. The line is arranged between the second conductor and the absolute resistance of the differential negative resistance element at _{the frequency fLC} of resonance due to the inductance of the line and the capacitance of the resonance portion. An element characterized by being a low-impedance line based on a value.

According to the element as one aspect of the present invention, the parasitic oscillation caused by the wiring structure can be reduced, and _{the terahertz wave having a desired oscillation frequency of f THz} can be oscillated or detected more stably.

The figure explaining the structure of the element of embodiment. The figure explaining the admittance characteristic of the element of embodiment. The figure explaining the structure of the element of Example 1. FIG. The figure explaining the structure of the element of the modification 1 described in Example 2. FIG. The figure explaining the structure of the element of the modification 2 described in Example 2. FIG. The figure explaining the admittance characteristic of the oscillator of the modification 2 which is described in Example 2. FIG. The figure explaining the structure of the element of the modification 3 described in Example 2. FIG. The figure explaining the admittance characteristic of the element of the modification 3 described in Example 2. FIG. The figure explaining the thickness dependence of the dielectric of the admittance characteristic of the element of embodiment. The figure explaining the line length dependence of the admittance characteristic of the element of embodiment. The figure explaining the structure of the conventional oscillator.

The element 100 according to this embodiment will be described with reference to FIG. The element 100 is an oscillator that oscillates an electromagnetic wave having _{an oscillation frequency of f THz} (resonance frequency f _THz). 1 (a) is a perspective view showing the appearance of the element 100 according to the present embodiment, FIG. 1 (b) is a sectional view taken along the line AA, and FIG. 1 (c) is a sectional view taken along the line BB. The element 100 is hereinafter referred to as an "oscillator 100".

First, the configuration of the oscillator 100 will be described. The oscillator 100 has a resonance portion 102, a line 108, and a bias circuit 120. The resonance portion 102 includes a terahertz wave band resonator including a patch conductor (first conductor) 103, a ground conductor (second conductor) 104, and a first dielectric 105a, and a differential negative resistance element 101. Have.

The differential negative resistance element 101 is an element in which a region in which the current decreases as the voltage increases, that is, a region having a negative resistance (differential negative resistance region) appears in the current-voltage characteristics. The differential negative resistance element 101 is typically a high-frequency element such as an RTD, an Esaki diode, a Gunn diode, or a transistor having one terminal terminated. Further, a tannet diode, an impat diode, a heterojunction bipolar transistor (HBT), a compound semiconductor FET, a high electron mobility transistor (HEMT) and the like may be used. Further, the differential negative resistance of the Josephson element using a superconductor may be used. In this embodiment, a case where a resonance tunnel diode (RTD), which is a typical differential negative resistance element operating in the terahertz wave band, is used for the differential negative resistance element 101 will be described as an example.

The resonance unit 102 is an active antenna in which a resonator and a differential negative resistance element 101 are integrated. The resonator is a terahertz wave resonator, and has a patch conductor 103, a ground conductor 104, and a dielectric 105a arranged between the patch conductor 103 and the ground conductor 104. A configuration in which a dielectric 105a is sandwiched between two conductors, a patch conductor 103 and a ground conductor 104, is well known as a microstrip resonator using a microstrip line having a finite length or the like. In this embodiment, a patch antenna is used as a terahertz wave resonator.

The patch antenna is set so that the width of the patch conductor 103 in the AA direction is a λ / 2 resonator. A differential negative resistance element 101 is arranged between the patch conductor 103 and the ground conductor 104. In the present embodiment, the resonance portion 102 is a configuration in which the differential negative resistance element 101 and a terahertz wave band resonator such as a patch antenna are integrated.

Here, the "dielectric" in the present specification is a substance having a dielectric property superior to that of a conductive material, and is a material that behaves as an insulator or a high resistor that does not conduct electricity with respect to a DC voltage. Typically, a material having a resistivity of 1 kΩ · m or more is suitable. Specific examples include plastic, ceramic, silicon oxide, silicon nitride and the like.

The resonance portion 102 is an active antenna in which a patch antenna and a differential negative resistance element 101 are integrated. _{Therefore, the oscillation frequency f THz} defined by the resonance portion 102 of the oscillator 100 is determined as the resonance frequency of the all-parallel resonance circuit in which the reactance of the patch antenna and the reactance of the differential negative resistance element 101 are combined. That is, the oscillator 100 oscillates a terahertz wave having _{an oscillation frequency of f THz.}

Specifically, Jpn. J. Apple. Phys. , Vol. 47, No. From the equivalent circuit of the RTD oscillator disclosed in 6,4375 (2008), _{for the resonant circuit combining the RTD} and the admittance of the antenna (Y RTD and Y _ANT ), the amplitude condition of the equation (1) and the phase of the equation (2). A frequency that satisfies the two conditions is determined as the _{oscillation frequency f THz.} Here, Re [Y _RTD ] is the admittance of the differential negative resistance element 101 and has a negative value.
Re [Y _RTD ] + Re [Y _ANT ] ≤ 0 (1)
Im [Y _RTD ] + Im [Y _ANT ] = 0 (2)

The bias circuit 120 for supplying the bias voltage to the differential negative resistance element 101 includes a resistor 110 connected in parallel with the differential negative resistance element 101, a capacitance 109 connected in parallel with the resistor 110, a power supply 112, and a wiring 111. including. Since the wiring 111 always has a parasitic inductance component, it is shown as an inductance in FIG. The power supply 112 supplies the current required to drive the differential negative resistance element 101 and adjusts the bias voltage. The bias voltage is typically selected from the differential negative resistance region of the differential negative resistance element 101.

The line 108 is a distributed constant line, and the bias voltage from the bias circuit 120 is supplied to the elementary differential negative resistor 101 via the line 108. The line 108 of this embodiment is a microstrip line. The line 108 has a strip conductor (third conductor) 106, a grounding conductor (fourth conductor) 104, a second dielectric 105b, and a fifth conductor 107, and is grounded with the strip conductor 106. The second dielectric 105b is sandwiched between the conductor 104 and the conductor 104. The strip conductor 106 and the patch conductor 103 are electrically connected via a fifth conductor 107.

The thickness of the second dielectric 105b is configured to be thinner than the thickness of the first dielectric 105a. Therefore, the fifth conductor 107 connects the step (height difference) between the first dielectric 105a and the second dielectric 105b, and electrically and physically connects the patch conductor 103 and the strip conductor 106. Has the role of a plug to connect to. As a result, the resonance portion 102 and the line 108 are DC-coupled. In the present embodiment, the second conductor of the resonance portion 102 and the fourth conductor of the line 108 are ground conductors 104, which are the same layer, but different conductors may be used.

The resistance 110 and the capacitance 109 of the bias circuit 120 are _{parasitic oscillations of a relatively low frequency resonance frequency f sp} (f _sp <f _LC <f _THz , typically a frequency band from DC to 10 GHz) caused by the bias circuit 120. Is suppressed. Here, the frequency f _{LC is the frequency of the LC resonance due to the inductance L of the line 108 and the capacitance CANT} of the resonance portion 102 including the differential negative resistance element 101 and the patch antenna. Details of this will be described later.

It is preferable that the value of the resistance 110 is selected to be equal to or slightly smaller than the absolute value of the differential negative resistance in the differential negative resistance region of the differential negative resistance element 101. Resistor 110 is arranged at a distance d ₂ away from the negative differential resistance element 101. Then, mainly in _{the wavelength band of 4 × d 2} or more, the bias circuit 120 outside the resistor 110 has a low impedance when viewed from the differential negative resistance element 101, that is, the absolute differential negative resistance of the differential negative resistance element 101. The impedance is low based on the value. In other words, it is preferable that the resistor 110 is set to have a low impedance when viewed from the differential negative resistance element 101 in the frequency band below _{f SP} (f _SP <f _LC <f _THz).

Like the resistor 110, the capacitance 109 is preferably set so that the impedance is equal to or slightly lower than the absolute value of the differential negative resistance of the differential negative resistance element 101. Generally, it is preferable that the capacity 109 is large, and in the present embodiment, it is about several tens of pF. The capacity 109 is a decoupling capacity directly connected to a microstrip line as a line 108. For example, using a MIM (Metal-insulator-Metal) structure in which a patch antenna 102 and a substrate (not shown) are used together. May be good.

In the conventional element configuration, the frequency f _LC (f _LC ≈ 1 / {2π√ (LC _ANT )} due to the inductance L of the line 108 and the capacitance (capacitance) C _{ANT of the differential negative resistance element 101 and the resonance portion 102,} An LC resonance of f _SP <f _LC <f _THz ) was sometimes formed. The frequency _fLC is mainly determined by the capacitance of the differential negative resistance element 101, the length and width of the line (microstrip line in this embodiment) 108, the area of the patch antenna, the thickness of the dielectric 105a, and the like. Typically, the range is 10 GHz or more and 500 GHz or less.

Due to the structure of the patch antenna, it is not easy to directly connect the bias circuit 120 including the resistor 110 and the capacitance 109 described above to the resonance portion 102 without interfering with the resonance electric field _{having an oscillation frequency of f THz.} Therefore, in order to supply the bias voltage to the differential negative resistance element 101, it is necessary to connect the bias circuit 120 and the resonance portion 102 via the line 108 which is a feeder line. Thus, line 108, since it is located closer to the negative differential resistance element 101 than the bias circuit 120, there is the oscillation of the parasitic frequency f _LC due to the gain of the negative differential resistance element 101 occurs.

In the oscillator 100 of the present embodiment, the resonance portion 102 including the differential negative resistance element 101 and the patch antenna and the bias circuit 120 including the resistor 110 and the capacitance 109 are connected by a line 108 which is a distributed constant line. Therefore, the line 108, which is a distributed constant line, is a line for connecting the capacitance 109, which is the decoupling capacitance of the bias circuit 120, and the resonance portion 102 including the patch antenna. The oscillator 100 of the present embodiment is an active patch antenna having a structure in which a differential negative resistance element 101 and a first dielectric 105a are sandwiched between two conductors, a patch conductor 103 and a ground conductor 104. The resistor 110 of the bias circuit 120 is a shunt resistor, and the capacitance 109 is a decoupling capacitance, which are connected in parallel with the differential negative resistance element 101 and the power supply 112. The line 108, which is a microstrip line, is connected between the patch antenna of the resonance portion 102 and the capacitance 109.

The oscillator 100 has a structure and a configuration such that the line (microstrip line) 108 has a low impedance when viewed from the differential negative resistance element 101 in the frequency band near the _{LC resonance frequency f LC.} The configuration in which the line 108 has a low impedance when viewed from the differential negative resistance element 101 is, in other words, a low impedance line in which the line 108 is based on the absolute value of the differential negative resistance of the differential negative resistance element 101. It is a composition.

Here, the low impedance line based on the absolute value of the differential negative resistance element 101 has a relatively high impedance when the absolute value of the differential negative resistance element 101 is used as a reference. Refers to low lines. A line having a relatively low impedance is typically a line having a value of 10 times or less, preferably less than or equal to the absolute value of the differential negative resistance of the differential negative resistance element 101. .. Generally, the absolute value of the differential negative resistance of the differential negative resistance element 101 depends on the structure, but is about 0.1Ω or more and 100Ω or less. Therefore, the impedance of the low impedance line is set in the range of about 1Ω or more and 1000Ω. Will be done.

To construct an impedance line 108 based on the absolute value of the differential negative resistance of the differential negative resistance element 101, for example, the thickness of the line 108 is adjusted to change the characteristic impedance of the line 108 to the differential negative resistance element. It is set to 10 times or less the absolute value of the differential negative resistance of 101. More preferably, the characteristic impedance of the line 108 is set to be equal to or less than the absolute value of the differential negative resistance of the differential negative resistance element 101. _{Further, the absolute value (| 1 / Re [Y RTD} ] |) of the real part of the impedance, which is the gain, may be considered in consideration of the frequency characteristic of the differential negative resistance element 101. In this case, the characteristic impedance is 10 times or less the absolute value (| 1 / Re [Y _RTD ] |) of the real part of the impedance of the differential negative resistance element 101, preferably the real part of the impedance of the differential negative resistance element 101. A line having an absolute value of (| 1 / Re [Y _RTD ] |) or less is used as the line 108.

With such a configuration, the line 108, the electromagnetic wave loss at a frequency near the frequency f _LC has become as large structure. Therefore, since the microstrip line as the line 108 is a lossy line, the loss of the parasitic LC resonance caused by the line 108 increases, so that the oscillation of the LC resonance cannot be oscillated or is reduced. Can be done.

Here, assuming that the length of the line 108 is d ₁ and the distance from the differential negative resistance element 101 to the resistor 110 is d ₂ , the frequency band near the frequency _{f LC is 4 × d 1} or more in terms of wavelength 4 It is a wavelength band of × d _{2 or less.} This is determined by the arrangement and structure of the line 108 and the resistor 110, and is typically in the range of several GHz or more and 500 GHz or less.

The "low impedance line in which the line 108 is based on the absolute value of the differential negative resistance of the differential negative resistance element 101" in the present specification can be simply designed by considering the characteristic impedance of the line 108. That is, the characteristic impedance of the line 108 is 10 times or less the absolute value of the differential negative resistance of the differential negative resistance element 101, preferably the characteristic impedance of the line 108 is the absolute value of the differential negative resistance of the differential negative resistance element 101. The line structure should be designed so that it is less than or equal to the value. At this time, the line 108 becomes a low impedance line based on the absolute value of the differential negative resistance of the differential negative resistance element 101.

If _{the impedance of the line 108 as seen from the differential negative resistance element 101 at the frequency fLC} is 10 times or less the absolute value of the differential negative resistance, the loss due to the line 108 is greater than the gain of the differential negative resistance element 101. Since the magnitude cannot be ignored, the oscillation of LC resonance can be suppressed. In particular, the _{lower the impedance of the line 108 as seen from the differential negative resistance element 101 at the frequency fLC,} the greater the loss of electromagnetic waves in this frequency band on the line 108, which is effective in suppressing the oscillation of LC resonance. If the impedance of the line 108 seen from the differential negative resistance element 101 at the frequency f _LC is equal to or less than the absolute value of the differential negative resistance element, the loss exceeds the gain of the differential negative resistance element 101, so that the LC resonance cannot oscillate. Can be done.

Further, if the impedance of the line 108 as seen from the differential negative resistance element 101 is designed to be low, a structure or configuration capable of obtaining the above-mentioned effect can be realized. Further, the closer the structure of the line 108 to the differential negative resistance element 101 with respect to the differential negative resistance element 101 to the differential negative resistance element 101 and the resonance portion 102, the more effective it is in suppressing the oscillation of LC resonance. Specifically, it is preferably arranged at a distance of _{λ THz} or less from the differential negative resistance element 101. Note that λ _THz is the wavelength of a terahertz wave having an oscillation frequency of f _THz.

Such a configuration is due to the effect of the loss of the line 108, and in _{the low frequency region of less than f THz} including the _{frequencies f sp and f LC} , the equation (1) is not satisfied and Re [Y _RTD ] + Re [Y _ANT ]> 0. It becomes. Further, in _{the frequency region near the oscillation frequency f THz} , a structure satisfying the equation (1) can be realized. Here, Re [Y _RTD ] has a negative value, and in DC, it gradually approaches the reciprocal of the differential negative resistance of the differential negative resistance element 101. Therefore, the oscillation of LC resonance is suppressed, and a terahertz wave having _{an oscillation frequency of f THz can be obtained.}

The main losses of a distributed constant line such as the line 108 include a conductor loss due to the skin effect and a dielectric loss due to the dielectric loss tangent (tan δ). On the equivalent circuit of the distributed constant line, it is represented by the inductance L, the resistor R connected in series, and the leakage conductance G connected in parallel with the capacitance, respectively. Of these, the leakage conductance G is expressed by G = tan δ × ω × C, so that the higher the frequency, such as a terahertz wave, the more the loss due to the dielectric loss cannot be ignored. Further, the loss due to the dielectric loss is an increasing function of the dielectric loss tangent tan δ and the capacitance C of the line.

Therefore, by choosing to select the material and structure of the line 108 and a dielectric loss tangent tanδ and capacitance C of the line 108 suitably increases the leakage conductance G of electromagnetic waves around a frequency _{f LC} of the line 108, around the frequency _{f LC} Dielectric loss can be increased. Specifically, in the _{vicinity of the frequency fLC} , the dielectric loss is 1/10 or more of the absolute value of the reciprocal of the differential negative resistance element 101, more preferably the absolute value of the reciprocal of the differential negative resistance. If it is the above, it is effective in suppressing the oscillation near the _{frequency fLC.}

For example, if sufficiently thin thickness of the dielectric 105b, the capacitance C of the microstrip line 108 is increased, the dielectric loss increases in the vicinity of f _LC, line 108 is the negative differential of the negative differential resistance element 101 It is a low impedance line based on the absolute value of resistance. Specifically, the thickness of the dielectric 105b depends on the type of the dielectric, but typically, if the thickness is 0.001 μm or more and 1 μm or less, the characteristics of the microstrip line 108 in the vicinity of _{the frequency fLC} The impedance is 1Ω or more and 100Ω or less. The absolute value of the differential negative resistance of the differential negative resistance element 101 depends on the structure, but is approximately 1 Ω or more and 100 Ω or less.

Therefore, the thickness of the line 108 can be adjusted to set the characteristic impedance to 10 times or less the absolute value of the differential negative resistance element 101 or to be less than or equal to the absolute value of the differential negative resistance. At this time, the impedance of the line 108 as seen from the differential negative resistance element 101 is 10 times or less the absolute value of the differential negative resistance or less than the absolute value of the differential negative resistance. _{As a result, the dielectric loss near the frequency f LC} of the microstrip line 108 increases, so that the oscillation of LC resonance can be suppressed or disabled.

By making the thickness of the second dielectric 105b thinner than the thickness of the first dielectric 105a of the resonance portion 102, the patch antenna, which is a terahertz wave resonator and radiator, has low loss and high radiation efficiency. It is possible to increase the loss of only the line 108 while maintaining the above. That is, the electromagnetic wave having the oscillation frequency f _THz can maintain low loss and high radiation efficiency in the resonance portion 102, and the loss _{of the electromagnetic wave having the frequency f LC} in the microstrip line 108 can be increased.

FIG. 9 is a graph plotting the results of analyzing the thickness dependence of the dielectric 105b of the admittance characteristic of the oscillator 100. Silicon nitride, which is often used in semiconductor manufacturing, was used as the second dielectric 105b. For conventional oscillator of the second dielectric 105b has a thickness of 3 [mu] m, the vicinity of the frequency _{f LC} (≒ 0.08THz) admittance is there a low loss (high impedance) consists of the following 10 mS. On the other hand, in the case of the oscillator 100, by reducing the film thickness from 1 μm to 0.1 μm, the change in admittance near the _{oscillation frequency f THz is minimal, and the admittance near the frequency f LC} is increased from about 10 mS to 100 mS or more. It can be adjusted arbitrarily.

In this way, the thickness of the second dielectric 105b is set so that the thickness of the second dielectric 105b of the line 108 is thinner than the thickness of the first dielectric 105a of the resonance portion 102. As a result, the difference in the equivalent refractive index between the line 108 and the resonance portion 102 becomes large, so that impedance mismatch occurs at the connection portion between the resonator 102 and the resonance portion 108. Therefore, when the dielectric 105b becomes thinner, LC resonance frequency _{f LC} due to the inductance of the capacitor and the line 108 of the resonator 102 including the antenna, a resonance due to the capacitance of the resonance part 102, the inductance of the line 108 It splits into two, the resulting resonance. As a result, two resonance points occur close in frequency before and after the frequency f _LC, band of low impedance occurs before and after the frequency f _LC.

Further, if a material having a large dielectric loss tang tan δ is used for the dielectric 105b, the dielectric loss of electromagnetic waves near the _{frequency fLC can be increased without changing the structural dimensions of the line 108.} The material with a large dielectric loss tangent tanδ at the frequency _{f LC,} silicon nitride, polyaramid, polyethylene terephthalate, PMMA, ABS, and the like polycarbonate. Further, it may be a substance formed by dispersing fine powder of metal filler or carbon graphite in a synthetic resin such as styrene or epoxy whose dielectric loss magnitude depends on the electric field strength. Furthermore, it may be a larger meta-material structure of the dielectric loss tangent tanδ at the frequency _{f LC.}

Here, the length d ₁ of the line 108 is preferably long as long as the conductor resistance is not noticeable, and _{is preferably λ THz} / 4 or more. Note that λ _THz is the wavelength of a terahertz wave having an oscillation frequency of f _THz.

FIG. 10 is a graph plotting the results of analyzing the length dependence of the line 108 of the admittance characteristic of the oscillator 100. The admittance characteristics of the oscillator 100 when the line lengths are set to 100 μm, 200 μm, 300 μm, 400 μm, and 500 μm are shown. _Assuming that λ THz is the wavelength of a terahertz wave having a frequency of f _{THz , 100 μm corresponds to λ THz} / 4. When the strip conductor 106 is long, the inductance L increases, and the frequency f _{LC of the} LC resonance shifts to the low frequency side. Therefore, it _{is suppressed that the loss near the frequency f LC} affects the electric field of the terahertz wave having the _{oscillation frequency f THz} in the resonance portion 102. On the other hand, if the strip conductor 106 is short, the inductance L decreases, so that the frequency f _{LC of the} LC resonance shifts to the high frequency side.

_{As a result, the influence of the terahertz wave having the oscillation frequency f THz} on the electric field cannot be ignored in terms of distance and frequency, which causes loss of the terahertz wave having the oscillation frequency f _THz and multi-mode. This is because the line 108 length and the low impedance frequency band near the _{frequency f LC are widened, and the frequency f LC} itself shifts to the low frequency side. Therefore, the length of the line 108 is preferably long, preferably λ _THz / 4 or more. However, since the line 108 causes a long series resistance, may be appropriately designed in consideration of the impedance in the vicinity of the series resistor and f _LC.

The widths of the conductor 107 and the line 108 are preferably such that they do not interfere with the resonant electric field in the resonant portion 102, and for example, λ _THz / 10 or less is preferable. Further, it is preferable that the conductor 107 and the line 108 are arranged in the node of the electric field of the terahertz wave having the _{oscillation frequency f THz which is fixed in the resonance portion 102.} At this time, the line 108 has a configuration in which the impedance is higher than the absolute value of the differential negative resistance of the differential negative resistance element 101 in the frequency band near the _{oscillation frequency f THz.} Therefore, it is suppressed that the loss of the electromagnetic wave in the vicinity of the frequency f _LC affects the electric field having _{the oscillation frequency f THz in the resonance portion 102.}

Here, "paragraph of the electric field of the terahertz wave of the oscillation frequency f _THz for standing in the resonance section 102" is the area where a substantial section of the electric field of the terahertz wave of the oscillation frequency f _THz for standing the resonance part 102 That is. That is, it can be said that it is a region that is a substantial node of the electric field of the terahertz wave having _{an oscillation frequency of f THz} that is fixed in the patch antenna that is the resonator of the terahertz wave. Specifically, the electric field intensity of the terahertz wave of the oscillation frequency f _THz for standing in the resonance part 102, that of about one order of magnitude lower region than the maximum electric field intensity of the terahertz wave of the oscillation frequency f _THz for standing the resonance part 102 Is. Desirably, the electric field intensity of the terahertz wave of the oscillation frequency _{f THz} is, 1 ^{/ e} 2 of the maximum electric field intensity of the terahertz wave of the oscillation frequency _{f THz} for standing the resonance section 102 (e is the base of natural logarithms) becomes less position Is preferable.

The analysis result of the admittance characteristic of the oscillator 200 of the first embodiment, which is a specific example of the present embodiment, will be described with reference to FIG. The configuration of the oscillator 200 will be described with reference to Example 1 described later.

FIG. 2 shows the analysis results of the admittance characteristics of the oscillator 200 disclosed in Example 1, and shows the admittance real part Re [Y _RTD ] of the RTD, which is a differential negative resistance element, and the admittance real part Re [Y] of the patch antenna. It is a graph in which _{ANT] is plotted.} The graph shows admittance _Re patch antenna with a line 208 which becomes low impedance in a frequency band near the frequency _{f LC} of Example 1 below _{[Y ANTwith 208]} also. _{In addition, the admittance Re [YANT} ] of a conventional patch antenna equipped with a microstrip line having a relatively high impedance as a line is shown.

In the oscillator 200, the oscillation frequency f _THz satisfying the equation (2) is estimated to be 0.45 THz, and the LC resonance frequency f _{LC is estimated to be 0.08 THz.} In the case of a conventional microstrip line having a relatively high impedance, since the oscillation condition of Eq. (1) is satisfied at both _{the oscillation frequency f THz} and the frequency f _{LC, parasitic oscillation including LC resonance may occur.}

On the other hand, when the line 208 having a low impedance in the frequency band near the frequency _{fLC of the first embodiment is provided, the impedance at the oscillation frequency f THz} becomes high and the oscillation condition is satisfied. However, in the low frequency region of DC or more and _{less than f THz including the frequency f LC} , the line 208 has a low impedance when viewed from the RTD 201, and does not satisfy the oscillation condition of the equation (1). Therefore, it can be seen that in the configuration of the first embodiment, the parasitic oscillation including the LC resonance is suppressed.

As described above, the oscillator of Example 1 has a high impedance in the vicinity _{of the desired oscillation frequency f THz} in the terahertz wave band, and is low in the frequency region of parasitic oscillation of _{DC or more and less than f THz} including the LC resonance of the _{frequency f LC.} An oscillation circuit that becomes an impedance is realized. The high impedance and the low impedance here are high impedance when viewed from RTD201 (that is, with reference to RTD201) and low impedance when viewed from RTD201 (that is, with reference to RTD201). Therefore, even in an oscillator using a microstrip type resonator, low-frequency parasitic oscillation caused by the bias circuit 120 and the feeding structure is suppressed, and a terahertz wave having _{a desired oscillation frequency of f THz defined by the resonator is stabilized.} It becomes possible to oscillate.

_{With such a configuration, the loss of the terahertz wave at the oscillation frequency f THz} can be reduced by arbitrarily designing the dimensions and materials of the structure of the microstrip resonator and the distributed constant line. In addition, with proper design, it is possible _{to minimize the loss of the terahertz wave at the oscillation frequency f THz} and maximize the loss at the frequency of parasitic oscillation.

The oscillators of the present embodiment and the first embodiment can reduce the parasitic oscillation caused by the wiring structure, which has been a problem in the microstrip type resonator including the patch antenna and the like. Specifically, it reduces the occurrence of new parasitic oscillation caused by the inductance of the strip conductor for bias-feeding the differential negative resistance element. Therefore, it is also possible to reduce or suppress the parasitic oscillation in the frequency region of less DC than f _THz microstrip type resonator may be more stable terahertz wave of a desired oscillation frequency f _THz resonator.

Further, since the terahertz wave having an oscillation frequency of f _THz can be stably obtained, the terahertz wave having a desired oscillation frequency f _THz in the microstrip type resonator can be obtained with a higher output.

(Example 1)
The configuration of the oscillator 200, which is an element that oscillates a terahertz wave according to this embodiment, will be described with reference to FIG. FIG. 3 is a diagram illustrating the configuration of the oscillator 200. The oscillator 200 is an element that oscillates _{an oscillation frequency f THz = 0.45 THz.} In this embodiment, a resonance tunnel diode (RTD) is used as the differential negative resistance element 201. The RTD201 used in this embodiment is configured with, for example, a multiple quantum well structure made of InGaAs / InAlAs and InGaAs / AlAs on an InP substrate 230 and an electrical contact layer made of n-InGaAs.

As the multiple quantum well structure, for example, a triple barrier structure is used. More specifically, it is composed of a semiconductor multilayer film structure of AlAs (1.3 nm) / InGaAs (7.6 nm) / InAlAs (2.6 nm) / InGaAs (5.6 nm) / AlAs (1.3 nm). Of these, InGaAs is a well layer, and lattice-matched InAlAs and unmatched AlAs are barrier layers. These layers are intentionally left as undoped layers that are not carrier-doped.

Such a multiple quantum well structure is sandwiched between an electrical contact layer made of n-InGaAs having an electron concentration of 2 × 10 ¹⁸ cm ^-3. In the current-voltage I (V) characteristics of the structure between the electrical contacts, the peak current density is 280 kA / cm ² , and the differential negative resistance region is from about 0.7 V to about 0.9 V. When the RTD201 has a mesa structure of about 2 μmΦ, a peak current of 10 mA and a differential negative resistance of -20 Ω can be obtained.

The resonance portion 202 includes a resonator having a patch conductor 203, a ground conductor 204, and a first dielectric 205a, and an RTD 201. The resonator includes a square patch antenna having a side of the patch conductor 203 of 200 μm. Between the patch conductor 203 and the ground conductor 204, 3 μm thick BCB (benzocyclobutene, manufactured by Dow Chemical Co., Ltd., ε _r = 2.4) and 0.1 μm thick silicon nitride are used as the first dielectric 205a. Have been placed.

An RTD 201 having a diameter of 2 μm is connected between the patch conductor 203 and the ground conductor 204, and the RTD 201 is arranged at a position shifted by 80 μm in the resonance direction from the center of gravity of the patch conductor 203. The single resonance frequency of the patch antenna is about 0.48 THz, but considering the reactance of the RTD which is the differential negative resistance element 201, the oscillation frequency (resonance frequency) f THz of the oscillator 200 is about 0.45 _THz. ..

The patch conductor 203 is connected to the strip conductor 206 of the microstrip line as the line 208 via the conductor 207. As a result, the patch antenna is connected to the MIM capacity 209 via the line 208. With such a configuration, the line 208 connects the bias circuit 220 and the resonance portion 202.

The size of the MIM capacity 209 was set to 100 pF in this example. Wiring 211 including wire bonding is connected to the MIM capacity 209, and the bias voltage of the differential negative resistance element 201 is adjusted by the power supply 212. _{The frequency f LC of the LC resonance formed by the inductance L 1} of the strip conductor 206 of the microstrip line 208 and the capacitance C of the resonance portion 202 is about 0.08 THz.

The line 208 has a strip conductor 206, a ground conductor 204, and a second dielectric 205b disposed between the strip conductor 206 and the ground conductor 204. The second dielectric 205b is silicon nitride having a thickness of about 0.1 μm. The impedance of the line 208 is lower than the absolute value of the differential negative resistance of the RTD 201 at the _{LC resonance frequency f LC.} The specific dimensions of the microstrip line as the line 208 are such that a line having a width a of about 6 μm and a length b of about 100 μm extends from the connection portion with the resonance portion 202, and further has a width c of about 20 μm and a total length of about 600 μm. The track is extending. A line having a width c of about 20 μm is connected to a MIM capacity 209. The line having a width c of about 20 μm extends by the length e in the direction opposite to the MIM capacity 209, and is configured to bend twice at a substantially right angle and extend the length e toward the MIM capacity 209. The length e of this embodiment is about 200 μm, and the length d is about 400 μm. The distance f between the microstrip line of length e and the microstrip line of length d is about 60 μm.

The patch conductor 203 is connected to the conductor 207 at a high-frequency electric field node resident in the resonance portion 202 at _{an oscillation frequency of f THz} (= 0.45 THz), and is connected to the line 208 and the resonance electric field of the terahertz wave having an _{oscillation frequency of f THz.} Interference is suppressed.

With such a configuration, as can be seen from the result of the admittance analysis shown in FIG. 2, the oscillation condition of Eq. (1) is satisfied at _{the oscillation frequency f THz} , but the low frequency of DC or more and _{less than f THz including the frequency f LC.} In the region, the impedance becomes low and the equation (1) is not satisfied. Therefore, the oscillator 200 suppresses low-frequency parasitic oscillation caused by the bias circuit 220 and the feeding structure, and stabilizes a terahertz wave having _{a desired oscillation frequency of f THz defined by the resonance portion 202 including the RTD 201 and the patch antenna.} It becomes possible to oscillate.

The oscillator 200 according to this embodiment is manufactured as follows. First, the next layer is epitaxially grown on the InP substrate 230 by a molecular beam epitaxy (MBE) method, an organometallic vapor phase epitaxy (MOVPE) method, or the like. That is, the resonance tunnel diode (RTD) 201 made of n-InP / n-InGaAs and InGaAs / InAlAs is epitaxially grown in this order. When an n-type conductive substrate is selected as the InP substrate 230, it may be epitaxially grown from n-InGaAs.

Next, the RTD201 is etched into an arc-shaped mesa having a diameter of 2 μm. For etching, EB (electron beam) lithography and dry etching by ICP (inductively coupled plasma) are used. Photolithography may be used. Subsequently, the ground conductor 204 is formed on the etched surface by the lift-off method. A 0.1 μm silicon nitride film is formed on the entire surface of the second dielectric 205b and the side wall protective film of the resonance tunnel diode. Further, the spin coating method and dry etching are used to embed the first dielectric 205a with BCB, and the Ti / Pd / Au patch conductor 203 is formed by the lift-off method.

Next, the BCB outside the patch antenna is removed by a dry etching method to expose a 0.1 μm silicon nitride film which is the second dielectric 205b. The upper electrodes of the conductor 207, the strip conductor 206, and the MIM capacity 209 are formed by the lift-off method. Finally, by the lift-off method, a Bi pattern is formed in the portion that becomes the resistor 210, the ground conductor 204 and the upper electrode of the MIM capacity 209 are connected, and the wiring 211 and the power supply 212 are connected by wire bonding or the like. The example oscillator 200 is completed. Power is supplied to the oscillator 200 from the bias circuit 220, and when a bias voltage, which is usually a differential negative resistance region, is applied to supply a bias current, it operates as an oscillator.

In this embodiment, as RTD201, a triple barrier resonance tunnel diode composed of InGaAs / InAlAs and InGaAs / AlAs grown on an InP substrate has been described. However, the semiconductor device of the present invention can be provided not only by these structures and material systems but also by combinations of other structures and materials. For example, a resonance tunnel diode having a double barrier quantum well structure or a resonance tunnel diode having four or more multiple barrier quantum wells may be used.

Moreover, each of the following combinations may be used as the material.
-GaAs / AlGaAs / and GaAs / AlAs, InGaAs / GaAs / AlAs formed on the GaAs substrate
-InGaAs / AlGaAsSb formed on the InP substrate
-InAs / AlAsSb and InAs / AlSb formed on the InAs substrate.
-SiGe / SiGe formed on the Si substrate

The above-mentioned structure and material may be appropriately selected according to a desired frequency and the like.

As described above, the oscillator of this embodiment has a high impedance in the vicinity _{of the desired oscillation frequency f THz} in the terahertz wave band, and the frequency region of the parasitic oscillation of _{DC or more and less than f THz} including the LC resonance of the _{frequency f LC is low.} An oscillation circuit that becomes an impedance is realized. Therefore, the oscillator of this embodiment using the microstrip type resonator suppresses low-frequency parasitic oscillation caused by the wiring structure such as the bias circuit and the feeding structure, and stabilizes the terahertz wave having _{a desired oscillation frequency of f THz.} Can oscillate.

Further, since the terahertz wave having an oscillation frequency of f _THz can be stably obtained, the terahertz wave having a desired oscillation frequency f _THz in the microstrip type resonator can be obtained with a higher output.

(Example 2)
In the second embodiment, the elements of the modified examples 1 to 3 of the oscillator 200 of the first embodiment will be described with reference to FIGS. 4 to 8. First, the configuration of the oscillator 300 as the element of the first modification will be described with reference to FIG. FIG. 4 is a diagram illustrating the configuration of the oscillator 300. The oscillator 300 is an oscillator that oscillates an oscillation frequency f _THz = 0.45 THz like the oscillator 200. The oscillator 300 differs from the oscillator 200 in the form of the resonance portion 302. Further, the oscillator 300 is different from the oscillator 200 of the first embodiment in the form of the line 308. The form of the line may be appropriately designed as needed. Since other configurations are the same as those of the oscillator 200, detailed description thereof will be omitted.

The resonance portion 302 has a patch conductor 203, a ground conductor 204, and a first dielectric 305a arranged between the patch conductor 203 and the ground conductor 204. A part of the side wall of the first dielectric 305a is not substantially perpendicular to the substrate 230, and the angle formed by the substrate 230 and the first dielectric 305a is about 60 °. The patch conductor 203 is connected to the strip conductor 206 of the microstrip line as the line 208 via a fifth conductor 207.

The fifth conductor 207 is arranged on a slope that has been tapered so that the angle formed by a part of the first dielectric 305a with the substrate 230 is about 60 °. The fifth conductor 207 is a plug for connecting a step (height difference) between the patch conductor 203 and the strip conductor 206. Such a configuration can reduce the increase in resistance due to poor metal coverage at the stepped portion between the first and second dielectrics 305a and 205b having different thicknesses.

The oscillator 400 of the second modification of the oscillator 200 will be described with reference to FIG. 5 (a) is a diagram for explaining the configuration of the oscillator 400, FIG. 5 (b) is a sectional view taken along the line DD, and FIG. 5 (c) is a sectional view taken along the line EE. The oscillator 400 is an oscillator that oscillates an oscillation frequency f _{THz = 0.45 THz. The} oscillator 400 uses a strip line as a distributed constant line as a line 408 connecting the resonance portion 402 and the bias circuit (not shown). Since other configurations are the same as those of the oscillator 200, only the necessary parts will be described below.

The line 408 of the oscillator 400 has a strip conductor (third conductor) 406, a second dielectric 405b, and a third dielectric 422. Strip conductor 406 is embedded in the dielectric layer comprising the second third of the dielectric 422 and the thickness _{t 2} of the thickness _{t 1} and a dielectric 405 b. Further, the dielectric layer has a structure in which the sixth conductor 421 and the second conductor (fourth conductor) 204 are sandwiched from above and below. The strip conductor 406 and the patch conductor 203 are connected via a fifth conductor 207.

The fifth conductor 207 is a plug for connecting a step (height difference) between the patch conductor 203 and the strip conductor 406, and the resonance portion 202 and the strip line 408 are DC-coupled. The sixth conductor 421 and the second conductor 204 may be electrically and physically connected to each other.

Here, the admittance characteristic of the oscillator 400 will be described with reference to FIG. FIG. 6 shows the analysis result of the admittance characteristic of the oscillator 400, and plots the admittance real part Re [Y _RTD ] of the RTD 401, which is a differential negative resistance element, and the admittance real part Re [Y _ANT ] of the patch antenna. It is a graph. In the graph, for the real part Re [ _YANT ] of the admittance of the antenna, the thickness of the dielectric layer containing the third dielectric 422 and the second dielectric 405b is t ₁ = t ₂ = 100 nm and t. The analysis results in the case of 2 types of ₁ = t _{2 = 500 nm are shown.} Further, the gain of RTD401, that is, the real part Re [Y _RTD ] of admittance, showed two kinds of analysis results when the diameters of RTD401 were 2 μm and 3 μm.

From the graph, it can be seen that the impedance of the frequency band near _fLC can be adjusted by adjusting the thickness of the dielectric layer (third dielectric 422 and the second dielectric 405b) of the line 408. Incidentally, the impedance in the frequency band near f _LC, other than the thickness of the dielectric layer is also adjustable by selecting the material of the dielectric 405 b, 422 size and the second and third line 408, It may be appropriately selected depending on the gain of RTD401. Configuration of the oscillator 400 is oscillated in the loss in the region of low frequencies below DC or f _THz is suppressed, it is possible desired oscillation at the frequency f _THz loss less oscillation condition is satisfied.

The oscillator 500 of the third modification of the first embodiment will be described with reference to FIG. 7 (a) is a diagram illustrating the configuration of the oscillator 500, FIG. 7 (b) is a sectional view taken along line GG, and FIG. 7 (c) is a sectional view taken along line FF. The oscillator 500 is an element that oscillates _{an oscillation frequency f THz = 0.45 THz.} The oscillator 500 uses a square coaxial line as the line 508 that connects the resonance portion 502 and the bias circuit (not shown). Since other configurations are the same as those of the oscillator 200, detailed description thereof will be omitted, and only necessary parts will be described below.

The square coaxial line as the line 508 is a distributed constant line. The line 508 has a strip conductor (third conductor) 506, a sixth conductor 521, a third dielectric layer 522, and a fourth conductor (ground conductor) 204. Strip conductor 506 is embedded in the dielectric layer and a second dielectric 205b of the third dielectric layer 522 having a thickness of h ₁ thickness h _2, the dielectric layer is the sixth conductor 521 It has a structure surrounded by a conductor layer including the fourth conductor 204 and the fourth conductor 204. The strip conductor 506 and the patch conductor 203 are connected via a fifth conductor 207, which is a plug for connecting a step (height difference). The line 508 is a low impedance line based on the absolute value of the differential negative resistance of the element 201.

Here, the admittance characteristic of the oscillator 500 will be described with reference to FIG. FIG. 8 is an analysis result of the admittance characteristic of the oscillator 500, and is a graph in which _{the real part Re [Y RTD ] of the admittance of the RTD which is the element 201 and the real part Re [Y ANT} ] of the admittance of the patch antenna are plotted. .. In the graph, for the real part Re [Y _ANT ] of the admittance of the antenna, the thickness of the dielectric layer containing the third dielectric 522 and the second dielectric 205b is h ₁ = h ₂ = 100 nm and h _1. The analysis results in the case of two types of = h _{2 = 500 nm are shown.} In addition, the gain of RTD, that is, the real part Re [Y _RTD ] of admittance, showed two kinds of analysis results when the diameter of RTD was 2 μm and 3 μm.

From the graph, the third dielectric 522 stripline 508 by adjusting the thickness of the second dielectric 505b, it is possible to adjust the impedance in the frequency band near f _LC. Incidentally, the impedance in the frequency band near f _LC, other than the thickness of the dielectric layer is also adjustable by selecting the length or the material of each dielectric 505b, 522 of the line 508, by the gain of the device 201 It may be selected as appropriate. Configuration of the oscillator 500 is oscillated in the loss in the region of low frequencies below DC or f _THz is suppressed, it is possible desired oscillation at the frequency f _THz loss less oscillation condition is satisfied.

The modified examples 1 to 3 of the oscillator 200 have been described above. Also in the above-described modifications 1 to 3, the line connecting the resonance portion and the bias circuit is configured to be a low impedance circuit with reference to the absolute value of the differential negative resistance of the differential negative resistance element. It is possible to reduce parasitic oscillation caused by the wiring structure. As a result, a terahertz wave having a desired oscillation frequency of f _THz can be oscillated more stably.

Although the preferred embodiments of the present invention have been described above, the present invention is not limited to these embodiments, and various modifications and modifications can be made within the scope of the gist thereof.

For example, in the above-described embodiments and examples, the case where the carrier is an electron is described, but the description is not limited to this, and holes may be used. .. The substrate and dielectric materials may be selected according to the application, including semiconductors such as silicon, gallium arsenide, indium arsenide, and gallium phosphide, and resins such as glass, ceramics, Teflon (registered trademark), and polyethylene terephthalate. Can be used.

Further, in the above-described embodiments and examples, a square patch is used as the resonator of the terahertz wave, but the shape of the resonator is not limited to this, and for example, a polygon such as a rectangle and a triangle, or a circle. , A resonator or the like having a structure using a patch conductor such as an ellipse may be used.

Further, the number of differential negative resistance elements integrated in the oscillator is not limited to one, and a resonator having a plurality of differential negative resistance elements may be used. The number of lines is not limited to one, and a plurality of lines may be provided.

The oscillators described in the above embodiments and examples can also be used as detectors for detecting terahertz waves. It is also possible to oscillate and detect terahertz waves using the oscillators described in the above-described embodiments and examples.

100 element 101 differential negative resistance element 102 resonance part 103 patch conductor (first conductor)
104 Ground conductor (second conductor)
105a First Dielectric (Dielectric)
105b Second Dielectric 108 Line 120 Bias Circuit

Claims (15)

An element that oscillates or detects terahertz waves.
A resonant portion having a differential negative resistance element, a first conductor, a second conductor, and a first dielectric,
A bias circuit that supplies a bias voltage to the differential negative resistance element and
It has a strip line connecting the resonance portion and the bias circuit.
The differential negative resistance element and the first dielectric are arranged between the first conductor and the second conductor.
The strip line includes a third conductor that is electrically connected to the first conductor, a fourth conductor that is electrically connected to the second conductor, and the third conductor and the fourth conductor. With a second dielectric placed between and
The material contained in the second dielectric is different from the material contained in the first dielectric.
The impedance of the resonance portion including the strip line is the differential negative resistance of the differential negative resistance element at the frequency FLC, where the frequency of resonance due to the inductance of the strip line and the capacitance of the resonance portion is fLC. An element characterized by being 10 times or less an absolute value. The element according to claim 1, wherein the thickness of the second dielectric is thinner than the thickness of the first dielectric. The element according to claim 1 or 2 , wherein the dielectric loss tangent of the material of the second dielectric is larger than the dielectric loss tangent of the material of the first dielectric. The element according to claim 2 or 3, wherein the first dielectric has at least a first layer and a second layer. The element according to claim 4, wherein the second dielectric has the first layer. The device according to claim 4 or 5, wherein the material of the first layer is silicon nitride, and the material of the second layer is benzocyclobutene. The invention according to any one of claims 1 to 6, wherein the impedance of the resonance portion including the strip line is equal to or less than the absolute value of the differential negative resistance of the differential negative resistance element at the frequency fLC. The element described. If the real part of the admittance of the differential negative resistance element is Re [YRTD] and the real part of the admittance of the resonance part including the strip line is Re [YANT], and the resonance frequency of the terahertz wave is fTHz, the resonance At the frequency fTHz
Re [YRTD] + Re [YANT] ≤ 0
At the frequency fLC
Re [YRTD] + Re [YANT]> 0
The element according to any one of claims 1 to 7, wherein the element satisfies. The element according to any one of claims 1 to 8, wherein the thickness of the first dielectric is 0.001 μm or more and 1 μm or less. The element according to any one of claims 1 to 9, wherein the length of the strip line is λTHz / 4 or more, where λTHz is the wavelength of the terahertz wave. The element according to any one of claims 1 to 10, wherein the resonance portion and the strip line are connected by a node of the electric field of the terahertz wave resident in the resonance portion. .. The element according to any one of claims 1 to 11, wherein the strip line is a microstrip line. The bias circuit has a shunt resistor and a decoupling capacitance.
The element according to any one of claims 1 to 12, wherein the strip line is connected between the resonance portion and the decoupling capacitance. The element according to any one of claims 1 to 13, wherein the strip line is a line that is lossy at the frequency fLC. The element according to any one of claims 1 to 14, wherein the width of the third conductor is narrower than the width of the first conductor.

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