JP6309173B2 - Motor drive device, heat pump device using the motor drive device, refrigeration air conditioner, and blower - Google Patents

Description

  The present invention relates to a motor drive device that drives a motor, and a heat pump device, a refrigeration air conditioner, and a blower device that use the motor drive device.

  Patent Document 1 below discloses a permanent magnet synchronous motor including a rotor provided with a permanent magnet and an armature provided with a winding, a magnetic pole position detector that detects the rotational position of the magnetic pole of the rotor, It consists of a speed detector that detects the rotational speed, a current detector that detects the current value flowing through the armature, and a control device that performs current control and speed control of the permanent magnet synchronous motor, and when the load torque is constant, A technique is disclosed in which the excitation current id is changed so that the excitation current id is changed in the direction in which the current decreases from the previous excitation current id and current information, and the phase approaches the current minimum.

JP 2001-8486 A

  However, in the technique shown in Patent Document 1, the energization phase that minimizes the current is not obtained in the control process, and it takes time to reach the current phase angle that minimizes the current. For this reason, in the method of Patent Document 1, the convergence to the energization phase where the current is minimized is not very good, an error occurs between the output voltage command value and the actual output voltage, and the output voltage from the inverter is ideal. There was a problem that the output voltage could deviate and high-power or high-efficiency operation might not be possible.

  The present invention has been made in view of the above, and a motor drive device that enables high-output or high-efficiency operation even when there is an error between the output voltage command value and the actual output voltage, and An object of the present invention is to provide a heat pump device, a refrigerating and air-conditioning device, and a blower using a motor driving device.

  In order to solve the above-described problems and achieve the object, a motor driving device according to the present invention includes a motor that drives a load, an inverter that applies a voltage to the motor, and an inverter control unit that controls a voltage output from the inverter. The voltage output from the inverter is increased in accordance with the increase in the number of revolutions of the motor so that the current flowing to the motor at the first number of revolutions within the range of the number of revolutions of the motor approaches the voltage value that minimizes the current. The voltage output from the inverter is controlled.

  According to the present invention, even when there is an error between the output voltage command value and the actual output voltage, there is an effect that high output or high efficiency operation becomes possible.

The block diagram which shows the structure of the motor drive device in this Embodiment Circuit diagram of the inverter in this embodiment The figure which shows an example of the relationship between the current phase angle β and the output torque in a certain operation state where the IPMSM is driven at a constant current The figure which shows the relationship between the electric current phase angle (beta) at the time of changing and driving the load torque of IPMSM on constant rotation frequency conditions, and an electric current value. Vector diagram when driving IPMSM under certain operating conditions The figure which shows the relationship between (delta) axis | shaft current i (delta) and exciting current command value I (gamma) ^* in this Embodiment. Vector diagram in consideration of voltage error ΔV due to dead time in the present embodiment The figure which shows the change of the voltage with respect to the rotation speed in this Embodiment. The figure which shows the change of the electric current with respect to the rotation speed in this Embodiment. Schematic configuration diagram when the motor drive device according to the present embodiment is applied to a heat pump device Circuit configuration diagram of heat pump device according to the present embodiment Mollier diagram about the state of the refrigerant of the heat pump device shown in FIG.

  A motor driving apparatus according to an embodiment of the present invention will be described below with reference to the accompanying drawings. In addition, this invention is not limited by embodiment shown below.

Embodiment.
First, the basic configuration and operation of the motor drive device in the present embodiment will be described. FIG. 1 is a block diagram showing a configuration of a motor drive device 1 according to the present embodiment, and FIG. 2 is a diagram showing a configuration example of an inverter 3 shown in FIG.

  As shown in FIG. 1, the motor drive device 1 in the present embodiment is a device that supplies AC power to a motor 5 that is an AC motor to rotationally drive the motor 5, and a DC power source that is a DC power supply source. 2, an inverter 3 that converts direct current into alternating current and applies an alternating voltage to the motor 5, and an inverter control unit 4 that controls the inverter 3.

  As shown in FIG. 2, the inverter 3 is parallel to the transistor element (IGBT (Insulated Gate Bipolar Transistor) in FIG. 2) and the transistor element, that is, in parallel so that the direction of current flow is opposite to the transistor element. Two switching elements 3a and 3b each having a connected diode element are connected in series to form a one-phase arm (a U-phase arm in FIG. 2). An inverter circuit is configured by being connected in parallel. In the inverter circuit configured as shown in FIG. 2, the switching element on the high voltage side located on the upper side is “upper arm” or “positive arm”, and the switching element on the low voltage side located on the lower side is “lower arm”. Or referred to as “negative arm”.

  Each connection point of the switching element 3a and the switching element 3d, the switching element 3b and the switching element 3e, and the switching element 3c and the switching element 3f is drawn from the inverter 3 and connected to the motor 5.

  The inverter control unit 4 drives the switching elements 3a to 3f of the inverter 3 in order to drive a gate drive signal (also referred to as “PWM signal”) that is pulse width modulated (hereinafter referred to as “PWM”). UP, UN, VP, VN, WP, WN are generated and output to the inverter 3. The AC voltage commanded by the gate drive signals UP, VP, WP, UN, VN, WN is applied to the motor 5 and the motor 5 rotates.

  Returning to FIG. 1, current detection units 6 a and 6 b for detecting a current flowing in the electrical wiring are provided in the electrical wiring connecting the inverter 3 and the motor 5. The current detection unit 6 a detects the U-phase current iu of the current flowing into and out of the motor 5, and the current detection unit 6 b detects the W-phase current iw of the current flowing into and out of the motor 5. In addition, arrangement | positioning of the electric current detection part 6a, 6b of illustration is an illustration, What is necessary is just to be able to detect the electric current of at least 2 phases among U phase, V phase, and W phase, and it should just be arrange | positioned in any 2 phases. The current detection units 6a and 6b convert the detected current into a current value or a current signal, and output the current value or current signal to the three-phase / two-phase conversion unit 7 described later.

  Next, the inverter control unit 4 will be described. As shown in FIG. 1, the inverter control unit 4 uses a current value or current signal detected by the current detection units 6 a and 6 b as a current value represented by an excitation current component and a torque current component using the electrical angle phase θ, or A three-phase to two-phase converter 7 for converting the current signal. In the present embodiment, the γ-axis current iγ and the δ-axis current iδ, which are current values on the γ-δ axis, are used for control as the excitation current component and the torque current component.

Further, the inverter control unit 4 obtains the frequency compensator 8 that calculates the frequency compensation amount ωd that is the compensation amount of the rotation frequency from the δ-axis current iδ obtained by the three-phase to two-phase conversion unit 7, and the frequency compensator 8. A primary frequency calculation unit 9 for obtaining a primary frequency ω1 which is a compensated frequency command value ω ^* by compensating the frequency command value ω ^* which is a target frequency given from the outside by using the frequency compensation amount ωd obtained, and a primary frequency calculation unit 9 is used to calculate the electrical angle phase θ based on the primary frequency ω1 obtained in step 9, the excitation current command value iγ ^* generated internally, the input γ-axis current iγ and the primary frequency ω1. On the basis of the voltage command values Vd ^* and Vq ^* for driving the motor 5, a voltage command value calculator 10 for calculating the voltage command values Vd ^* and Vq ^*, and γ-δ axis voltage command values Vγ ^* and Vδ obtained by the voltage command value calculator 10 ^* the electrical angle phase Using an electrical angle phase θ obtained by the calculation unit 13, the three-phase voltage command values in a three-phase coordinate system ^{Vu *,} Vv *, and two-to-three phase conversion unit 11 for converting the Vw ^*, two-to-three phase conversion A PWM signal generation unit 12 that generates a PWM signal for on / off control of the switching elements 3a to 3f in the inverter 3 based on the three-phase voltage command values Vu ^* , Vv ^* , Vw ^* obtained from the unit 11; , And is configured.

  The primary frequency calculation unit 9 can be configured using an adder / subtractor as shown in the figure. Further, the electrical angle phase calculation unit 13 can be configured using an integrator as shown in the figure.

In addition, the voltage command value calculation unit 10 is based on the δ-axis current iδ, and the excitation current for calculating the optimum excitation current command value iγ ^* when it is controlled so as to approach the condition where the efficiency of the motor 5 is maximized. A command value calculation unit 14 is provided.

  Next, in the motor drive device 1 configured as described above, a series of operations for generating a PWM signal for controlling the inverter 3 after detecting a current flowing into and out of the motor 5 will be described with reference to FIG. This will be described with reference to the drawings.

  First, U-phase current iu and W-phase current iw, which are phase currents for two phases of the phase current flowing into and out of motor 5 that is the driving load of inverter 3, are detected by current detectors 6a and 6b. The detected U-phase current iu and W-phase current iw are input to the three-phase / two-phase conversion unit 7 of the inverter control unit 4. The three-phase / two-phase converter 7 converts the U-phase current iu and the W-phase current iw detected by the current detectors 6a and 6b into a γ-axis current iγ and a δ-axis current iδ that are current values on the γ-δ axis. To do. Of the converted γ-axis current iγ and δ-axis current iδ, the δ-axis current iδ is input to the frequency compensator 8 and the voltage command value calculation unit 10, and the γ-axis current iγ is input to the voltage command value calculation unit 10. .

The frequency compensator 8 calculates a frequency compensation amount ωd for compensating the frequency command value ω ^* given from the outside of the inverter control unit 4 based on the δ-axis current iδ. The frequency compensation amount ωd calculated by the frequency compensator 8 is input to the primary frequency calculation unit 9. The primary frequency calculation unit 9 calculates a primary frequency ω1 obtained by compensating the frequency command value ω ^* with the frequency compensation amount ωd. The calculated primary frequency ω1 is input to the voltage command value calculation unit 10 as a compensated frequency command value.

The voltage command value calculator 10 uses the primary frequency ω1 that is the compensated frequency command value to drive the motor 5, and the γ-axis voltage command value Vγ ^* and δ-axis that are output voltage command values on the γ-δ axis. The voltage command value Vδ ^* is obtained by calculation. The voltage command value calculation unit 10 further uses the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* , which are output voltage command values on the γ-δ axis, as a d-axis, which is an output voltage command value on the dq axis. An operation for converting the voltage command value Vd ^{* into the} q-axis voltage command value Vq ^* is performed. Here, the voltage command values Vd ^* and Vq ^* are voltage commands on the dq axis which is an orthogonal axis different from the γ-δ axis representing the excitation current component and the torque current component. The difference between the γ-δ axis and the dq axis will be described later. The d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* calculated by the voltage command value calculation unit 10 are input to the two-phase / three-phase conversion unit 11.

The excitation current command value calculation unit 14 in the voltage command value calculation unit 10 calculates the optimum excitation current command value iγ ^* by calculation based on the δ-axis current iδ. Here, the optimum excitation current command value iγ ^* is a command value of the γ-axis current that has the highest efficiency value when controlled based on given conditions. In other words, under a given condition, the command value of the γ-axis current at which the output torque becomes the maximum or more than a desired value, or the γ-axis current at which the current flowing through the motor 5 becomes the minimum or less than the desired value. Command value.

In FIG. 1, the excitation current command value iγ ^* is obtained based on the δ-axis current iδ that is a torque current component. However, based on the γ-axis current iγ, the frequency command value ω ^* , and the primary frequency ω1, A known method for obtaining the excitation current command value iγ ^* may be used. A method for obtaining the excitation current command value iγ ^* will be described later.

  Moreover, although the case where the current detection unit is provided is described in the present embodiment, it can be realized even when the current detection unit is not provided. For example, when performing motor control based on a position signal corresponding to the rotor rotational position of the motor, the rotor rotational speed can be detected by the period of the position signal, etc., so the excitation current command value is obtained based on the rotational speed detection value. Alternatively, there is a method of storing the excitation current command value according to the number of rotations by a table or the like. Further, the torque current based on the rotation speed detection value may also be stored as a table, and the excitation current command value may be set from the torque current.

The two-phase / three-phase converter 11 converts the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^*, which are output voltage command values in a two-phase coordinate system, into a U-phase voltage, which is an output voltage command value in a three-phase coordinate system. A command value Vu ^* , a V-phase voltage command value Vv ^*, and a W-phase voltage command value Vw ^* are generated. The generated U-phase voltage command value Vu ^* , V-phase voltage command value Vv ^*, and W-phase voltage command value Vw ^* are input to the PWM signal generation unit 12.

The PWM signal generator 12 generates a PWM signal based on the output voltage command values (Vu ^* , Vv ^* , Vw ^* ) in the three-phase coordinate system, and controls the switching elements 3a to 3f in the inverter 3 to be turned on or off. To do. With this PWM signal, the switching elements 3a to 3f in the inverter 3 are turned on or off, the DC voltage Vdc of the DC power supply 2 is converted into three-phase AC, and applied to the motor 5 that is a driving load. Driven.

  Next, referring to FIG. 3, the current phase angle β in an operating state in which the motor 5 as a driving load uses an interior permanent magnet synchronous motor (hereinafter referred to as “IPMSM”) The relationship of output torque will be described. FIG. 3 is a diagram showing an example of the relationship between the current phase angle β and the output torque in a certain operating state in which the IPMSM is driven at a constant current. In FIG. 3, the horizontal axis represents the phase angle β of the motor current with respect to the voltage applied to the motor 5, and the vertical axis represents the output torque.

  Also, the figure shows the magnet torque generated by the IPMSM rotor magnet, the reluctance torque generated by the difference between the d-axis inductance Ld and the q-axis inductance Lq, and the output torque that is the combined torque of the magnet torque and the reluctance torque. Yes. The horizontal axis represents the range where the current phase angle β is in the range of 0 to 90 [deg], and each characteristic of the magnet torque, the reluctance torque, and the combined torque represents the relationship with the current advance angle.

  As shown in FIG. 3, in the IPMSM, the output torque changes depending on the current phase angle β even under the same current condition. For this reason, by controlling the current phase angle β even under the same current condition, it becomes possible to set the output torque to a certain value or more or a maximum value, and a high output device can be obtained.

  Therefore, under the condition where the load torque is constant, the current value required to obtain the same output torque as the load torque can be controlled to a certain value or less or a minimum value by appropriately controlling the current phase angle β. It becomes possible, and a highly efficient apparatus can be obtained.

  Further, if the output torque is controlled to a certain value or more or the maximum value under a constant current condition, the current value becomes a certain value or less or a minimum value under a constant load torque. For this reason, if the output torque is controlled to a certain value or more or the maximum value under a constant current condition, the current value becomes a certain value or less or a minimum value when the load torque is constant, and the current value becomes a certain value or less The value can be controlled, and a high-power or high-efficiency device can be obtained. Hereinafter, the method of controlling the current value to be a certain value or less or the minimum value is appropriately referred to as “current value minimum control method” or “current value minimum control”.

  FIG. 4 is a diagram showing the relationship between the current phase angle β and the current value when driven by changing the load torque of the IPMSM under a constant rotational frequency condition. In FIG. 4, the horizontal axis represents the current phase angle β, and the vertical axis represents the current value. In the figure, τn (n = 0, 1, 2, 3, 4) indicates the output torque of the IPMSM, and the magnitude relationship is τ4> τ3> τ2> τ1> τ0.

  In FIG. 4, a black circle is a point where the current is minimized. That is, when the IPMSM is driven under the same rotational frequency condition, if the load torque changes (τ0 → τ1 → …… → τ4), the current phase angle β at which the current value becomes the minimum according to the change of the load torque is also obtained. Change. However, even when the load torque changes, as shown in FIG. 4, it is possible to control the current phase angle β to minimize the current value, and to drive the IPMSM with the current value being minimum. Is possible.

  If the IPMSM is driven with the minimum current value, the copper loss of the IPMSM can be minimized, and if the IPMSM is driven with the minimum current value, the conduction loss of the switching element of the inverter can be minimized. The IPMSM can be operated with high efficiency. Further, if at least one of the switching elements of the inverter is a wide band gap semiconductor (hereinafter referred to as “WBG semiconductor”), further loss reduction is possible.

  Examples of the WBG semiconductor include silicon carbide (SiC), gallium nitride (GaN), and diamond. A switching element and a diode element formed of a WBG semiconductor have a high voltage resistance and a high allowable current density. Therefore, it is possible to reduce the size of the switching element and the diode element. By using these reduced switching element and diode element, it is possible to reduce the size of the semiconductor module incorporating these elements.

  In addition, the switching element and the diode element formed of such a WBG semiconductor have high heat resistance. As a result, the heat sink fins of the heat sink can be miniaturized and the water cooling part can be air cooled, so that the semiconductor module can be further miniaturized.

  Furthermore, the switching element and the diode element formed by such a WBG semiconductor have low power loss. As a result, the inverter can be operated more efficiently.

  Next, a “current value minimum control method” that embodies the main operation of the motor drive device according to the present embodiment will be described.

  First, the voltage equation in the dq axis coordinates when the motor is IPMSM in particular is generally given by the following equation (1). Here, the d-axis and the q-axis are defined as the q-axis, where the magnetic pole direction of the rotor of the motor is the d-axis, and the axis that makes a phase advanced from the d-axis in the rotation direction by a predetermined angle (90 degrees in the case of two poles). doing.

  In the above equation (1), Vd is the IPMSM d-axis voltage, Vq is the IPMSM q-axis voltage, R is the IPMSM resistance per phase, Ld is the IPMSM d-axis inductance, Lq is the IPMSM q-axis inductance, Φf represents the induced voltage constant of IPMSM, id represents the d-axis current of IPMSM, iq represents the q-axis current of IPMSM, ω represents the rotational frequency of IPMSM, and p represents the differential operator according to time t. If the number of pole pairs of the IPMSM is Pm, the output torque τm of the IPMSM is generally given by the following equation (2).

  Further, the d-axis current id, the q-axis current iq, and the current value Ia that is the magnitude of the current are defined as the following equation (3).

Substituting i _d and i _q in the above equation (3) into i _d and i _q in the above equation (2), the torque τm in the above equation (2) is expressed as in the following (4).

  Here, the current phase angle at which the output torque is maximized or the current phase angle at which the motor current is minimized is βm, and the procedure for obtaining the current phase angle βm will be described below.

  First, the above equation (4) is partially differentiated with respect to the current phase angle β, and the equation obtained by partial differentiation is 0, that is, the partial differential equation = 0 on the right side is given, thereby giving the maximum torque or the current at which the current is minimized. The following equation (5) is obtained as the phase angle βm.

  However, in the calculation of the current phase angle βm, as shown in the above equation (5), the current that causes the output torque to become a certain value or more without calculating the value that maximizes the output torque or minimizes the current. The phase angle or the current phase angle at which the current becomes a certain value or less may be βm, and an approximate expression such as the following expression (6) may be used.

  Here, the above formula (6) is obtained by expanding the above formula (5) to the second order term by expanding the Macrolin series. By using the above equation (6), which is an approximate expression, the output torque is not maximized and the current is not minimized, but the output torque can be increased to a desired value or more, and the current is desired. Therefore, it is possible to obtain a motor drive device with high output and low current.

  Further, as described above, since the above expression (6) is an approximate expression and simplified, the calculation load on the microcomputer can be reduced. For this reason, when the calculation formula (6) is implemented, the calculation time can be shortened. Further, when the arithmetic expression (6) is implemented, it is possible to cope with a low-cost microcomputer, and the degree of freedom in selecting the microcomputer can be increased. Further, when the arithmetic expression (6) is implemented, the convergence of the calculation is improved by shortening the calculation time, so that the followability can be improved in the control.

  Even when a method using an approximate expression is employed, if the control accuracy needs to be improved, the accuracy of the approximate expression may be improved by increasing the number of expansions of the Macrolin series. Of course, the arithmetic expression (5) may be used as it is.

  Next, the relationship between the dq axis coordinate system and the γ-δ coordinate system will be described. In the present embodiment, it is assumed that the synchronous motor is driven without using a position sensor for detecting the rotor position. In the case of driving without using a position sensor, it is actually difficult to detect the dq axis of an interior permanent magnet (IPM) rotor. For this reason, in this embodiment, the γ-δ axes as shown in FIG. 5 are used as coordinate axes for control, that is, control axes. In the following description, the control axis is a case where the γ-δ axis, in which the current component is easily detected, is used. However, the dq axis is based on the rotor position of the IPMSM without using the γ-δ axis. It goes without saying that may be used.

  FIG. 5 is a vector diagram when the IPMSM is driven under a certain operating condition. The horizontal axis represents the γ-axis and the vertical axis represents the δ-axis. In FIG. 5, the d-axis and the q-axis are shown at angular positions rotated by Δθ with respect to the γ-axis and the δ-axis, respectively.

  In FIG. 5, iγ is the γ-axis current of IPMSM, and iδ is the δ-axis current of IPMSM. Ia is a combined vector of iγ and iδ, and represents the magnitude of the current flowing through the IPMSM, that is, the motor current. As shown in FIG. 5, the angle formed between the motor current Ia and the q axis is the current phase angle β, and the phase of the motor current Ia is advanced by the current phase angle β with respect to the q axis.

  The δ-axis current iδ or the q-axis current iq is generally referred to as a torque current as a component representing a torque component of a current component. The δ-axis current iδ or the q-axis current iq has a magnitude corresponding to the output torque of the motor 5.

  In FIG. 5, the motor current Ia is shown as a combined vector of the excitation current iγ and the torque current iδ, but the excitation current id of the dq axis coordinate system and the excitation current iq of the dq axis coordinate system are shown. It can also be expressed as a composite vector.

  Further, in FIG. 5, Φf is a magnetic flux vector generated by the rotor magnet and is located on the d axis. Since Φf is located on the d-axis, the d-axis coordinate is the magnitude of the magnetic flux vector. The magnitude of the magnetic flux vector is referred to as “induced voltage constant”. Lq · iδ is a δ-axis component of the stator magnetic flux, Ld · iγ is a γ-axis component of the stator magnetic flux, and Φ0 is a combined magnetic flux of the stator magnetic flux and the rotor magnetic flux. That is, the combined magnetic flux Φ0 is a combined magnetic flux vector of the rotor magnetic flux Φf, the δ-axis component Lq · iδ of the stator magnetic flux, and the γ-axis component Ld · iγ of the stator magnetic flux. That is, the resultant magnetic flux vector Φ0 is generated by the stator-side magnetic flux (Ld · iγ + Lq · iδ) generated by the excitation current iγ and the torque current iδ and the rotor Φf that is the magnetic flux of the rotor magnet.

  Furthermore, in FIG. 5, ω · Φ0 is a voltage vector generated by the combined magnetic flux, R · Ia is a voltage vector that compensates for the voltage effect caused by the resistance of the motor, and Va is the voltage vector ω · Φ0 and the voltage vector R · It is a combined vector with Ia and represents the voltage applied to the IPMSM, that is, the motor applied voltage.

  Next, the excitation current iγ will be described. The exciting current iγ is a control amount for driving at the current phase angle β at which the current is minimized in the “current value minimum control” of the present embodiment.

  When the output voltage vector Va is changed according to the operation state of the IPMSM, the exciting current iγ changes. As the excitation current iγ changes, the current phase angle β changes. Since the combined vector of the excitation current iγ and the torque current iδ is the motor current Ia, the current of the current vector Ia can be controlled by controlling the excitation current iγ and controlling the output voltage vector Va among the excitation current iγ and the torque current iδ. The phase angle β can be changed.

  In this embodiment, since the γ-δ axis is used as the control axis, the value of the γ-axis inductance Lγ should be used as the excitation axis inductance, and the value of the δ-axis inductance Lδ should be used as the torque axis inductance. However, for simplification of control, it is handled as Lγ≈Ld and Lδ≈Lq. Even if they are handled as Lγ≈Ld and Lδ≈Lq, there is no problem in control. Note that handling as Lγ≈Ld and Lδ≈Lq corresponds to approximating the phase difference Δθ between the dq axis and the γ−δ axis as Δθ≈0.

Further, in the present embodiment, even in a control system that does not have a speed controller, that is, control that does not use the torque current command value (iδ ^* ), the minimum current phase angle βm by “current value minimum control” is obtained. The output voltage vector Va is controlled by controlling only the γ-axis excitation current iγ. The minimum current phase angle βm is an angle formed by the current vector Ia with respect to the q axis of the dq axis coordinate system. FIG. 5 shows a state where the minimum current phase angle βm is obtained by the “current value minimum control”, and the illustrated current phase angle β corresponds to the minimum current phase angle βm.

Further, in the control method without the speed controller and the control method without the position sensor, the actual dq axis position is unknown. Therefore, in this embodiment, the magnitude of the voltage vector is used as a variable used for control so that the influence of the phase difference Δθ between the control axis (γ−δ axis) and the motor axis (dq axis) can be ignored. A certain motor voltage Va, voltage command value Va ^* , and motor current Ia which is the magnitude of the current vector are used.

In the present embodiment, the excitation current command value calculation unit 14 calculates the “current value minimum control” even in a control method in which the synchronous motor driving means does not have a speed control loop or a current control loop. The excitation current command value iγ ^* is obtained as follows.

In FIG. 4, it can be seen that the higher the output torque τn, the larger the current phase angle β at which the current is minimized. Here, the δ-axis current iδ is a component relating to the torque of the current component, and has a magnitude corresponding to the output torque of the motor. That is, the minimum current phase angle βm and the motor current Ia are theoretically determined according to the torque. If the minimum current phase angle βm and the motor current Ia are obtained, the d-axis current id and the q-axis current iq can be obtained from the above-described equation (3). When the phase difference Δθ between the dq axis and the γ-δ axis is Δθ≈0, iq = iδ and id = iγ ^* can be obtained. Therefore, as shown in FIG. 6, it is possible to obtain the excitation current command value Iγ ^* for the δ-axis current iδ. In FIG. 6, the horizontal axis represents the δ-axis current iδ, and the vertical axis represents the excitation current command value Iγ ^* . The relationship shown in FIG. 5 is implemented in the form of an arithmetic expression or in the form of a table. Thus, the excitation current command value calculation unit 14 can be realized.

If the phase difference Δθ between the dq axis and the γ-δ axis is not negligible, the excitation current command value Iγ ^* for the δ-axis current iδ should be obtained in consideration of the influence of Δθ in advance. In other words, this can be dealt with by preparing a relational expression or table between the δ-axis current iδ and the excitation current command value Iγ ^* in consideration of the influence of Δθ.

The voltage command value calculation unit 10 obtains a γ-axis voltage command value Vγ ^* and a δ-axis voltage command value Vδ ^* , which are voltage command values of the γ-δ axis, based on the obtained excitation current command value Iγ ^* . The calculation formulas for obtaining the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* vary depending on the control method of the synchronous motor, but the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^**. As long as is obtained, any arithmetic expression may be used.

Here, an example of an equation for calculating the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* , which are outputs of the voltage command value calculation unit 10, is shown in the following equation (7).

The above expression (7) represents an arithmetic expression in the case of using a synchronous motor position sensorless control method by primary magnetic flux control, for example. In the equation (7), φγ ^* is a primary magnetic flux command value in primary magnetic flux control, Φγerr is a primary magnetic flux error, Kγ is a γ-axis control gain, and Kδ is a δ-axis control gain.

The γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* are converted into the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* by the voltage command value calculation unit 10, and the d-axis voltage command value Vd ^*. The q-axis voltage command value Vq ^* is converted into a three-phase voltage command value Vu ^* , Vv ^* , Vw ^* by the two-phase / three-phase converter 11. The PWM signal generation unit 12 generates a PWM signal for on / off control of the switching elements 3a to 3f of the inverter 3 based on the three-phase voltage command values Vu ^* , Vv ^* , Vw ^* . When generating a PWM signal, it is a well-known technical matter that a dead time, which is a short-circuit prevention time, is provided in order to prevent the switching elements 3a to 3f of the inverter 3 from being vertically short-circuited.

  Further, when the switching elements 3a to 3f of the inverter 3 are controlled using the dead time, it is known that a voltage error ΔV represented by the following equation (8) occurs with respect to the output voltage command value.

In the above equation (8), Td is a dead time, fc is a carrier frequency for generating a PWM signal, and Vdc is a DC voltage applied to the inverter 3 by the DC power supply 2. Due to this voltage error ΔV, even when the excitation current command value i γ ^* for enabling the minimum current control is used, the voltage output from the inverter 3 decreases, the excitation increases, and the current increases. The problem of end up arises.

  Therefore, when the motor 5 is operated in an overload state, the operation of the inverter 3 may be stopped by an overcurrent cutoff measure for preventing an increase in current flowing through the motor 5. In reality, it is necessary to perform control to reduce the load so that such an overcurrent interruption measure does not work, and as a result, the maximum output is reduced. For example, in the case of a motor applied to an air conditioner, There is a possibility that problems such as a decrease in cooling capacity or heating capacity may occur.

Therefore, in order to output the motor applied voltage Va having the maximum efficiency, the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* in consideration of the voltage error ΔV due to the dead time Td are used as shown in FIG. Become. FIG. 7 is a vector diagram when the voltage error ΔV due to the dead time Td is taken into consideration. In FIG. 7, the same or similar vectors as those shown in the vector diagram of FIG.

  Here, when the currents on the dq axis at the minimum current phase angle βm in the case of controlling by “current value minimum control” are idm and iqm, the d-axis voltage Vdm that is the d-axis component of the motor voltage Va at the rotational frequency ω and The q-axis voltage Vqm, which is the q-axis component of the motor voltage Va, is expressed by the following equation (9).

On the other hand, the d-axis voltage command value Vd ^* and the q-axis voltage finger Vq ^* in consideration of the influence of the voltage error ΔV due to the dead time Td are as follows when the voltage errors on the dq axis due to the dead time Td are ΔVd, ΔVq 10) It can express with a formula.

  However, the voltage errors ΔVd and ΔVq on the dq axis are expressed by the following equation (11).

  In the above equation (11), “A” is a proportionality constant.

When the dead time Td is provided, the electrical energy supplied to the motor 5 is reduced by the amount of time for which the voltage dead time Td is provided. Therefore, a d-axis voltage command value Vd ^* and a q-axis voltage finger Vq ^{* obtained} by adding the d-axis voltage error ΔVd and the q-axis voltage error ΔVq, respectively, are used in order to compensate the electrical energy corresponding to the shaved portion. By using the d-axis voltage command value Vd ^* and the q-axis voltage finger Vq ^* obtained by adding the d-axis voltage error ΔVd and the q-axis voltage error ΔVq, it is possible to apply the motor voltage Va that achieves the maximum efficiency to the motor 5. Become.

In the primary magnetic flux control, the primary magnetic flux command value Φγ ^* is controlled by the exciting current command value Iγ ^* to be expressed by the following equation (12), and the γ-axis component of the primary magnetic flux command value Φγ ^* is the combined magnetic flux vector. It is controlled so as to coincide with the γ-axis component of Φ0.

If the primary magnetic flux error Φγerr is controlled to be 0 in the above equation (7), the γ-axis voltage command value Vγ ^* and the δ-axis voltage command value Vδ ^* are expressed by the following equation (13). Is done.

Therefore, the excitation current command value Iγ ^* is calculated under the condition that the scalar value of the above equation (10) and the scalar value of the above equation (13) coincide, that is, the condition that the following equation (14) is satisfied, and is obtained by measurement. By performing control so that the obtained γ-axis excitation current iγ coincides with the excitation current command value Iγ ^* , “current value minimum control” can be realized.

The calculation formula of the excitation current command value Iγ ^* can be expressed by the following formula (15).

  In the above equation (15), αm is an angle formed by the δ-axis current iδ and the q-axis, that is, the phase angle of the δ-axis current viewed from the q-axis. This αm is also the angle formed between the γ-axis current iγ and the d-axis, that is, the phase angle of the γ-axis current viewed from the d-axis.

Further, as shown in the above equation (15), since the excitation current command value Iγ ^* depends on the set values of the dead time Td and the carrier frequency fc, the excitation current command value Iγ ^* is matched with these set values in advance ^. It goes without saying that setting is desirable.

Since the DC voltage Vdc and the rotational frequency ω of the DC power supply 2 are expected to fluctuate during operation, the DC voltage Vdc is detected to control the excitation current command value Iγ ^* or the frequency command value ω ^*. Alternatively, the excitation current command value Iγ ^* may be controlled according to the primary frequency ω1. These control units are not shown, but can be configured in the excitation current command value calculation unit 14 of FIG. If such control is performed, the influence of the voltage error ΔV due to the dead time Td can be further reduced, and the operation in the “current value minimum control” can be performed more reliably.

Further, in the method of this embodiment, as compared to gradually control the exciting current command value i? ^* So that the minimum current, for controlling the excitation current command value i? ^* According to the above value, the load Since unstable operation due to fluctuations or rotation speed fluctuations can be avoided, the time until control converges can be shortened. This effect is an effect not seen in the prior art.

As described above, the calculation method of the excitation current command value Iγ ^* in the primary magnetic flux control has been described. However, in the case of a control system having an estimation unit that estimates the electrical angle of the motor 5 or a detection unit that detects the electrical angle. Operates so that the dq axis and the γ-δ axis substantially coincide with each other, and therefore, coincides with the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* shown in the above equation (10), respectively. The excitation current command value Iγ ^* may be selected, and the same effect as that of the present embodiment can be obtained.

  8 and 9 are examples of characteristic curves when the “current value minimum control method” according to the present embodiment is applied. FIG. 8 is a diagram illustrating a change in voltage with respect to the rotational speed, and FIG. It is a figure which shows the change of the electric current with respect to rotation speed. The horizontal axis may be read as load torque.

  8 and 9, the waveform (1) indicated by a broken line is an ideal waveform when there is no dead time Td, the waveform (2) indicated by a thin solid line sets the dead time Td, and the conventional method is used. It is a waveform when applied. As shown in the figure, a voltage error (see FIG. 8) or a current error (see FIG. 9) due to the dead time Td occurs in both waveforms.

  On the other hand, in FIG. 8 and FIG. 9, the waveform (3) indicated by the bold solid line is a waveform when the dead time Td is set and the method of the present embodiment is applied. As is clear from the comparison of the waveforms of (1) and (3), the voltage error or current error is large when the rotational speed is small, and the voltage error or current error decreases as the rotational speed increases. Yes.

  The voltage error ΔV due to the dead time Td is a fixed value when the carrier frequency fc is constant and the DC voltage Vdc does not fluctuate as shown in the above equation (8). Therefore, the ratio of the voltage error ΔV to the output voltage is high in the low speed region where the output voltage is small, and as shown in FIGS. 8 and 9, the voltage error or current error is low in the low speed region where the compensation amount is relatively large. Becomes larger.

  Therefore, when applying the method of the present embodiment, it is preferable to control so that the compensation amount is reduced when the rotational speed is small and the compensation amount is increased as the rotational speed approaches the rated rotational speed. If such control is performed, stable operation in a low speed range is possible. Further, if such control is performed, an increase in current near the rated speed can be suppressed, so that the operation range can be prevented from being reduced due to the operation current limit, and a wide operation range can be secured.

  Therefore, when applying the method of the present embodiment, it is preferable to control so that the compensation amount is reduced when the rotational speed is small and the compensation amount is increased as the rotational speed approaches the rated rotational speed. If such control is performed, stable operation in a low speed range is possible. Further, if such control is performed, an increase in current near the rated speed can be suppressed, so that the operation range can be prevented from being reduced due to the operation current limit, and a wide operation range can be secured. In this embodiment, an example in which the target rotation speed, that is, the first rotation speed is set as the rated rotation speed is shown. However, not only the rated rotation speed but also a rotation speed lower than the rated rotation speed has similar characteristics. The music can be obtained.

  As a method of compensating for the voltage error ΔV, there is a method of detecting the current flowing through the motor 5 by the current detection units 6a and 6b and compensating the output voltage according to the polarity of the detection value of the current detection units 6a and 6b. . However, in the case of this method, when the DC voltage Vdc is high, the current flowing through the inverter 3 becomes small. For this reason, when the detection accuracy of the current detectors 6a and 6b is low, an error occurs in the detection polarity due to the influence of noise or the like, and the voltage error ΔV is back-compensated. On the contrary, the voltage error ΔV increases. There is a problem that the operation of the motor 5 becomes unstable.

On the other hand, in the method of the present embodiment, since the excitation current command value Iγ ^* is only selected in consideration of the voltage error ΔV regardless of the polarity of the current, the unstable operation of the motor 5 due to reverse compensation of dead time correction is performed. It is possible to achieve “minimum current value control” while avoiding this.

  However, it is possible to set the dead time correction time within a range where the motor 5 does not become unstable. For example, in the above equation (11), 3 μs, which is the difference between the actual dead time (for example, 5 μs) and the dead time correction time (for example, 2 μs), may be set as Td in the above equation (11). . By performing such control, it is possible to realize “minimum current value control” while performing the minimum dead time correction.

  Further, when driving the motor 5 that is a driving load, when the calculation process by the excitation current command value calculation unit 14 described above is executed during the period from the start until reaching a certain low speed, the motor at the time of control execution 5 may not be able to maintain synchronous operation due to fluctuations in the output torque. For this reason, the execution of the calculation process by the excitation current command value calculation unit 14 may be temporarily stopped until the motor 5 reaches the set rotation speed or more. By bypassing low-speed arithmetic processing, unintended step-out can be avoided and a highly reliable device can be obtained.

  Further, in the region where the PWM signal is overmodulated, it becomes difficult to control the output voltage due to overmodulation, so that the effect of executing the calculation process by the excitation current command value calculation unit 14 is almost eliminated. For this reason, in the overmodulation region, the execution of the calculation process by the excitation current command value calculation unit 14 may be temporarily stopped to simplify the control. In particular, when integral control is used for excitation current control, etc., the operation of the integrator becomes abnormal during overmodulation. An efficient and reliable device can be obtained.

In addition, if the calculation process by the excitation current command value calculation unit 14 is executed during operation in an unsteady state such as start-up, acceleration operation, or deceleration operation, the control may become unstable or step out. There is also a case. Therefore, such a non-steady operation state, strengthened by applying the excitation control method, the above (15) to set the excitation current command value i? ^* To be exciting current command value i? ^* Be stronger than the excitation obtained by equation Thus, it is possible to avoid a situation where the control becomes unstable or step out, and it is possible to obtain a device with high control stability and reliability.

  Note that “higher excitation” generally refers to a case where an excitation current is passed so that the resultant magnetic flux vector Φ0 intensifies the magnetic flux vector Φf by the rotor magnet (when id> 0). On the contrary, the case where the excitation current flows so that the combined magnetic flux vector Φ0 weakens the magnetic flux vector Φf by the rotor magnet (when id <0) is called weakening excitation.

  The motor driving device 1 described in the present embodiment is an example of a sensorless driving method that does not use a position sensor that detects the rotor position of the motor 5 that is a driving load. A highly reliable device can be obtained. Note that the method of the present embodiment can also be applied to a driving method for detecting the rotor position, and the same effect can be obtained.

In the motor drive device 1 described in the present embodiment, in the calculation of the excitation current command value Iγ ^* , the resistance value R per phase of the IPMSM, the d-axis inductance Ld of the IPMSM, the q-axis inductance Lq of the IPMSM, The calculation is performed using a motor constant indicating the characteristics of the motor, such as the induced voltage constant Φf of IPMSM. For this reason, when the temperature of the motor changes due to the heat generation of the motor, the ambient environment of the motor, or the like, the motor constant changes, and an error may occur between the actual motor constant and the motor constant used for the calculation. When this error occurs, it may be difficult to control at a desired operating point.

  In order to more effectively realize the “current value minimum control” according to the present embodiment even for such a change in motor constant, for example, the voltage command value calculation unit 10 described in the present embodiment, etc. In addition, motor constant measuring means for measuring a motor constant using a current value flowing through the motor 5 and an applied voltage value to the motor 5 and a control constant changing means for changing control constants such as a control gain and a frequency compensation gain are provided. The motor constant is measured by the motor constant measuring means, the measured motor constant is corrected with respect to changes in the ambient environment such as temperature, and the control constant is changed by the control constant changing means according to the corrected motor constant. You just have to do it. The measurement of the motor constant may be performed online or offline.

  If the control as described above is performed, the motor constant can be changed following the change in the motor characteristics due to the temperature change, and the control constant can be changed. Even if it occurs, the “current value minimum control” according to the present embodiment can be more effectively executed.

  In the present embodiment, the IPMSM is exemplified as the motor 5, but the same effect can be obtained with any motor as long as it is a synchronous motor. For example, if it is a synchronous motor having a permanent magnet such as a permanent magnet synchronous motor (PMSM) or a brushless DC motor (Brushless Direct Current Motor: BLDCM), the same formula as this embodiment should be used. Can be realized. In particular, in PMSM, in a surface permanent magnet synchronous motor (SPMSM), the d-axis inductance Ld and the q-axis inductance Lq in each equation may be set to Ld = Lq, and a reluctance motor (Reluctance Motor: (RM), Synchronous Reluctance Motor (SynRM), Switched Reluctance Motor (SRM), and other synchronous motors that do not have permanent magnets (1) to (15) The induced voltage constant Φf inside may be calculated as Φf = 0.

Further, when operating in the relationship between the δ-axis current iδ and the excitation current command value iγ ^* shown in FIG. 6, the δ-axis current iδ is removed by using a low-pass filter to reduce the excitation current. It is possible to prevent a sudden change in the command value Iγ ^* , and it is possible to drive the motor 5 stably.

Furthermore, by storing and referring to the relationship between the δ-axis current iδ and the excitation current command value iγ ^* in advance as table data, processing such as a microcomputer used in the inverter control unit 2 can be minimized. Cost reduction is possible by using an inexpensive microcomputer.

  In addition, if many values are stored in the table data, the necessary memory of the microcomputer increases. Therefore, the number of necessary data is set to the minimum, and various interpolation methods such as linear interpolation are used for the values between the data. The number of required memories can be reduced, and the cost can be reduced by using an inexpensive microcomputer.

  The motor drive device according to the present embodiment is used as a motor drive device that drives a compressor such as an air conditioner, a heat pump water heater, a refrigerator, or a refrigerator, and a motor drive device that drives a motor of a blower. be able to. Among these application examples, the case where the motor drive device according to the present embodiment is applied to a heat pump device will be described below.

  FIG. 10 is a schematic configuration diagram when the motor drive device according to the present embodiment is applied to a heat pump device. In the heat pump apparatus 100 shown in FIG. 10, a compressor 51 having a motor 5 and a compression mechanism 50, a four-way valve 59, a heat exchanger 52, an expansion mechanism 61, and a heat exchanger 57 are sequentially connected by piping. It is configured. Hereinafter, the heat pump device 100 will be described in more detail.

  FIG. 11 is a circuit configuration diagram of the heat pump device 100 according to the present embodiment, and FIG. 12 is a Mollier diagram regarding the state of the refrigerant of the heat pump device 100 shown in FIG. 11. In FIG. 12, the horizontal axis represents specific enthalpy and the vertical axis represents refrigerant pressure.

  In the heat pump device 100, a compressor 51, a heat exchanger 52, an expansion mechanism 53, a receiver 54, an internal heat exchanger 55, an expansion mechanism 56, and a heat exchanger 57 are sequentially connected by piping, Is provided with a main refrigerant circuit 58 that circulates. In the main refrigerant circuit 58, a four-way valve 59 is provided on the discharge side of the compressor 51 so that the refrigerant circulation direction can be switched. A fan 60 is provided in the vicinity of the heat exchanger 57. The heat pump device 100 also includes an injection circuit 62 that connects between the receiver 54 and the internal heat exchanger 55 to the injection pipe of the compressor 51 by piping. An expansion mechanism 61 and an internal heat exchanger 55 are sequentially connected to the injection circuit 62.

  A water circuit 63 through which water circulates is connected to the heat exchanger 52. In addition, the water circuit 63 is connected to a device that uses water such as a water heater, a radiator, a radiator such as floor heating, and the like.

  First, the operation | movement at the time of the heating operation of the heat pump apparatus 100 is demonstrated. During the heating operation, the four-way valve 59 is set in the solid line direction. The heating operation includes not only heating used for air conditioning, but also hot water supply that heats water to make hot water.

  The gas-phase refrigerant (point 1 in FIG. 12) that has become high-temperature and high-pressure in the compressor 51 is discharged from the compressor 51 and is heat-exchanged and liquefied by a heat exchanger 52 that is a condenser and a radiator. Point 2). At this time, the water circulating in the water circuit 63 is warmed by the heat radiated from the refrigerant and used for heating and hot water supply.

  The liquid-phase refrigerant liquefied by the heat exchanger 52 is decompressed by the expansion mechanism 53 and becomes a gas-liquid two-phase state (point 3 in FIG. 12). The refrigerant in the gas-liquid two-phase state by the expansion mechanism 53 is heat-exchanged with the refrigerant sucked into the compressor 51 by the receiver 54, and is cooled and liquefied (point 4 in FIG. 12). The liquid phase refrigerant liquefied by the receiver 54 branches and flows into the main refrigerant circuit 58 and the injection circuit 62.

  The liquid-phase refrigerant flowing through the main refrigerant circuit 58 is heat-exchanged by the internal heat exchanger 55 with the refrigerant flowing through the injection circuit 62 that has been decompressed by the expansion mechanism 61 into a gas-liquid two-phase state, and further cooled (FIG. 12). Point 5). The liquid-phase refrigerant cooled by the internal heat exchanger 55 is decompressed by the expansion mechanism 56 and becomes a gas-liquid two-phase state (point 6 in FIG. 12). The refrigerant that has become a gas-liquid two-phase state by the expansion mechanism 56 is heat-exchanged with the outside air by the heat exchanger 57 serving as an evaporator and heated (point 7 in FIG. 12). Then, the refrigerant heated by the heat exchanger 57 is further heated by the receiver 54 (point 8 in FIG. 12) and sucked into the compressor 51.

  On the other hand, as described above, the refrigerant flowing through the injection circuit 62 is decompressed by the expansion mechanism 61 (point 9 in FIG. 12), and is heat-exchanged by the internal heat exchanger 55 (point 10 in FIG. 12). The gas-liquid two-phase refrigerant (injection refrigerant) heat-exchanged by the internal heat exchanger 55 flows into the compressor 51 from the injection pipe of the compressor 51 in the gas-liquid two-phase state.

  In the compressor 51, the refrigerant sucked from the main refrigerant circuit 58 (point 8 in FIG. 12) is compressed and heated to an intermediate pressure (point 11 in FIG. 12). The injection refrigerant (point 10 in FIG. 12) joins the refrigerant compressed and heated to the intermediate pressure (point 11 in FIG. 12), and the temperature decreases (point 12 in FIG. 12). And the refrigerant | coolant (point 12 of FIG. 12) which temperature fell further is compressed and heated, becomes high temperature high pressure, and is discharged (point 1 of FIG. 12).

  When the injection operation is not performed, the opening degree of the expansion mechanism 61 is fully closed. That is, when the injection operation is performed, the opening degree of the expansion mechanism 61 is larger than the predetermined opening degree. However, when the injection operation is not performed, the opening degree of the expansion mechanism 61 is more than the predetermined opening degree. Make it smaller. Thereby, the refrigerant does not flow into the injection pipe of the compressor 51. Here, the opening degree of the expansion mechanism 61 is controlled electronically by a control unit such as a microcomputer.

  Next, the operation of the heat pump device 100 during the cooling operation will be described. During the cooling operation, the four-way valve 59 is set in a broken line direction. The cooling operation includes not only cooling used in air conditioning but also making cold water by taking heat from water and freezing.

  The gas-phase refrigerant (point 1 in FIG. 12) that has become high-temperature and high-pressure in the compressor 51 is discharged from the compressor 51 and is heat-exchanged and liquefied by a heat exchanger 57 that is a condenser and a radiator. Point 2). The liquid-phase refrigerant liquefied by the heat exchanger 57 is decompressed by the expansion mechanism 56 and becomes a gas-liquid two-phase state (point 3 in FIG. 12). The refrigerant in the gas-liquid two-phase state by the expansion mechanism 56 is heat-exchanged by the internal heat exchanger 55, cooled and liquefied (point 4 in FIG. 12). In the internal heat exchanger 55, the refrigerant that has become a gas-liquid two-phase state by the expansion mechanism 56 and the liquid-phase refrigerant that has been liquefied by the internal heat exchanger 55 have been decompressed by the expansion mechanism 61, and have become a gas-liquid two-phase state. Heat is exchanged with the refrigerant (point 9 in FIG. 12). The liquid refrigerant (point 4 in FIG. 12) heat-exchanged by the internal heat exchanger 55 branches and flows into the main refrigerant circuit 58 and the injection circuit 62.

  The liquid-phase refrigerant flowing through the main refrigerant circuit 58 is heat-exchanged with the refrigerant sucked into the compressor 51 by the receiver 54 and further cooled (point 5 in FIG. 12). The liquid-phase refrigerant cooled by the receiver 54 is decompressed by the expansion mechanism 53 and becomes a gas-liquid two-phase state (point 6 in FIG. 12). The refrigerant in the gas-liquid two-phase state by the expansion mechanism 53 is heat-exchanged and heated by the heat exchanger 52 serving as an evaporator (point 7 in FIG. 12). At this time, when the refrigerant absorbs heat, the water circulating in the water circuit 63 is cooled and used for cooling and freezing. Then, the refrigerant heated by the heat exchanger 52 is further heated by the receiver 54 (point 8 in FIG. 12) and sucked into the compressor 51.

  On the other hand, as described above, the refrigerant flowing through the injection circuit 62 is decompressed by the expansion mechanism 61 (point 9 in FIG. 12), and is heat-exchanged by the internal heat exchanger 55 (point 10 in FIG. 12). The gas-liquid two-phase refrigerant (injection refrigerant) heat-exchanged by the internal heat exchanger 55 flows from the injection pipe of the compressor 51 in the gas-liquid two-phase state. The compression operation in the compressor 51 is the same as in the heating operation.

  When the injection operation is not performed, the opening degree of the expansion mechanism 61 is fully closed so that the refrigerant does not flow into the injection pipe of the compressor 51 as in the heating operation.

  In the above description, the heat exchanger 52 has been described as a heat exchanger such as a plate heat exchanger that exchanges heat between the refrigerant and the water circulating in the water circuit 63. The heat exchanger 52 is not limited to this and may exchange heat between the refrigerant and the air. Further, the water circuit 63 may be a circuit in which other fluid circulates instead of a circuit in which water circulates.

  The heat pump device 100 described above can be used in a heat pump device using an inverter compressor such as an air conditioner, a heat pump water heater, a refrigerator, or a refrigerator. For this reason, in the air conditioner, the heat pump water heater, the refrigerator, and the refrigerator that use the heat pump device, the motor drive device 1 according to the present embodiment described above is applied to the heat pump device. You can enjoy the effect.

  In addition, if a WBG semiconductor is used for the switching element constituting the inverter in the above-described embodiment and the diode element connected in parallel to the switching element, switching at a higher frequency is possible. The current flowing to the inverter can be reduced by reducing the winding current by increasing the winding impedance of the compressor motor, and a more efficient heat pump device can be obtained. Furthermore, since it is easy to increase the frequency, it is possible to set the frequency to be higher than the audible frequency band, and there is an advantage that noise countermeasures are easy.

  Although both the switching element and the diode element are preferably formed of a WBG semiconductor, any one of the elements may be formed of a WBG semiconductor, and the effects in the above-described embodiments can be obtained. .

  In addition to the WBG semiconductor, a similar effect can be obtained by using a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) having a super junction structure, which is known as a highly efficient switching element.

  Note that the configurations shown in the above embodiments are examples of the contents of the present invention, and can be combined with other known techniques, and can be combined without departing from the gist of the present invention. It is also possible to omit or change a part of.

  DESCRIPTION OF SYMBOLS 1 Motor drive device, 2 DC power supply, 3 Inverter, 4 Inverter control part, 5 Motor, 6a, 6b Current detection part, 7 3 phase 2 phase conversion part, 8 Frequency compensator, 9 Primary frequency calculation part, 10 Voltage command value Calculation unit, 11 Two-phase three-phase conversion unit, 12 PWM signal generation unit, 13 Electrical angle phase calculation unit, 14 Excitation current command value calculation unit, 50 Compression mechanism, 51 Compressor, 52, 57 Heat exchanger, 53, 56 , 61 expansion mechanism, 54 receiver, 55 internal heat exchanger, 58 main refrigerant circuit, 59 four-way valve, 60 fan, 62 injection circuit, 63 water circuit, 100 heat pump device.

Claims (12)

A motor driving the load;
An inverter for applying a voltage to the motor;
An inverter control unit for controlling the voltage output by the inverter,
The voltage output from the inverter is increased in accordance with the increase in the rotation speed of the motor so that the current flowing through the motor becomes close to the voltage value at which the current flowing through the motor becomes the minimum at the first rotation speed within the range of the drive rotation speed of the motor. Controlling the voltage output by the inverter ;
During execution of weak excitation control, execution of control to compensate by increasing the voltage output by the inverter is stopped.
Motor drive. A motor driving the load;
An inverter for applying a voltage to the motor;
An inverter control unit for controlling the voltage output by the inverter,
The voltage output from the inverter is increased in accordance with the increase in the rotation speed of the motor so that the current flowing through the motor becomes close to the voltage value at which the current flowing through the motor becomes the minimum at the first rotation speed within the range of the drive rotation speed of the motor. Controlling the voltage output by the inverter;
When the rotational speed of the motor is small, the compensation amount of the voltage error due to the dead time of the inverter is reduced, and the compensation amount is controlled to increase as the rotational speed of the motor approaches the first rotational speed.
Motor drive. A current detector for detecting a current flowing in the prior SL motor,
The motor drive device according to claim 1 or 2 for controlling the voltage which the inverter outputs based on the output of the current detector. Includes a coordinate conversion means for converting the output of the previous SL current detector to the exciting current and the torque current,
The motor driving device according to claim 3 , wherein a voltage output from the inverter is controlled based on the excitation current command. Excitation magnetic axis current, torque current, based on the frequency command value or primary frequency, an arithmetic unit for generating an excitation current command value,
The motor driving device according to claim 4 , wherein a voltage output from the inverter is controlled based on the excitation current command. During operation in non-steady conditions including time and deceleration operation start and during acceleration operation, any one of claims 1 to 5 for stopping the execution of the control to compensate by increasing the voltage which the inverter outputs 1 The motor drive device according to item. During operation in non-steady conditions including time and deceleration operation start and during acceleration operation, it stops the execution of the control to compensate by increasing the voltage the inverter output, a stronger excitation control excitation current The motor drive device according to claim 1, wherein control is executed by generating a command value. In the region where the PWM signal that are generated becomes overmodulation, the motor driving apparatus according to any one of claims 1 5 for stopping the execution of the control to compensate by increasing the voltage which the inverter outputs. At least one motor driving apparatus according to any one of claims 1 formed by the wide band gap semiconductor 8 of the switching elements or diodes constituting the inverter. A heat pump device equipped with the motor drive device according to any one of claims 1 to 9 . A refrigerating and air-conditioning apparatus equipped with the motor drive device according to any one of claims 1 to 9 . The air blower which mounts the motor drive device of any one of Claim 1 to 9 .

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