JP5473289B2 - Control device and control method for permanent magnet type synchronous motor - Google Patents

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor, and more particularly to a device and method for making an inverter output voltage proportional to a voltage command value even in an overmodulation control region.

  The torque τ of the permanent magnet type synchronous motor is generally expressed by the following equation (1).

Here, Λ _d is an induced voltage coefficient, and i _d and i _q are d-axis and q-axis armature currents, respectively. L _d and L _q are the inductances of the d axis and the q axis, respectively, and n is the number of pole pairs. The d axis is an axis provided in the field direction of the motor rotor, and the q axis is an axis whose phase is advanced 90 degrees from the d axis.

The first term of equation (1) represents the torque (magnet torque) generated by the q-axis current flowing in a direction perpendicular to the field direction by the magnet. Further, the second term of the same expression represents the torque (reluctance torque) generated when the reluctance is different between the d-axis direction and the q-axis direction. In a surface magnet type motor (SPM) where L _d = L _q , the second term can be ignored. Even in the case of an internal magnet type motor (IPM), if the difference between L _d and L _q is small, the second term is often ignored.

For this reason, when controlling a permanent magnet type synchronous motor, if it can be controlled to I _d = 0, torque can be generated with the least current, and the efficiency becomes high. When the sensorless control does not detect the field direction of the motor, and a method of calculating the d-axis current i _d from inverter δ axis and γ-axis current can be calculated based on the motor constant (induced voltage constant) is known Yes.

Based on the above, Patent Document 1 proposes a control device for a permanent magnet type synchronous motor that has a simple configuration and performs high-efficiency control with a minimum current. As shown in FIG. 15, the basic configuration of this apparatus is a general V / f control configuration, but the current of the terminals of a permanent magnet type synchronous motor (hereinafter simply referred to as a motor) 8 is supplied to a three-phase / The power factor angle setting means 10 is set so that the rotation axis used for the two-phase conversion is not θ but θ−φ. In order to obtain φ, first, I is obtained using the δ-axis and γ-axis currents fed back. When I is obtained, the constant 9 is multiplied by a constant so that φ is subtracted from the rotation axis θ of the three-phase / two-phase conversion by means 9 for obtaining φ. The current that has undergone the three-phase / two-phase conversion with θ−φ as the rotation axis is negatively fed back to the voltage command, and the speed command nω _m ^* is corrected so as to match the power factor angle φ of the motor 8 so as to be the minimum current. The

Japanese Patent Laying-Open No. 2006-197789 (FIG. 1)

By the way, in the conventional control device, the maximum value VMMAX of the output voltage (voltage between the v _u and v _v lines) of the inverter 7 is limited to the value of the following equation (2) or less. However, in Equation (2), Vdc is a converter voltage in the inverter 7.

VMMAX is limited to a value equal to or less than the value of equation (2) because the region where the relationship between the voltage command value v ^* _γ and the output voltage of the inverter is proportional (hereinafter, v / f control region) is shown in FIG. This is because the value is up to the value of Expression (2) as shown. Of course, it is possible to make the voltage command value v ^* _γ larger than the value given by the equation (1). However, as shown in FIG. 16, the voltage command value v ^* _γ and the inverter output voltage are not proportional to each other. Until now, an overmodulation control region that exceeds the v / f control region has never been used.
The present invention has been made based on such a technical problem. A control device and a control method for a permanent magnet type synchronous motor capable of making an inverter output voltage proportional to a voltage command value even in an overmodulation control region. The purpose is to provide.

For this purpose, the control device for a permanent magnet type synchronous motor according to the present invention sets the voltage command value II that sets the voltage command value II proportional to the motor frequency in two axes of the δ axis and the γ axis of the rotation rectangular coordinate system. Means, two-phase / three-phase conversion means for converting the coordinate command of the two-axis voltage command value II into a three-phase voltage command value III, an inverter for converting the power of the three-phase voltage command value III and applying it to the motor, A means for feeding back the motor terminal current to the voltage command value II without detecting the field direction of the motor, a power factor angle determining means for determining the power factor angle from the feedback current, and a motor terminal obtained in three phases In the overmodulation control region, the voltage factor III and the output voltage from the inverter are proportional to each other in the power factor angle adjusting means that adjusts the power factor angle to the rotation angle used when converting the current into Cartesian coordinates. Correction means for correcting the command value II. That. The correcting means corrects only one of the voltage command values v ^* _δ and v ^* _γ set by the voltage command value setting means between the voltage command value setting means and the two-phase / three-phase conversion means. Features.

Also, in the permanent magnet synchronous motor control device of the present invention preferably has a PWM waveform means for converting the voltage command values III of the three-phase pulse width modulation waveform.

The present invention can be regarded as a control method of a permanent magnet type synchronous motor, and this control method sets a voltage command value II proportional to the motor frequency on two axes of a δ axis and a γ axis of a rotation rectangular coordinate system. Step (a), step (b) for coordinate conversion of the voltage command value II for the two axes to a voltage command value III for three phases, and power conversion of the voltage command value III for three phases by an inverter and applying it to the motor Step (c), step (d) of feeding back the motor terminal current without detecting the field direction of the motor, step (e) of determining the power factor angle from the current in step (d) of feeding back A step (f) of adding or subtracting the power factor angle to the rotation angle used when converting the motor terminal current obtained in three phases into orthogonal coordinates ; and the voltage command value III and the output voltage from the inverter in the overmodulation control region Is proportional In so that comprises a step (g) for correcting the voltage command value II, and in step (g), between steps (a) and step (b), the voltage command set in step (a) Only one of the values v ^* _δ and v ^* _γ is corrected .

Also, in the control method of the present invention, converts the voltage command value III of the three-phase pulse width modulation waveform, it is preferable to have a step (h) is supplied to the inverter.

  According to the present invention, the inverter output voltage can be made proportional to the voltage command value even in the overmodulation control region.

  Hereinafter, embodiments of a permanent magnet type synchronous motor control device according to the present invention will be described in detail with reference to the drawings.

FIG. 1 is a functional block diagram showing a controller 1 for a permanent magnet type synchronous motor according to an embodiment of the present invention. In FIG. 1, ω _m is a rotational speed command (value), n is the number of pole pairs of the motor, ω ₁ is an inverter output frequency (primary frequency), v ^* _δ and v ^* _γ are δ-axis and γ-axis voltage commands ( _^value), v ^* _u, v * ^{v, v} _{* w} each u-phase, v-phase voltage command of the w-phase _(value), v _u, v v, _{v w} respectively u phase, v phase, and w-phase The output voltages, i _u , i _v , and i _w are the u-phase, v-phase, and w-phase output currents, i _δ and i _γ are the δ-axis and γ-axis inverter output currents, respectively, and i _δ is the excitation current. The component i _γ is a torque current component. θ is the output voltage phase, and φ is the power factor angle.

  The control device 1 includes a speed command setting unit 2, a voltage command value setting unit 3, a voltage command value correction unit 4, a two-phase / three-phase conversion unit 5, a PWM (Pulse Wide Modulation) waveform unit 6, and an inverter 7. It comprises a three-phase / two-phase conversion means 9 and a power factor angle setting means 10. A permanent magnet type synchronous motor (hereinafter simply referred to as a motor) 8 is finally driven by the voltage applied from the inverter 7. The control device 1 performs sensorless control that does not detect the field direction of the motor.

In the voltage command value setting means 3, the currents _{u u} , i _v , i _w flowing through the three phases of the u, v, w 3 phases flowing by applying the output voltages v _u , v _v , v _w of the inverter 7 to the motor 8 are converted into three phases. Using the δ-axis current i _δ , the inverter output frequency ω ₁ , the proportionality constant K, the d-axis induced voltage coefficient Λ _d , and the proportional gain K after the three-phase / two-phase conversion by the / two-phase conversion means 9 The δ-axis voltage command value v ^* _δ and the γ-axis voltage command value v ^* _γ are obtained by the calculations of equations (3-1) and (3-2).

  Below, the control when the voltage command value setting means 3 is made into said Formula (3-1), (3-2) is demonstrated. First, in order to help understanding, a case where the power factor angle φ (i) is 0 will be described. Since φ is a power factor angle, when the phase difference between the voltage applied to the motor 8 and its current is 0, the power factor represented by cos φ is 1. Therefore, this control is so-called power factor 1 control.

In the voltage command value calculation by the above formulas (3-1) and (3-2), for example, if the γ-axis voltage command value v ^* _γ is smaller than the terminal voltage of the motor 8, a δ-axis inverter is used to weaken the field. The output current i _δ becomes negative. On the contrary, if v ^* _γ is large, the field becomes stronger in accordance with it, and i _δ becomes positive.
Specifically, in the calculation of Expression (3-1), the δ-axis voltage command value v ^* _δ is negatively proportionally controlled so that the δ-axis current i _δ becomes zero. Now, if there is fluctuation of the positive and negative i _[delta], the voltage command value so as to eliminate immediately wander i _[delta] by the gain K? V ^* _[delta] is determined.

The γ-axis voltage command value v ^* _γ is a voltage value that compensates for the voltage of Λ _d ω ₁ corresponding to the induced voltage, and in addition, in order to perform an operation for setting the δ-axis current i _δ to 0, Integration control for integrating i _δ is performed. That is, in the calculation of Expression (3-2), if the δ-axis current i _δ accumulates with either positive or negative value, a voltage value that returns to the original value with a larger value is determined. Although integral control is used here, proportional control, proportional integral control, or the like may be used depending on control responsiveness.

By doing so, i _δ becomes 0, and when i _δ becomes 0, v _δ also becomes 0. Eventually, the current flowing through the motor 8 and the motor applied voltage are only γ-axis components, and the phase coincides, so that the power factor is 1. This alone minimizes the product of voltage and current and enables power-saving control.

However, in the IPM (internal magnet type) motor, the reluctance torque that is the second term of the formula (1) can also be used. Therefore, the current can be further reduced by using both the magnet torque and the reluctance torque in a balanced manner. . When the motor current (absolute value obtained by combining the δ axis and the γ axis) I flows with an advance angle θ with respect to the induced voltage, the d axis current i _d and the q axis current i _q are expressed by the equations (4-1) and (4 -2).

  The condition for maximizing the motor torque τ with respect to the motor current I is that τ (expression (5-1)) obtained by substituting expressions (4-1) and (4-2) into expression (1). The derivative with respect to is zero. When this is expressed as equation (5-2), it is as follows.

When I · sin θ and I · cos θ in the above equation (5-2) are expressed by i _d and i _q in equations (4-1) and (4-2), the following equation (6) is obtained.

Equation (6) as _{A = Λ d / 2 (L} d -L q), an expression that represents the relationship of ^{_{(i d + A) 2 -i}} q 2 = A 2, the vertical axis a _{i d,} i _{If q} is on the horizontal axis, the graph is as shown in FIG.

Therefore, the figure shows the combination of i _d and i _q when the minimum current is reached, but the relationship between i _d and i _q includes a square root. A technique such as making a table is required.

  Therefore, control is performed based on the phase difference between the motor current I and the motor applied voltage V. First, since φ is the difference between the phase of the motor applied voltage V and the phase of the motor current I, it is expressed by the following equation (7).

As described above, the relationship between i _d and i _{q at} the minimum current is ( _id + A) ² −i _q ² = A ² , which is a hyperbola, so that the hyperbolic function (sinh, cosh) It can be expressed as equations (8-1) and (8-2).

When α is sufficiently small and can be regarded as 0, it can be approximated as i _d = Aα ² and i _q = Aα, respectively. Further, v _d and v _q obtained by capturing the voltage applied to the motor on the d-axis and the q-axis can be regarded as the following expressions (9-1) and (9-2) from an equivalent expression of the motor circuit. This is because resistance is usually negligible at the operating speed.

  Using these equations, the phase difference φ in equation (7) can be approximated as in the following equation (10).

Here, since the motor current I can be expressed by the following equation (11), the phase difference φ is eventually expressed by the following equation (12). That is, the phase difference φ is substantially proportional to the current. Further, i _δ = 0 by control, and the motor current I is proportional to i _γ , so the phase difference φ is also proportional to i _γ .
FIG. 3 shows the relationship between the phase difference φ and the motor current I obtained by plotting using the d-axis current and the q-axis current without approximation as described above. Even in such a case, it is almost proportional, and the validity of control using the proportional relationship can be understood.

As described above, the control of the minimum current necessary for generating the torque can be realized by positively controlling the power factor angle (phase difference) instead of φ = 0. Therefore, in the control device 1, as shown in FIG. 1, the rotation axis used when the current of the terminal of the motor 8 is three-phase / two-phase converted by the three-phase / two-phase conversion means 9 is θ− instead of θ−. A power factor angle setting means 10 for φ is provided. Since φ is a function of the motor current I, the motor current I is first obtained using the fed back γ-axis current (i _δ is controlled to 0) as shown in FIG. When the motor current I is obtained, φ is obtained by multiplying it by a constant. Thus, φ is subtracted from the rotation axis θ of the three-phase / two-phase conversion.

Then, the current subjected to the three-phase / two-phase conversion by the three-phase / two-phase conversion means 9 with θ-φ as the rotation axis is negatively fed back to the speed command nω _m ^* which is a frequency multiplied by the gain Kω, and is a permanent magnet type. The command nω _m ^* is corrected so that the synchronous motor 8 operates stably. This is because the load increases and the current increases when the rotor position is delayed relative to the rotating magnetic field. Conversely, the load decreases and the current decreases as the rotor position advances relative to the rotating magnetic field. It stabilizes by correcting. Since it is controlled so that v _δ = I _δ = 0 by the equations (3-1) and (3-2), it is used for current three-phase / 2-phase conversion and voltage two-phase / 3-phase conversion. If the phases to be operated are different, the difference φ is controlled to be the power factor angle.

Since the relationship between the motor current I and the phase difference φ is proportional, φ may be derived by calculation by selecting an appropriate proportionality constant, or accuracy in a large current value range is obtained. If there is, the relationship between the motor current I and φ may be tabulated. Even in the case of adopting a table configuration, there is an advantage that it is not necessary to take fine table data because it is almost proportional.
Even in this case, unlike referring to a two-dimensional table that is a combination of i _δ and i _γ , since the motor current I is obtained by actual feedback, φ can be uniquely determined and is always appropriate. The phase difference φ can be obtained.

  As described above, according to the control device 1, the torque of the permanent magnet type synchronous motor 8 can be controlled with the minimum current in consideration of the reluctance torque of the IPM motor. In addition, since the torque control with the minimum current can be performed only by changing the rotation axis angle at the time of the three-phase / two-phase conversion without requiring a complicated calculation, the configuration becomes simple and the maintainability is improved. Further, as shown in FIG. 4, the torque range in which the V / f control is appropriate is expanded with respect to the case of the power factor 1 control, and the current is adjusted while the basic V / f control is performed from the low speed rotation to the high speed rotation. It is possible to operate with minimum.

Next, the control device 1 has voltage command value correction means 4 for correcting the voltage command value v ^* _γ . The voltage command value correction means 4 is located between the voltage command value setting means 3 and the two-phase / three-phase conversion means 5, and the voltage command value v ^* _δ and the voltage command value generated by the voltage command value setting means 3. Among the v ^* _γ , the voltage command value v ^* _γ is corrected. The voltage command value v ^* _δ may be corrected, but according to the study by the present inventors, it is sufficient to correct only the voltage command value v ^* _γ .

As described with reference to FIG. 16, conventionally, there is no control within the overmodulation control region. However, the control device 1 enables proportional control of the overmodulation control region by correcting the voltage command value v ^* _γ . FIG. 5 is a graph (table) showing an example of the contents of the correction, where the horizontal axis indicates the voltage command value v ^* _γ , and the vertical axis indicates the corrected voltage command value obtained by correcting the voltage command value v ^* _γ. v ^* _γ ^* is shown. In other words, the correction is performed by multiplying the voltage command value v ^* _γ by a corresponding value on the correction function curve L so that the basic component of the actual PWM waveform output generated by the voltage command value is proportional to the voltage command value III. To be output. The correction is performed in the overmodulation control region. In FIG. 5, “100” on the horizontal axis is the start point of the overmodulation control region, that is, the correction start point. Further, the correction is applied in a range of Vdc × 1.12 (“112” on the horizontal axis) or less. The reason why the correction is Vdc × 1.12 or less is that if the value exceeds the correction, pulse width modulation cannot be performed.

FIG. 6 shows the relationship between the voltage command value v ^* _γ (v ^* _γ ^* ) and the output voltage of the inverter 7 when the correction voltage command value v ^* _γ ^* is applied to the overmodulation control region. Even in the overmodulation control region following the v / f control where proportional control is possible, the output voltage of the inverter 7 with respect to the voltage command value v ^* _γ (v ^* _γ ^* ) up to a range of 1.12 times the converter voltage Vdc. Proportional control of Vm becomes possible. Therefore, according to the control device 1, the maximum value of the output voltage of the inverter can be increased to 1.12 × VMMAX as compared with the conventional control device that does not use the overmodulation control region.

Since the fundamental wave component of the actual PWM waveform output is proportionally output to the overmodulation control region with respect to the voltage command value by the voltage command value correcting means 4, the following effects can be obtained when the motor 8 is driven.
(1) By driving the motor 8 having the same number of turns as the motor according to the conventional control that does not use the overmodulation control region, the output voltage of the inverter 7 is increased to 1.12 times that of the conventional one. Therefore, if the motor 7 having the same number of turns as in the conventional control is used, as shown in FIG. 7, the control can be performed up to the number of revolutions 1.12 times the conventional maximum number of revolutions.
(2) If the motor 8 having a larger number of turns than the motor according to the conventional control is driven, as shown in FIG. 8, the maximum rotational speed is the same level as the conventional one, but the output current of the inverter 7 is 1 / Since it is reduced to 1.12, the operation efficiency of the motor 8 is improved.

The control device 1 includes PWM waveform means 6. In the v / f control region, the PWM waveform means 6 performs two-phase modulation in which one of the three phases continuously stops switching for about 60 degrees in electrical angle, and in the overmodulation control region, three-phase modulation is performed. The modulation is performed by superimposing the third-order harmonic on the modulation.
By providing such a PWM waveform means 6, a sine wave can be converted into a PWM waveform, and the number of times of switching during control can be reduced to improve the driving efficiency of the motor 8.

  Further, after the output voltage of the inverter 7 reaches the maximum value (1.12 × VMMAX) in the overmodulation control region, the control device 1 weakens the control so that the maximum value voltage is constant as shown in FIG. Field control can be implemented. By doing so, the rotational speed control of the motor 8 can be continued even after the maximum voltage of the inverter 7 is reached.

If the control described above becomes possible, it can contribute to miniaturization and power saving of industrial products. For example, since a motor for driving a compressor of an air conditioner is a device that consumes a large amount of power together with a request for downsizing, there is a strong demand for power saving. The permanent magnet type synchronous motor has been adopted as one of the manifestations, but the control device 1 meets the above requirements.
In the control device 1, the voltage command value correction unit 4 is provided between the voltage command value setting unit 3 and the two-phase / three-phase conversion unit 5. Further, the control device 1 of the voltage command value v ^* _[delta] and the voltage command value v ^* _gamma, are corrected only the voltage command value v ^* _gamma. As described above, the control device 1 shows the correction while minimizing the change of the configuration.

Next, a motor step-out detection method performed by the control device 1 for the motor 8 according to the embodiment of the present invention described above will be described.
As described above, the permanent magnet type synchronous motor according to the present embodiment is controlled so that v _δ = 0 as shown in the equations (3-1) and (3-2). In the control method according to the present embodiment, for stabilization of control, the deviation from the δ-axis current command value i _δ ^* is integrated, and the voltage command values v _δ and v _γ are determined so as to cancel the deviation. doing. Therefore, in the normal operation state, the γ-axis voltage offset value V _γ ofs (= K∫i _δ dt) converges to a substantially constant value.
FIG. 10 shows a waveform at the time of starting the permanent magnet type synchronous motor in a normal state. In FIG. 10, the γ-axis voltage offset value V _γ ofs, the δ-axis current i _δ , and the motor current waveform (U phase) are shown from the top. As shown in this figure, during normal operation, both the γ-axis voltage offset value V _γ ofs and the δ-axis current i _δ converge to zero.

On the other hand, when the step-out occurs, the δ-axis current i _δ increases and a strong field state is obtained. FIG. 11 shows a waveform when the permanent magnet type synchronous motor 8 is started in the step-out state. In FIG. 11, the γ-axis voltage offset value V _γ ofs, the δ-axis current i _δ , and the motor current waveform (U phase) are shown from the top. As shown in this figure, it can be seen that in the step-out state, the γ-axis voltage offset value V _γ ofs and the δ-axis current i _δ are increased.
Therefore, in the step-out detection method of the control device 1, step-out is detected by detecting an increase in the δ-axis current i _δ or the γ-axis voltage offset value V _γ ofs.

Specifically, the step-out is detected when a state where the δ-axis current i _δ is equal to or greater than a predetermined value continues for a predetermined period. This makes it possible to detect the motor step-out very easily. Here, the specified value and the predetermined period are values that can be arbitrarily set according to the characteristics of the motor.
Further, the step-out may be detected by using the γ voltage offset value V _γ ofs, which is an integral value of the δ-axis current, instead of the δ-axis current i _δ . Specifically, if the gamma voltage offset value V _gamma ofs becomes a specified value or higher, preferably, when the gamma voltage offset value V _gamma ofs is that specified value or more states continues for a predetermined period of time, detects the loss of synchronism You may make it do.

Generally, in a synchronous motor, a pull-in operation is performed by forced synchronous operation when starting up. During the forced synchronous operation, the δ-axis current i _δ does not converge to zero, and therefore, if only the δ-axis current i _δ is monitored as described above, there is a possibility of erroneous detection of step-out. Even in such a case, by using the γ voltage offset value V _γ ofs, which is an integral value of the δ-axis current i _δ , it is possible to prevent erroneous detection of step-out and improve step-out detection accuracy. it can.

As described above, according to the control device of the present embodiment, the motor is removed using at least one of the γ-axis voltage offset value V _γ ofs and the δ-axis current i _δ used for motor control. Since tone detection is performed, it is possible to realize step-out detection very easily.

Furthermore, in the step-out detection method according to the present embodiment, the specified value may be updated according to the temperature in consideration of a change in the magnetic flux amount due to the motor temperature. For example, when the temperature rises, the amount of magnetic flux decreases, and thus the δ-axis current i _δ decreases. Conversely, when the temperature decreases, the amount of magnetic flux increases, and thus the δ-axis current i _δ increases. Therefore, by setting the specified value high when the temperature rises and setting the specified value low when the temperature decreases, it is possible to further increase the accuracy of step-out detection.

Note that the γ-axis voltage offset value V _γ ofs also varies depending on the change in the magnetic flux amount of the motor. FIG. 12 to FIG. 14 show changes in each state quantity at the start-up when the amount of magnetic flux is changed in a normal state, that is, in a state where no step-out has occurred. FIG. 12 shows each state quantity when the magnetic flux amount is normal, FIG. 14 shows the case where the magnetic flux amount is increased, and FIG. 13 shows each state quantity when the magnetic flux amount is intermediate between FIGS. In FIGS. 12 to 14, waveforms of the γ-axis voltage offset value V _γ ofs, the δ-axis current i _δ , and the motor current waveform (U phase) are shown from the top.

Thus, even if no step-out occurs, the γ-axis voltage offset value V _γ ofs increases or decreases according to the amount of magnetic flux. Therefore, if only the change in the γ-axis voltage offset value V _γ ofs is detected, it is possible to distinguish whether the change is caused by a step-out or a change in the amount of magnetic flux. It can be difficult. However, even in such a case, the step-out and the electromagnetic state can be distinguished by the combination of the γ-axis voltage offset value V _γ ofs and the inverter output power.

In other words, when the amount of magnetic flux of the motor increases, even if the γ-axis voltage offset value V _γ ofs increases, the inverter output takes a value corresponding to the operation state at that time, whereas at the time of out-of-step occurrence When the γ-axis voltage offset value V _γ ofs is increased, the inverter does not work, so the output power is decreased and becomes equal to or less than the specified value.
Therefore, when the state where the γ-axis voltage offset value V _γ ofs is equal to or greater than the specified value continues for a predetermined period and the state where the inverter output power is equal to or less than the specified value continues for a predetermined period, it is determined that the step-out is caused It is possible to avoid erroneous detection due to fluctuations in the amount of magnetic flux. In addition to the inverter output power, it is also possible to make a determination based on the effective value of the motor current or the like.

  The control device 1 is applied to a motor for driving a compressor of an air conditioner, but the present invention is not limited to this and can be widely applied to control of a permanent magnet type synchronous motor.

It is a block diagram which shows the system configuration | structure of the control apparatus which concerns on this Embodiment. It is a graph which shows the relationship between the d-axis current and the q-axis current at the minimum current, the vertical axis is the d-axis current, and the horizontal axis is the q-axis current. It is a graph which shows the relationship between the electric current when it becomes the minimum electric current, and a phase difference, a vertical axis | shaft is a phase difference and a horizontal axis is an electric current. It is a graph which shows the current-torque characteristic in the case of power factor 1 control, and the case of this invention, a vertical axis | shaft is an electric current and a horizontal axis is a torque. 6 is a graph showing a correction completion curve for correcting a voltage command value v ^* _γ to a corrected voltage command value v ^* _γ ^* in the overmodulation control region. It is a graph which shows the relationship between voltage command value v ^* _γ (v ^* _γ ^* ) and the output voltage of an inverter when correction voltage command value v ^* _γ ^* is applied to the overmodulation control region. It is a graph which shows the effect when using the control apparatus which concerns on this Embodiment, and driving the motor with the same number of turns. It is a graph which shows the effect when using the control apparatus which concerns on this Embodiment, and driving a motor with many turns. It is a graph which shows performing field-weakening control after the output voltage of an inverter reaches | attains the maximum value in an overmodulation control area | region using the control apparatus which concerns on this Embodiment. The waveform at the time of starting of a permanent magnet type synchronous motor in a motor normal state is shown. The waveform at the time of starting of a permanent magnet type synchronous motor in a motor step-out state is shown. The waveform at the time of starting of a permanent magnet type | mold synchronous motor in the case of a motor normal state and a magnetic flux amount in a normal state is shown. The waveform at the time of starting of a permanent magnet type synchronous motor when a motor is in a normal state and the amount of magnetic flux is slightly increased from a normal state is shown. The waveform at the time of starting of a permanent-magnet-type synchronous motor in the case of a motor normal state and when the amount of magnetic flux is further increased from the amount of magnetic flux of FIG. It is a block diagram which shows the system configuration | structure of the conventional control apparatus disclosed by patent document 1. FIG. It is a graph which shows the relationship with the output voltage of a voltage command value inverter in the conventional control apparatus.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Control apparatus 2 ... Speed command setting means 3 ... Voltage command value setting means 4 ... Voltage command value correction means 5 ... 2-phase / 3-phase conversion means 6 ... PWM waveform means 7 ... Inverter 8 ... Permanent magnet type synchronous motor (motor) )
9 ... 3-phase / 2-phase conversion means 10 ... Power factor angle setting means

Claims (4)

Voltage command value setting means for setting a voltage command value II proportional to the frequency of the motor on two axes of the δ axis and the γ axis of the rotation rectangular coordinate system;
2-phase / 3-phase conversion means for coordinate-converting the voltage command value II of the two axes into a voltage command value III of three phases;
An inverter that converts the voltage command value III of the three phases into power and applies it to the motor;
Means for feeding back the terminal current of the motor to the voltage command value II without detecting the field direction of the motor;
Power factor angle determining means for determining a power factor angle from the current by the feedback;
Power factor angle adjusting means for adjusting the power factor angle to a rotation angle used when converting the motor terminal current obtained in three phases into orthogonal coordinates;
Correction means for correcting the voltage command value II so that the voltage command value III and the output voltage from the inverter are proportional to each other in the overmodulation control region;
The correction means includes
Only one of the voltage command values v ^* _δ and v ^* _γ set by the voltage command value setting unit is corrected between the voltage command value setting unit and the two-phase / three-phase conversion unit. Control device for permanent magnet type synchronous motor.   The control apparatus for a permanent magnet type synchronous motor according to claim 1, further comprising PWM waveform means for converting the voltage command value III of three phases into a pulse width modulation waveform. A step (a) of setting a voltage command value II proportional to the frequency of the motor in two axes of the δ axis and the γ axis of the rotation rectangular coordinate system;
A step (b) of converting the voltage command value II of the two axes into a voltage command value III of three phases;
(C) applying power to the motor by converting the voltage command value III of three phases by an inverter;
Feedback the terminal current of the motor without detecting the field direction of the motor (d);
Determining a power factor angle from the current from the feedback step (d) (e);
(F) adding or subtracting the power factor angle to a rotation angle used when converting the motor terminal current obtained in three phases into orthogonal coordinates;
In the overmodulation control region, so that the output voltage from said voltage command value III and inverter proportional, have a, and step (g) correcting the voltage command value II,
In step (g),
A permanent magnet type synchronous motor characterized in that only one of the voltage command values v ^* _δ and v ^* _γ set in step (a) is corrected between step (a) and step (b). Control method. The method for controlling a permanent magnet type synchronous motor according to claim 3 , further comprising a step ( h ) of converting the voltage command value III of three phases into a pulse width modulation waveform and supplying the pulse width modulation waveform to the inverter.

Images