JP5228435B2 - Inverter control device and control method thereof - Google Patents

Description

  The present invention relates to an inverter control device for driving a permanent magnet type synchronous motor having a permanent magnet as a rotor without using a speed / position detector, and a control method therefor.

An inverter control device that drives a conventional permanent magnet type synchronous motor without using a speed / position detector is a method using an induced voltage of the synchronous motor, and a γ axis and π / 2 [rad] which are designated magnetic axes of the motor. in a coordinate system consisting of advanced δ-axis, the distribution gain K ₂ made larger as the absolute value of the smaller distribution gain K ₁ and the rotor speed command omega _RREF greater the absolute value of the rotor speed command omega _RREF becomes large increases In addition, K ₁ is added to the rotor speed command ω _RREF, and the rotor speed estimated value ω _RP obtained from the induced voltage of the synchronous motor or the estimated voltage estimated value is multiplied by K _2. The rotor speed ω _Rγ of the −δ axis is determined (see, for example, Patent Document 1). In addition, as a method of using the saliency of the synchronous motor, a high frequency signal is superimposed on the estimated magnetic flux axis, and an error signal of the magnetic flux position is extracted from a voltage and current detection signal having the same frequency component as the high frequency signal to determine the magnitude and position of the magnetic flux. Some estimate and perform stable control over the entire drive range including the zero speed and zero output frequency regions (see, for example, Patent Document 2). In addition, there is a type in which q-axis inductance L _q that changes during operation is identified using a transient identification model, and is used in a position estimator that estimates speed and position to improve position estimation accuracy (for example, non-patent) Reference 1).

  Generally, an induced voltage model of a synchronous motor used when estimating a speed and a position using an induced voltage generated by the synchronous motor is constructed in a γ-δ coordinate system, and the model is expressed by the equations (1) and (2). Indicated by

  here,

  Further, the phase difference Δθ between the dq axis and the γ-δ axis can be obtained by the equation (3).

Here, it is known that the d-axis inductance L _d hardly changes with respect to the load, but the q-axis inductance L _q changes greatly.

In FIG. 4, 101 is a distributed gain generator, 102 is a speed controller, 103 is a γ-axis current command generator, 104 is a δ-axis current controller, 105 is a γ-axis current controller, 106 is a vector control circuit, 107 is an inverter circuit, 108 is a synchronous motor, 109 is a three-phase to two-phase converter, 110 is a γ-δ axis current / induced voltage estimator, 111 is a proportional control speed calculator, 112 is a magnetic axis calculator, 113 is integral, and magnetic axis rotation Speed calculator.
The distribution gain generator 101 creates distribution gains K ₁ and K ₂ in response to the speed command ω _RREF , and the γ-axis command i _γREF includes i _γREFL : low-speed command, i _γREFH : high-speed command and K ₁ and K ₂ as γ This is input to the shaft current command generator 103 and created. The speed controller 102 outputs a δ-axis current i _δREF (k + 1), outputs δ and γ-axis voltage commands v _δREF and v _γREF from the δ-axis current controller 104 and the γ-axis current controller 105 and inputs them to the vector control circuit 106. The The vector control circuit 106 receives θ _est (k + 1) from the magnetic axis calculator 112 and inputs the magnitude and position angle of the voltage to the inverter circuit 107. The inverter circuit 107 supplies current to the synchronous motor 108. Iu and iw are input to the three-phase to two-phase converter 109 to create a rotor coordinate system γ-δ axis current. This is input to the γ-δ axis current / induced voltage estimator 110 and voltage commands v _δREF and v _γREF are also input. The induced voltage estimated values ε _δest (k + 1) and ε _γest (k + 1) are input to the proportional control speed calculator 111 by the γ-δ axis current / induced voltage estimator 110, and the command control speed ω _RTest is output. On the other hand, i _γest (k + 1) and i _δest (k + 1) are input to the δ-axis current controller 104 and the γ-axis current controller 105 to create a voltage command. ω _RTest , ω _RREF , and distribution gains K ₁ and K ₂ are input to the integral / magnetic axis rotation speed calculator 113 to output the rotation speed ω _{Rγest of the} magnetic axis.
The induced voltage estimator 110 _calculates the induced voltage values ε _γest and ε _δest by the equation (4) and the speed estimated value ω _RTest by the equation (5).

Where ω _{Rφest is a} temporary rotor speed estimation value, K _{p is a} proportional gain, T _{S is a} calculation period, and K _{1 to} K _{8 are} observer gains.
Thus, the conventional inverter controller estimates and calculates the induced voltage of the synchronous motor, and estimates and calculates the rotor speed and magnetic pole position of the synchronous motor based on the induced voltage model formula built on the γ-δ axes. is there.
JP-A-10-174499 (page 7, FIG. 2) Japanese Patent Laying-Open No. 2003-299381 (page 7, FIG. 1) 2006 IEEJ Industrial Application Conference, No. 1-106, p. 533-536 (2006-3)

An inverter control device for driving a conventional permanent magnet type synchronous motor without using a speed / position detector includes a stator resistance R, a d-axis inductance L _d , a q-axis inductance L _q and an induced voltage included in the induced voltage model formula. If there is an error in the motor parameter such as the constant φ, there is an error in the estimated induced voltage value, resulting in an error in the estimated rotor speed value or vibration of the motor. Further, when a high frequency signal is superimposed, torque vibration is generated by the superimposed signal, and there is a problem that a switching device is required for the adaptive adjuster at the low speed and the high speed depending on the speed. Further, when the transient identification model is used, there is a problem that torque pulsation occurs because a q-axis current with an instantaneous change is required as the identification signal.
The present invention has been made in view of such problems, and an inverter based on position sensorless control that can suppress a position estimation error even when the motor parameter changes and can stably drive without torque vibration or torque pulsation. An object is to provide a control device.

In order to solve the above problem, the present invention is configured as follows.
An inverter control apparatus according to the present invention includes an inverter unit that supplies a voltage to a permanent magnet type synchronous motor having a permanent magnet as a rotor, a current detector that detects a motor current flowing in the motor, and a magnetic axis of the motor. The phase difference between the dq axis composed of q advanced by π / 2 [rad] from the d axis and the γ-δ axis composed of γ advanced from the specified magnetic axis of the motor by γ and π advanced by π / 2 [rad] from the γ axis. In an inverter control apparatus provided with a position speed estimator for estimating and calculating the magnetic pole position and rotor speed of the electric motor so that Δθ is zero, it is delayed or advanced by π / 4 [rad] with respect to the γ-δ axis . set the gamma _m - [delta _m axis, provided the induced voltage model of the electric motor to the gamma _m - [delta _m-axis, using a voltage command value and the current detection value of gamma _m - [delta _m-axis, gamma _m - [delta _m-axis and the induced voltage estimator for calculating the EMF value of the electric motor estimated the And it is to include an axial error calculator for calculating the phase difference Δθ by the inverse tangent function calculation using the ratio of [delta] _m-axis component to the gamma _m-axis component of the induced voltage value.
In addition, the present invention is, before Symbol induced voltage model,

  However,

The approximate induced voltage model indicated by
Further, the present invention is pre-Symbol induced voltage estimator, the voltage command value of gamma _m - [delta _m coordinate system, using the current detection value and the rotor speed, estimates and calculates the front Ki誘 electromotive voltage It is a state observer.
Further, the present invention is pre-Symbol error calculator is an input the induced voltage value,

Is used to calculate the phase difference Δθ.

In order to solve the above problem, the present invention is as follows.
The present invention relates to an inverter for supplying voltage to a permanent magnet type synchronous motor having a permanent magnet as a rotor, a current detector for detecting a motor current flowing in the motor, and a magnetic axis of the motor from d and d axes. The phase difference Δθ between the dq axis composed of q advanced by π / 2 [rad] and the specified magnetic axis of the motor is γ, and the γ-δ axis composed of δ advanced from the γ axis by π / 2 [rad] is zero. In the control method of the inverter control apparatus including the position speed estimator for estimating and calculating the magnetic pole position and the rotor speed of the electric motor, the γ delayed or advanced by π / 4 [rad] with respect to the γ-δ axis set _m - [delta _m axis, provided the induced voltage model of the electric motor to the gamma _m - [delta _m-axis, with said voltage command value and the current detection value, the motor induced induction of the gamma _m - [delta _m-axis The voltage value is estimated and calculated, and δ _{m with} respect to the γ _m- axis component of the induced voltage value The procedure of calculating the phase difference Δθ by the arc tangent function calculation using the ratio of the axis components was taken.

According to the present invention, it is possible to suppress a position estimation error and realize a stable motor drive without limiting a control target or an operation state.
Further, according to the present invention, the phase difference Δθ can be calculated from the detected current value by a simple calculation.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

FIG. 1 is a control block diagram of a position sensorless inverter control apparatus of the present invention. In the figure, a speed controller 1 inputs a rotor speed command value ω ^* and a rotor speed estimated value ω ^ to be described later so that the deviation becomes a rotational speed as commanded by proportional / integral (PI) control. A torque current command i _δ ^* is created. gamma-axis and [delta]-axis current controller 2 and 3, the current command value i _gamma ^* of the respective ^axes, i _[delta] ^* and the current value i _gamma described later, as input i _[delta], respectively, as a current as commanded voltage Commands v _γ ^* and v _δ ^* are created. The vector control circuit 4 receives voltage commands v _γ ^* and v _δ ^* in the γ-δ coordinate system and a magnetic pole estimated position θ ^ described later, and obtains a voltage amplitude value and a voltage phase to be applied to the synchronous motor 6. The inverter unit 5 receives the voltage amplitude value and the voltage phase, converts them into voltage commands v _u ^* , v _v ^* , and v _w ^* for each phase and applies them to the synchronous motor 6. The synchronous motor 6 is connected to the inverter unit 5 as a control target. The current detector 13 detects phase currents i _u , i _v , and i _w .

The coordinate converter 7 advances the detected phase currents i _u , i _v , and i _w by α axis which is one phase of the stator of the synchronous motor 6 and π / 2 [rad] from the α axis. The current values i _α and i _β in the α-β coordinate system composed of the β axis are converted. The coordinate converter 8 converts the current values i _α and i _β into current values i _γ and i _δ in the γ-δ coordinate system. In the conversion, an estimated value θ ^ of the magnetic pole position described later is used. Coordinate converter 9 is -π / 4 [rad] transducer rotates, the current value of the gamma-[delta] coordinate system i _gamma, the current value of the i _δ γ _m -δ _m coordinate system i _{γ ^m,} i _δ ^The voltage command values v _γ ^* and v _δ ^* are converted into v _γ ^m and v _δ ^m . The induced voltage estimator 10 includes an induced voltage model of the synchronous motor 6 configured on the γ _m -δ _m axis that is delayed by π / 4 [rad] with respect to the γ-δ axis, and includes voltage command values v _γ ^m , v The current value and the induced voltage estimated value of the synchronous motor 6 are calculated using _δ ^m and the current detection values i _γ ^m and i _δ ^m . The axis error calculator 11 calculates the phase difference Δθ using the induced voltage estimated value estimated by the induced voltage estimator 10. Further, the position / speed estimator 12 receives the phase difference Δθ, and calculates and outputs the rotor speed estimated value ω ^ and the magnetic pole position estimated value θ ^.
The operations of the induced voltage estimator 10, the axis error calculator 11, and the position / speed estimator 12 will be described in detail later.
The present invention is different from the prior art in that a coordinate converter 9 is added and an induced voltage estimator 10 and an axis error calculator 11 are further modified.

Next, the control block diagram shown in FIG. 1 will be described.
The coordinate converter 9 is a converter that converts current values i _γ and i _δ in the γ-δ coordinate system into current values i _γ ^m and i _δ ^m in the coordinate system γ _m -δ _m coordinate system. The transformation matrix T ₄₅ is shown in the equation (6). Further, the voltage command values v _γ ^* and v _δ ^* in the γ-δ coordinate system are also used for the voltage command values v _{γ in the} coordinate system γ _m -δ _m coordinate system using the same transformation matrix T _45. ^m and v _δ ^m are obtained.

The induced voltage estimator 10 calculates current values i _γ ^m and i _δ ^m and voltage command values v _γ ^m and v _δ in the γ _m -δ _m coordinate system at time t = k · T _s (T _s : calculation period). ^m is input, the current values i _γ ^m ^ and i _δ ^m ^ in the γ _m -δ _m coordinate system at t = (k + 1) · T _{s according to the} equation (7), the induced voltage according to the equation (8) ε _γ ^m ^ and ε _δ ^m ^ are estimated and calculated.

Here, K ₁₁ , K ₁₂ , K ₂₁ , K ₂₂ , K ₃₁ , K ₃₂ , K ₄₁ , K ₄₂ : Observer gain.

The induced voltages ε _γ ^m and ε _δ ^m are called extended induced voltages and are expressed by the equation (10).

The axis error calculator 11 receives the estimated induced voltages ε _γ ^m ^ and ε _δ ^m ^, performs an arc tangent function calculation using ε _γ ^m and ε _δ ^m shown in equation (11), and d− The phase difference Δθ between the q axis and the γ-δ axis is calculated.

FIG. 2 is a detailed view of the position / speed estimator 12. In the figure, reference numeral 21 denotes a speed estimator composed of proportional / integral elements. When the phase difference Δθ calculated by the above equation (11) is inputted, the phase difference Δθ is controlled to be 0 and rotated. The child speed estimated value ω ^ is output. A magnetic pole position calculator 22 multiplies the rotor speed estimated value ω ^ by the calculation period T _s to calculate a phase update amount per calculation period T _s and integrates it to obtain an estimated value θ ^ of the magnetic pole position. Output.
Thus, the induced voltage model of the synchronous motor is configured on the γ _m -δ _m axis delayed by π / 4 [rad] from the γ-δ axis, and the voltage command values v _γ ^m , v _δ ^m and the current detection value i _{γ are formed.} ^The induced voltage estimated value is calculated using ^{m 1} and i _δ ^m to estimate and calculate the phase difference Δθ, the rotor speed estimated value ω ^, and the magnetic pole position estimated value θ ^.

Next, how the position estimation error can be suppressed by estimating the position on the γ _m -δ _m coordinate even if the motor parameter changes will be described.
FIG. 3 is an impedance locus diagram in light load and heavy load in the γ-δ coordinate system and the γ _m -δ _m coordinate system. In the figure, 31 is an impedance locus at light load, 32 is an impedance locus at heavy load, 33 is a permanent magnet rotor, 34 is a change in impedance due to load fluctuation in the δ axis, and 35 is in the δ _m axis. It represents the change in impedance due to load fluctuation.
Here, the relative relationship of each coordinate system is summarized. The α-β coordinate system is a fixed coordinate system based on one phase of the stator winding, and the dq coordinate system is a coordinate system in a phase that is advanced by θ from the α-β coordinate system, and the rotor magnetic axis is referenced. It is said. The γ-δ coordinate system is delayed by Δθ with respect to the dq coordinate system, and the γ _m -δ _m coordinate system is delayed by π / 4 [rad] with respect to the γ-δ coordinate system.
In FIG. 3, the impedance locus of a motor having saliency, such as an embedded permanent magnet motor (IPM), is an ellipse 31 having a large diameter in a direction orthogonal to the magnetic pole direction. However, the inductance of a motor with saliency decreases as the current flowing through the motor increases, and the q-axis inductance is much smaller than the d-axis inductance. For this reason, assuming that the stator resistance (R) is constant, the impedance locus that was elliptical 31 at the time of light load approaches the circle 32.

For an electric motor having such an impedance locus, when an induced voltage model is constructed in the γ-δ coordinate system as in the prior art, when calculating the phase difference Δθ, the parameter error of the q-axis inductance L _q is between the numerator and the denominator. The change is not the same ratio but an imbalance. For this reason, if the phase difference Δθ is calculated by the equation (3) described in the prior art, a phase error occurs.

However, when the induced voltage model is constructed in the γ _m -δ _m coordinate system as in the present invention, i _γ ^m and i _δ ^m are substantially equal, and therefore the parameter error of the q-axis inductance L _q is the estimated induced voltage ε _It has the same influence on _γ ^m and ε _δ ^m . Therefore, parameter error of the q-axis inductance L _q becomes possible to suppress the error of the phase difference Δθ, which does not give an arctan calculation in the influence of the right side, to the calculation of (11) (number 12).
In other words, when an induced voltage model is constructed in the γ _m -δ _m coordinate system, the inductance change 35 on the δ _m axis can be suppressed smaller than the inductance change 34 on the δ axis due to the load fluctuation. The same effect as suppressing the parameter error of the inductances L _d and L _{q in} the induced voltage model is obtained.

As for the induced voltage constant φ, if an induced voltage model is constructed in the γ-δ coordinate system, the parameter error of the induced voltage constant φ affects only ε _δ , but if constructed in the γ _m -δ _m coordinate system, π Since the sin and cos values of / 4 [rad] are equal, the parameter error of the induced voltage constant φ is equally affected by ε _γ ^m and ε _δ ^m when the phase difference Δθ in the normal control state is near 0. Thus, no error occurs in the calculated phase difference Δθ.

As described above, the position estimation can be performed while suppressing the deterioration of the estimation performance of the phase difference Δθ with respect to the change of the q-axis inductance L _q and the induced voltage constant φ, and furthermore, the superposition of the high frequency signal and the identification signal are input. Therefore, it is possible to realize a stable motor drive without torque vibration.

The control block diagram of the inverter control apparatus which shows 1st Example of this invention Detailed view of the position speed estimator 12 of the present invention Impedance locus diagrams at light load and heavy load in γ-δ coordinate system and γ _m -δ _m coordinate system for explaining the operation of the present invention Control block diagram of a conventional inverter control device

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Speed controller 2 d-axis current controller 3 q-axis current controller 4 Vector control circuit 5 Inverter part 6 Synchronous motor 7, 8, 9 Coordinate converter 10 Induced voltage estimator 11 Axis error calculator 12 Position speed estimator 13 Current detector 21 Speed estimator 22 Magnetic pole position calculator 31 Light load impedance locus 32 Heavy load impedance locus 33 Permanent magnet 34 δ-axis impedance change due to load 35 δ _m- axis impedance change due to load 101 Distribution gain generator 102 Speed controller 103 γ-axis current command generator 104 δ-axis current controller 105 γ-axis current controller 106 Vector control circuit 107 Inverter circuit 108 Synchronous motor 109 Three-phase two-phase converter 110 γ-δ-axis current / induced voltage estimator 111 Proportional control Speed calculator 112 Magnetic axis calculator 113 Integration, magnetic axis rotation Speed calculator

Claims (5)

An inverter unit for supplying a voltage to a permanent magnet type synchronous motor having a permanent magnet as a rotor, a current detector for detecting a motor current flowing through the motor, and a magnetic axis of the motor from d and d axis to π / 2 [ rad] The phase difference Δθ between the dq axis composed of advanced q and the designated magnetic axis of the motor is γ, and the γ-δ axis composed of δ advanced by π / 2 [rad] from the γ axis is set to zero. In an inverter control device comprising a position speed estimator for estimating and calculating the magnetic pole position and rotor speed of the motor,
Set π / 4 [rad] delayed or advanced gamma _m - [delta _m-axis with respect to gamma-[delta] axes, it provided the induced voltage model of the electric motor to the gamma _m - [delta _m-axis, the gamma _m - [delta _m-axis and the induced voltage estimator for estimating the EMF value of the motor of gamma _m - [delta _m-axis by using the voltage command value and the current detection value,
Inverter control apparatus characterized by comprising an axial error calculator for calculating the phase difference Δθ by the inverse tangent function calculation using the ratio of [delta] _m-axis component to the gamma _m-axis component of the induced voltage value. The induced voltage model is
The inverter control device according to claim 1, wherein the inverter control device is an approximated induced voltage model represented by: The induced voltage estimator, the voltage command value of gamma _m - [delta _m coordinate system, using the current detection value and the rotor speed, and wherein the pre-Symbol induced voltage is a state observer for estimating The inverter control device according to claim 1 or 2. The axis error calculator has the induced voltage value as input,
The inverter control apparatus according to claim 1, wherein the phase difference Δθ is calculated using An inverter unit for supplying a voltage to a permanent magnet type synchronous motor having a permanent magnet as a rotor, a current detector for detecting a motor current flowing through the motor, and a magnetic axis of the motor from d and d axis to π / 2 [ rad] The phase difference Δθ between the dq axis composed of advanced q and the designated magnetic axis of the motor is γ, and the γ-δ axis composed of δ advanced by π / 2 [rad] from the γ axis is set to zero. In the control method of the inverter control device including the position speed estimator for estimating and calculating the magnetic pole position and the rotor speed of the electric motor,
Set π / 4 [rad] delayed or advanced gamma _m - [delta _m-axis with respect to gamma-[delta] axes, it provided the induced voltage model of the electric motor to the gamma _m - [delta _m-axis, the gamma _m - [delta _m-axis using voltage command value及beauty current detection value, and estimates and calculates the EMF value of the motor of gamma _m - [delta _m-axis,
Control method for an inverter control device, characterized in that the treatment with the procedure of calculating the phase difference Δθ by the inverse tangent function calculation using the ratio of [delta] _m-axis component to the gamma _m-axis component of the induced voltage value.