JP3795845B2 - Array antenna control method, reception signal signal-to-noise ratio calculation method, and radio receiver adaptive control method - Google Patents

Description

【００２３】
エスパアンテナ装置１００の構造は巡回的な対称性を有しているため、この行列Ｚの４９個の要素のうち独立な要素は６個の要素となる。これらはその物理的意味からそれぞれ以下のように呼ばれるべき複素パラメータである。
【００２４】
【表１】
―――――――――――――――――――――――――――――――――――
ｚ_００：励振素子の自己入力インピーダンス
ｚ_０１：励振素子と非励振素子との間の結合インピーダンス
ｚ_１１：非励振素子の自己入力インピーダンス
ｚ_１２：互いに隣接する２つの非励振素子間の結合インピーダンス
ｚ_１３：次に隣接する（１つ間をおいて隣接する）２つの非励振素子間の結合インピーダンス
ｚ_１４：互いに対向する２つの非励振素子間の結合インピーダンス
―――――――――――――――――――――――――――――――――――
【００２５】
なお、後述する実施例で用いた各インピーダンス値は以下の通りである。
（ａ）ｚ_００＝＋５２．０−５．７ｊ
（ｂ）ｚ_０１＝＋２３．９−２９．２ｊ
（ｃ）ｚ_１１＝＋６４．０−３．４ｊ
（ｄ）ｚ_２１＝＋２９．７−２９．８ｊ
（ｅ）ｚ_３１＝−１３．９−２７．６ｊ
（ｆ）ｚ_４１＝−２６．０−１６．７ｊ
ここで、インピーダンス値の単位は全てΩである。バラクタダイオードであるリアクタンス素子１２−１乃至１２−６のリアクタンス値をｘ_１，ｘ_２，…，ｘ_６とすると、エスパアンテナ装置１００の指向性（アレーファクタ）Ｄ_ａ（θ，φ）は次式で表される（例えば、従来技術文献２参照）。
【００２６】
【数２】
Ｄ_ａ（θ，φ）
＝ａ（θ，φ）^Ｔｉ（ｘ_１，ｘ_２，…，ｘ_６）
【００２７】
ここで、ａ（θ，φ）は、エスパアンテナ装置１００の位相中心を中央の非励振素子Ａ０にとった場合のステアリングベクトルであり、仰角θと方位角φの関数として次式で表される。
【００２８】
【数３】

【００２９】
ここで、ｄは半径ｒに等しい素子間隔であり、βは自由空間中の伝播定数である。また、ｉ（ｘ_１，ｘ_２，…，ｘ_６）はエスパアンテナの等価ウエイトベクトルであり、次式で表される。
【数４】
ｉ（ｘ_１，ｘ_２，…，ｘ_６）
＝Ｚ^−１（ｖ_ｓｕ_０−Ｘｉ）
＝ｖ_Ｓ（Ｚ＋Ｘ）^−１ｕ_０
ここで、ｕ_０は次式で表される単位ベクトルである。
【数５】
ｕ_０＝［１，０，…，０］^Ｔ
また、Ｘは、ＲＦ受信機の入力インピーダンスｚ_ｓと可変リアクタンス素子１２−１乃至１２−６のリアクタンス値を成分とする次式の対角行列であるリアクタンス行列である。
【数６】
Ｘ＝ｄｉａｇ［ｚ_ｓ，ｊｘ_１，ｊｘ_２，…，ｊｘ_６］
【００３０】
従って、エスパアンテナ装置１００の指向性（アレーファクタ）Ｄ_ａ（θ，φ）は次式で表される。
【数７】
Ｄ_ａ（θ，φ）
＝ｚ_ｓａ(θ，φ)^Ｔ｛Ｚ＋ｄｉａｇ[ｚ_ｓ，ｊｘ_１，ｊｘ_２，・・・，ｊｘ_６]｝^−１ｕ_０
【００３１】
複数の信号波が到来する場合にはそれらの信号波形を成分とするベクトル
【数８】
ｓ（ｔ）＝［ｓ_１（ｔ），ｓ_２（ｔ），…，ｓ_Ｋ（ｔ）］
を定義する。ここで、Ｋは信号の数である。これらを同時に受信した場合のエスパアンテナ装置１００の出力信号は次式で表される。
【数９】
ｙ（ｔ）＝ｉ（ｘ_１，ｘ_２，…，ｘ_６）^ＴＡ（θ，Φ）ｓ（ｔ）＋ｎ（ｔ）
ここで、Ａ（θ，Φ）はアレーマニホールドであり、次式で表される。
【数１０】
Ａ（Θ，Φ）
＝［ａ（θ_１，φ_１），ａ（θ_２，φ_２），…，ａ（θ_Ｋ，φ_Ｋ）］
ここで、
【数１１】
Θ＝｛θ_１，θ_２，…，θ_Ｋ｝
【数１２】
Φ＝｛φ_１，φ_２，…，φ_Ｋ｝
であり、ｎ（ｔ）は加算的雑音波形である。
【００３２】
従って、エスパアンテナ装置１００の受信信号ｙ（ｔ）は次式で表される。
【数１３】

【００３３】
ここで、ｓ_ｋ（ｔ）とθ_ｋ，φ_ｋはそれぞれ、第ｋ番目の信号の時間波形と、到来方向である。
【００３４】
次いで、本実施形態で用いる「ブラインド適応ビーム形成」について説明する。適応ビーム形成の目的は上記数１３で導出したアンテナ受信出力信号ｙ（ｔ）に含まれる信号対干渉雑音の電力比ＳＩＮＲを実質的に最大化することである。ブラインド制御とは所望波に含まれる信号情報を全く参照することなくアンテナ可変パラメータ（一般的にはウエイトベクトル：ここでは、可変リアクタンス素子１２−１乃至１２−６のリアクタンス値）を更新することである。
【００３５】
適応的にビームを形成するためには、（１）送信パケットのヘッダに参照信号を含ませておき、（２）受信側で予めこの参照信号系列を知っておき、（３）参照信号の同期タイミングを検出して、（４）アレーの重み係数をトレーニングする、というプロセスが通常用いられる。例えばエスパアンテナ装置１００の適応ビーム形成法として受信信号と参照信号との相互相関係数を最大化するアルゴリズム「ＭＣＣＣ：Maximum Cross Correlation Coefficient」がある（例えば、従来技術文献３「神谷ほか，“エスパアンテナの基本検討：適応制御に基づくＳＩＲ特性の統計的規範”，電子情報通信学会技術報告，ＡＰ２０００−１７５，ＳＡＮＥ２０００−１５６，ｐｐ．１７−２４，２００１年１月」参照。）。これに対して、ブラインド適応ビーム形成とは参照信号なしで適応的にビームを形成する機能であり、上記（１）〜（３）のプロセスを省略することができる。
【００３６】
本実施形態においては、ｍ相ＰＳＫ変調信号に特有の性質に着目し、これを利用したブラインド規範を提案する。ここで着目する性質とは、「ｍ相ＰＳＫ変調信号は変調データに関わらずこれをｍ乗すると一定の複素数値となる」ことである。通信路で雑音あるいは干渉を被るとこの一定の複素数値からのゆらぎが受信側で観測される。このゆらぎが小さいほど所望信号が高い純度で抽出できていると考えられる。そこで、上述のごとく導出した受信アンテナの出力信号の第ｍ次モーメントを用いた、次式の規範関数とすることを提案する。
【００３７】
【数１４】

【００３８】
ここで、Ｅ［・］は引数・のアンサンブル平均（所定の時間における平均値）を表す。分母はｍ乗された信号の平均電力を表している。規範関数Ｊ_ｍ（ｙ（ｔ））の物理的解釈は詳細後述する。この規範関数の利点は、上述の「一定の複素数値」を含んでいないことである。すなわち、この値を受信側で予め知っている必要が無い。このことは、アンテナ及び受信回路系の絶対利得や固定的な位相回転量に左右されないことを意味しており、実際の無線システムに用いる際の重要な利点である。
【００３９】
次いで、上記規範関数を用いた適応ビーム形成について説明する。「適応ビーム形成」とは、上記数１３で導出した、エスパアンテナ装置１００の受信信号ｙ（ｔ）に含まれる信号対干渉雑音電力比ＳＩＮＲ＝Ｓ／（Ｎ＋Ｉ）を実質的に最大化するようにアンテナ可変パラメータ（エスパアンテナ装置１００では各可変リアクタンス素子１２−１乃至１２−６のリアクタンス値）を更新することである。上述の規範関数に基づいてリアクタンス値を反復更新することにより、アンテナ指向性が出力ＳＩＮＲ最大となる最適ビームパターンすなわち所望波の方向へメインビームが形成され干渉波の方向へヌルが形成されるビームパターンとなる。
【００４０】
すなわち、上記規範関数Ｊには目標値Ｃが含まれておらず、受信信号ｙ（ｔ）のみであって、しかも受信信号のｍ乗｛（ｙ（ｔ））^ｍ｝を用いて表されて記述されている。この場合、目標値が未知の状態で制御できることが大きなメリットである。この規範に基づいてリアクタンス値を、例えば最急勾配法などの非線形計画法における反復的な数値解法を用いて反復更新することにより、アンテナ出力の信号対干渉雑音電力比（ＳＩＮＲ）が実質的に最大となるように、すなわち、エスパアンテナ装置１００の主ビームを所望波の方向に向けかつ干渉波の方向にヌルを向けるように最適ビームが形成される。
【００４１】
次いで、最急勾配法を用いたアンテナビームの適応制御について説明する。最急勾配法を用いるときの可変リアクタンス素子１２−１乃至１２−６のリアクタンス値のセット（リアクタンスベクトル）ｘに対する漸化式は次式で表される。
【００４２】
【数１５】
ｘ（ｎ＋１）＝ｘ（ｎ）＋μ∇Ｊｎ
【数１６】

【００４３】
ここで、ｎはｘの更新の次数、パラメータμは試行錯誤的に定められるステップサイズである。ここで、最急勾配法は、最急降下法を含む方法の概念であるが、本実施形態では、規範関数の値を実質的に最大するように最適解を求める方法を用いる。
【００４４】
さらに、最急勾配法による具体的な、最適解を求める手順について説明する。上記数１５を用いた最急勾配法によって規範関数Ｊｎを可能な限り大きくするような良好なリアクタンスベクトルｘを発見するためには、以下の手順を用いる。
（ｉ）最初に、反復数パラメータｎ（すなわち、ｎ回目の反復）を１に設定し、予め決められたリアクタンスベクトルの初期値ｘ（１）（例えば、当該エスパアンテナ装置１００をオムニアンテナに設定にするときのリアクタンスベクトル）によって処理を開始する。
（ｉｉ）次いで、この初期値（ｎ＝１のとき）又は現在の推定値（ｎ≧２のとき）を使用して、反復数パラメータｎ（すなわち、ｎ回目の反復）における規範関数Ｊｎの勾配ベクトル∇Ｊｎを計算する。
（ｉｉｉ）勾配ベクトル∇Ｊｎの方向と同一の方向に初期値又は現在の推定値を変更することで、リアクタンスベクトルｘにおける次の推定値を計算する。
（ｉｖ）反復数パラメータｎを１だけインクリメントし、ステップ（ｉｉ）に戻って処理を繰り返す。この繰り返し処理は、リアクタンスベクトルｘが実質的に収束する反復数まで実行される。
【００４５】
図４は、図１の適応制御型コントローラ２０によって実行される、最急勾配法によるより具体的な適応制御処理を示すフローチャートである。
【００４６】
図４のステップＳ１において、まず、反復数パラメータｎを１にリセットし、リアクタンスベクトルｘ（１）にその初期値を設定挿入し、ステップＳ２において素子パラメータｋを１にリセットする。次いで、ステップＳ３において受信信号ｙ（ｔ）を測定し、ステップＳ４において数１４を用いて規範関数Ｊの値を計算し、Ｊ^（０）に設定挿入する。さらに、ステップＳ５においてリアクタンス値ｘ_ｋに所定の摂動値Δｘ_ｋを加算し、その加算値をリアクタンス値ｘ_ｋとして設定した後、ステップＳ６において受信信号ｙ（ｔ）を測定し、ステップＳ７において数１４を用いて規範関数Ｊの値を計算する。そして、ステップＳ８においてＪ−Ｊ^（０）の値を計算して∂Ｊｎ／∂ｘ_ｋに代入し、ステップＳ９においてリアクタンス値ｘ_ｋに所定の摂動値Δｘ_ｋを減算しその減算値をリアクタンス値ｘ_ｋとして設定することにより摂動前の値に戻した後、ステップＳ１０において素子パラメータｋはＫ（＝６）以上であるか否かが判断される。ステップＳ１０でＮＯであれば、ステップＳ１１で素子パラメータｋを１だけインクリメントしてステップＳ５に戻り、上述の処理を繰り返す。一方、ステップＳ１０でＹＥＳであるときは、ステップＳ１２において、数１５の漸化式を用いて、リアクタンスベクトルｘの次の推定値ｘ（ｎ＋１）を計算した後、ステップＳ１３において反復数パラメータｎが所定の反復数Ｎに到達しているか否かを判断し、ＮＯであれば、ステップＳ１４において反復数パラメータｎを１だけインクリメントした後、ステップＳ２からの処理を繰り返す。一方、ステップＳ１３でＹＥＳであるときは、十分に収束しているものと判断し、計算されたリアクタンスベクトルｘの値を有するリアクタンス値信号を可変リアクタンス素子１２−１乃至１２−６に出力して設定する。
【００４７】
以上説明したように、本実施形態によれば、適応制御型コントローラ２０は、エスパアンテナ装置１００の励振素子Ａ０によって受信された受信信号ｙ（ｔ）に基づいて、非線形計画法における反復的な数値解法である、例えば、最急勾配法を用いて、上記受信信号ｙ（ｔ）のみであって受信信号ｙ（ｔ）のｍ乗を用いて表された規範関数（数１４）の値が実質的に最大となるように、上記エスパアンテナ装置１００の主ビームを所望波の方向に向けかつ干渉波の方向にヌルを向けるための各可変リアクタンス素子１２−１乃至１２−６のリアクタンス値を計算して設定する。従って、参照信号を必要とせずに、アレーアンテナの指向性を所望波の方向に主ビームを向けかつ干渉波の方向にヌルを向けるように適応制御することができる。ここで、参照信号を必要としないので、当該装置の構成を簡単化できる。また、規範関数Ｊは受信信号ｙ（ｔ）のみで記述されているので、適応制御コントローラ２０の計算処理をきわめて簡単に実行できる。
【００４８】
以上の実施形態においては、６本の非励振素子Ａ１乃至Ａ６を用いているが、その本数は少なくとも複数本あれば、当該アレーアンテナ装置の指向特性を電子的に制御することができる。それに代わって、６個よりも多くの非励振素子を備えてもよい。また、非励振素子Ａ１乃至Ａ６の配置形状も上記の実施形態に限定されず、励振素子Ａ０から所定の距離だけ離れていればよい。すなわち、各非励振素子Ａ１乃至Ａ６に対する間隔は一定でなくてもよい。
【００４９】
以上の実施形態においては、最急勾配法を用いて各可変リアクタンス素子１２のリアクタンス値を計算しているが、本発明はこれに限らず、以下に示す順次ランダム法、ランダム法、高次元二分法などの非線形計画法における反復的な数値解法を用いてもよい。
【００５０】
なお、順次ランダム法においては、以下の手順を用いる。
（ｉ）最初に、反復数パラメータｎ（すなわち、ｎ回目の反復）を１に設定し、リアクタンスベクトルの所定の初期値ｘ（１）（例えば、当該エスパアンテナ装置１００をオムニアンテナに設定にするときのリアクタンスベクトル）によって処理を開始する。
（ｉｉ）次いで、この初期値（ｎ＝１のとき）又は現在の推定値（ｎ≧２のとき）を使用して、反復数パラメータｎ（すなわち、ｎ回目の反復）における推定値への加算値を所定の存在範囲内で乱数を発生させて計算する。
（ｉｉｉ）計算された加算値を上記推定値に加算することにより、リアクタンスベクトルにおける次の推定値を計算する。
（ｉｖ）反復数パラメータｎを１だけインクリメントし、ステップ（ｉｉ）に戻って処理を繰り返す。この繰り返し処理は、規範関数Ｊの値が所定のしきい値（例えば０．９）以上になるまで実行される。
【００５１】
また、ランダム法においては、以下の手順を用いる。
（ｉ）最初に、リアクタンスベクトルの所定の初期値ｘ（１）（例えば、当該エスパアンテナ装置１００をオムニアンテナに設定にするときのリアクタンスベクトル）によって処理を開始する。
（ｉｉ）次いで、この初期値を使用して、当該初期値への加算値を所定の存在範囲内で乱数を発生させて計算する。
（ｉｉｉ）計算された加算値を上記初期値に加算することにより、リアクタンスベクトルにおける推定値を計算する。
（ｉｖ）計算された推定値における規範関数Ｊの値が所定のしきい値（例えば０．９）以上であれば、当該推定値を設定すべきリアクタンスベクトルとするが、ＮＯであれば、ステップ（ｉｉ）に戻って処理を繰り返す。
【００５２】
さらに、高次元二分法においては、以下の手順を用いる。
（ｉ）最初に、反復数パラメータｎ（すなわち、ｎ回目の反復）を１に設定して処理を開始する。
（ｉｉ）次いで、リアクタンスベクトルの各リアクタンス値の所定の存在範囲（なお、２回目以降は、前に選択された推定値の存在範囲）を均等に二分し、二分された各存在範囲の平均値（各可変リアクタンス素子１２−１乃至１２−６に対して２つの平均値）を計算する。
（ｉｉｉ）この２つの平均値に対する規範関数Ｊの値を計算し、規範関数Ｊの値が大きい方を、リアクタンスベクトルにおける次の推定値とする。
（ｉｖ）反復数パラメータｎを１だけインクリメントし、ステップ（ｉｉ）に戻って処理を繰り返す。この繰り返し処理は、規範関数Ｊの値が所定のしきい値（例えば０．９）以上になるまで実行される。
【００５３】
以上の実施形態においては、規範関数Ｊを適応制御のためのリアクタンス値を求めるための規範関数とし、それを実質的に最大となるようにリアクタンスベクトルの最適解を計算しているが、本発明はこれに限らず、規範関数Ｊの逆数を適応制御のためのリアクタンス値を求めるための規範関数とし、それを実質的に最小となるようにリアクタンスベクトルの最適解を計算してもよい。
【００５４】
以上の実施形態においては、６本の非励振素子Ａ１乃至Ａ６と、それに対応した可変リアクタンス素子１２−１乃至１２−６とを備えているが、本発明はこれに限らず、少なくとも１つの非励振素子Ａ１と、それに対応した可変リアクタンス素子１２−１を備えてもよい。また、それぞれの個数は複数でもよい。
【００５５】
＜第２の実施形態＞
図５は、本発明に係る第２の実施形態であるアレーアンテナの制御装置の構成を示すブロック図である。
【００５６】
この実施形態では、アレーアンテナ５０の各アンテナ素子５１−１乃至５１−Ｐで受信した信号を、可変移相器５３−１乃至５３−Ｐと加算器である合成器５４によって構成されたＲＦ帯のＢＦＮ(Beam Forming Network)回路で合成する構成を採用し、このアレーアンテナの制御装置は、複数Ｐ個のアンテナ素子５１−１乃至５１−Ｐが互いに所定の間隔で配置されてなるアレーアンテナ５０（例えば、リニアアレーであり、２次元形状又は３次元形状で配置されてもよい。）のビームを制御するための適応制御型制御装置であり、適応制御型コントローラ６０を備えたことを特徴としている。ここで、送信された無線信号はｍ相ＰＳＫ変調され（ｍは２以上の整数である。）、適応制御型コントローラ６０は、合成後の受信信号に基づいて、非線形計画法における反復的な数値解法である、例えば、最急勾配法を用いて、上記受信信号ｙ（ｔ）のｍ乗を用いて表された規範関数（数１４）の値が実質的に最大となるように、当該アレーアンテナ５０の主ビームを所望波の方向に向けかつ干渉波の方向にヌルを向けるための可変移相器５３−１乃至５３−Ｐの移相量に対応する各移相制御電圧ｖ_ｐ（ｐ＝１，２，…，Ｐ）を計算して設定することを特徴としている。
【００５７】
以下、図５に示すアレーアンテナの制御装置の構成について説明する。図５において、複数Ｐ個のアンテナ素子５１−１乃至５１−Ｐが互いに所定の間隔で１直線上に並置されてなるアレーアンテナ５０によって無線信号が受信され、各アンテナ素子５１−１乃至５１−Ｐで受信された無線信号はそれぞれ、低雑音増幅器（ＬＮＡ）５２−１乃至５２−Ｐを介して可変移相器５３−１乃至５３−Ｐに入力される。各可変移相器５３−１乃至５３−Ｐはそれぞれ、入力される無線信号を、適応制御型コントローラ６０から出力される各移相制御電圧ｖ_ｐ（ｐ＝１，２，…，Ｐ）に対応した各移相量だけ移相した後、合成器５４に出力する。合成器５４は入力されるＰ個の無線信号を電力合成して、合成後の無線信号を、受信信号ｙ（ｔ）として図１の無線受信機１０と同様の構成を有する無線受信機１０に出力する。
【００５８】
次いで、無線受信機１０は、入力される合成された受信信号ｙ（ｔ）に基づいて、図１の無線受信機１０と同様に、互いに直交した受信信号からの２つのデジタルベースバンド信号を得て、適応制御型コントローラ６０に出力する。適応制御型コントローラ６０は、入力される２つのデジタルベースバンド信号に基づいて、非線形計画法における反復的な数値解法である、例えば、最急勾配法を用いて、図４の適応制御処理と同様の処理を実行することにより、上記受信信号ｙ（ｔ）のみであって受信信号ｙ（ｔ）のｍ乗を用いて表された規範関数（数１４）の値が実質的に最大となるように、当該アレーアンテナ５０の主ビームを所望波の方向に向けかつ干渉の方向にヌルを向けるための可変移相器５３−１乃至５３−Ｐの移相量に対応する各移相制御電圧ｖ_ｐ（ｐ＝１，２，…，Ｐ）を計算し、それを可変移相器５３−１乃至５３−Ｐに印加することにより対応する各移相量を設定する。
【００５９】
この実施形態に係る適応制御型コントローラ６０においても、第１の実施形態に係る適応制御型コントローラ２０と同様に、参照信号を必要とせずに、アレーアンテナの指向性を所望波の方向に主ビームを向けかつ干渉波の方向にヌルを向けるように適応制御することができる。ここで、参照信号を必要としないので、当該装置の構成を簡単化できる。また、規範関数Ｊは受信信号ｙ（ｔ）のみで記述されているので、適応制御コントローラ６０の計算処理をきわめて簡単に実行できる。
【００６０】
以上の実施形態においては、最急勾配法を用いて各可変移相器５３−１乃至５３−Ｐの移相量に対応する移相制御電圧ｖ_ｐを計算しているが、本発明はこれに限らず、上述の順次ランダム法、ランダム法、高次元二分法などの非線形計画法における反復的な数値解法を用いてもよい。また、規範関数Ｊはその逆数を用いてもよい。
【００６１】
図６は、図１のエスパアンテナ装置１００を用いて実行された、ブラインド適応ビーム形成のシミュレーションのフローを示す図である。このシミュレーションでは、上述の定式化モデルと同様に、励振素子Ａ０として半波長ダイポールアンテナを用い、非励振素子Ａ１乃至Ａ６として円形アレー配列された６本のダイポールアンテナを用いる。また、エスパアンテナ装置１００に到来する所望波と干渉波の到来方向は未知（適応制御）とし、トレーニング信号も用いないこと（ブラインド処理）とする。
【００６２】
図６のシミュレーションフローにおいては、干渉波ステアリングベクトル、所望波ステアリングベクトル、アンテナ構造のパラメータ、到来波信号、及び雑音に基づいて、ステップＳＳ１乃至ＳＳ５の処理を行うことにより当該アンテナビームの適応制御を行い、最後に、指向性アレーファクタ及び出力ＳＩＮＲを計算して出力する（ステップＳＳ６，ＳＳ７）。このステップＳＳ１乃至ＳＳ７の処理では、受信信号ｙ（ｔ）を受信し（ステップＳＳ１）、受信された受信信号ｙ（ｔ）に基づいて規範関数Ｊ_ｍ｛ｙ（ｔ）｝を計算し（ステップＳＳ２）、リアクタンス行列の更新を行って（ステップＳＳ３）リアクタンス行列を計算した後（ステップＳＳ４）、等価ウエイトベクトルを計算する（ステップＳＳ５）。そして、等価ウエイトベクトルから指向性アレーファクタを計算する（ステップＳＳ６）一方、受信信号ｙ（ｔ）と雑音ｎ（ｔ）とから出力ＳＩＮＲを計算している（ステップＳＳ７）。
【００６３】
このシミュレーションにおいては、エスパアンテナ装置１００に到来する所望波と干渉波の到来方向は未知（適応制御）とし、トレーニング信号も用いないこと（ブラインド処理）とする。所望波に加えて干渉波も同時に到来する環境でシミュレーションする。所望波、干渉波ともにＱＰＳＫランダム変調信号、雑音は加算的ガウス雑音とする。これら所望波、干渉波、雑音は相互に無相関とする。簡単のため伝送路における帯域制限フィルタ、遅延広がり、角度広がり、フェージング、ドップラ効果、同期誤差を全て無視する。この条件下で、上述した規範関数に基づいて６個の可変リアクタンス素子１２−１乃至１２−６のリアクタンス値を制御する。シミュレーションに用いることができるアンテナ構造パラメータの一例は、被制御素子数：６、素子間隔：全て１／４波長、各ダイポール半径：１／１００波長、素子の長さ方向の波長短縮率：０．９２６である。また、エスパアンテナ装置１００に接続されるＲＦ送受信機の内部インピーダンスはｚ_ｓ＝５０Ωとする。最適化アルゴリズムとしては、純粋ランダム探索法、最急勾配法、高次元二分法、順次ランダム法、回帰ステップ法、ハミルトン力学による方法などを用いることができる。
【００６４】
以上説明したように、本実施形態によれば、エスパアンテナ装置１００はハードウエア構成が簡単であるにもかかわらず、適切な規範と帰還制御によりブラインドビーム形成が可能であるということをｍ相ＰＳＫ波受信の場合について示した。
【００６５】
以上の実施形態においては、上記数１４の規範関数を用いているが、上記数１４における時間平均Ｅ（・）は、例えば周波数分割多重方式で送信されたデータ信号を一時に受信して並列処理する場合に、例えば１シンボルなどの所定の期間において複数のデータ信号についての平均値であってもよい。
【００６６】
＜第３の実施形態＞
図７は本発明に係る第３の実施形態であるアレーアンテナの制御装置の構成を示すブロック図である。本実施形態に係るアレーアンテナの制御装置は、図１の第１の実施形態に比較して以下の点が異なる。
（１）無線受信機１０に代えて、ｍ相ＰＳＫ信号を受信する無線受信機１０において波形等化器６−１，６−２をさらに備えた無線受信機１０ａを備えたこと。
（２）適応制御型コントローラ２０に代えて、適応制御型コントローラ２０において、上述の規範関数の値を計算し、計算された規範関数に基づいて上記規範関数と信号対雑音電力比との関係を示す式を用いて上記受信信号の信号対雑音電力比を計算し、上記計算された信号対雑音比が実質的に最大となるように波形等化器６−１，６−２を適応制御する適応制御型コントローラ２０ａを備えたこと。
【００６７】
図７の詳細説明する前に、規範関数である汎関数の定義、及び信号対雑音電力比の計算方法について以下に説明する。
【００６８】
無線受信機において最適受信するために、可変の信号波形等化器や信号フィルタやリニアライザを適応フィードバック制御するには、信号対雑音比の推定が有力な手段となる。特に、トレーニング参照信号や信号レプリカを用いない、図１や図７の無線受信装置においてはブラインド動作のＳＮ比推定技術を確立する必要がある。これまで、受信データの統計的期待値や分散を用いたブラインド推定関数が例えば、従来技術文献４「T. A. Summers et al., "SNR mismatch and online estimation in turbo decoding", IEEE Transaction on Communications, Vol. COM-46, No. 4, pp.421-423, April. 1998」、従来技術文献５「A. Ramesh et al, "SNR estimation in generalized fading channels and its application to turbo decoding", Proceeding of. IEEE ICC 2001, Helsinki, June 2001」及び従来技術文献６「滝沢ほか，“並列組合せＳＳ通信方式における通信路状態の推定方法に関する一検討（２）”，電子情報通信学会総合全国大会，Ａ−５−６，ｐｐ．１８８，２００２年３月」において提案されている。これらは変調方式としてＢＰＳＫを想定し、かつ、復調においては完全に同期検波が確立したものと仮定している。また、これらの従来技術文献４，５，６では、雑音は実数として扱われているため、雑音による位相ゆらぎが考慮されていない。
【００６９】
本実施形態ではより実用的な無線システムの観点から、多相ＰＳＫに適用でき、かつ、完全同期が確立していない「準同期」状態でも動作するブラインド推定法を提案する。まず、ＰＳＫ信号の高次モーメントについて説明する。
ｍ相ＰＳＫ信号を準同期状態で標本化すると、次式の複素変数ｓが観測される。
【００７０】
【数１７】

【００７１】
ここで、
ψ＝２πｄ／ｍ；
ｄ∈｛０，１，２，…，（ｍ−１）｝
である。また、ａ_０は初期振幅、φ_０は初期位相、ｄは情報データ、δ_０は同期ずれによる周波数偏差である。複素変数ｓを確率変数とみなすと、その第ｋ次モーメントは次式で表される。
【００７２】
【数１８】

【００７３】
ここで、準同期すなわち周波数偏差は平均化操作の時間Ｔに比べて小さいとしてδ_０の２次項以降は無視し、周波数変動と情報データは無相関であり、かつ、情報データｄが０からｍ−１までの一様分布であるとすると、次式を得る。
【００７４】
【数１９】

【００７５】
一方、絶対値についてはｍ値にかかわらず、｜ｓ｜＝ａ_０であるので、絶対値の高次モーメントは単純に次式で表される。
【００７６】
【数２０】

【００７７】
ここで、ＳはＰＳＫ信号の平均電力である。
【００７８】
次いで、ガウス雑音の高次モーメントについて説明する。ガウス雑音は振幅と位相が独立であり、位相は０から２πの間で一様に分布する。従って、そのモーメントは任意の次数ｐについて次式で表される。
【００７９】
【数２１】

【００８０】
また、信号と雑音は互いに独立であること、ならびに雑音のモーメントが零であることから、これらの結合モーメントも次式で表される。
【００８１】
【数２２】

ここで、

である
【００８２】
一方、ガウス雑音の絶対値の偶数次モーメントについては、次式の漸化式を用いることにより、下記の式を得る。
【００８３】
【数２３】

【数２４】

【００８４】
ここで、Ｎはガウス雑音の平均電力である。
【００８５】
さらに、ブラインド汎関数について定義する。上述のｍ相ＰＳＫ信号とガウス雑音の高次モーメントの特徴に着目し、これらの和である受信信号
【数２５】
ｙ＝ｓ＋ｎ
が受信されるシステムにおいて、受信信号ｙについてのｍ次のモーメントを用いた次式の汎関数を定義する。
【００８６】
【数２６】

【００８７】
この汎関数は受信した信号ｙだけで定義されているので、信号と雑音を分離することなく、かつ、送信信号レプリカを用いることなくＳＮ比を盲的に推定することができる。この汎関数の物理的意味を以下に述べる。
【００８８】
まず、上記数２６の分子を二項展開し、信号と雑音が無相関であることを用いると、次式を得る。
【００８９】
【数２７】

【００９０】
これに上述で得た高次モーメントの式を適用すると次式を得る。
【００９１】
【数２８】

【００９２】
次に、上記数２６の分母を展開すると、次式を得る。
【００９３】
【数２９】

【００９４】
ここで、上付きの記号＊は複素共役を示す。上記数２９の第１項と第３項はそれぞれ信号と雑音の平均電力を意味し、第２項はこれらの結合モーメントなので零となる。従って、次式を得る。
【００９５】
【数３０】

【００９６】
これらを上述の汎関数に代入すると、次式を得る。
【数３１】

【００９７】
これを変形すれば，次式を得る。
【００９８】
【数３２】

【００９９】
これらの式は汎関数とＳＮ比との関係を示す式であり、受信信号レベルを検出することにより、数２６を用いて汎関数の値を計算し、当該汎関数の値を数３１又は数３２に代入することによりＳＮ比に関する高次方程式となり、これを例えば、ニュートン法などの方程式の数値解法を用いて解であるＳＮ比を計算できる。
【０１００】
さらに、上記の汎関数を用いた無線受信機における適応制御方法について図７を参照して説明する。
【０１０１】
図７の無線受信機１０ａにおいて、波形等化器６−１が乗算器２−１とＡ／Ｄ変換機５−１との間に挿入され、波形等化器６−２が乗算器２−２とＡ／Ｄ変換機５−２との間に挿入される。各波形等化器６−１，６−２は例えば公知のトランスバーサルフィルタであり、異なる複数の遅延量で遅延された受信信号を所定の乗算パラメータで乗算することによりＰＳＫ受信信号の波形を制御して等化するものである。適応制御型コントローラ２０ａは、図１の適応制御型コントローラ２０の処理に加えて、Ａ／Ｄ変換器５−１，５−２の出力信号に基づき受信信号レベルを検出し、数２６を用いて汎関数の値を計算し、当該汎関数の値を数３１又は数３２に代入することによりＳＮ比に関する高次方程式となり、これを例えば、ニュートン法などの方程式の数値解法を用いて解であるＳＮ比を計算する。次いで、適応制御型コントローラ２０ａは、計算したＳＮ比に基づいて、そのＳＮ比が実質的に最大となるように、各波形等化器６−１，６−２の乗算パラメータを適応制御する。なお、複数の乗算パラメータを制御する方法については、例えば、上述の最急勾配法、順次ランダム法、ランダム法、高次元二分法などの非線形計画法における反復的な数値解法を用いることができる。
【０１０２】
以上の実施形態においては、アナログ波形等化器６−１，６−２を用いているが、本発明はこれに限らず、デジタル波形等化器を用いてもよい。この場合、アナログ波形等化器６−１，６−２に代えて、Ａ／Ｄ変換器５−１と適応制御型コントローラ２０ａとの間にデジタル波形波形等化器を挿入し、Ａ／Ｄ変換器５−２と適応制御型コントローラ２０ａとの間にデジタル波形波形等化器を挿入する。
【０１０３】
以上の実施形態においては、受信信号のＳＮ比に基づいた適応制御方法の対象として波形等化器６−１，６−２を用いているが、本発明はこれに限らず、受信信号のＳＮ比に対して影響を与える無線受信機の信号等化器、信号ろ波器、リニアライザ、同調器などの信号処理手段を用いてもよい。ここで、例えば、信号ろ波器は、上記アナログ波形等化器６−１，６−２又はデジタル波形等化器の位置に挿入され、所定の帯域の信号ろ波処理を実行する。また、リニアライザは、上記アナログ波形等化器６−１，６−２又はデジタル波形等化器の位置に挿入され、所定の線形等化処理を実行する。さらに、同調器は、例えば適応制御型コントローラ２０ａの制御動作に含まれ、計算されたＳＮ比に基づいて、ＳＮ比が実質的に最大となるように、局部発振器３の局部発振周波数を制御することにより、当該無線受信機１０ａの受信周波数を、所望波の信号周波数に実質的に同一になるように同調する。
【０１０４】
以上の実施形態においては、複素ガウス雑音ならびに多相ＰＳＫ信号のモーメントを高次まで定式化し、ＰＳＫ変調に特有な信号点配置に着目することにより汎関数を定義し、これが信号対雑音比の推定指標となることを上記のモーメントの定式を用いて解析的に示した。さらに、有限データ長の信号と加算的ガウス雑音が混在する系における本汎関数の統計的振舞を計算機シミュレーションにより示した。平均化のデータ数が少ないと特に低ＳＮ比の領域で分散が大きい。データ数を多くしていくと解析的に導出した単調増加曲線に漸近し、高い精度でＳＮ比を推定して計算することができる。本汎関数は演算が容易であり、同期検波も必要としないので、簡易な民生コンシューマ向の適応受信システム等のブラインド制御規範としての利用することができる。
【０１０５】
以上の実施形態においては、６本の非励振素子Ａ１乃至Ａ６と、それに対応した可変リアクタンス素子１２−１乃至１２−６とを備えているが、本発明はこれに限らず、少なくとも１つの非励振素子Ａ１と、それに対応した可変リアクタンス素子１２−１を備えてもよい。また、それぞれの個数は複数でもよい。
【０１０６】
【発明の効果】
以上詳述したように本発明に係るアレーアンテナの制御方法によれば、従来技術のエスパアンテナやリニアアレーアンテナなどのアレーアンテナの制御方法において、当該アレーアンテナによって受信された受信信号に基づいて、非線形計画法における反復的な数値解法を用いて、所定の期間における上記受信信号のｍ乗値の平均値の絶対値の（１／ｍ）乗値を上記受信信号の２乗値の絶対値の平均値の（１／２）乗値で除算した関数である規範関数の値が実質的に最大となるように、上記アレーアンテナの主ビームを所望波の方向に向けかつ干渉波の方向にヌルを向けるための各可変リアクタンス素子のリアクタンス値を計算して設定する。従って、参照信号を必要とせずに、アレーアンテナの指向性を所望波の方向に主ビームを向けかつ干渉波の方向にヌルを向けるように適応制御することができる。ここで、参照信号を必要としないので、当該装置の構成を簡単化できる。また、規範関数は受信信号のｍ乗値のみで記述されているので、適応制御処理の計算処理をきわめて簡単に実行できる。
【０１０７】
また、本発明に係る受信信号の信号対雑音比の計算方法によれば、ｍ相ＰＳＫ変調された（ここで、ｍは２以上の整数である。）無線信号を受信信号として受信する無線受信機において、所定の期間における上記受信信号のｍ乗値の平均値の絶対値の（１／ｍ）乗値を上記受信信号の２乗値の絶対値の平均値の（１／２）乗値で除算した関数である規範関数の値を計算し、計算された規範関数に基づいて上記規範関数と信号対雑音比との関係を示す式を用いて上記受信信号の信号対雑音比を計算する。従って、きわめて簡単な方法で受信信号の信号対雑音比を、リアルタイムで高精度で計算することができる。
【０１０８】
さらに、本発明に係る無線受信機の適応制御方法によれば、上記受信信号の信号対雑音比の計算方法により受信信号の信号対雑音比を計算し、上記計算された信号対雑音比が実質的に最大となるように、無線受信機の信号等化器又は信号ろ波器である信号処理手段を適応制御する。従って、上記計算した信号対雑音比に基づいて、無線受信機の信号処理手段をリアルタイムでかつ高精度で適応制御できる。
【図面の簡単な説明】
【図１】 本発明に係る第１の実施形態であるアレーアンテナの制御装置の構成を示すブロック図である。
【図２】 図１のエスパアンテナ装置１００の詳細な構成を示す断面図である。
【図３】 図１のエスパアンテナ装置１００の非励振素子Ａｎと可変リアクタンス素子１２−ｎの接続点付近の回路を示す回路図である。
【図４】 図１の適応制御型コントローラ２０によって実行される、最急勾配法による適応制御処理を示すフローチャートである。
【図５】 本発明に係る第２の実施形態であるアレーアンテナの制御装置の構成を示すブロック図である。
【図６】 図１のエスパアンテナ装置１００を用いて実行された、ブラインド適応ビーム形成のシミュレーションのフローを示す図である。
【図７】 本発明に係る第３の実施形態であるアレーアンテナの制御装置の構成を示すブロック図である。
【符号の説明】
Ａ０…励振素子、
Ａ１乃至Ａ６…非励振素子、
１…低雑音増幅器（ＬＮＡ）、
２−１，２−２…混合器、
３…局部発振器、
４…９０度移相器、
５−１，５−２…Ａ／Ｄ変換器、
６−１，６−２…波形等化器、
８…給電用同軸ケーブル、
１１…接地導体、
１２−１乃至１２−６…可変リアクタンス素子、
１３…低域通過フィルタ、
１４…抵抗、
１５…キャパシタ、
２０，２０ａ…適応制御型コントローラ、
５０…アレーアンテナ、
５１−１乃至５１−Ｎ…アンテナ素子、
５２−１乃至５２−Ｎ…低雑音増幅器（ＬＮＡ），
５３−１乃至５３−Ｎ…可変移相器、
５４…合成器、
６０…適応制御型コントローラ、
１００…エスパアンテナ装置。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an array antenna control method capable of changing the directivity of an array antenna apparatus composed of a plurality of antenna elements, and more particularly, to an electronically controlled waveguide array antenna apparatus (ESPAR) antenna; Hereinafter, the present invention relates to an array antenna control method capable of adaptively changing the directivity characteristic of ESPAR antenna. The present invention also relates to a method of calculating a signal-to-noise ratio of a wireless receiver that calculates a signal-to-noise ratio of a received signal received by the wireless receiver, and an adaptive control method of the wireless receiver using the calculation method.
[0002]
[Prior art]
Prior art ESPAR antennas are disclosed in, for example, prior art document 1 “T. Ohira et al.,“ Electronically steerable passive array radiator antennas for low-cost analog adaptive beamforming, ”2000 IEEE International Conference on Phased Array System & Technology pp. 101. -104, Dana point, California, May 21-25, 2000 "and Japanese Patent Application Laid-Open No. 2001-24431. The ESPAR antenna is connected to an excitation element to which a radio signal is supplied, at least one non-excitation element that is provided at a predetermined interval from the excitation element and to which no radio signal is supplied, and the non-excitation element. A directivity characteristic of the array antenna can be changed by providing an array antenna including a variable reactance element and changing a reactance value of the variable reactance element.
[0003]
In general, the following method is used as a method of adaptively controlling the ESPAR antenna on the receiving side. That is, a learning sequence signal is included in advance at the beginning of each wireless packet data on the transmission side, the same signal as the learning sequence signal is generated on the reception side, and the received learning sequence signal on the reception side, Based on the fact that the cross-correlation with the generated learning sequence signal is substantially maximized (standard reference), the reactance value of the variable reactance element is changed to change its directivity. Thereby, the directivity of the ESPAR antenna is set to the optimum pattern, that is, a pattern in which the main beam is directed in the direction of the desired wave and null is formed in the direction of the interference wave.
[0004]
[Problems to be solved by the invention]
However, in this conventional example, a reference signal such as a learning sequence signal is necessary, and it is necessary to match this reference signal in advance on both the transmission side and the reception side. There was a problem that the circuit of this became complicated.
[0005]
In addition, in order to adaptively control the signal equalizer and signal filter in a radio receiver, it is necessary to estimate and calculate the signal-to-noise power ratio, but to calculate the received signal received in real time. I couldn't.
[0006]
The object of the present invention is to solve the above-mentioned problems, and can perform adaptive control so that the main beam of the array antenna is directed to the desired wave and the null is directed to the interference wave without requiring a reference signal. It is to provide a method for controlling an array antenna.
[0007]
Another object of the present invention is to solve the above-described problems. For example, in order to adaptively control the signal equalizer and the signal filter in a radio receiver, the signal-to-noise ratio of the received signal is set. It is an object of the present invention to provide a method for calculating a signal-to-noise ratio of a received signal that can be estimated and calculated, and an adaptive control method for a radio receiver using the calculation method.
[0008]
[Means for Solving the Problems]
  An array antenna control method according to a first invention includes an excitation element for receiving a radio signal,
  At least one non-excitation element provided at a predetermined distance from the excitation element;
  A variable reactance element connected to the non-excitation element,
  In the array antenna control method of changing the directivity characteristics of the array antenna by operating the non-excited element as a director or a reflector by changing a reactance value of the variable reactance element,
  the aboveReceiveThe received radio signal is m-phase PSK modulated (where m is an integer greater than or equal to 2), and based on the received signal received by the excitation element, using an iterative numerical solution in nonlinear programming The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. The reactance value of the variable reactance element for directing the main beam of the array antenna in the direction of the desired wave and directing the null in the direction of the interference wave is calculated so that the value of the reference function that is the function is substantially maximized. And a setting step.
[0009]
  Further, the array antenna control method according to the second invention is such that a plurality of P received signals received by an array antenna in which a plurality of P antenna elements are juxtaposed with each other at a predetermined interval are respectively converted into P phase shifts. In a method for controlling an array antenna that outputs a received signal after synthesis after phase-shifting by a predetermined phase shift amount by means,
  the aboveReceiveThe radio signal is m-phase PSK modulated (where m is an integer equal to or greater than 2), and based on the combined received signal, a repetitive numerical solution in a nonlinear programming method is used. A function obtained by dividing the absolute value of the average value of the m-th power value of the received signal in the period by the (1/2) power value of the average value of the absolute value of the square value of the received signal. The amount of phase shift of each phase shift means for directing the main beam of the array antenna in the direction of the desired wave and the null in the direction of the interference wave is calculated so that the value of a certain reference function is substantially maximized. And a step of setting.
[0010]
Furthermore, the signal-to-noise ratio calculation method for the received signal according to the third aspect of the invention is a wireless reception for receiving a radio signal that is m-phase PSK modulated (where m is an integer of 2 or more) as a received signal. In the machine
The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. Calculating a value of a reference function that is a function, and calculating a signal-to-noise ratio of the received signal using an expression indicating a relationship between the reference function and a signal-to-noise ratio based on the calculated reference function It is characterized by that.
[0011]
Still further, an adaptive control method for a wireless receiver according to a fourth aspect of the present invention is to receive a wireless signal that is m-phase PSK modulated (where m is an integer of 2 or more) as a received signal, and the received signal In a wireless receiver provided with signal processing means for processing
The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. Calculating a value of a reference function that is a function, and calculating a signal-to-noise ratio of the received signal using an expression indicating a relationship between the reference function and the signal-to-noise ratio based on the calculated reference function;
Adaptively controlling the signal processing means so that the calculated signal-to-noise ratio is substantially maximized. Here, the signal processing means is preferably a signal equalizer, a signal filter, a linearizer or a tuner of a radio receiver.
[0012]
DETAILED DESCRIPTION OF THE INVENTION
Embodiments according to the present invention will be described below with reference to the drawings.
[0013]
<First Embodiment>
FIG. 1 is a block diagram showing a configuration of an array antenna control apparatus according to a first embodiment of the present invention. As shown in FIG. 1, the array antenna control apparatus according to this embodiment includes an ESPAR antenna apparatus 100 including one excitation element A0 and six non-excitation elements A1 to A6, a radio receiver 10, and the like. And an adaptive control type controller 20.
[0014]
Here, the transmitted radio signal is m-phase PSK modulated (where m is an integer equal to or greater than 2), and the adaptive control controller 20 is configured by a digital computer such as a computer, for example, and the ESPAR antenna apparatus 100 Based on the received signal y (t) received by the excitation element A0, the received signal y (t) is raised to the m-th power using, for example, the steepest gradient method, which is an iterative numerical method in nonlinear programming. Is set so that the main beam of the ESPAR antenna apparatus 100 is directed in the direction of the desired wave and in the direction of interference so that the value of the normative function (for example, Equation 14 described below) is maximized. It is characterized in that the reactance values of the variable reactance elements 12-1 to 12-6 to be directed are calculated and set.
[0015]
In FIG. 1, an ESPAR antenna device 100 includes an excitation element A0 and non-excitation elements A1 to A6 provided on a ground conductor 11, and the excitation element A0 includes six excitation elements A0 provided on a circumference of a radius r. They are arranged so as to be surrounded by the non-excitation elements A1 to A6. Preferably, the non-exciting elements A1 to A6 are provided at equal intervals on the circumference of the radius r. The lengths of the excitation elements A0 and the non-excitation elements A1 to A6 are configured to be, for example, about λ / 4 (where λ is the wavelength of the desired wave), and the radius r is λ / 4. Configured. The feeding point of the excitation element A0 is connected to the low noise amplifier (LNA) 1 via the coaxial cable 8, and the non-excitation elements A1 to A6 are connected to the variable reactance elements 12-1 to 12-6, respectively. The reactance values of the reactance elements 12-1 to 12-6 are set by reactance value signals from the adaptive control type controller 20.
[0016]
FIG. 2 is a longitudinal sectional view of the ESPAR antenna device 100. The excitation element A0 is electrically insulated from the ground conductor 11, and the non-excitation elements A1 to A6 are grounded with respect to the ground conductor 11 via the variable reactance elements 12-1 to 12-6. The operation of the variable reactance elements 12-1 to 12-6 will be described. For example, when the longitudinal lengths of the excitation element A0 and the non-excitation elements A1 to A6 are substantially the same, for example, the variable reactance element 12-1 Is inductive (L property), the variable reactance element 12-1 becomes an extension coil, and the electrical lengths of the non-excitation elements A1 to A6 are longer than that of the excitation element A0, and function as a reflector. On the other hand, for example, when the variable reactance element 12-1 has capacitance (C-type), the variable reactance element 12-1 becomes a shortening capacitor, and the electrical length of the non-excitation element A1 becomes shorter than that of the excitation element A0. Acts as a director. The non-excitation elements A2 to A6 connected to the other variable reactance elements 12-2 to 12-6 operate in the same manner.
[0017]
Therefore, in the ESPAR antenna apparatus 100 of FIG. 1, the plane directivity characteristics of the ESPAR antenna apparatus 100 are changed by changing the reactance values of the variable reactance elements 12-1 to 12-6 connected to the non-excitation elements A1 to A6. Can be changed.
[0018]
  In the array antenna control apparatus of FIG. 1, the excitation element A0 of the ESPAR antenna apparatus 100 receives a radio signal y (t), and the received signal y (t), which is the received radio signal, passes through the coaxial cable 8. Input to the wireless receiver 10, the wireless receiver 10 performs BPSK demodulation processing on the received signal y (t) to obtain two digital baseband signals from the BPSK demodulated received signals orthogonal to each other. . That is, in the radio receiver 10, the received signal y (t) is first subjected to high frequency amplification by the low noise amplifier (LNA) 1 and then divided into two, and one of the two divided received signals y (t) is mixed by the mixer 2-2. 1 is mixed with the local oscillation signal from the local oscillator 3, and the I signal after direct conversion is A / D converted by the A / D converter 5-1 to obtain a digital baseband I signal. On the other hand, the other received signal y (t) divided into two is mixed by the mixer 2-2 with the local oscillation signal phase shifted by 90 degrees from the local oscillation signal by the 90 degree phase shifter 4 and then directly. The converted Q signal is A / D converted by the A / D converter 5-2 to obtain a digital baseband Q signal. These two digital baseband signals are output as data signals and also output to the adaptive control type controller 20. Next, the adaptive control type controller 20 uses the steepest gradient method, for example, based on the two digital baseband signals representing the received signal y (t) received by the excitation element A0 of the ESPAR antenna apparatus 100. The main beam of the ESPAR antenna apparatus 100 is such that only the signal y (t) and the value of the reference function (Expression 14) expressed using the m-th power of the received signal y (t) is substantially maximized. To the direction of the desired waveInterference waveReactance value x of each variable reactance element 12-1 to 12-6 for directing null in the direction of_k(K = 1, 2,..., 6) is calculated, and reactance value signals indicating the calculated values are output to the variable reactance elements 12-1 to 12-6, so that the reactance value x_kSet.
[0019]
FIG. 3 is a circuit diagram showing a circuit near the connection point between the non-excitation element An and the variable reactance element 12-n of the ESPAR antenna apparatus 100 of FIG. In FIG. 3, a DC bias voltage, which is a reactance value signal from the adaptive control type controller 20, passes through an L-type low-pass filter 13 including a resistor 14 and a capacitor 15, for example, a variable reactance element 12 that is a varactor diode. −n (n = 1, 2,..., 6), whereby the reactance values x of the variable reactance elements 12-1 to 12-6 are applied._k(K = 1, 2,..., 6) is controlled.
[0020]
Next, the ESPAR antenna device 100 is formulated. In this formulation model, a half-wave dipole antenna is used as the non-exciting element A0, and six dipole antennas arranged in a circular array are used as the non-exciting elements A1 to A6. All the element intervals are λ / 4, and each dipole is a conductive cylinder having a radius λ / 100. The wavelength shortening rate in the length direction of the element is 0.926. Varactor diodes, which are variable reactance elements 12-1 to 12-6, are loaded in series with respect to the central portion of each of the non-excitation elements A1 to A6, and the directivity is determined by the combination of these reactance values. .
[0021]
The mutual coupling between elements is obtained from the structural parameters of the antenna by using electromagnetic field analysis by the method of moments, and this is expressed by the impedance matrix Z as shown in the following equation (for example, the prior art document 2 “Takashi Ohira,“ Espa. A method for easily obtaining reactance for forming a main beam of an antenna into a desired method: quasi-synchronous synthesis and steepest gradient method ", IEICE Technical Report, AP2001-48, pp.1-6, 20017 See "Month".)
[0022]
[Expression 1]

[0023]
Since the structure of the ESPAR antenna device 100 has cyclic symmetry, the independent elements of the 49 elements of the matrix Z are 6 elements. These are complex parameters that should be called as follows from their physical meanings.
[0024]
[Table 1]
―――――――――――――――――――――――――――――――――――
z₀₀: Self-input impedance of excitation element
z₀₁: Coupling impedance between excitation element and non-excitation element
z₁₁: Self-input impedance of non-excited element
z₁₂: Coupling impedance between two non-excited elements adjacent to each other
z₁₃: Next, the coupling impedance between two non-excited elements (adjacent to each other)
z₁₄: Coupling impedance between two non-excited elements facing each other
―――――――――――――――――――――――――――――――――――
[0025]
In addition, each impedance value used in the Example mentioned later is as follows.
(A) z₀₀= + 52.0-5.7j
(B) z₀₁= + 23.9-29.2j
(C) z₁₁= + 64.0-3.4j
(D) z₂₁= + 29.7-29.8j
(E) z₃₁= -13.9-27.6j
(F) z₄₁= -26.0-16.7j
Here, the unit of the impedance value is Ω. The reactance values of the reactance elements 12-1 to 12-6, which are varactor diodes, are expressed as x.₁, X₂, ..., x₆Then, the directivity (array factor) D of the ESPAR antenna device 100_a(Θ, φ) is expressed by the following equation (see, for example, Prior Art Document 2).
[0026]
[Expression 2]
D_a(Θ, φ)
= A (θ, φ)^Ti (x₁, X₂, ..., x₆)
[0027]
Here, a (θ, φ) is a steering vector when the phase center of the ESPAR antenna device 100 is taken as the central non-excitation element A0, and is expressed by the following equation as a function of the elevation angle θ and the azimuth angle φ. .
[0028]
[Equation 3]

[0029]
Here, d is an element interval equal to the radius r, and β is a propagation constant in free space. I (x₁, X₂, ..., x₆) Is an equivalent weight vector of the ESPAR antenna and is expressed by the following equation.
[Expression 4]
i (x₁, X₂, ..., x₆)
= Z^-1(V_su₀-Xi)
= V_S(Z + X)^-1u₀
Where u₀Is a unit vector expressed by the following equation.
[Equation 5]
u₀= [1, 0, ..., 0]^T
X is the input impedance z of the RF receiver._sAnd a reactance matrix that is a diagonal matrix of the following equation having reactance values of the variable reactance elements 12-1 to 12-6 as components.
[Formula 6]
X = diag [z_s, Jx₁, Jx₂, ..., jx₆]
[0030]
Therefore, directivity (array factor) D of ESPAR antenna apparatus 100_a(Θ, φ) is expressed by the following equation.
[Expression 7]
D_a(Θ, φ)
= Z_sa (θ, φ)^T{Z + diag [z_s, Jx₁, Jx₂, ..., jx₆]}^-1u₀
[0031]
When multiple signal waves arrive, a vector whose components are those signal waveforms
[Equation 8]
s (t) = [s₁(T), s₂(T), ..., s_K(T)]
Define Here, K is the number of signals. The output signal of ESPAR antenna apparatus 100 when these are received simultaneously is expressed by the following equation.
[Equation 9]
y (t) = i (x₁, X₂, ..., x₆)^TA (θ, Φ) s (t) + n (t)
Here, A (θ, Φ) is an array manifold and is expressed by the following equation.
[Expression 10]
A (Θ, Φ)
= [A (θ₁, Φ₁), A (θ₂, Φ₂), ..., a (θ_K, Φ_K]]
here,
## EQU11 ##
Θ = {θ₁, Θ₂, ..., θ_K}
[Expression 12]
Φ = {φ₁, Φ₂, ..., φ_K}
And n (t) is an additive noise waveform.
[0032]
Therefore, the received signal y (t) of the ESPAR antenna apparatus 100 is expressed by the following equation.
[Formula 13]

[0033]
Where s_k(T) and θ_k, Φ_kAre the time waveform and direction of arrival of the kth signal, respectively.
[0034]
Next, “blind adaptive beamforming” used in the present embodiment will be described. The purpose of adaptive beamforming is to substantially maximize the signal-to-interference noise power ratio SINR included in the antenna reception output signal y (t) derived by the above equation (13). Blind control is to update antenna variable parameters (generally weight vectors: here reactance values of variable reactance elements 12-1 to 12-6) without referring to any signal information included in the desired wave. is there.
[0035]
In order to form a beam adaptively, (1) the reference signal is included in the header of the transmission packet, (2) the reference signal sequence is known in advance on the receiving side, and (3) the reference signal is synchronized. The process of detecting timing and (4) training the array weighting factors is typically used. For example, there is an algorithm “MCCC: Maximum Cross Correlation Coefficient” that maximizes the cross-correlation coefficient between a received signal and a reference signal as an adaptive beam forming method of the ESPAR antenna apparatus 100 (for example, the conventional art document 3 “Kamiya et al.,“ ESPAR (Refer to "Basic Study of Antenna: Statistical Standard of SIR Characteristics Based on Adaptive Control", IEICE Technical Report, AP2000-175, SANE2000-156, pp.17-24, January 2001)). On the other hand, blind adaptive beam forming is a function of adaptively forming a beam without a reference signal, and the processes (1) to (3) can be omitted.
[0036]
In this embodiment, paying attention to the characteristic peculiar to the m-phase PSK modulation signal, a blind standard using this is proposed. The property to be noted here is that “the m-phase PSK modulation signal becomes a constant complex value when raised to the mth power regardless of the modulation data”. When receiving noise or interference on the communication channel, fluctuations from this constant complex value are observed on the receiving side. It is considered that the desired signal can be extracted with higher purity as the fluctuation is smaller. Therefore, it is proposed to use a standard function of the following equation using the m-th moment of the output signal of the receiving antenna derived as described above.
[0037]
[Expression 14]

[0038]
Here, E [•] represents an ensemble average (an average value at a predetermined time) of the argument •. The denominator represents the average power of the m-th signal. Normative function J_mThe physical interpretation of (y (t)) will be described in detail later. The advantage of this normative function is that it does not include the “constant complex value” described above. That is, it is not necessary for the receiving side to know this value in advance. This means that it does not depend on the absolute gain and fixed phase rotation amount of the antenna and the receiving circuit system, and is an important advantage when used in an actual radio system.
[0039]
Next, adaptive beam forming using the reference function will be described. The “adaptive beamforming” is to maximize the signal-to-interference noise power ratio SINR = S / (N + I) included in the received signal y (t) of the ESPAR antenna apparatus 100 derived from the above equation (13). In addition, the antenna variable parameters (reactance values of the variable reactance elements 12-1 to 12-6 in the ESPAR antenna apparatus 100) are updated. By repetitively updating the reactance value based on the above-described normative function, an optimum beam pattern in which the antenna directivity is maximum in output SINR, that is, a beam in which a main beam is formed in the desired wave direction and a null is formed in the interference wave direction. It becomes a pattern.
[0040]
That is, the normative function J does not include the target value C, only the received signal y (t), and the mth power of the received signal {(y (t))^m} Is used to express the description. In this case, it is a great merit that the target value can be controlled in an unknown state. Based on this criterion, the reactance value is repeatedly updated using an iterative numerical method in a nonlinear programming method such as the steepest gradient method, so that the signal-to-interference noise power ratio (SINR) of the antenna output is substantially reduced. The optimum beam is formed so as to be maximum, that is, to direct the main beam of the ESPAR antenna apparatus 100 in the direction of the desired wave and the null in the direction of the interference wave.
[0041]
Next, adaptive control of the antenna beam using the steepest gradient method will be described. The recurrence formula for the set of reactance values (reactance vectors) x of the variable reactance elements 12-1 to 12-6 when using the steepest gradient method is expressed by the following equation.
[0042]
[Expression 15]
x (n + 1) = x (n) + μ∇Jn
[Expression 16]

[0043]
Here, n is the update order of x, and the parameter μ is a step size determined by trial and error. Here, the steepest gradient method is a concept of a method including the steepest descent method, but in the present embodiment, a method for obtaining an optimal solution so as to substantially maximize the value of the normative function is used.
[0044]
Further, a specific procedure for obtaining the optimum solution by the steepest gradient method will be described. In order to find a good reactance vector x that makes the normative function Jn as large as possible by the steepest gradient method using Equation 15, the following procedure is used.
(I) First, the iteration number parameter n (that is, the nth iteration) is set to 1, and an initial value x (1) of a predetermined reactance vector (for example, the ESPAR antenna device 100 is set as an omni antenna). The process is started by the reactance vector).
(Ii) This initial value (when n = 1) or current estimate (when n ≧ 2) is then used to determine the slope of the normative function Jn in the iteration number parameter n (ie, the nth iteration) Calculate the vector ∇Jn.
(Iii) The next estimated value in the reactance vector x is calculated by changing the initial value or the current estimated value in the same direction as the direction of the gradient vector ∇Jn.
(Iv) The iteration number parameter n is incremented by 1, and the process returns to step (ii) to repeat the process. This iterative process is executed up to the number of iterations at which the reactance vector x substantially converges.
[0045]
FIG. 4 is a flowchart showing more specific adaptive control processing by the steepest gradient method, which is executed by the adaptive control type controller 20 of FIG.
[0046]
In step S1 of FIG. 4, first, the iteration number parameter n is reset to 1, the initial value is set and inserted in the reactance vector x (1), and the element parameter k is reset to 1 in step S2. Next, the received signal y (t) is measured in step S3, and the value of the normative function J is calculated using equation 14 in step S4.⁽⁰⁾Insert set to. Further, in step S5, the reactance value x_kA predetermined perturbation value Δx_kAnd add the value to reactance value x_kAfter that, the received signal y (t) is measured in step S6, and the value of the normative function J is calculated using equation 14 in step S7. In step S8, JJ⁽⁰⁾To calculate ∂Jn / ∂x_kAnd reactance value x in step S9_kA predetermined perturbation value Δx_kAnd subtract the subtracted value to reactance value x_kAfter returning to the value before the perturbation, it is determined in step S10 whether or not the element parameter k is K (= 6) or more. If “NO” in the step S10, the element parameter k is incremented by 1 in a step S11, the process returns to the step S5, and the above-described processing is repeated. On the other hand, if YES in step S10, the next estimated value x (n + 1) of the reactance vector x is calculated in step S12 using the recurrence formula of equation 15, and then the iteration number parameter n is determined in step S13. It is determined whether or not the predetermined number of iterations N has been reached. If NO, the number of iterations parameter n is incremented by 1 in step S14, and then the processing from step S2 is repeated. On the other hand, if YES in step S13, it is determined that the signal has sufficiently converged, and a reactance value signal having the calculated reactance vector x value is output to the variable reactance elements 12-1 to 12-6. Set.
[0047]
  As described above, according to the present embodiment, the adaptive control type controller 20 is based on the received signal y (t) received by the excitation element A0 of the ESPAR antenna apparatus 100, and is an iterative numerical value in nonlinear programming. For example, using the steepest gradient method, the value of the normative function (Expression 14) represented by the mth power of the received signal y (t) is substantially equal to the received signal y (t). The main beam of the ESPAR antenna device 100 is directed in the direction of the desired wave andInterference waveThe reactance values of the variable reactance elements 12-1 to 12-6 for directing null in the direction are calculated and set. Therefore, the directivity of the array antenna can be adaptively controlled so that the main beam is directed in the direction of the desired wave and null is directed in the direction of the interference wave without requiring a reference signal. Here, since no reference signal is required, the configuration of the apparatus can be simplified. Further, since the normative function J is described only by the received signal y (t), the calculation process of the adaptive controller 20 can be executed very easily.
[0048]
In the above embodiment, six non-excitation elements A1 to A6 are used. However, if there are at least a plurality of the non-excitation elements A1 to A6, the directivity of the array antenna apparatus can be electronically controlled. Alternatively, more than six non-exciting elements may be provided. Further, the arrangement shape of the non-excitation elements A1 to A6 is not limited to the above-described embodiment, and it is only necessary to be away from the excitation element A0 by a predetermined distance. In other words, the intervals with respect to the non-excitation elements A1 to A6 may not be constant.
[0049]
In the above embodiment, the reactance value of each variable reactance element 12 is calculated using the steepest gradient method, but the present invention is not limited to this, and the following sequential random method, random method, high-dimensional bisection An iterative numerical method in nonlinear programming such as the method may be used.
[0050]
In the sequential random method, the following procedure is used.
(I) First, the iteration number parameter n (that is, the nth iteration) is set to 1, and a predetermined initial value x (1) of the reactance vector (for example, the ESPAR antenna apparatus 100 is set to an omni antenna). The process starts with the reactance vector at the time.
(Ii) The initial value (when n = 1) or the current estimate (when n ≧ 2) is then used to add to the estimate in the iteration number parameter n (ie, the nth iteration) The value is calculated by generating a random number within a predetermined range.
(Iii) The next estimated value in the reactance vector is calculated by adding the calculated added value to the estimated value.
(Iv) The iteration number parameter n is incremented by 1, and the process returns to step (ii) to repeat the process. This iterative process is executed until the value of the normative function J becomes a predetermined threshold value (for example, 0.9) or more.
[0051]
In the random method, the following procedure is used.
(I) First, processing is started by a predetermined initial value x (1) of a reactance vector (for example, a reactance vector when the ESPAR antenna device 100 is set to an omni antenna).
(Ii) Next, using this initial value, a value added to the initial value is calculated by generating a random number within a predetermined existence range.
(Iii) The estimated value in the reactance vector is calculated by adding the calculated addition value to the initial value.
(Iv) If the value of the normative function J in the calculated estimated value is equal to or greater than a predetermined threshold value (for example, 0.9), the estimated value is set as the reactance vector to be set. Return to (ii) and repeat the process.
[0052]
Further, in the high-dimensional bisection method, the following procedure is used.
(I) First, the iteration number parameter n (that is, the nth iteration) is set to 1 and the process is started.
(Ii) Next, a predetermined existence range of each reactance value of the reactance vector (in the second and subsequent times, the existence range of the estimated value selected before) is equally divided into two, and an average value of the two existence ranges divided into two (Two average values for each variable reactance element 12-1 to 12-6) is calculated.
(Iii) The value of the normative function J for these two average values is calculated, and the one with the larger value of the normative function J is set as the next estimated value in the reactance vector.
(Iv) The iteration number parameter n is incremented by 1, and the process returns to step (ii) to repeat the process. This iterative process is executed until the value of the normative function J becomes a predetermined threshold value (for example, 0.9) or more.
[0053]
In the above embodiment, the reference function J is used as a reference function for obtaining the reactance value for adaptive control, and the optimum solution of the reactance vector is calculated so as to be substantially maximized. However, the present invention is not limited to this, and the inverse of the normative function J may be used as a normative function for obtaining a reactance value for adaptive control, and an optimum solution of the reactance vector may be calculated so as to be substantially minimized.
[0054]
In the above embodiment, the six non-excitation elements A1 to A6 and the corresponding variable reactance elements 12-1 to 12-6 are provided. However, the present invention is not limited to this, and at least one non-excitation element is included. An excitation element A1 and a variable reactance element 12-1 corresponding to the excitation element A1 may be provided. Moreover, the number of each may be plural.
[0055]
<Second Embodiment>
FIG. 5 is a block diagram showing the configuration of the array antenna control apparatus according to the second embodiment of the present invention.
[0056]
  In this embodiment, signals received by the antenna elements 51-1 to 51-P of the array antenna 50 are converted into RF bands formed by variable phase shifters 53-1 to 53-P and a combiner 54 that is an adder. The BFN (Beam Forming Network) circuit is used to synthesize, and the array antenna control device has a plurality of P antenna elements.51-1 to 51-PAre adaptive control controllers for controlling the beams of array antennas 50 (for example, linear arrays, which may be arranged in a two-dimensional shape or a three-dimensional shape) arranged at predetermined intervals from each other, An adaptive control type controller 60 is provided. Here, the transmitted radio signal is m-phase PSK modulated (m is an integer equal to or greater than 2), and the adaptive control type controller 60 repeats numerical values in nonlinear programming based on the synthesized received signal. For example, by using the steepest gradient method, which is a solution method, the value of the reference function (Expression 14) expressed using the m-th power of the received signal y (t) is substantially maximized. Direct the main beam of the antenna 50 in the direction of the desired wave andInterference waveEach phase shift control voltage v corresponding to the phase shift amount of the variable phase shifters 53-1 to 53-P for directing null in the direction of_p(P = 1, 2,..., P) is calculated and set.
[0057]
The configuration of the array antenna control apparatus shown in FIG. 5 will be described below. In FIG. 5, a radio signal is received by an array antenna 50 in which a plurality of P antenna elements 51-1 to 51-P are arranged in parallel on a straight line at a predetermined interval, and each antenna element 51-1 to 51-P is received. The radio signals received by P are input to variable phase shifters 53-1 to 53-P through low noise amplifiers (LNA) 52-1 to 52-P, respectively. Each of the variable phase shifters 53-1 to 53-P receives the input radio signal from each of the phase shift control voltages v output from the adaptive control type controller 60._pAfter the phase shift amount corresponding to (p = 1, 2,..., P), the phase is output to the synthesizer 54. The synthesizer 54 synthesizes the power of the input P radio signals, and the synthesized radio signal is received signal y (t) to the radio receiver 10 having the same configuration as the radio receiver 10 in FIG. Output.
[0058]
Next, the wireless receiver 10 obtains two digital baseband signals from the received signals orthogonal to each other based on the input combined received signal y (t), similarly to the wireless receiver 10 of FIG. To the adaptive control type controller 60. The adaptive control type controller 60 is an iterative numerical method in nonlinear programming based on two input digital baseband signals, for example, using the steepest gradient method, similar to the adaptive control processing of FIG. By executing the process of (5), only the received signal y (t) and the value of the reference function (Expression 14) expressed using the mth power of the received signal y (t) is substantially maximized. In addition, each phase shift control voltage v corresponding to the amount of phase shift of the variable phase shifters 53-1 to 53-P for directing the main beam of the array antenna 50 in the direction of the desired wave and directing null in the direction of interference._p(P = 1, 2,..., P) is calculated and applied to the variable phase shifters 53-1 to 53-P to set the corresponding phase shift amounts.
[0059]
Also in the adaptive control type controller 60 according to this embodiment, as in the adaptive control type controller 20 according to the first embodiment, the directivity of the array antenna is set in the direction of the desired wave without requiring a reference signal. Can be adaptively controlled so that the null is directed toward the interference wave. Here, since no reference signal is required, the configuration of the apparatus can be simplified. Further, since the normative function J is described only by the received signal y (t), the calculation process of the adaptive controller 60 can be executed very easily.
[0060]
In the above embodiment, the phase shift control voltage v corresponding to the amount of phase shift of each of the variable phase shifters 53-1 to 53-P using the steepest gradient method._pHowever, the present invention is not limited to this, and an iterative numerical method in the above-described nonlinear programming method such as the sequential random method, the random method, or the high-dimensional bisection method may be used. The normative function J may use the reciprocal thereof.
[0061]
  FIG. 6 is a diagram showing a simulation flow of blind adaptive beamforming executed using the ESPAR antenna apparatus 100 of FIG. In this simulation, similar to the formulation model described above,Excitation elementA half-wave dipole antenna is used as A0, and six dipole antennas arranged in a circular array are used as the non-excitation elements A1 to A6. Further, it is assumed that the arrival directions of the desired wave and the interference wave arriving at the ESPAR antenna apparatus 100 are unknown (adaptive control) and that no training signal is used (blind processing).
[0062]
In the simulation flow of FIG. 6, adaptive control of the antenna beam is performed by performing steps SS1 to SS5 based on the interference wave steering vector, the desired wave steering vector, the antenna structure parameter, the incoming wave signal, and the noise. Finally, the directivity array factor and the output SINR are calculated and output (steps SS6 and SS7). In the processing of steps SS1 to SS7, the reception signal y (t) is received (step SS1), and the norm function J is based on the received reception signal y (t)._m{Y (t)} is calculated (step SS2), the reactance matrix is updated (step SS3), the reactance matrix is calculated (step SS4), and then the equivalent weight vector is calculated (step SS5). Then, the directivity array factor is calculated from the equivalent weight vector (step SS6), while the output SINR is calculated from the received signal y (t) and the noise n (t) (step SS7).
[0063]
In this simulation, it is assumed that the arrival directions of the desired wave and the interference wave arriving at the ESPAR antenna apparatus 100 are unknown (adaptive control) and that no training signal is used (blind processing). Simulation is performed in an environment in which interference waves simultaneously arrive in addition to desired waves. QPSK random modulation signal and noise are both additive and Gaussian noise for both the desired wave and the interference wave. These desired wave, interference wave, and noise are uncorrelated with each other. For simplicity, the band limiting filter, delay spread, angular spread, fading, Doppler effect, and synchronization error are all ignored in the transmission path. Under this condition, the reactance values of the six variable reactance elements 12-1 to 12-6 are controlled based on the above-described normative function. Examples of antenna structure parameters that can be used for the simulation include: number of controlled elements: 6, element spacing: all 1/4 wavelength, each dipole radius: 1/100 wavelength, wavelength shortening rate in the length direction of the element: 0. 926. The internal impedance of the RF transceiver connected to the ESPAR antenna device 100 is z_s= 50Ω. As an optimization algorithm, a pure random search method, a steepest gradient method, a high-dimensional bisection method, a sequential random method, a regression step method, a method based on Hamiltonian dynamics, or the like can be used.
[0064]
As described above, according to the present embodiment, the ESPAR antenna apparatus 100 is capable of blind beam formation by an appropriate standard and feedback control even though the hardware configuration is simple. The case of wave reception is shown.
[0065]
In the above embodiment, the normative function of the above equation 14 is used. The time average E (•) in the above equation 14 is obtained by receiving a data signal transmitted by, for example, a frequency division multiplexing method in parallel and performing parallel processing. In this case, for example, an average value for a plurality of data signals in a predetermined period such as one symbol may be used.
[0066]
<Third Embodiment>
FIG. 7 is a block diagram showing the configuration of the array antenna control apparatus according to the third embodiment of the present invention. The array antenna control apparatus according to this embodiment differs from the first embodiment of FIG. 1 in the following points.
(1) Instead of the wireless receiver 10, the wireless receiver 10 that further includes waveform equalizers 6-1 and 6-2 is provided in the wireless receiver 10 that receives the m-phase PSK signal.
(2) Instead of the adaptive control type controller 20, the adaptive control type controller 20 calculates the value of the above reference function, and based on the calculated reference function, the relationship between the reference function and the signal-to-noise power ratio is obtained. The signal-to-noise power ratio of the received signal is calculated using the equation shown, and the waveform equalizers 6-1 and 6-2 are adaptively controlled so that the calculated signal-to-noise ratio is substantially maximized. An adaptive control type controller 20a is provided.
[0067]
Prior to the detailed description of FIG. 7, the definition of a functional that is a reference function and a method for calculating a signal-to-noise power ratio will be described below.
[0068]
For adaptive feedback control of variable signal waveform equalizers, signal filters, and linearizers for optimal reception at a wireless receiver, signal-to-noise ratio estimation is an effective means. In particular, it is necessary to establish an SN ratio estimation technique for blind operation in the wireless receivers of FIGS. 1 and 7 that do not use a training reference signal or a signal replica. Up to now, blind estimation functions using statistical expectation values and variance of received data have been described in, for example, the prior art document 4 “TA Summers et al.,“ SNR mismatch and online estimation in turbo decoding ”, IEEE Transaction on Communications, Vol. COM-46, No. 4, pp.421-423, April. 1998 ”, Prior Art Document 5“ A. Ramesh et al, “SNR estimation in generalized fading channels and its application to turbo decoding”, Proceeding of. IEEE ICC 2001, Helsinki, June 2001 ”and prior art document 6“ Takizawa et al., “A Study on Estimating Channel State in Parallel Combination SS Communication System (2)”, IEICE General Conference, A-5-6 , Pp. 188, March 2002 ”. These assume that BPSK is assumed as a modulation method, and that synchronous detection is completely established in demodulation. In these prior art documents 4, 5, and 6, noise is handled as a real number, and therefore phase fluctuation due to noise is not considered.
[0069]
In the present embodiment, from the viewpoint of a more practical wireless system, a blind estimation method that can be applied to multiphase PSK and that operates even in a “quasi-synchronization” state in which perfect synchronization is not established is proposed. First, the high-order moment of the PSK signal will be described.
When the m-phase PSK signal is sampled in a quasi-synchronized state, a complex variable s of the following equation is observed.
[0070]
[Expression 17]

[0071]
here,
ψ = 2πd / m;
dε {0, 1, 2,..., (m−1)}
It is. A₀Is the initial amplitude, φ₀Is the initial phase, d is the information data, δ₀Is a frequency deviation due to a synchronization shift. When the complex variable s is regarded as a random variable, the kth moment is expressed by the following equation.
[0072]
[Formula 18]

[0073]
Here, it is assumed that quasi-synchronization, that is, the frequency deviation is smaller than the time T of the averaging operation₀If the second and subsequent terms are ignored, the frequency fluctuation and the information data are uncorrelated, and the information data d has a uniform distribution from 0 to m−1, the following equation is obtained.
[0074]
[Equation 19]

[0075]
On the other hand, the absolute value | s | = a regardless of the m value.₀Therefore, the absolute high-order moment is simply expressed by the following equation.
[0076]
[Expression 20]

[0077]
Here, S is the average power of the PSK signal.
[0078]
Next, the high-order moment of Gaussian noise will be described. Gaussian noise is independent in amplitude and phase, and the phase is uniformly distributed between 0 and 2π. Therefore, the moment is expressed by the following equation for an arbitrary order p.
[0079]
[Expression 21]

[0080]
Further, since the signal and the noise are independent from each other and the moment of the noise is zero, these coupling moments are also expressed by the following equation.
[0081]
[Expression 22]

here,

Is
[0082]
On the other hand, for the even-order moment of the absolute value of Gaussian noise, the following formula is obtained by using the recurrence formula of the following formula.
[0083]
[Expression 23]

[Expression 24]

[0084]
Here, N is the average power of Gaussian noise.
[0085]
Furthermore, the blind functional is defined. Paying attention to the characteristics of the above-mentioned m-phase PSK signal and the high-order moment of Gaussian noise, the received signal that is the sum of these characteristics
[Expression 25]
y = s + n
In the system in which is received, the functional of the following equation using the m-th moment for the received signal y is defined.
[0086]
[Equation 26]

[0087]
Since this functional is defined only by the received signal y, the signal-to-noise ratio can be estimated blindly without separating the signal and noise and without using a transmission signal replica. The physical meaning of this functional is described below.
[0088]
First, when the numerator of Equation 26 is expanded into two terms and the fact that the signal and noise are uncorrelated is obtained,
[0089]
[Expression 27]

[0090]
Applying the high-order moment formula obtained above to this gives the following formula.
[0091]
[Expression 28]

[0092]
Next, when the denominator of the above equation 26 is expanded, the following expression is obtained.
[0093]
[Expression 29]

[0094]
Here, the superscript symbol * indicates a complex conjugate. The first and third terms of Equation 29 above mean the average power of the signal and noise, respectively, and the second term is zero because of the coupling moment. Therefore, the following equation is obtained.
[0095]
[30]

[0096]
Substituting these into the above-mentioned functional yields:
[31]

[0097]
If this is modified, the following equation is obtained.
[0098]
[Expression 32]

[0099]
These expressions are expressions showing the relationship between the functional and the SN ratio. By detecting the received signal level, the value of the functional is calculated using Equation 26, and the value of the functional is expressed by Equation 31 or By substituting into 32, a high-order equation relating to the S / N ratio is obtained, and the S / N ratio as a solution can be calculated using a numerical solution of an equation such as Newton's method.
[0100]
Further, an adaptive control method in the wireless receiver using the above functional will be described with reference to FIG.
[0101]
In the radio receiver 10a of FIG. 7, the waveform equalizer 6-1 is inserted between the multiplier 2-1 and the A / D converter 5-1, and the waveform equalizer 6-2 is the multiplier 2-. 2 and the A / D converter 5-2. Each of the waveform equalizers 6-1 and 6-2 is, for example, a known transversal filter, and controls the waveform of the PSK reception signal by multiplying the reception signal delayed by a plurality of different delay amounts by a predetermined multiplication parameter. And equalize. The adaptive control type controller 20a detects the received signal level based on the output signals of the A / D converters 5-1 and 5-2 in addition to the processing of the adaptive control type controller 20 of FIG. By calculating the value of the functional and substituting the value of the functional into Equation 31 or Equation 32, a higher-order equation relating to the S / N ratio is obtained, which is a solution using a numerical solution of an equation such as Newton's method, for example. Calculate the signal-to-noise ratio. Next, the adaptive control type controller 20a adaptively controls the multiplication parameters of the waveform equalizers 6-1 and 6-2 based on the calculated S / N ratio so that the S / N ratio is substantially maximized. As a method for controlling a plurality of multiplication parameters, for example, an iterative numerical solution method in the nonlinear programming method such as the steepest gradient method, the sequential random method, the random method, or the high-dimensional bisection method can be used.
[0102]
In the above embodiment, the analog waveform equalizers 6-1 and 6-2 are used. However, the present invention is not limited to this, and a digital waveform equalizer may be used. In this case, instead of the analog waveform equalizers 6-1 and 6-2, a digital waveform waveform equalizer is inserted between the A / D converter 5-1 and the adaptive control type controller 20a, and the A / D A digital waveform waveform equalizer is inserted between the converter 5-2 and the adaptive control type controller 20a.
[0103]
In the above embodiment, the waveform equalizers 6-1 and 6-2 are used as targets of the adaptive control method based on the SN ratio of the received signal. However, the present invention is not limited to this, and the SN of the received signal is not limited thereto. Signal processing means such as a signal equalizer, a signal filter, a linearizer, and a tuner that affect the ratio may be used. Here, for example, the signal filter is inserted at the position of the analog waveform equalizers 6-1 and 6-2 or the digital waveform equalizer, and executes signal filtering processing of a predetermined band. The linearizer is inserted at the position of the analog waveform equalizers 6-1 and 6-2 or the digital waveform equalizer, and executes a predetermined linear equalization process. Further, the tuner is included in the control operation of the adaptive control type controller 20a, for example, and controls the local oscillation frequency of the local oscillator 3 based on the calculated SN ratio so that the SN ratio is substantially maximized. Thus, the reception frequency of the radio receiver 10a is tuned so as to be substantially the same as the signal frequency of the desired wave.
[0104]
In the above embodiment, the complex Gaussian noise and the moment of the polyphase PSK signal are formulated to a higher order, and the functional is defined by focusing on the signal point arrangement peculiar to PSK modulation, which is an estimation of the signal-to-noise ratio. It was analytically shown to be an index using the above moment formula. Furthermore, the statistical behavior of this functional in a system with a finite data length signal and additive Gaussian noise is shown by computer simulation. When the number of data to be averaged is small, the variance is large particularly in the low SN ratio region. As the number of data is increased, the curve gradually approaches an analytically derived monotonically increasing curve, and the SN ratio can be estimated and calculated with high accuracy. Since this functional is easy to calculate and does not require synchronous detection, it can be used as a blind control standard for a simple adaptive reception system for consumer consumers.
[0105]
In the above embodiment, the six non-excitation elements A1 to A6 and the corresponding variable reactance elements 12-1 to 12-6 are provided. However, the present invention is not limited to this, and at least one non-excitation element is included. An excitation element A1 and a variable reactance element 12-1 corresponding to the excitation element A1 may be provided. Moreover, the number of each may be plural.
[0106]
【The invention's effect】
As described above in detail, according to the array antenna control method according to the present invention, in the array antenna control method such as the ESPAR antenna or the linear array antenna of the prior art, based on the received signal received by the array antenna, Using an iterative numerical method in nonlinear programming, the absolute value of the average value of the m-th power value of the received signal in a predetermined period is the (1 / m) power value of the absolute value of the square value of the received signal. The main beam of the array antenna is directed in the direction of the desired wave and null in the direction of the interference wave so that the value of the normative function, which is a function obtained by dividing the average value by the (½) power, is substantially maximized. The reactance value of each variable reactance element for directing is calculated and set. Therefore, the directivity of the array antenna can be adaptively controlled so that the main beam is directed in the direction of the desired wave and null is directed in the direction of the interference wave without requiring a reference signal. Here, since no reference signal is required, the configuration of the apparatus can be simplified. Further, since the normative function is described only with the m-th power value of the received signal, the calculation process of the adaptive control process can be executed very easily.
[0107]
Further, according to the method for calculating the signal-to-noise ratio of a received signal according to the present invention, wireless reception for receiving a radio signal that has been m-phase PSK modulated (where m is an integer of 2 or more) as a received signal. In the machine, the (1 / m) power of the absolute value of the mean value of the m-th power value of the received signal in a predetermined period is the (1/2) power value of the average value of the absolute value of the square value of the received signal. The value of the normative function, which is a function divided by, is calculated, and the signal-to-noise ratio of the received signal is calculated using an equation indicating the relationship between the normative function and the signal-to-noise ratio based on the calculated normative function. . Therefore, the signal-to-noise ratio of the received signal can be calculated with high accuracy in real time by a very simple method.
[0108]
Furthermore, according to the adaptive control method of a radio receiver according to the present invention, the signal-to-noise ratio of the received signal is calculated by the signal-to-noise ratio calculation method of the received signal, and the calculated signal-to-noise ratio is substantially equal. Therefore, the signal processing means which is a signal equalizer or a signal filter of the wireless receiver is adaptively controlled so as to be maximized. Accordingly, the signal processing means of the wireless receiver can be adaptively controlled in real time and with high accuracy based on the calculated signal-to-noise ratio.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of an array antenna control apparatus according to a first embodiment of the present invention;
FIG. 2 is a cross-sectional view showing a detailed configuration of the ESPAR antenna device 100 of FIG. 1;
3 is a circuit diagram showing a circuit near a connection point between a non-excitation element An and a variable reactance element 12-n of the ESPAR antenna apparatus 100 of FIG. 1;
4 is a flowchart showing an adaptive control process by the steepest gradient method, which is executed by the adaptive control type controller 20 of FIG. 1. FIG.
FIG. 5 is a block diagram showing a configuration of an array antenna control apparatus according to a second embodiment of the present invention;
FIG. 6 is a diagram showing a flow of blind adaptive beam forming simulation executed using the ESPAR antenna apparatus 100 of FIG. 1;
FIG. 7 is a block diagram showing a configuration of an array antenna control apparatus according to a third embodiment of the present invention.
[Explanation of symbols]
A0: Excitation element,
A1 to A6 ... non-excited elements,
1 ... Low noise amplifier (LNA),
2-1, 2-2 ... mixer,
3 ... Local oscillator,
4 ... 90 degree phase shifter,
5-1, 5-2 ... A / D converter,
6-1, 6-2 ... waveform equalizer,
8 ... Coaxial cable for feeding,
11: Ground conductor,
12-1 to 12-6 ... variable reactance element,
13 ... Low-pass filter,
14 ... resistance,
15: Capacitor,
20, 20a ... adaptive control type controller,
50 ... Array antenna,
51-1 to 51-N ... antenna elements,
52-1 to 52 -N: Low noise amplifier (LNA),
53-1 to 53 -N ... variable phase shifter,
54. Synthesizer,
60: Adaptive control type controller,
100: ESPAR antenna device.

Claims (8)

An excitation element for receiving a radio signal;
At least one non-excitation element provided at a predetermined distance from the excitation element;
A variable reactance element connected to the non-excitation element,
In the array antenna control method, by changing the reactance value of the variable reactance element, the non-excited element is operated as a director or a reflector, respectively, and the directivity characteristics of the array antenna are changed.
The received radio signal is m-phase PSK modulated (where m is an integer greater than or equal to 2), and an iterative numerical solution in nonlinear programming is performed based on the received signal received by the excitation element. The (1 / m) power of the absolute value of the average value of the m-th power value of the received signal in a predetermined period is used as the (1/2) power value of the average value of the absolute value of the square value of the received signal. The reactance value of the variable reactance element for directing the main beam of the array antenna in the direction of the desired wave and the null in the direction of the interference wave so that the value of the reference function, which is a function divided by A method for controlling an array antenna, comprising the step of calculating and setting A plurality of P received signals received by an array antenna in which a plurality of P antenna elements are juxtaposed at a predetermined interval are phase-shifted by a predetermined amount of phase by P phase-shifting means, and then synthesized. In the array antenna control method for outputting the combined received signal,
The received radio signal is m-phase PSK modulated (where m is an integer greater than or equal to 2), and based on the combined received signal, using an iterative numerical solution in nonlinear programming, The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. The phase shift amount of each phase shift means for directing the main beam of the array antenna in the direction of the desired wave and the null in the direction of the interference wave is set so that the value of the reference function as a function is substantially maximized. An array antenna control method comprising a step of calculating and setting. In a radio receiver that receives an m-phase PSK modulated radio signal (where m is an integer greater than or equal to 2) as a received signal,
The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. Calculating a value of a reference function that is a function, and calculating a signal-to-noise ratio of the received signal using an expression indicating a relationship between the reference function and a signal-to-noise ratio based on the calculated reference function A method for calculating a signal-to-noise ratio of a received signal. In a radio receiver including a signal processing unit that receives an m-phase PSK-modulated radio signal (where m is an integer of 2 or more) as a received signal and processes the received signal.
The absolute value of the average value of the m-th power value of the received signal in the predetermined period is divided by the (1/2) power value of the average value of the absolute value of the square value of the received signal. Calculating a value of a reference function that is a function, and calculating a signal-to-noise ratio of the received signal using an expression indicating a relationship between the reference function and the signal-to-noise ratio based on the calculated reference function;
Adaptive control of the signal processing means such that the calculated signal-to-noise ratio is substantially maximized.   5. The wireless receiver adaptive control method according to claim 4, wherein the signal processing means is a signal equalizer of the wireless receiver.   5. The wireless receiver adaptive control method according to claim 4, wherein the signal processing means is a signal filter of the wireless receiver.   5. The wireless receiver adaptive control method according to claim 4, wherein the signal processing means is a linearizer of the wireless receiver.   5. The radio receiver adaptive control method according to claim 4, wherein the signal processing means is a radio receiver tuner.

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