JP3747415B1 - Multiplex transmission apparatus and multiple transmission method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To multiplex data with high efficiency with respect to a multiplex transmission apparatus and a multiplex transmission method for data transmission or recording / reproduction.
In a multiplex transmission apparatus and multiplex transmission method including one or both of a multiplexing processing unit and a demultiplexing processing unit, signal point generating means (signal for generating a signal point for modulating data) The signal point generated by the point generator 14) and the signal point generator is converted into a signal on the time axis by the inverse fast Fourier transform unit 15, arranged on the time axis at Nyquist time intervals, and on the frequency axis This includes a configuration and a processing process in which a plurality of carrier frequencies are arranged and multiplexed at Nyquist frequency intervals.
[Selection] Figure 1

Description

  The present invention relates to a multiplex transmission apparatus and a multiplex transmission method for multiplexing data, and belongs to the data transmission field and the data processing field, and can be applied to various wired transmission systems and various wireless transmission systems as data transmission. In the data processing field, the present invention can be applied to a recording / reproducing system equivalent to a multiplexed data transmission path.

  In the data transmission field and the recording / reproduction field in data processing, the transmission efficiency and the recording efficiency are improved by multiplexing the data. Moreover, since the power line carrier system in the data transmission field uses a power line as a data transmission path, reflected waves are randomly generated due to a large number of branch paths, and noise components generated from various electric devices are generated. Overlaying the data component causes a data error. In such a data transmission field, for example, a QAM method / SS method / OFDM method / Wavelet-OFDM method is known as a multiplexing method.

  The above-mentioned QAM (Quadrature Amplitude Modulation) method is the fastest method capable of transmitting data without intersymbol interference, and is based on Nyquist transmission. In this Nyquist transmission, the transfer function is a sequence orthogonal to (010) on the time axis, and high-speed transmission is possible with no waste on the time axis. However, since the Nyquist filter is used as the waveform shaping filter, there is a waste on the frequency axis due to the roll-off rate. In addition, it is not possible to reduce leakage by narrowing a specific band.

  The SS (Spread Spectrum) method is said to be resistant to noise, but even when 31PN is applied, the S / N improvement amount is about 10 * LOG (31) = about 15 dB at most, and the power line carrier system It is not enough to apply as it is. In addition, as with QAM transmission, it is not easy to reduce leakage in a specific band.

  In addition, OFDM (Orthogonal Frequency Division Multiplexing) system has good frequency efficiency by orthogonally multiplexing the signals of each channel on the frequency axis. However, since it corresponds to multipath etc., the guard time is increased on the time axis. Since it is provided, the transmission efficiency on the time axis is not so good. Moreover, since the unit function is used for the transfer function, the leakage reduction in the specific band is only about 13 dB. Also, unnecessary band removal is about 33 dB at maximum.

  The Wavelet-OFDM method is a method that realizes time axis orthogonality / frequency axis orthogonality using Wavelet waveforms, and is the method with the highest data transmission efficiency at present, but in this method that is actually realized. The noise suppression level of the unnecessary band is about 35 dB, which is not always sufficient for application to the power line carrier system. Also, since there is no guard time as in OFDM, transmission efficiency is good, but it is said that there is a weak aspect with respect to the multipath associated with the long-distance branch line of the access system.

  Various means have already been proposed as a data transmission system. For example, a modulation / demodulation system that always reduces the bit error rate of received data to an optimal value as the reference data value for demodulation regardless of the reception form of the receiving device or the fluctuation state of the transmission path. Has been proposed (see, for example, Patent Document 1). Also, on the frequency axis, input signals are transmitted using carriers that do not interfere with each other, and carrier signals that are affected by noise on the receiving side are removed or the synthesis distribution ratio is reduced on the receiving side, thereby reducing the effects of noise as much as possible. A communication system is known (see, for example, Patent Document 2).

  Also, in a wireless transmission system, avoiding distortions such as local drop in the spectrum of the entire band, selecting a fine beam more finely, improving the S / N of the output signal and reducing the amount of calculation of the weight coefficient An array antenna control method and apparatus that can be reduced are known (see, for example, Patent Document 3). In addition, the transmitting side inserts a zero point between signal points and transmits, and the receiving side extracts a noise component on the zero point and interpolates between the zero points, and uses the noise component predicted by the interpolation. There has been proposed a noise removal method and apparatus that removes noise on signal points and realizes stable data transmission (see, for example, Patent Document 4). A means for performing multiplex transmission using orthogonal sequences is also known (see, for example, Patent Document 5).

The redundant signal point is transmitted so that the EOR value of the signal point becomes a specific value on the transmission side, and the EOR value is calculated from the received signal point on the reception side, and this calculation result is different from the predetermined set value. In this case, error correction means for confirming the transmission quality of each transmission path and performing error correction on the receiving side so that the EOR value becomes a predetermined value for the received data of the channel having the highest probability of error occurrence. It is known (see, for example, Patent Document 6). The transmission side uses a spread spectrum system, and the reception side uses a correlation filter so that the time response waveform of the output of the correlation filter is 1 for the center and all zeros for the others. There is known a means for interpolating and predicting the noise superimposed on the portion 1 to remove the noise (see, for example, Patent Document 7). There is also known a means for improving the reception performance by improving the estimation accuracy of the transmission path characteristics on the receiving side in the digital terrestrial broadcasting system to which the OFDM modulation / demodulation system is applied (see, for example, Patent Document 8).
JP-A-7-321766 JP-A-11-163807 JP-A-11-234025 JP 2002-164801 A WO02 / 47304 Japanese Patent Laid-Open No. 2003-134096 JP 2003-324360 A JP 2004-96703 A

Even if various means of the multiplex transmission system in the above-mentioned conventional example are applied to a multiplex transmission system having a poor data transmission environment such as a power line carrier system, it is not easy to obtain sufficient characteristics. . Thus, the problems to be solved by the present invention include the following first to sixth. The first is the realization of highly efficient data transmission. Improvement of transmission efficiency is indispensable for realizing high-speed data transmission, and transmission efficiency of at least 95% or more is desirable.
Second, it is easy to reduce leakage in a specific band. For example, in the 2 MHz to 30 MHz band, there are a large number of existing radio stations, and there are also radio stations receiving up to the limit of reception sensitivity. For this reason, it is desirable to reduce leakage of these radio stations by at least about 30 dB so as not to cause interference.

The third is noise suppression in unnecessary bands. When applied to a power line carrier system, there are noise from household electrical appliances connected to the power line, and flying noise of radio waves emitted from various radio stations. Most of these noises are narrow-band, large-amplitude tone noise groups. In the power line carrier system, it is necessary to realize stable data transmission with respect to these narrow band tone noise groups, and it is desirable that the suppression of adjacent noise is at least 70 dB or more.
The fourth is in-band noise cancellation. The out-of-band tone noise seen from each channel can be suppressed by a noise suppression filter or the like. However, tone noise that falls within the same band cannot be suppressed, and it is necessary to remove the noise by noise canceling means. The noise cancellation gain is desirably at least 50 dB.

The fifth is multipath compatible. Since the power line is connected to various home appliances by a large number of branch connections, a multipath is generated on the transmission line with these branches. In the power line carrier system, it is necessary to have sufficient strength against these multipaths.
The sixth is timing phase synchronization. A number of channels are multiplexed and transmitted on the frequency axis. The group delay characteristic of the line is not necessarily flat. For this reason, not only frequency synchronization of timing but also timing phase synchronization of individual channels is indispensable.

  An object of the present invention is to solve the above first to sixth problems in data multiplexing.

  The multiplex transmission apparatus of the present invention is a multiplex transmission apparatus having one or both of a data multiplex processing unit and a demultiplexing processing unit, wherein the multiplex processing unit is a signal for modulating the data. A configuration including point generating means and means for multiplexing the signal points generated by the signal point generating means by arranging a plurality of carrier frequencies at Nyquist time intervals on the time axis and at Nyquist frequency intervals on the frequency axis It is what has.

  The multiplexing means divides the signal point of the data generated by the signal point generating means into a real part and an imaginary part, and one of the real part and the imaginary part is 1 / It has a configuration in which the waveform is synthesized by shifting the time length by 2 Nyquist.

  The multiplexing means obtains a copy of the inverse fast Fourier transform output signal of the signal point of the data generated by the signal point generating means over a plurality of times, and multiplies the time response waveform of the transmission Nyquist filter as a window function. And means for sequentially adding the output signals of the means on the time axis.

  The multiplexing processing unit sequentially distributes the signal points of the data generated by the signal point generating means to even channels and odd channels, and sets window functions for the even channels and odd channels to each other by ½ Nyquist time. And a means for synthesizing the waveforms by multiplying each of them by the time difference.

  The multiplexing processing unit includes a means for selecting and multiplexing the data generated by the signal point generating means with respect to the adjacent channel so that the data signal waveform and the interference waveform of the adjacent channel are orthogonal to each other. Is.

  In the multiplex transmission apparatus having one or both of the data multiplexing processing unit and the demultiplexing processing unit, the demultiplexing processing unit is configured to multiply the time response waveform of the received Nyquist filter as a window function; A first means for performing fast Fourier transform and adding to the output signal of the means at a Nyquist time interval; a second means for multiplying the window function by shifting the length by 1/2 Nyquist time; And means for extracting a real part and an imaginary part from the output signals of the first and second means and performing signal point determination.

  The demultiplexing processing unit multiplies the time response waveform of the reception Nyquist filter with a time difference of ½ Nyquist time for each of the even channel and the odd channel as a window function, and provides a high Nyquist time interval for each multiplication output. It has a configuration including means for performing Fourier transform and adding and performing signal point determination corresponding to the even channel and the odd channel.

  Further, the means for multiplying the window function divides the window function into a region of a central portion of the time response waveform and both side portions of the central portion, the window function of the central portion region is a rectangular window function, and the both side portions The final window function is a coefficient obtained by multiplying the window function of the region by the Hanning window function or a window function similar to the Hanning window function by the time response waveform of the Nyquist filter.

  Further, the signal point generating means of the multiplexing processing unit has zero point inserting means for inserting a zero point between signal points, and the means for determining the signal point of the demultiplexing processing unit is on the zero point. And a means for interpolating and predicting the noise component on the signal point to remove the noise component on the signal point.

  Further, the demultiplexing processing unit has a configuration including a time equalizer corresponding to a channel for equalizing the group delay characteristics.

  The multiplexing processing unit includes means for transmitting a signal to be transmitted by spreading it on one or both of the frequency axis and the time axis, and the demultiplexing processing unit has a signal point corresponding to the signal of the spread channel. It has the structure provided with the means which performs determination, adds, and performs a signal point determination again with respect to the result of this addition.

  The multiplexing processing unit includes means for transmitting a signal to be transmitted by spreading it on one or both of the frequency axis and the time axis, and the demultiplexing processing unit has a signal point corresponding to the signal of the spread channel. In this configuration, there is provided a means for performing determination, multiplying and adding a coefficient corresponding to the transmission quality corresponding to the channel, and performing signal point determination again on the result of the addition.

  The multiplex processing unit includes means for adding redundancy according to a frequency axis or a time axis, and the demultiplexing processing unit includes a channel-compatible transmission quality detecting unit, and a channel-compatible transmission quality detecting unit. Means for correcting the error using the transmission quality and the redundancy corresponding to the frequency axis or the time axis is provided.

  The demultiplexing processing unit has means for extracting the timing phase of the signal corresponding to the channel subjected to reception demodulation and fast Fourier transform, and adjusting the timing phase.

  The multiplex transmission method of the present invention is a multiplex transmission method that performs either or both of data multiplexing processing and demultiplexing processing by a configuration including either or both of a multiplexing processing unit and a demultiplexing processing unit. A signal point for modulating the data is generated by the signal point generating means of the multiplexing processing unit, and the signal point is a Nyquist time interval on the time axis and a plurality of carrier frequencies on the frequency axis. It includes a process of arranging and multiplexing at Nyquist frequency intervals.

  The signal point of the data generated by the signal point generating means is divided into a real part and an imaginary part, and either one of the real part and the imaginary part is shifted by 1/2 Nyquist time length with respect to the other. This includes a process of synthesizing the waveforms and performing multiplexing processing so as to be orthogonal to the time axis and orthogonal to the frequency axis.

  Further, a copy of the inverse fast Fourier transform output signal of the signal point of the data generated by the signal point generating means is obtained over a plurality of times, the time response waveform of the transmission Nyquist filter is multiplied as a window function, and the output signal resulting from the multiplication is timed. It includes the process of adding sequentially on the axis.

  The signal points of the data generated by the signal point generating means are sequentially distributed to the even and odd channels, and the window functions for the even and odd channels are respectively multiplied by the time difference of 1/2 Nyquist time. This includes the process of waveform synthesis.

  Further, it includes a process of selecting and multiplexing the data generated by the signal point generating means with respect to the adjacent channel so that the data signal waveform and the interference waveform of the adjacent channel are orthogonal to each other.

  In the multiplex transmission method for performing either or both of data multiplexing processing and demultiplexing processing by a configuration including either or both of a multiplexing processing unit and a demultiplexing processing unit, A processing unit fast Fourier transforms the output signal obtained by multiplying the time response waveform of the received Nyquist filter as a window function at a Nyquist time interval and adds it by a first means, and the window function is ½ Nyquist time. Including a step of multiplying by shifting, fast Fourier transform, adding by a second means, extracting a real part and an imaginary part from the output signals of the first and second means, and performing signal point determination. It is a waste.

  In the demultiplexing process, the time response waveform of the reception Nyquist filter with a time difference of ½ Nyquist time is multiplied as a window function for the even channel and the odd channel, respectively, and the Nyquist time interval for each multiplication output is multiplied. This includes a process of performing fast Fourier transform and adding, and performing signal point determination corresponding to the even channel and the odd channel.

  In the process of multiplying the window function, the window function is divided into a central portion of the time response waveform and regions on both sides of the central portion, and the time response waveform of the central portion region is a rectangular window function. The final window function is a coefficient obtained by multiplying the window function of the both side regions by the Hanning window function or a window function similar to the Hanning window function by the time response waveform of the Nyquist filter.

  Also, in the multiplexing process, a zero point is inserted between signal points from the signal point generating means of the multiplexing processing unit, and in the demultiplexing process, a noise component on the zero point is extracted. The method includes interpolation prediction of the noise component on the signal point and removing the noise component on the signal point.

  The demultiplexing process includes a process of providing a time equalizer corresponding to the channel to equalize the group delay characteristic corresponding to the channel.

  In the multiplexing process, the transmission signal has a process of spreading the signal to be transmitted on either the frequency axis or the time axis or both, and in the demultiplexing process, the signal of the spread channel is handled. And performing signal point determination and adding signal points, and performing signal point determination again on the result of the addition.

  In the multiplexing process, the transmission signal includes a process of spreading the signal to be transmitted on one or both of the frequency axis and the time axis, and sending the signal to the spread channel in the demultiplexing process. This includes a step of performing signal point determination, multiplying and adding a coefficient corresponding to the transmission quality corresponding to the channel, and performing signal point determination again on the result of the addition.

  In the multiplexing process, there is a process of adding a redundancy corresponding to the frequency axis, and in the demultiplexing process, the transmission quality corresponding to the channel is detected, and the transmission quality and the frequency axis are detected. And a process of correcting an error using redundancy according to.

  The demultiplexing process includes a step of extracting a timing phase of a channel-corresponding signal that has been received and demodulated and subjected to fast Fourier transform, and adjusting the timing phase.

  The above-mentioned conventional QAM, SS, OFDM, Wavelet-OFDM, and Nyquist-OFDM of time-axis orthogonal / frequency-axis orthogonal multiplex transmission by arranging signal points at the Nyquist time interval and the Nyquist frequency interval of the present invention, 39 is compared with items 1 to 5 of high-efficiency data transmission, specific band leakage reduction, unnecessary band noise suppression, in-band noise cancellation, and multipath. That is, the target value for multiplexed transmission is 95% or more for item 1, 30 dB or more for item 2, 70 dB or more for item 3, 50 dB or more for item 4, and item 5 is acceptable. Is shown by adding a circle, and Nyquist-OFDM of the present invention can be satisfied with respect to the target value.

  The multiplex transmission apparatus of the present invention will be described with reference to FIG. 1. In the multiplex transmission apparatus that multiplexes data and transmits the data from the transmission side to the reception side, signal point generating means (signal point for modulating data) Generator 14) and means for multiplexing the signal points generated by the signal point generating means by arranging Nyquist time intervals on the time axis and a plurality of carrier frequencies on the frequency axis at Nyquist frequency intervals (reverse high speed). Fourier transform unit (IFFT)).

  The multiplex transmission method of the present invention is a multiplex transmission method in which data is multiplexed and transmitted from the transmission side to the reception side. Signal points for modulating the data by signal point generation means (signal point generation unit 14) are obtained. And a process of multiplexing the signal points by arranging a plurality of carrier frequencies at Nyquist frequency intervals on the time axis and Nyquist frequency intervals on the frequency axis.

  FIG. 1 is an explanatory diagram of a multiplex transmission apparatus according to a first embodiment of the present invention, and includes both configurations of a transmission side multiplexing processing unit and a reception side demultiplexing processing unit when applied to a power line carrier system. 1 is a digital unit, 2 is an analog unit, 3 is a power supply unit, 4 is a common mode choke coil (CMC) for suppressing a leakage electric field, 5 is 10BASE-T, 100BASE-TX, etc. 1 shows a LAN (indoor local area network) connection device. In the digital unit 1, 11 is a bridge circuit having a filtering function for discarding unnecessary data of transmission data and reception data by filtering, 12 is a scramble circuit (SCR), 13 is a summing circuit, 14 is A signal point generating unit as a signal point generating unit, 15 is an inverse fast Fourier transform unit (IFFT) constituting a main part of a multiplexing unit, 16 is a low-pass filter (LPF1), 17 is a modulating unit (MOD), 18 is A transmission carrier generation unit (transmission CRR), 22 is a descrambling circuit (DSCR), 23 is a difference circuit, 24 is a signal point determination unit as means for determining a signal point, 25 is a fast Fourier transform unit (FFT), and 26 is Low-pass filter (LPF4), 27 is a demodulation unit (DEM), 28 is a reception carrier generation unit (reception CRR), 29 is a timing synchronization unit (T MPLL) shows the. The configuration of reference numerals 12 to 18 constitutes the main part of the multiplexing processing section, and the configuration of reference numerals 22 to 29 constitutes the demultiplexing processing section. The functions of each part of the digital part 1 can also be realized by the arithmetic processing function of the processor.

  In the analog section 2, 31 is a DA converter (D / A), 32 is a low-pass filter (LPF2), 33 is a transmission driver circuit (DV), 34 is a transformer section (TR), and 35 is an AD converter ( A / D), 36 is a low-pass filter (LPF3), 37 is a gain switch unit (GSW), 38 is a high-pass filter (HPF), and 39 is a voltage-controlled crystal oscillator (VCXO). In the power supply unit 3, reference numeral 41 denotes a power output unit that supplies operating power of, for example, a voltage of 5 V to each unit, and 42 denotes a power supply filter.

  Transmission data input from the 10BASE-T or 100BASE-TX side via the connection device 5 is filtered in the bridge circuit 11 and input to the scrambler circuit 12, and the data is randomized to stabilize the transmission spectrum. Realization of stabilization of electric field / leakage electric field. Then, it is input to the summing circuit 13 and phase summing is performed to withstand line fluctuations. After this phase sum processing, transmission signal points of a plurality of channels are generated by the signal point generator 14 as signal point generating means. The signal point generator 14 can be configured by a ROM or the like, and can be configured to generate notches, spread spectrum, and insert a zero point for noise cancellation.

  The information on the frequency axis is converted into information on the time axis by the inverse fast Fourier transform unit 15, unnecessary band components are removed by the low-pass filter 16, and input to the modulation unit 17. Modulated by the transmission carrier. That is, a means for multiplexing the signal points at the Nyquist time interval on the time axis and at the Nyquist frequency interval on the frequency axis is configured. The modulation signal from the modulation unit 17 is input to the DA converter 31 of the analog unit 2 and converted into an analog signal. After the unnecessary band on the analog signal is removed by the low-pass filter 32, the transmission driver circuit 33 It is amplified and transmitted to the power line, for example, the AC 100V indoor distribution line side or the indoor power line side via the transformer unit 34 and the common mode choke coil 4. In this case, a plurality of carrier frequencies are arranged at Nyquist frequency intervals on the time axis and Nyquist frequency intervals on the frequency axis, and multiplexed data transmission is performed by time axis orthogonality / frequency axis orthogonality.

  The demultiplexing process on the reception side is the reverse process of the multiplexing process on the transmission side, and the received signal input via the common mode choke coil 4 and the transformer unit 34 is not required by the high-pass filter 38. After the low-frequency component is removed, the received signal is amplified to a predetermined level by the gain switch unit 37, and then the unnecessary high-frequency band component is removed by the low-pass filter 36. Then, it is converted into a digital signal by the AD converter 35 and input to the digital unit 1.

  The reception signal input to the digital unit 1 is demodulated in the demodulation unit 27 based on the carrier signal from the reception carrier generation unit 28 to become a baseband signal, and the unnecessary band is removed by the low-pass filter 26. The time axis information is converted into frequency axis information by the fast Fourier transform unit 25. Then, the reception signal point is determined by the signal point determination unit 24, the phase difference is obtained by the difference circuit 23, and then the original transmission data is reproduced by the descrambling circuit 22. Further, the data is transferred to a terminal (not shown) via the connection device 5 via the bridge circuit 11.

  The phase difference process described above is configured to be performed after the determination in the signal point determination unit 24, but may be configured to perform the phase difference process before the signal point determination. In the synchronous modem, it is necessary to synchronize the reception clock with the transmission clock. This synchronization signal transmits a reference signal for timing at a plurality of specific frequencies on the transmission side, and this synchronization signal is extracted on the reception side. In this way, synchronization with transmission is established. The synchronization signal may be extracted at a passband, baseband, or after fast Fourier transform (FFT), but synchronization can be performed by extracting a signal from a place where efficient processing can be performed. In FIG. 1, extraction is possible from both the output signal of the low-pass filter 26 and the output signal of the fast Fourier transform unit 25. Then, the phase synchronization unit 29 can control the voltage controlled crystal oscillator 39 to establish desired synchronization.

  The power supply unit 3 has a configuration including a power output unit 41 and a power filter 42, and forms a DC power voltage such as DC5V necessary for the operation of each unit from an AC voltage of AC100V by a switching power supply configuration or the like. In the switching power supply configuration, since switching noise is generated, the power supply filter 42 is configured to prevent the switching noise from leaking to the common mode filter 4 side. Further, it is necessary to minimize the common mode current from the power supply unit so that an unnecessary leakage electric field is not generated on the line side. Furthermore, by connecting the power supply unit 3 to the line, the LCL (ground to ground balance) in the transmission band or the like can be prevented so as not to deteriorate the ground balance, or to prevent a minute signal from being lost due to low impedance. It is necessary to set the normal mode impedance to a desired value or more.

  2 shows the main parts of the multiplexing processing unit and the demultiplexing processing unit of the multiplex transmission apparatus. FIG. 2 is an explanatory diagram of the main part of the digital part 1 of the multiplex transmission apparatus shown in FIG. 1, a transmission signal generating circuit corresponding to the signal point generating unit 14, 52 corresponds to the inverse fast Fourier transform unit 15 in FIG. 1, and a transmission IFFT unit constituting a means for multiplexing, 53 is a signal in FIG. A reception signal point determination circuit 54 corresponding to the point determination unit 24 and a reception FFT unit corresponding to the fast Fourier transform unit 25 in FIG. 55 is a real part inverse Fourier transform unit (Real-part IFFT), 56 is an imaginary part inverse Fourier transform unit (Imag-part IFFT), 57 and 58 are time axis copy window function multiplication units, and 59 is a 1/2 Nyquist time. A delay unit, 60 is a waveform synthesis circuit, 61 is a signal extraction / synthesis circuit of a real part and an imaginary part, 62 is a synthesis circuit (Σ), 63 is a fast Fourier transform unit (FFT), and 64 is a window function multiplication circuit.

  The transmission data is input to the transmission signal point generating circuit 51 to be used as a transmission signal point as a vector signal, and is divided into a real part (Real) and an imaginary part (Imag) of the signal point. The imaginary part is input to the Fourier transform unit 55, and the imaginary part is input to the imaginary part inverse fast Fourier transform unit 56 to perform inverse fast Fourier transform, respectively, and input to the time axis copy window function multiplication units 57 and 58. Time axis copy window function multipliers 57 and 58 include means for obtaining a copy of a signal on the time axis over a plurality of times and multiplying the time response waveform of the transmission Nyquist filter as a window function, and a 1/2 Nyquist time length delay In section 59, either the real part side or the imaginary part side is time-shifted by a ½ Nyquist time length, and then the waveform synthesis circuit 60 synthesizes the waveform of the real part and the imaginary part. Then, the signal is input to the modulation unit 17 in FIG. 1 through the low pass filter 16, modulated by the transmission carrier, and input to the analog unit 2.

  In the reception FFT unit 54, the demodulated reception signal is input, and the window function multiplication circuit 64 multiplies the window function corresponding to the time response waveform of the reception Nyquist filter to cut out the waveform at the Nyquist time interval. The fast Fourier transform unit 63 converts the frequency information into the frequency information, convolution integration by the synthesis circuit 62, and the real part and the imaginary part are shifted on the transmission side so as to be ½ Nyquist time length intervals, respectively. In the extraction / synthesis circuit 61, the signal is simply synthesized and input to the reception signal point determination circuit 53.

  The configuration of the first embodiment of the present invention is shown in FIG. 1 and FIG. 2, and the first problem among the first to sixth problems described above is the realization of high-efficiency data transmission. The key to realizing data transmission is to eliminate waste on the time axis / frequency axis. One example of such implementation is the conventional Wavelet-OFDM scheme described above. This Wavelet-OFDM method is a method that realizes time axis orthogonality / frequency axis orthogonality. However, the actual problem and the realized noise suppression level are about 35 dB in the conventional example, and noises from various signal sources, etc. Is not necessarily enough. This conventional Wavelet-OFDM method realizes time-axis orthogonal / frequency-axis orthogonal by limiting to scalar transmission, but as another method for realizing efficient data transmission on the time axis, Nyquist transmission There is a method.

  The transfer function of the Nyquist transmission method is (0, 1, 0), which is the method that can transmit at the highest speed without intersymbol interference, but at the same time, transmission that realizes time axis orthogonality equivalently on the time axis. It is a method. The present invention realizes multiplex transmission of time-axis orthogonal / frequency-axis orthogonal by applying this Nyquist transmission method and an OFDM method capable of orthogonalization on the frequency axis. Therefore, on the transmitting side, for example, the real part of the signal point is decomposed into the imaginary part, the real part is transmitted first, and then the imaginary part is transmitted after ½ Nyquist time length, so that the adjacent channel is transmitted. High-efficiency data transmission is possible without intersymbol interference.

  The second problem is reduction of leakage in a specific band. This can be realized by dividing the Nyquist filter into transmission and reception. In order to realize deeper leakage reduction with a smaller number of taps, it is possible to reduce the side lobe by multiplying the cosine filter on the transmission side by a unique window function.

  The third problem is noise suppression. Similar to the transmission side, the reception side is a cosine filter, and a unique window function is multiplied similarly to the transmission side, so that noise suppression exceeding 70 dB can be achieved.

  The fourth problem is noise cancellation. This is because a zero point is periodically inserted on the transmission side, the data signal point is transmitted between the zero point and the zero point, and the noise component on the zero point transmitted on the transmission side is transmitted on the reception side. Can be realized by performing interpolation prediction and canceling noise superimposed on the signal point.

  The fifth problem is multipath support. In order to reduce the occurrence of errors caused by superimposition of delayed signal components due to multipath due to branch circuits, etc., for example, by using a decision feedback equalizer, stable multipath removal on the reception side Is possible.

  The sixth problem is timing synchronization. There are two types of timing synchronization: frequency synchronization and phase synchronization. Regarding frequency synchronization, it is sufficient to perform frequency synchronization from synchronization signals obtained from a plurality of channels. By providing a filter and shifting the time phase, the timing phase is adjusted, or the equalizer on the receiving side is not a 1-tap complex equalizer but a double sampling equalizer is provided, and the timing phase is Can be applied.

  FIG. 3 shows a time response of a transmission path (filter). When an input signal is input to the transmission path (filter) as an impulse, the output signal is a time response by band limitation corresponding to the transmission path (filter) characteristics. It becomes a waveform. When data (various impulse waveforms) is continuously added to the input side of this transmission line (filter), these time response waveforms are output in an overlapping manner on the output side.

  FIG. 4 is an explanatory diagram of waveforms in Nyquist transmission. As shown in the figure, if the response waveform on the time axis is a waveform that passes through zero points at equal intervals, impulses may be transmitted continuously. The data transmission is possible at high speed without interference between the codes. This is the aforementioned Nyquist transmission. That is, the time response of the Nyquist transmission line is (0, 1, 0), which is equivalent to a series that is orthogonal on the time axis.

  FIG. 5 shows normalized frequency characteristics of the Nyquist transmission line. The filter characteristic of the Nyquist filter shows a cos square characteristic, and there is an element generally called a roll-off rate. In FIG. The case where the rate is 100% is shown.

  FIG. 6 is an image diagram of orthogonal frequency division multiplexing, in which each carrier frequency has an integer multiple relationship, and the carriers are orthogonal to each other. For this reason, although the spectrums overlap each other on the frequency axis, they are orthogonal to each other on the frequency axis, so that frequency decomposition can be performed by fast Fourier transform on the receiving side. On the transmission side, information on the frequency axis is converted to information on the time axis by inverse fast Fourier transform and transmitted.

  High-efficiency multiplex transmission is possible by multiplexing and transmitting the signals having the waveform orthogonal to each other on the time axis shown in FIG. 4 as signals having waveforms orthogonal to each other on the frequency axis shown in FIG. In this case, multiplexing is performed at the Nyquist time interval on the time axis, and multiplexing is performed at the Nyquist frequency interval on the frequency axis.

  FIG. 7 shows a time response waveform when the Nyquist transmission line (cos square characteristic) is divided into transmission and reception, etc., where the transmission filter is cos filter characteristic, the reception filter is also cos filter characteristic, and the transmission line is cos square characteristic. Show the case. As described above, the reason for dividing the filter characteristics into transmission and reception is to optimize the noise tolerance.

  FIG. 8 shows the time response characteristic of the cos filter, which is a response characteristic of (0, 1, 1, 0) at 1/2 Nyquist time intervals, and this is converted into a cos square as shown in FIG. The time response waveform of the filter [(0, 1, 2, 1, 0) at 1/2 Nyquist time interval, (0, 1, 0) at Nyquist time interval] can be obtained. High-speed data transmission is possible without intersymbol interference.

  FIG. 10 is an explanatory diagram of interference between adjacent channels, and shows a frequency spectrum when three channels are multiplexed. As shown in the figure, three channels of CH-1 / CH0 / CH + 1 are multiplexed on the frequency axis, but the frequency spectrum of channel CH0 is indicated by the frequency spectrum of channel CH-1 / CH + 1 and the hatching area. It overlaps with. This area should cause interference on the time axis / frequency axis on both sides.

Since the frequency characteristic of the channel CH0 is a cos filter on the transmission side, if the frequency characteristic of the channel CH0 is F [0] (f), f is in the range of −1 to 1 (Hz).
F [0] (f) = cos (f * π / 2) (001)
When the frequency characteristic of the channel CH + 1 is F [+1] (f), f is in the range of 0 to 2 (Hz).
F [+1] (f) = sin (f * π / 2) (002)
Therefore, the frequency spectrum F [0 + 1] (f) in the right hatched area in FIG. 10 is in the range of f = 0 to 1 (Hz).
F [0 + 1] (f) = cos (f * π / 2) * sin (f * π / 2) · (003)
= 1/2 (sin (2 * f * π / 2))
= 1/2 (sin (f * π)) (004)
It becomes.

Similarly, the interference area between channel CH-1 and channel CH0 is such that f is −2 to 0 (Hz) and F [−] when the frequency characteristic of channel CH-1 is F [−1] (f). 1] (f) = − sin (f * π / 2) (005)
Therefore, the frequency spectrum F [0-1] (f) in the left hatched area in FIG. 10 is in the range of f = −1 to 0 (Hz).
F [0-1] (f) = cos (f * π / 2) * (− sin (f * π / 2))
.... (006)
= −1 / 2 (sin (2 * f * π / 2))
= −1 / 2 (sin (f * π)) (... 007)
It becomes. Although both have different polarities, they have the same sin filter in terms of power spectrum. Since the transmission side transmits with a 100% cos filter and the reception side also transmits with a 100% cos filter, when the transmission / reception filter is convoluted, the frequency spectrum (interference spectrum) between adjacent channels has a 100% sin filter characteristic.

  11 and 12 are explanatory diagrams of interference between adjacent channels, where the vertical axis indicates amplitude, but each is shown with an offset added to the amplitude value, and the offset value is shown on the right side. Further, the cos response waveform is 8, the cos carrier waveform is 6, the sin carrier waveform is 4, the real part (Real) waveform is 2, and the image part (Imega) waveform is 0. This interference spectrum is a spectrum in which the bandwidth of the cos filter is halved and the frequency axis is shifted to the left and right by a ½ Nyquist frequency interval. Therefore, as shown in FIG. 11, when impulses are input on the real part side, the transfer function is (0, 1, -1, 0) on the real part side and (0, 0, 0, 0) on the imaginary part side. ) As shown in FIG. 12, when an impulse is input to the imaginary part side, the imaginary part side is (0, 1, -1, 0) and the real part side is (0, 0, 0, 0).

  However, these are observed at the Nyquist time interval. When viewed at the data sampling point on the receiving side, there is no interference on the real part side when an impulse is input on the real part side, but the imaginary part There will be interference on the side. Conversely, when an impulse is input on the imaginary part side, there is no interference on the imaginary part side, but interference occurs on the real part side. Care must be taken because the real part side and the imaginary part side of these interference components are shifted from each other by ½ Nyquist time (the in-phase component is not a problem, but for the negative-phase component, the zero cross point is the Nyquist interval. To itself).

  FIG. 13 is an explanatory diagram of an interference waveform after a ½ Nyquist time length shift. Each waveform is shown with the offset value shown on the right added to the amplitude of the vertical axis, and one solid line The waveforms of the cos response, cos carrier, sin carrier, and real part (Real) corresponding to PH1 and the waveform of the cos response, cos carrier, sin carrier, and real part (Real) corresponding to PH2 of the other chain line are shown.

  In this case, the real part component of the vector signal point is transmitted as it is through the transmission Nyquist filter. Next, the imaginary part component is processed in the same manner. However, the signal is shifted by ½ Nyquist time length with respect to the real part side signal and added to the real part side signal for transmission. This makes it possible to zero-cross the interference waveform between adjacent channels at every Nyquist time interval. Ultimately, a condition for accurately matching the timing phase and the carrier phase is required, but high-speed data transmission is possible by multiplexing in time axis orthogonal / frequency axis orthogonal without interference between adjacent channels.

  FIG. 14 shows a single-carrier-compatible transmission modulation section, where 71 is a transmission low-pass filter (transmission LPF), 72 is a transmission modulation section (transmission MOD), 73 is a transmission carrier generation section (transmission CRR), and 74 is zero. An insertion unit, 75 is an addition unit (Σ), T is a delay circuit, and shows a configuration corresponding to the configuration of the low-pass filter 16, the modulation unit 17, and the transmission carrier generation unit 18 in FIG. Xm + n and Xm-n indicate signals before nT time and after nT time with respect to Xm + 0, and Cn,... C0,.

  Signals input at the Nyquist rate are usually converted to an integral multiple of the Nyquist rate and transmitted. The input data signal is first converted from a Nyquist rate to a sampling rate (an integer multiple of the Nyquist rate) by a zero insertion unit 74 of the transmission low-pass filter 71 and a filter unit including a delay circuit and an addition unit 75. The transmission low-pass filter 71 shapes the waveform of the data signal so that it can be transmitted at high speed without intersymbol interference. Then, the transmission modulation unit 72 multiplies the carrier signal from the transmission carrier generation unit 73 and shifts the frequency to a desired frequency band.

When this is observed on the time axis by focusing on one impulse, the impulse, the filter output, the carrier signal, and the modulation signal are as shown in FIG. First, if the input impulse is Xk, the output F of the transmission low-pass filter 71 is
F = Xk * Cn ... Kx * C + n (008)
It becomes. Next, the output F of the transmission low-pass filter 71 is multiplied by the carrier signal E (jωt) = cos θ + jsin θ, where S is the modulated signal after multiplication.
S = F * E = Xk * Cn * E (jω (tp))... Kx * C + n * E (jω (t + p)) (009)
It becomes. This indicates that a sequence obtained by multiplying the input impulse Xk by the carrier signal E (jωt) is calculated, and the result is multiplied by the time response waveform of the cos filter as a window function. Since the input impulses are sequentially input in time series, the filter output multiplied by the window function is also sequentially output. In the filter calculation, convolution processing on the time axis is performed, but it is only necessary to perform simple addition on the time axis for the transmission signal finally output as a modulated waveform. It also indicates that the signals of the real part and the imaginary part should be shifted and added by a ½ Nyquist time length in order to eliminate interference between adjacent channels.

  FIG. 16 shows a main part of the transmission IFFT unit, and shows a main part of the multiplex processing unit in FIG. In FIG. 16, 51 is a transmission signal generation circuit corresponding to the signal point generation unit 14 in FIG. 1, 52 is a transmission IFFT unit corresponding to the inverse fast Fourier transform unit 15 in FIG. Part inverse Fourier transform unit (Real-part IFFT), 56 is an imaginary part inverse Fourier transform unit (Imag-part IFFT), 57 and 58 are time axis copy window function multiplication units, 59 is a 1/2 Nyquist time delay unit, 60 Indicates a waveform synthesis circuit.

  The transmission data subjected to the scramble processing and the summation processing is input to the transmission signal point generation circuit 51, and the transmission signal point as a vector signal is decomposed into a real part (Real) and an imaginary part (Imag) of the signal point. The real part is input to the real part inverse fast Fourier transform unit 55, the imaginary part is input to the imaginary part inverse fast Fourier transform unit 56, and each of the real parts is subjected to inverse fast Fourier transform, and the converted output signal is converted into a time axis copy window. In the function multipliers 57 and 58, the signal on the time axis is copied, multiplied by the window function according to the time response waveform of the transmission Nyquist filter, and in the 1/2 Nyquist time length delay unit 59, the real part side And the imaginary part side are time-shifted by 1/2 Nyquist time length, and then the waveform synthesis circuit 60 Combines the Rupert and imaginary part, enter the transmission modulator 72 through the transmission low-pass filter 71 shown in FIG. 14, multiplying the transmission carrier from the transmission carrier generating unit 73.

  FIG. 17 shows the relationship between the real part inverse fast Fourier transform unit (Real-part IFFT) 55, the imaginary part inverse fast Fourier transform unit (Imag-part IFFT) 56 and the waveform synthesis circuit 60 in FIG. It is explanatory drawing of the function of the time-axis copy window function multiplication parts 57 and 58 and the 1/2 Nyquist time length delay part 59, and, as mentioned above, the real part and imaginary of the vector signal point from the transmission signal point generation circuit 51 The parts are input to the real part inverse fast Fourier transform unit 55 and the imaginary part inverse fast Fourier transform unit 56, respectively, converted into signal components on the time axis, and multiplied by the time response waveform of the transmission Nyquist filter as a window function. Then, the real part side and the imaginary part side are time-shifted by ½ Nyquist time length. This state is indicated by a symbol array of IFFT having a Nyquist time length and an impulse response waveform. The waveform synthesis circuit 60 outputs a synthesized signal by performing vector addition on the real part side and the imaginary part side. In addition, since the continuously input transmission data is delayed by one Nyquist time length on the time axis, it is added to the previous waveform in the form shifted by one Nyquist time length, and the addition output is the transmission base. It becomes a band signal.

  18 shows a main part including the modulation processing means in the multiplex processing unit in FIG. 1, 74 is a signal point generation circuit, 75 is a transmission IFFT unit, 76 is a transmission LPF unit, 77 is a transmission MOD unit, Reference numeral 78 denotes a transmission CRR unit, which has a configuration corresponding to the signal point generation unit 14, the inverse fast Fourier transform unit 15, the low-pass filter 16, the modulation unit 17, and the transmission carrier generation unit in FIG. Further, the signal point generation circuit 74 corresponds to the transmission signal point generation circuit 51 of FIG. 16, and the transmission IFFT unit 75 corresponds to the transmission IFFT unit 52 of FIG. As described above, the transmission data input to the signal point generation circuit 74 is separated into a real part and an imaginary part, converted into a baseband time waveform by the transmission IFFT unit 75, and an unnecessary band by the transmission LPF unit 76. In the transmission MOD unit 77, the signal is modulated by the carrier frequency signal from the transmission CRR unit 78 to be a transmission signal input to the analog unit 2 (see FIG. 1).

  19 shows the main part including the demodulation processing means in the demultiplexing processing unit in FIG. 1, 84 is a signal point determination circuit, 85 is a reception FFT unit, 86 is a reception LPF unit, and 87 is a reception DEM unit. , 88 denote reception CRR units, which respectively correspond to the signal point determination unit 24, fast Fourier transform unit 25, low-pass filter 26, demodulation unit 27, and reception carrier generation unit 28 in FIG. The reception signal converted into a digital signal from the analog unit 2 (see FIG. 1) is input to the reception DEM unit 87, demodulated by the carrier signal from the reception CRR unit 88, unnecessary band is removed by the reception LPF unit 86, and reception is performed. The signal is subjected to Fourier transform by the FFT unit 85 to be a frequency domain signal, the signal point is determined by the signal point determination circuit 84, and is received data, which is input to the difference circuit 23 (see FIG. 1). A difference process opposite to the sum process is performed.

For reception demodulation, it is originally demodulated by each carrier signal E (jωt), and a reception signal point is obtained via a waveform shaping filter. This calculation is performed by converting a reception signal sequence (impulse sequence) When R (k−m),... R (k + m), first, carrier signals E (jω (tp)),..., E (jω (t + p)) are multiplied,
R (k−m) * E (jω (tp)),..., R (k + m) * E (jω (t + p))
Further, the waveform shaping filter coefficients C + n,..., C-n are multiplied to obtain a filter output F represented by the following equation.
F = Σ [R (k−m) * E (jω (tp)) * C + n +... + R (k + m) * E
(Jω (t + p)) * C−n] (010)

  The above equation is obtained by decomposing the signal sequence obtained by multiplying the received signal sequence R by the window function based on the time response waveform C of the waveform shaping filter on the frequency axis by fast Fourier transform, and adding it on the time axis (convolution integration) If this is implemented, it is shown that the received waveform shaping filter process can be processed very easily. On the transmitting side, the imaginary component is transmitted in a form shifted by ½ Nyquist time length. Therefore, on the receiving side, if the output calculation is performed at the double Nyquist frequency interval in the reception FFT processing unit 85. The received data can be reproduced. Specifically, it can be processed as a waveform shown in FIG.

  20 and 21 are explanatory diagrams of the reception FFT unit. The same reference numerals as those in FIG. 2 denote the same parts, and reference numeral 89 in FIG. 21 denotes a window function multiplication circuit / FFT / Σ. As described above, this is a functional block for explaining the operation of the synthesis circuit 62, the fast Fourier transform unit 63, and the window function multiplication circuit 64 (means for multiplying the window function) in FIG. The received signal is multiplied by a window function (time response waveform of the received Nyquist filter) in the window function multiplication circuit 64, and the result of multiplication with this window function is cut out at a Nyquist time interval and added, and fast Fourier transform is performed. The unit 63 performs FFT processing to obtain individual frequency information. Then, in the synthesis circuit 62, the FFT output is added (convolution integration of the filter) for the time length of the Nyquist filter to obtain a desired filter output. Since the real part and the imaginary part are each shifted by ½ Nyquist time length, the multiplication of the received signal and the window function is performed at an interval twice the Nyquist frequency (time axis of the window function). Are shifted by 1/2 Nyquist time interval). As a result, after the FFT, in order to obtain a desired real part signal / imaginary part signal in the synthesis circuit 62, these are simply synthesized in the signal extraction / synthesis circuit 61 to obtain a desired received signal point. As described above, the demultiplexing processing unit is a means for multiplying the time response waveform of the received Nyquist filter as a window function, and a first means for performing fast Fourier transform on the output signal of this means at a Nyquist time interval and adding it. And the second means for multiplying the window functions by shifting the window length by 1/2 Nyquist time, adding by performing fast Fourier transform, and extracting the real part and the imaginary part from each added output signal to determine the signal point. Means.

FIG. 22 is an explanatory diagram of the relationship between the number of channels and the frequency. In the frequency band of 6 channels, 5 of channels CH-2 to CH- + 2 with a Nyquist frequency interval centered on the channel CH-0. It shows that multiplexing for channels is possible. Therefore, the transmission efficiency Ea in this case is
Ea = (5/6) = 83.3 [%] (011)
Similarly, when 99 channels are multiplexed,
Ea = 99/100 = 99.0 [%] (012)
Thus, by increasing the number of multiplexing, highly efficient data transmission becomes possible.

  FIG. 23 is an explanatory diagram of specific band leakage reduction. In the case of interference prevention for a specific band in a frequency band by multiple channel multiplexing, for example, at least two channels are notch ( (Leakage reduction) can be performed.

  FIG. 24 is an explanatory diagram of unnecessary band suppression. The unnecessary band of each channel is cut (suppressed) by the Nyquist filter on the receiving side, whereby noise components due to the unnecessary band can be suppressed. This amount of noise suppression is determined by the filter characteristics (filter coefficient and number of taps), but can be optimized according to the requirements on the system side.

  FIG. 25 is an explanatory diagram of interference cancellation between adjacent channels, which can be applied, for example, when performing interference cancellation between adjacent channels at the time of training. For example, the even channels CH + 0, CH-2, and CH + 2 include ( (1, 1, 1, -1), and the odd channels CH + 1 and CH-1 are transmitted in the sequence (1, -1, 1, 1). The channel transmitted in the sequence of (1, -1) is received at (1, 1, 1, -1), and the channel transmitted in (1, -1, 1, 1) is (1, -1). , 1, 1). That is, since the adjacent channels can be transmitted orthogonally, the reception side can restore the received signal without interference between adjacent channels. When this means is applied, the transmission speed will be reduced by half, but it is possible to stably extract timing signals, carrier signals, etc. mainly by applying it to training signals transmitted and received prior to data transmission. can do.

  FIG. 26 shows an equivalent circuit of a low-pass filter, which can be applied to the above-described transmission low-pass filter and reception low-pass filter, where T is a delay circuit, Σ is an adder circuit, Cn,... C0,. C + n represents a tap coefficient.

  27 shows the transmission modulation section (MOD) 17 in FIG. 1, FIG. 28 shows the reception demodulation section (DEM) 27 in FIG. 1, cos θ and −sin θ indicate the center carrier, and the transmission modulation section In this case, the result of multiplication of the real part (input Real) and cos θ, and the result of multiplication of the imaginary part (input Imag) and −sin θ are combined into a modulation output signal. The reception demodulator multiplies the input signal by cos θ and −sin θ, respectively, and outputs a real part (output Real) and an imaginary part (output Imag).

  On the transmission side, the window function processing is performed by multiplying the IFFT output by the time response waveform of the transmission Nyquist filter as it is. On the reception side, the window function processing on the reception side is performed by multiplying the reception signal by the time response waveform of the reception Nyquist filter as it is, and then performing FFT processing. FIG. 29 shows an outline of the transmission / reception filter characteristics in this case. In the figure, the vertical axis represents amplitude characteristics, the horizontal axis represents frequency, and the Nyquist frequency interval. SBFRM indicates a subframe, and this time length is made equal to the Nyquist time length. That is, 2SBFRM indicates a filter characteristic in which the time response waveform length of the filter is set to a time length twice that of Nyquist. 8SBFRM has a filter characteristic having a Nyquist time length of 8 times. For this reason, if the number of SBFRM is large, the filter characteristics are good, but the processing becomes heavy as the number of taps increases. As is apparent from the figure, it is shown that the processing of the time length of 8SBFRM is insufficient to achieve the target of 70 dB.

  In general, if the filter function is subjected to time axis window function processing, components outside the unnecessary band can be improved. Typical window functions include square wave / triangular wave / Hanning window / Humming window / Blackman window / flat top window. Among these, those having excellent out-of-band characteristics include Hanning window / Blackman window / flat top window. Therefore, there are two main purposes: the purpose of multiplex transmission being Nyquist transmission and the reduction / removal of components outside unnecessary bands as much as possible. This is the first data transmission part. This is because it is sufficient that the 1st peak component is about 40 dB or less in the transmission / reception combining characteristic even in the case of 1024-value transmission. From the characteristics shown in (2), it is sufficient if there is a filter having a time length of 2SBFRM. Therefore, regarding the time portion exceeding the time length of 2SBFRM, for example, it is considered that the unnecessary band can be efficiently reduced / removed by multiplying the coefficient of the Hanning window.

  FIG. 30 is an explanatory diagram of the window function, where the vertical axis represents normalized amplitude, the horizontal axis represents frequency, the Nyquist time interval centered on 0, the window function time waveform, and the filter coefficient before and after the window function multiplication. Indicates. During the ± 1.5 Nyquist time length, the window function is multiplied by a value of 1.0 to ensure characteristics as a transmission line. For the portion where the window function exceeds ± 1.5 Nyquist time length, in order to reduce / remove unnecessary out-of-band components, the Hanning window function is multiplied to reduce / remove out of the unnecessary band. In this case, the central portion of the time response waveform is divided into regions of the central portion and both sides of the central portion, and the central partial region is a square window function, and the both side partial regions are a Hanning window function or a similar window function.

  FIG. 31 is an explanatory diagram of filter characteristics depending on the presence or absence of window function multiplication. Like FIG. 29, the vertical axis represents amplitude characteristics, the horizontal axis represents frequency, and the Nyquist frequency interval. In the case of only a rectangular window, the characteristic is a thin line, and the characteristic of a thick line is obtained by applying the function of the unique window shown in FIG. Therefore, in the vicinity of the Nyquist frequency interval 2, the target of 70 dB is achieved. Accordingly, it is possible to reduce leakage in a specific band on the transmission side and to suppress noise in the case of huge tone noise on the reception side.

  Noise suppression can be quite effective with respect to unnecessary components outside the band as seen from individual channels. However, there is no power for huge tone noise mixed in the same band. In this case, noise cancellation is performed by applying noise cancellation or the like with respect to the narrow-band giant tone noise mixed in the band.

  FIG. 32 is an explanatory diagram of a main part to which the noise canceling means is applied. The same reference numerals as those in FIGS. 18 and 19 denote the same parts, 91 is a signal point generator, 92 is a transmission zero point insertion circuit, and 93 is A reception noise cancellation circuit 94 indicates an FFT unit. On the transmission side, the signal point generation circuit 74 includes a signal point generation unit 91 and a transmission zero point insertion circuit 92. After the transmission signal point is generated by the signal point generation unit 91, Then, a zero point is inserted between the signal points, and a transmission signal is obtained through the transmission IFFT unit 75, the transmission LPF unit 76, and the transmission MOD unit 77 by the above-described means.

  On the reception side, the reception FFT unit 85 is configured by a reception noise cancellation circuit 93 and an FFT unit 94, and a received signal demodulated via the reception DEM unit 87 and the reception LPF unit 86 is input to the reception FFT unit 85. The Fourier transform is performed by the FFT unit 94, and the noise component on the zero point is extracted in the reception noise cancellation circuit 93, the noise component on the signal point between the zero points is interpolated and predicted, and the noise component on the signal point is calculated. The signal is removed and input to the signal point determination circuit 84. Basic means for performing noise cancellation processing on the receiving side by inserting the zero point is described in detail in the above-mentioned Patent Document 4 (Japanese Patent Laid-Open No. 2002-164801), and redundant description is omitted. In the present invention, as described above, in the method of orthogonal transmission on the time axis and the frequency axis, zero points are inserted alternately on the real part side and the imaginary part side. Similarly, on the receiving side, signal points appear alternately and zero points appear alternately. In the reception cancel circuit 93, noise thinning and interpolation prediction are performed in consideration of such points. Will be processed.

  FIG. 33 is an explanatory diagram of a main part to which a multipath countermeasure has been taken. The same reference numerals as those in FIG. 32 denote the same name parts, and 95 denotes a decision feedback type automatic equalizer. This decision feedback type automatic equalizer 95 removes the noise component on the signal point by the reception noise cancellation circuit 93 and inputs it, and feeds back the decision information of the signal point decision circuit 84 to perform equalization processing. . In various data transmission paths, reception distortion may occur due to multipath of the transmission paths. With respect to this multipath, the OFDM scheme provides a guard time to implement a multipath countermeasure. In ISDN, since the Nyquist time length is shorter than the multipath time length, a countermeasure is implemented by using a decision feedback type automatic equalizer. In PHS, since the Nyquist time length is sufficiently longer than the multipath time length, no particular measures are taken.

  As described above, the present invention is based on Nyquist transmission and is time-axis orthogonal / frequency-axis orthogonal. Therefore, providing a guard time as in OFDM is not a good measure for performing highly efficient data transmission. When the Nyquist time interval is larger than the multipath time interval (for example, the multipath time length in a megahertz band PLC (Power Line Communication) is about 2 μs at the maximum, the Nyquist time length is set to 2 Even when a decision feedback type automatic equalizer is provided (when doubled 4 μs), the tap coefficient does not grow (there is no value that can be grown). For this reason, it is conceivable to set the Nyquist time length sufficiently longer than the multipath time length as one means for countermeasures against multipath. In the case where the multi-value conversion rate is increased, and in addition, when considerable accuracy is required, it is preferable to provide a decision feedback type automatic equalizer 95 as shown in FIG.

  When multiple channels are multiplexed on the frequency axis, the timing frequency is determined by the transmission timing of the master station modem, so it may be one, but the timing phase depends on the group delay characteristics of individual transmission paths. Strictly speaking, time equalization is required. This time equalization can be received irrespective of the timing phase by shifting the LPF coefficient on the time axis to adjust the timing phase, or using a double sampling type automatic equalizer that is earlier than the Nyquist interval, for example. Either of these can be applied. For example, a time equalizer corresponding to a channel can be provided to equalize the group delay characteristics.

  FIG. 34 is an explanatory diagram of a main part to which means for adjusting the timing phase is applied. The same reference numerals as those in FIG. 33 denote the same names, 96 is a time equalization circuit, and 97 is a TIP (timing interpolation) phase. An adjustment unit 98 is a TIM (timing) extraction unit. This time equalization circuit 96 is provided between the FFT unit 94 and the reception noise cancellation circuit 93. The timing phase corresponding to the channel of the output of the FFT unit 94 is extracted by the TIM extraction unit 98, and the phase adjustment is performed in the TIP phase adjustment unit 97 so that the extraction result becomes a predetermined phase. The TIP phase adjustment unit 97 can be configured by a transversal filter similar to that shown in FIG. 26, for example. The timing phase is adjusted by moving the filter coefficient with time. Thereby, time equalization with respect to the group delay distortion of the transmission line can be performed. The detailed description of this time equalization is described in the above-mentioned Patent Document 7 (Japanese Patent Application Laid-Open No. 2003-324360), and thus a duplicate description is omitted.

  FIG. 35 is an explanatory diagram of a main part to which the error correction means is applied. The same reference numerals as those in FIGS. 32 to 34 denote the same name parts, 99 is a transmission error correction unit, 100 is a signal point determination unit, and 101 is reception. Indicates an error correction section. The transmission error correction unit 99 is provided on the transmission side, and the reception error correction unit 101 is provided on the reception side. In the power line carrier system, amplitude characteristics / group delay characteristics / losses are shown. The characteristic / signal-to-noise characteristic changes greatly along the frequency axis, and the data transmission quality has a large correlation with the frequency when the transmission path is determined. For this reason, data is transmitted with redundancy depending on frequency on the transmission side, and data transmission quality detection depending on individual frequencies (channels) is made on the reception side using redundancy added on the transmission side. By providing the means (SQD circuit), it is possible to perform strong error correction on the receiving side. The operation in the data transmission by the transmission error correction unit 99 and the reception error correction unit 101 in this case is described in, for example, the above-mentioned Patent Document 6 (Japanese Patent Laid-Open No. 2003-134095), and thus redundant description. Is omitted.

  In the power line carrier system, there are multipath / transmission path loss / group delay distortion associated with multi-branch connection. In addition, there is noise associated with incoming radio waves from home appliances / existing radio stations. These deterioration factors for data transmission change from moment to moment depending on the connection state of connected home appliances and the operating conditions, and therefore transmission at a specific frequency may not be guaranteed. For this reason, in order to realize stable data transmission, one solution is to perform information transmission over a plurality of frequencies. As a specific means, there is a spread spectrum. Since the transmission quality using the power line as a transmission line has a strong correlation with the frequency, it is a good idea to spread the spectrum without depending on the frequency.

  Also, switching noise generated from home appliances such as inverters is often a large number of harmonic groups. For this reason, when performing spread spectrum, it is desirable to arrange the selected frequencies irregularly (randomly) rather than regularly (for example, integer multiple intervals). Further, since the distortion of the transmission line covers a wide band, it is advantageous to arrange it over a wide range, not locally. For example, when there are 49 channels and 7 times the spread spectrum is performed, the frequency is roughly divided in units of 7 units, and among these 7 units, further selected by 7PN, the frequency axis is random and almost over a wide band. It is a good idea to achieve uniform transmission and improve transmission quality.

  FIG. 36 is an explanatory diagram of a main part for transmitting and receiving data by performing spectral dispersion on a plurality of channels with the multiplexing unit on the transmission side, and FIG. 37, respectively, with the demultiplexing unit on the reception side. For example, the signal A from the signal point generation unit 91 of the signal point generation circuit 74 described above is spread-modulated and transmitted in a state of being dispersed in the channels CH0, CH5, CH10, CH15, CH16, CH21, CH26, and CH31. In this case, if there are four types of modulation points MOD0 to MOD3, the modulation points are also made different depending on the channel.

  In FIG. 37, the received signal point determination & SQD (signal quality) part of the signal point determination circuit 84 for the signals A of the received channels CH0, CH5, CH10, CH15, CH16, CH21, CH26, and CH31. The determination result is weighted, added by the addition unit (Σ), and determined by the signal point determination unit 100 to be received data. The signal quality (SQD) in this case differs depending on the channel conditions CH0 to CH31 having different frequencies due to different transmission path conditions including noise and the like. The better the signal quality (SQD), the higher the weighting. It is possible to dramatically improve the transmission quality. In this case, the multiplexing processing unit multiplexes the signal to be transmitted on either or both of the frequency axis and the time axis in a spread state, and the demultiplexing processing unit performs signal point determination corresponding to the spread channel. A means for performing signal point determination again on the addition result or a signal point determination corresponding to the spread channel, and multiplying each coefficient as a weight corresponding to transmission quality (signal quality SQD) and adding the result And it can be set as the structure which has a means to perform signal point determination again with respect to the addition result.

  FIG. 38 is an explanatory diagram of Embodiment 2 of the present invention, showing the main parts of the multiplexing processing section and the demultiplexing processing section, and 51 for generating a transmission signal corresponding to the signal point generating section 14 in FIG. 1, 52 is a transmission IFFT unit corresponding to the inverse fast Fourier transform unit 15 in FIG. 1, 53 is a reception signal point determination circuit corresponding to the signal point determination unit 24 in FIG. 1, and 54 is in FIG. A reception FFT unit corresponding to the fast Fourier transform unit 25 is shown. Reference numerals 111 and 112 denote IFFT units, 113 denotes a time axis copy window function multiplication unit, 114 denotes a waveform synthesis circuit, 115 and 116 denote convolution synthesis units (Σ-convolution), 117 and 118 denote FFT units, and 119 denotes a window function multiplication circuit. Show.

  In the configuration shown in FIG. 2, the transmission signal point generation circuit 51 shows a case where the signal points corresponding to the transmission data are processed separately for the real part and the imaginary part. In this case, the even channel and the odd channel are divided and input to the IFFT units 111 and 112 of the transmission IFFT unit 52, respectively, and the signal on the frequency axis is converted into the signal on the time axis, and the time axis copy window function multiplication is performed. Input to the unit 113, the signals of the even channel and the odd channel are multiplied by the same window function as in the first embodiment, and one of them is delayed by ½ Nyquist time length, The waveform synthesis circuit 114 synthesizes and outputs the signals of the even channel and the odd channel.

  In the demultiplexing processing unit on the reception side, signals of the even channel and odd channel are multiplied by a window function corresponding to the window function on the transmission side by the window function multiplication circuit 115, and the even channel and odd channel are In order to restore the process in which either one of the above is delayed by 1/2 Nyquist time length on the transmission side, 1/2 Nyquist time length delay is performed, and signals on the time axis are respectively obtained by the FFT units 117 and 118. Is converted into a signal on the frequency axis, synthesized by the convolutional synthesis units 115 and 116, and the reception signal point determination circuit 53 determines each signal point of the even channel and the odd channel to obtain reception data.

  Due to the time shift of 1/2 Nyquist time length between the even channel and the odd channel, there is no interference between FIG. 13 as in the case of the time shift of 1/2 Nyquist time length for the real part and the imaginary part. Thus, multiplex transmission is possible.

  In the first and second embodiments of the present invention, the multiplex transmission apparatus can be configured to include only one of the multiplexing processing unit and the demultiplexing processing unit. Thus, a multiplex transmission apparatus having a transmission side multiplexing processing unit as a main part or a multiplex transmission apparatus having a reception side demultiplexing processing part as a main part can be provided. Similarly, only one of the multiplex transmission methods can be applied.

It is explanatory drawing of Example 1 of this invention. It is principal part explanatory drawing of Example 1 of this invention. It is explanatory drawing of the time response waveform of a transmission line. It is waveform explanatory drawing of Nyquist transmission. It is frequency characteristic explanatory drawing of a Nyquist transmission line. It is image explanatory drawing of orthogonal frequency division multiplexing. It is time response waveform explanatory drawing of a transmission / reception filter. It is time response waveform explanatory drawing of a cos filter. It is time response waveform explanatory drawing of a cos square filter. It is interference explanatory drawing between adjacent channels. It is interference explanatory drawing between adjacent channels. It is interference explanatory drawing between adjacent channels. It is interference explanatory drawing by 1/2 Nyquist time length shift. It is explanatory drawing of a transmission modulation part. It is waveform explanatory drawing of a transmission modulation part. It is explanatory drawing of a transmission IFFT part. It is function explanatory drawing of a transmission IFFT part. It is principal part explanatory drawing of the transmission side. It is principal part explanatory drawing on the receiving side. It is explanatory drawing of a reception FTT part. It is function explanatory drawing of a reception FTT part. It is explanatory drawing of transmission efficiency. It is explanatory drawing of specific band leak reduction. It is explanatory drawing of noise suppression. It is explanatory drawing of the interference removal between adjacent channels. It is explanatory drawing of a low-pass filter. It is explanatory drawing of a transmission modulation part. It is explanatory drawing of a reception demodulation part. It is filter characteristic explanatory drawing in the case of no window function. It is explanatory drawing of a window function and a filter coefficient. It is filter characteristic explanatory drawing in the case of window function multiplication. It is principal part explanatory drawing on the transmission / reception side to which the noise cancellation means is applied. It is principal part explanatory drawing of the transmission / reception side to which the multipath countermeasure is applied. It is principal part explanatory drawing of the transmission / reception side to which timing phase adjustment is applied. It is principal part explanatory drawing of the transmission / reception side to which error correction is applied. It is explanatory drawing of the frequency spreading | diffusion on the transmission side. It is explanatory drawing of the frequency spreading | diffusion on the receiving side. It is principal part explanatory drawing of Example 2 of this invention. It is explanatory drawing of a target specification.

Explanation of symbols

1 Digital part 2 Analog part 3 Power supply part 4 Common mode choke coil (CMC)
11 Bridge circuit 12 Scramble circuit (SCR)
13 Summing Circuit 14 Signal Point Generating Unit 15 Inverse Fast Fourier Transform Unit (IFFT)
16 Low-pass filter (LPF1)
17 Modulator (MOD)
18 Transmission carrier generator (transmission CRR)
22 Descramble circuit (DSCR)
23 Difference circuit 24 Signal point determination unit 25 Fast Fourier transform unit (FFT)
26 Low-pass filter (LPF4)
27 Demodulator (DEM)
28 Received carrier generator (received CRR)
29 Timing synchronization unit (TIMPLL)

Claims (20)

In a multiplex transmission apparatus having a data multiplexing processing section or a multiplex transmission apparatus having a data multiplexing processing section and a demultiplexing processing section ,
The multiplexing processing unit divides a signal point generation unit for modulating the data, and a signal point generated by the signal point generation unit into a real part and an imaginary part, and has a Nyquist time interval on the time axis and On the frequency axis, each is arranged and multiplexed at Nyquist frequency intervals, and the waveform is obtained by shifting one of the multiplexed output signals of the real part and the imaginary part by ½ Nyquist time length with respect to the other. A multiplexing transmission apparatus comprising means for combining . The means for multiplexing of the multiplexing processing unit includes means for multiplying the inverse fast Fourier transform output signal of the signal point of the data generated by the signal point generating means with the time response waveform of the transmission Nyquist filter as a window function, 2. The multiplex transmission apparatus according to claim 1 , further comprising means for sequentially adding the output signals of the means on the time axis . In a multiplex transmission apparatus having a data multiplexing processing section or a multiplex transmission apparatus having a data multiplexing processing section and a demultiplexing processing section,
The multiplexing processing unit includes a signal point generating unit for modulating the data, and a signal point generated by the signal point generating unit, with a Nyquist time interval on the time axis and a plurality of carrier frequencies on the frequency axis. The signal points of the data generated by the signal point generating means are sequentially distributed to the even channel and the odd channel, and the window functions for the even channel and the odd channel are ½ Nyquist time mutually. And a means for synthesizing a waveform by multiplying each by a time difference.
A multiplex transmission apparatus characterized by that . The multiplexing processing unit has a configuration including means for selecting and multiplexing the data generated by the signal point generating means with respect to the adjacent channel so that the data signal waveform and the interference waveform of the adjacent channel are orthogonal to each other. The multiplex transmission apparatus according to any one of claims 1 to 3 . In a multiplex transmission apparatus having a data demultiplexing processing section or a multiplex transmission apparatus having a data multiplexing processing section and a demultiplexing processing section,
The demultiplexing processing unit includes means for multiplying the time response waveform of the received Nyquist filter as a window function, first means for performing fast Fourier transform on the output signal of the means at a Nyquist time interval, and adding the window. A second means for multiplying a function by shifting the function by 1/2 Nyquist time, adding by performing a fast Fourier transform, and extracting a real part and an imaginary part from the output signals of the first and second means to obtain a signal point And a means for making a determination
A multiplex transmission apparatus characterized by that . In a multiplex transmission apparatus having a data demultiplexing processing section or a multiplex transmission apparatus having a data multiplexing processing section and a demultiplexing processing section,
The demultiplexing processing unit multiplies the time response waveform of the reception Nyquist filter with a time difference of ½ Nyquist time for each of the even-numbered channel and the odd-numbered channel as a window function, and performs a fast Fourier with a Nyquist time interval for each multiplication output. It has a configuration including means for performing conversion and adding and performing signal point determination corresponding to the even channel and the odd channel
A multiplex transmission apparatus characterized by that . The means for multiplying the window function divides the window function into regions of a central portion of the time response waveform and both side portions of the central portion, the window function of the central portion region is a rectangular window function, and the both side partial regions The final window function is a coefficient obtained by multiplying a window function similar to the Hanning window function or a window function similar to the Hanning window function by the time response waveform of the Nyquist filter. The multiplex transmission apparatus described. The multiplexing processing unit includes means for spreading and transmitting a signal to be transmitted on one or both of a frequency axis and a time axis, and the demultiplexing processing unit determines a signal point corresponding to the signal of the spread channel. 8. The multiplex transmission apparatus according to claim 1, further comprising means for performing signal addition and performing signal point determination on the addition result again . The multiplexing processing unit includes means for spreading and transmitting a signal to be transmitted on one or both of a frequency axis and a time axis, and the demultiplexing processing unit determines a signal point corresponding to the signal of the spread channel. 9. The method of claim 1, further comprising means for multiplying and adding a coefficient corresponding to the transmission quality corresponding to the channel and performing signal point determination again on the result of the addition . 2. A multiplex transmission apparatus according to item 1. The multiplexing / demultiplexing unit according to claim 5 or 6, further comprising means for extracting a timing phase of a channel-corresponding signal subjected to reception demodulation and fast Fourier transform, and adjusting the timing phase. Transmission equipment. In a multiplex transmission method using a multiplex transmission device having a data multiplexing processing unit or a multiplex transmission device having a data multiplexing processing unit and a demultiplexing processing unit,
A signal point for modulating the data is generated by the signal point generating means of the multiplexing processing unit, the signal point is divided into a real part and an imaginary part, and the time axis is the Nyquist time interval and the frequency axis Are arranged at Nyquist frequency intervals and multiplexed, and either one of the output signals multiplexed by the real part and the imaginary part is shifted by 1/2 Nyquist time length with respect to the other to synthesize the waveform. Includes a process of multiplexing so that the axis is orthogonal and the frequency axis is orthogonal
And a multiplex transmission method. The process of multiplying the inverse fast Fourier transform output signal of the signal point of the data generated by the signal point generating means by using the time response waveform of the transmission Nyquist filter as a window function and sequentially adding the output signals by the multiplication on the time axis 12. The multiplex transmission method according to claim 11, further comprising: In a multiplex transmission method using a multiplex transmission device having a data multiplexing processing unit or a multiplex transmission device having a data multiplexing processing unit and a demultiplexing processing unit,
A signal point for modulating the data is generated by the signal point generation means of the multiplexing processing unit, and the signal point is divided into Nyquist time intervals on the time axis and a plurality of carrier frequencies on the frequency axis. The signal points of the data generated by the signal point generating means are sequentially distributed to the even channel and the odd channel, and the window functions for the even channel and the odd channel are mutually divided by a time difference of 1/2 Nyquist time. Includes the process of waveform synthesis by multiplying each
And a multiplex transmission method. 14. The method of claim 11, further comprising selecting and multiplexing the data generated by the signal point generating means so that the data signal waveform and the interference waveform of the adjacent channel are orthogonal to each other. The multiplex transmission method according to any one of the preceding claims. In a multiplex transmission method using a multiplex transmission apparatus having a data demultiplexing processing unit or a multiplex transmission apparatus having a data multiplexing processing unit and a demultiplexing processing unit,
The output signal obtained by multiplying the time response waveform of the received Nyquist filter as a window function by the demultiplexing process in the demultiplexing processing unit is fast Fourier transformed at the Nyquist time interval and added by the first means, and The window function is multiplied by 1/2 Nyquist time length, multiplied by a fast Fourier transform and added by a second means, and a real part and an imaginary part are extracted from the output signals of the first and second means. A multiplex transmission method comprising the step of performing signal point determination . In a multiplex transmission method using a multiplex transmission apparatus having a data demultiplexing processing unit or a multiplex transmission apparatus having a data multiplexing processing unit and a demultiplexing processing unit,
In the demultiplexing process in the multi-division processing unit, the time response waveform of the reception Nyquist filter having a time difference of 1/2 Nyquist time is multiplied as a window function for each of the even channel and the odd channel, Including a process of performing a fast Fourier transform of Nyquist time intervals on the multiplication output and performing addition, and performing signal point determination corresponding to the even channel and the odd channel
And a multiplex transmission method. In the process of multiplying the window function, the window function is divided into a central portion of the time response waveform and regions of both sides of the central portion, and the time response waveform of the central portion region is a square window function, 14. The final window function is a coefficient obtained by multiplying a window function of the both side partial regions by a Hanning window function or a window function similar to the Hanning window function by a time response waveform of a Nyquist filter. Or a multiplex transmission method according to 15 or 16 ; In the multiplexing process, a signal to be transmitted is spread and transmitted on one or both of the frequency axis and the time axis, and in the demultiplexing process, the signal of the spread channel is handled. 18. The multiplex transmission method according to claim 11, further comprising a step of performing signal point determination and adding, and performing signal point determination again on the result of the addition . In the multiplexing process, the process includes a step of spreading and transmitting a signal to be transmitted on one or both of the frequency axis and the time axis, and in the demultiplexing process, a signal corresponding to the signal of the spread channel is transmitted. 19. The method of performing point determination, multiplying and adding a coefficient corresponding to transmission quality corresponding to the channel, and performing signal point determination again on the result of the addition. 2. A multiplex transmission method according to claim 1. 17. The demultiplexing process includes a step of extracting a timing phase of a signal corresponding to a channel subjected to reception demodulation and fast Fourier transform, and adjusting the timing phase . Multiplex transmission method.

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