JP3416863B2 - Power supply - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for generating an AC voltage from a DC power supply using a semiconductor element and supplying power to a load side through a small high frequency transformer or a coupling coil. Personnel, especially AV such as TV and video
The present invention relates to a power supply device suitable for use in equipment, cordless telephones, communication equipment such as communication equipment, small electric appliances such as electric toothbrushes and electric shavers, electric tools and outdoor goods, and in-vehicle equipment.

[0002]

2. Description of the Related Art Conventional Example (Japanese Patent Laid-Open No. 4-295284) As shown in FIG. 60, a capacitor C2 is charged from a power source 3 through a resistor R8, and this voltage is made equal to or higher than a threshold voltage of FET1 to start oscillation. While FET1 is close to conduction,
The electric charge of the capacitor C2 is transferred to the resistor R210 and the diode D1.
10, by discharging through FET1, power supply 3
Even if the voltage, the coupling coil, the gap of the transformer T, or the like changes, the class C self-excited oscillation operation is always performed stably. This enables conventional commercial power supply transformation, rectification,
Compared with a method such as DC output by smoothing, a self-excited oscillation operation is performed at a high frequency, so that a simple and small power supply device is realized with a small number of parts.

Conventional example (JP-A-6-70461) As shown in FIG. 61, in the conventional circuit of FIG. 60, a resistor R is further provided between the source of the FET1 and the ground line.
310 and R320 are connected in series, and the transistor Q1 is connected between the gate of the FET1 and the ground line.
The base of the transistor Q1 is connected to the connection point of the resistors R310 and R320. In the conventional example, the ON time of the FET1 becomes long when the power is turned on, and the exciting current I _D flowing during that time also becomes large. Therefore, when the FET1 is turned off, a high level flyback is generated due to the energy accumulated in the primary winding L1. There is a risk that a voltage is generated and the FET 1 is destroyed. However, the above configuration prevents this.

The operation of the conventional example will be described. The bias voltage obtained by charging the capacitor C2 is FET1.
When the threshold voltage of is reached, FET1 is turned on.
When the bias voltage V _G2 becomes higher than the drain voltage V _{F of the} FET1 during the ON period of the FET1, the bias control circuit including the series circuit of the resistor R210 and the diode D110 causes the capacitor C to pass through the bias control circuit and the FET1.
The electric charge of 2 is discharged to stabilize the bias voltage V _G2 and stabilize the oscillation operation. When the voltage between the resistors R310 and R320 rises due to the exciting current I _D and the base current is supplied to the transistor Q1, the transistor Q1 turns on, which lowers the gate voltage V _G of the FET1 and the FET1. Is turned off. In this way, the on-time of the FET1 when the power is turned on can be greatly shortened, so that the stored energy due to the exciting current I _D can be suppressed to an appropriate amount and the flyback voltage generated after the FET1 is turned off can be reduced.

[0005]

Problems of the Prior Art Bias voltage V _G2 is controlled by being discharged through resistor R210 and diode D110 when the level is higher than voltage V _F. During a transition such as power-on or a sudden change in load level, the amount of energy stored in the resonant circuit changes significantly, so that the level of the voltage V _F changes greatly, but the voltage V _F is not controlled.
The level of the bias voltage V _G2 also greatly changes accordingly. Therefore, the bias voltage V _G2 drops too much and the FET
In some cases, it becomes impossible to turn on 1 and resonance cannot be maintained, resulting in intermittent oscillation.

For example, during the first ON period of the FET1 when the power is turned on, the bias voltage V _G2 is discharged through the resistor R210 and the diode D110, and the voltage V _G induced in the feedback winding L3 is equal to or lower than the threshold voltage V _{th of the} FET1. FET1 continues to be turned on until. Since the ON period is long, the bias voltage V _G2 is excessively decreased as shown in FIG. 62, and the voltage V _G is not sufficiently increased and the voltage V _F is attenuated when the FET 1 is turned on next time. , Stable oscillation cannot be sustained. Further, when a commercial power supply is used as the input power supply, the peak voltage of the voltage V _F becomes several hundred volts to 1,000 volts, so that the diode D110 is used.
Therefore, it is necessary to use a high breakdown voltage element, which makes the element expensive.

Further, the circuit of the conventional example shown in FIG.
Since it is a voltage-current resonance type self-excited oscillation circuit, there is a problem in that the output greatly changes with changes in the power supply voltage. However, in general electric equipment,
It is desirable to obtain an almost constant output. The commercial power supply voltage is 100 V, 120 V, 200 V, 240 V, etc., and the voltage values thereof are greatly different in each country of the world. Therefore, in order to make an electric device using a power supply device obtained by rectifying and smoothing a commercial power supply to obtain a DC voltage available in the world, the fluctuation range of the power supply is usually designed to be ± 10%. The voltage stabilizing circuit and the like have to be added, resulting in a large increase in production cost, and the merit of the simple circuit is lost. Further, in recent years, electric toothbrushes and the like which are non-contact and transmit electric power by electromagnetic induction have been put into practical use in order to improve poor contact and to be used around water. It is desired to downsize and stabilize the output of such an electric device which is separated and attached in a non-contact manner and used by a particularly simple circuit configuration.

Problem of Conventional Example The resistance value of the resistor R32 is set so that the transistor Q1 is turned on when the exciting current I _D is below a predetermined level so that the excessive exciting current I _D does not flow and the FET 1 is destroyed when the power is turned on. ing. Therefore, the transistor Q1 does not operate during the oscillation period in the steady state.
On the other hand, a bias voltage V _{G2 is} generated by the bias control circuit.
However, as in the case of the conventional example, when the drain voltage V _F changes, the bias voltage V _G2 also changes accordingly, so that the FET1 cannot be turned on and the resonance state cannot be maintained. However, there was a problem that the oscillation became unstable.

By the way, in an electric device which can be detached and attached in a non-contact manner and used, the presence or absence of a load, the presence of a regular load and the presence of a foreign substance are discriminated from each other to ensure reliable and low cost. It is desired to perform electric power transmission, and it is also necessary to take preventive measures against abnormal heat generation when foreign matter is placed.

An object of the present invention is to solve the above problems and to provide a low-cost power supply device that stably oscillates.

It is another object of the present invention to provide a power supply device that is small in size, has a reduced number of parts, is low in cost, and keeps its output substantially constant even when the input voltage changes significantly.

It is another object of the present invention to provide a power supply device that keeps the oscillation amplitude constant and does not exceed the withstand voltage of the switching element even with respect to temperature fluctuations, load changes, and the like.

It is another object of the present invention to provide a non-contact type power supply device capable of discriminating the presence / absence of a load, a normal load and a foreign substance, and performing proper power transmission and prevention of power loss. To do.

[0014]

To achieve the above object of the Invention The present invention has a switching element and a feedback winding,
Connected to a switching control terminal of the switching element via the feedback winding, and a self-excited oscillation circuit for self-oscillating a resonance circuit composed of a primary winding of a transformer and a capacitor connected in series to the switching element. And a bias voltage generating circuit for supplying power to a load connected to the secondary side of the transformer when the power source is connected, a discharge loop is formed when the switching means is turned on.
To maintain a stable bias voltage,
All discharge current flowing through the switching means.
It is those (claim 1).

Further, the bias voltage generating circuit comprises a capacitor, and the switching means discharges the electric charge accumulated in the capacitor when turned on (claim 2).

The switching means is connected to a switching control terminal of the switching element (claim 3).

Further, the present invention is the power supply device according to claim 3, further comprising a feedback circuit for generating a feedback signal from an electric signal in one circuit and outputting it to the switching means (claim 4). .

Further, the feedback circuit is for generating a feedback signal according to the voltage level of the power supply (claim 5).

Further, the feedback circuit is for generating a feedback signal according to an electric signal generated on the primary side of the transformer (claim 6).

Further, the feedback circuit is for generating a feedback signal according to an electric signal generated on the secondary side of the transformer (claim 7).

The feedback circuit comprises at least one of a resistor and a constant voltage element (claim 8).

According to the present invention, in the power supply device according to the fourth aspect, a current detecting resistor for detecting a current flowing through the switching element, and a current detecting resistor interposed between the current detecting resistor and the switching control end of the switching means. The feedback circuit is configured so that the feedback signal is output between the switching control end of the switching means and the resistor (claim 9).

The feedback circuit keeps the resonance voltage of the resonance circuit constant (claim 1).
0).

The feedback circuit is an automatic control circuit that combines proportional control and integral control (claim 11).

Further, the feedback circuit has
It comprises a detection winding magnetically coupled to the next winding, a capacitor for smoothing the induced voltage in the detection winding, and a resistor connected in parallel with this capacitor.
2).

Further, the feedback circuit is the same as the above-mentioned 1
A detection winding magnetically coupled to the next winding is provided, and the feedback signal is output from the ON side of the detection winding (claim 13).

Further, according to the present invention, in the power supply device according to claim 1, the current control circuit is provided, and the current control circuit is provided with a circuit for feeding back a part of information of the device so as to make the output substantially constant. (Claim 14).

According to the present invention, in the power supply device according to claim 14, a detection circuit for detecting a state signal on the load side,
And a signal processing circuit for performing signal processing on the detected signal and guiding the signal to the feedback circuit (claim 15).

The signal processing circuit transmits the detected signal to the primary side via the transformer (claim 16).

The detection circuit detects the output current to the secondary side load (claim 17), the output voltage to the secondary side load (claim 18), or the secondary side. The output to the load may be detected (claim 19).

Further, the signal processing circuit includes a circuit for voltage-frequency converting the detection signal detected by the detection circuit,
The frequency of the signal transmitted via the transformer is
And a circuit for converting voltage (claim 2).
0).

Further, the signal transmission of the detection signal via the transformer is performed in parallel with the power transmission performed via the transformer (claim 21).

According to the present invention, in the power supply device according to the sixteenth aspect of the present invention, the period of signal transmission performed through the transformer for the detection signal and the period of electric power transmission performed through the transformer are alternately switched. A switching circuit is provided (claim 22).

Further, the transformer is detachable between the primary side and the secondary side having a load, and the electric power is transmitted in a contactless manner (claim 23).

According to the present invention, in the power supply unit according to the twenty-third aspect, a judging circuit for detecting whether or not the placed load is regular is provided, and the secondary circuit is provided only when the regular load is placed. Power is supplied to the side (claim 24).

The discriminating circuit may discriminate the load based on the information on the primary side (claim 25) or may discriminate the load based on information from the secondary side (claim 2).
6).

Further, it is preferable that the information on the primary side uses a change in inductance (claim 27), a change in resonance frequency (claim 28) and a change in current (claim 2).
9), or a change in voltage or the like (claim 30).

Further, the switching element is composed of a transistor, and a self-bias circuit for applying a self-bias between the base of the transistor and the ground is provided (claim 31).

Further, the self-bias circuit is provided between the emitter of the transistor and the current detection resistor connected to the emitter (claim 3).
2).

The self-biasing circuit is connected in series to the base of the transistor (claim 33).

Further, the self-bias circuit is a constant voltage element (claim 34).

The constant voltage element is a constant voltage diode (claim 35).

Further, the constant voltage element comprises a plurality of diodes connected in series (claim 36).

Further, according to the present invention, the switching element is composed of a transistor, and the ON period of the transistor is controlled by controlling the charging current of the bias circuit of the self-excited oscillation circuit (claim 37).

The control of the charging current of the bias circuit is performed by switching the resistance value (claim 38).

The control of the charging current of the bias circuit is performed by a current control element inserted in series with the bias circuit (claim 39).

Further, the current control element controls the charging current by an external control power source (claim 40).

Further, the external control power source is generated by an induced electromotive force of the transformer (claim 41).

Further, the feedback circuit is for generating a feedback signal for preventing an excessive current from flowing into the switching means when the power is turned on (claim 42).

Further, the feedback circuit generates a feedback signal for intermittently oscillating the oscillation of the switching element while the load is open (claim 43).

[0051]

According to the first aspect of the invention, when the power supply is connected and the bias voltage is applied to the switching control end of the switching element, self-excited oscillation is started and the load connected to the secondary side of the transformer is started. Is powered. And
When the current flowing through the switching element reaches a predetermined level, the switching means turns on and a discharge loop is formed.
The switching loop that is turned on has a discharge loop.
All the discharge current that flows flows. In this way , the bias voltage of the bias voltage generating circuit is held at a stable voltage.

According to the second aspect of the invention, when the switching means is turned on, the electric charge accumulated in the capacitor is discharged and the bias voltage is held at a stable voltage.

According to the third aspect of the invention, since the switching element is turned off when the switching means is turned on, the current flowing through the switching element does not exceed a predetermined level.

According to the fourth aspect of the invention, the feedback signal is generated from the electric signal in the one circuit and is output to the switching means.

According to the fifth aspect of the invention, since the feedback signal according to the voltage level of the power source is output to the switching means, a constant power is output regardless of the fluctuation of the voltage level of the power source.

Further, according to the invention of claim 6, since the feedback signal according to the electric signal generated on the primary side of the transformer is output to the switching means, it is constant regardless of the fluctuation of the power supply voltage or the level of the load. Electric power is output.

Further, according to the invention of claim 7, since the feedback signal according to the electric signal generated on the secondary side of the transformer is output to the switching means, it is constant regardless of the level fluctuation of the power supply voltage or the load. Electric power is output.

According to the invention described in claim 8, since the feedback circuit is composed of at least one of a resistor and a constant voltage element, a power supply device having a simple circuit configuration can be obtained.

According to the ninth aspect of the invention, the voltage across the current detecting resistor due to the current flowing through the switching element is applied to the switching control end of the switching means via the resistor, and the applied voltage exceeds a predetermined level. When,
The switching means is turned on and the switching element is turned off. On the other hand, since the feedback signal is output between the switching control end of the switching means and the resistor, ON / OFF of the switching means is controlled by the feedback signal.

Further, according to the tenth aspect of the invention, since the resonance voltage of the resonance circuit is made constant, a constant electric power is output irrespective of the power supply voltage and load level fluctuations.

According to the eleventh aspect of the invention, the output proportional to the resonance voltage of the resonance circuit is integrated and output to the switching means. For example, when a voltage proportional to the resonance voltage of the resonance circuit is output, the proportional output is compared with a reference value, the difference is output, and the output of the comparison means is integrated and output to the switching means. You may do it.

According to the invention of claim 12, 1
A voltage proportional to the induced voltage in the next winding is induced in the detection winding, and the induced voltage in the detection winding is smoothed by the capacitor. At this time, when the induced voltage in the primary winding decreases due to an increase in load level or the like, the charge in the capacitor is discharged by the resistor. Therefore, a feedback signal having a level corresponding to the induced voltage in the primary winding is output to the switching means without delaying the decrease in the induced voltage in the primary winding.

According to the invention of claim 13, 1
A voltage proportional to the induced voltage in the next winding is induced in the detection winding. Since this induced voltage is output to the switching means from the ON side of the detection winding, the induced voltage of the detection winding is output to the switching means when the switching element is turned on. Therefore, when the power is turned on, the switching means is turned on earlier than when the detection winding is output from the off side, so that the on time of the switching element is not prolonged. As a result, stable oscillation is maintained.

According to the fourteenth aspect of the present invention, a part of the information of the power supply device is detected, and the detected information is fed back to the current control circuit to make the output substantially constant.

According to the fifteenth aspect of the present invention, the state signal on the load side is detected by the detection circuit, and the detected signal is subjected to signal processing by the signal processing circuit and then guided to the feedback circuit. .

According to the sixteenth aspect of the invention, the signal detected by the detection circuit is transmitted to the primary side via the transformer.

According to the seventeenth to nineteenth aspects of the invention, the detection circuit detects the output current to the secondary load, the output voltage to the secondary load, or the secondary load. The output to the side load is detected, and the detection results are fed back, so that the current, voltage and output are made constant.

According to the invention of claim 20, 2
The detection signal detected by the detection circuit on the secondary side is voltage-frequency converted, transmitted to the primary side via a transformer, and then frequency-voltage converted on the primary side to be returned to the original, The detection signal is transmitted via the transformer.

According to the invention as set forth in claim 21, power transmission from the primary side of the transformer to the secondary side and signal transmission of the detection signal from the secondary side of the transformer to the primary side are performed simultaneously. Be seen.

According to the twenty-second aspect of the present invention, the power transmission from the primary side of the transformer to the secondary side and the signal transmission of the detection signal from the secondary side of the transformer to the primary side are switched circuits. Alternately according to the switching by.

According to a twenty-third aspect of the present invention, the transformer has a primary side having a power supply section and a primary winding, and a secondary side.
The secondary winding and the secondary side having a load can be attached and detached, and the secondary side casing is placed on a predetermined position of the primary side casing to enable electric power transmission without contact.

According to the invention of claim 24, 1
It is determined whether or not the object placed on the predetermined position of the casing on the secondary side is a regular load, and if it is a regular load, power is supplied to the secondary side.

According to the twenty-fifth and twenty-sixth aspects of the present invention, the load is determined based on the information on the primary side or based on the information from the secondary side.

According to the twenty-seventh aspect of the present invention, the information on the change of the inductance is used as the information on the primary side, specifically, the change of the resonance frequency as in the twenty-eighth aspect of the invention. Information relating to a change in current or a change in voltage is adopted, and whether or not the load is appropriate is determined based on this information.

According to the thirty-first aspect of the invention, a transistor is used as the switching element.
In this case, the self-bias provided by the self-bias circuit provided between the base of the transistor and the ground raises the operating voltage to make it equivalent to the FET, and functions as a switching element.

According to the thirty-second aspect of the invention, since the self-bias circuit is provided between the emitter of the transistor and the current detection resistor connected to this emitter, the operating voltage of the transistor becomes high. .

Further, according to the invention of claim 33, since the self-bias circuit is connected in series to the base of the transistor, the operating voltage of the transistor becomes high.

According to the thirty-fourth aspect of the invention, the operating voltage of the transistor increases by the voltage of the constant voltage element.

According to the invention as set forth in claim 35, the operating voltage of the transistor is increased by the voltage of the constant voltage diode.

According to the thirty-sixth aspect of the invention, the operating voltage of the transistor is increased by the voltage obtained by adding the forward voltages of the plurality of diodes.

According to the thirty-seventh aspect of the invention, the ON period of the transistor used as the switching element is set by controlling the charging current from the bias circuit provided in the self-excited oscillation circuit.

According to the thirty-eighth aspect of the present invention, the charging current flowing through the bias circuit is controlled by switching the resistance value that regulates the amount of current, whereby the ON period of the transistor is set.

According to the thirty-ninth aspect of the invention, the charging current flowing through the bias circuit is controlled by the current control element inserted in series with the bias circuit, whereby the ON period of the transistor is set.

According to the invention as set forth in claim 40, the current control element controls the charging current based on an external control power source, whereby the ON period of the transistor is set.

According to the forty-first aspect of the invention, the external control power source is generated by utilizing the induced electromotive force of the transformer, and the charging current is controlled thereby.

According to the invention of claim 42, the feedback signal generated when the power is turned on (power is turned on) causes an excessive current to flow into the switching means (apply an overvoltage to the switching means) during that time. To be prevented.

Further, according to the invention of claim 43, the oscillation of the switching element is intermittently operated by the feedback signal while the load is open.

[0088]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A first embodiment of the power supply device according to the present invention will be described below with reference to FIGS. FIG. 1 is a circuit diagram showing a first embodiment of a power supply device.

The power source E rectifies an AC input from a commercial power source and supplies a DC power source to this power source device, and is connected between the power source terminals A and B.
This power supply device is wound on the same magnetic circuit as the resonance circuit including the primary winding L1 and the capacitor C1 of the transformer, the switching FET1 and the primary winding L1 for self-excited oscillation, and the mutual inductance M3. Feedback winding L to be coupled
3 and the like, and outputs the output to the secondary winding L2.

Between the power supply terminals A and B, a resonance circuit composed of the primary winding L1 and the capacitor C1, a parallel circuit composed of a diode and a resistor, a switching FET1 and resistors R3 and R4 are connected in series. And FET1
The current flowing into the primary winding L1 is switched by turning on and off, and the secondary winding L2 and the feedback winding L are thereby switched.
A voltage is induced in 3.

Further, a starting resistor R8 and a bias voltage generating capacitor C2 are connected in series between the power supply terminals A and B so that the power supply E charges the capacitor C2 via the resistor R8. Has become. The connection point between the resistor R8 and the capacitor C2 is connected to the FET1 through the feedback winding L3.
When the bias voltage due to the charging voltage of the capacitor C2 reaches the threshold voltage of the FET1, the FET1 is turned on.

In the transistor Q1, the base is connected to the connection point of the resistors R3 and R4, the emitter is connected to the power supply terminal B, the collector is connected to the cathode of the diode D4, and the anode of the diode D4 is connected to the gate of the FET1. .

On the other hand, the load E2 is connected to the secondary winding L2 of the transformer via the diodes D1 and D2.
The electric power induced in the secondary winding L2 is supplied to the diodes D1 and D2.
With this, full-wave rectification is performed and DC power is supplied to the load E2. For example, if the load E2 is a secondary battery, a charging current is supplied.

Next, the operation of the circuit configured as described above will be described with reference to the waveform chart of FIG.

When the power source E is connected to the power source terminals A and B,
First, when the voltages V _G and V _G2 increase and the voltage V _G reaches the threshold value of the FET1, the FET1 starts to turn on and the current I _D starts to flow. The current I _D initially flows as the charging current of the capacitor C1 and then becomes equal to the coil current I _L1 and gradually increases. At this time, the primary winding L1 and the feedback winding L3
The mutual inductance M3 between the feedback winding L3 causes M
An electromotive force of 3 · dI _L1 / dt is generated, the voltage V _G rapidly rises, the FET 1 is completely turned on, and the current I _D further increases.

[0096] Then, the resistance R4 (the resistance value is R ₄₎
When the voltage I _D · R ₄ across the threshold voltage of the transistor Q1 exceeds the threshold value of the transistor Q1, the transistor Q1 is turned on and the current I _S3 is applied during the period T1.
Flows. As a result, the level of the voltage V _G is lowered and F
When the voltage falls below the threshold of ET1, FET1 starts to turn off and the coil current I _L1 also starts to decrease. As a result, the voltage V _G sharply drops due to the voltage induced in the feedback winding L3, and the FET1 is completely turned off.

When the FET1 is turned off, the capacitor C1
The coil circuit I _L1 and the voltage V _C on the primary side of the transformer are free oscillations due to the resonance circuit composed of the primary winding L ₁ and the voltage V _G is again FE due to the coil current I _L1 .
Begin to exceed the threshold of T1.

In this way, the FET1 is repeatedly turned on and off, and the bias voltage V _G2 of the capacitor C2 is made up of the charging current due to the current I _S1 and the discharging current due to the current I _S3 flowing when the transistor Q1 is turned on. When the equilibrium state is reached, the stable oscillation state is reached.

By the way, since the base voltage V _{B of the} transistor Q1 is equal to the voltage I _D · R ₄ across the resistor R4, the transistor Q1 is turned on when this voltage reaches the threshold of the transistor Q1, and FE
T1 turns off immediately after the transistor Q1 turns on.

That is, the current I _D turns off at I _DMAX2 where the peak value is almost constant. Since the current I _D is almost equal to the coil current I _L1 , immediately after the FET1 is turned off, the primary winding L1
The energy P _L stored in is

[0101]

## EQU1 ## P _L = 1 / 2L ₁ · I _DMAX2 ² However, the inductance of the primary winding L1 is L
_Set to ₁ .

On the other hand, the energy P _C stored in the capacitor C1 immediately before the FET1 is turned off is about

[0103]

## EQU2 ## It is represented by P _C = 1 / 2C ₁ · E ₀ ² . However, the capacity of the capacitor C1 is C ₁ , and the voltage of the power source E is E ₀ .

Therefore, free oscillation due to resonance in the resonance circuit formed by the capacitor C1 and the primary winding L1 after the FET1 is turned off is performed by energy P = P _L + P _C. This energy P is applied to the transformer 2 during free vibration.
Since it is consumed by the amount transferred to the load E2 on the next side and the amount converted into heat by loss, the amount decreases when the FET1 is turned on next time. However, when FET1 turns on,
Energy is _supplied to the primary winding L1 again by the current I _DMAX2, so that stable energy supply can be realized.

As described above, according to the first embodiment, the FET
Since the OFF state of 1 is determined by the voltage I _D · R ₄ across the resistor R4, that is, the current I _D , an excessive current I _D does not flow even during a transition. Therefore, the bias voltage V _G2
Is stable without dropping too much, and oscillation can be stably maintained.

Further, since the transistor Q1 and the like are used around the gate of the FET1, it is possible to use an inexpensive and small low withstand voltage element.

Next, a second embodiment of the power supply device according to the present invention will be described with reference to FIGS. FIG. 3 is a circuit diagram showing a second embodiment of the power supply device. In the second and subsequent embodiments, the same parts as those in the first embodiment are designated by the same reference numerals and the description thereof will be omitted.

In the second embodiment, the level of the current I _DMAX2 is changed by feeding back the voltage change of the power source E, and the resistor R19,
R20 is connected, and the connection point of the resistors R19 and R20 is connected to the base of the transistor Q1. A resistor R5 is provided between the base of the transistor Q1 and the connection point of the resistors R3 and R4.

Next, the operation will be described with reference to the waveform chart of FIG. When the current I _D is 0, the voltage V _B is proportional to the voltage level of the power source E, that is, the voltage of the power source E is the combined resistance of the resistor R20 and the resistors R4 and R5 connected in parallel, and the resistor R19. Voltage value V divided by
_{It is B1} . If the voltage value V _B1 is less than or equal to the threshold value of the transistor Q1, the transistor Q1 is kept off. Then, FET1 is turned on, the current I _D starts to flow, the voltage _{V B = V B1 + I D} · R 4. When this voltage V _B reaches the threshold value of the transistor Q1, the transistor Q1 turns on.

Since the voltage value V _B1 is proportional to the voltage level of the power source E, the voltage value V _B1 increases as the voltage level of the power source E increases, so that the transistor Q1 turns on with a smaller current I _D , When the voltage level of the power source E becomes lower, the voltage value V _B1 decreases, so that the power source E turns on with a larger current _ID . Therefore, as P _C increases, it operates to reduce P _L. By setting the resistance values of the resistors R19, R20, R4, and R5 to appropriate values, the energy P = P _L +
P _C can be made substantially constant, and as a result, the output current I _O can also be made substantially constant.

As described above, according to the second embodiment, the power source E
Even if the voltage level of is changed, the output level of the transformer secondary side can be made substantially constant.

As shown in FIG. 5, a Zener diode Z12 may be connected between the power supply terminal A and the resistor R19. In this case, the change amount of the voltage value V _B1 can be increased with respect to the change of the voltage level of the power source E. Therefore, when the voltage level of the power source E is high,
The output level can be easily suppressed.

Further, as shown in FIG. 6, the diode D4
The anode of may be connected to the connection point of the resistor R8 and the capacitor C2 instead of the gate of the FET1. Also in this case, the same operation as in the case of FIG. 3 is performed, and the same effect is obtained.

Here, the shape of the transformer will be described. In the second embodiment, as shown in FIG. 3, coils are wound around the primary core K1 and the secondary core K2, each of which is a U-shaped core, to form the primary side and the secondary side. It is arranged. When the primary core K1 and the secondary core K2 are attached to each other so as to face each other, electric power is transferred from the primary side to the secondary side by electromagnetic induction through an insulator such as a casing of each device, air, or water. Communication is possible.

Next, a third embodiment of the power supply device according to the present invention will be described with reference to FIGS. FIG. 7 is a circuit diagram showing a third embodiment of the power supply device.

In the third embodiment, the voltage on the primary side of the transformer, that is, the resonance voltage V _C is kept constant. In this circuit, in addition to the first embodiment, a new winding L4 is formed on the same magnetic path as the primary winding L1 of the transformer, and a diode D9 and a capacitor for rectifying and smoothing the induced voltage of this winding L4. C71 and resistors R1 and R2 for dividing the rectified and smoothed DC voltage are provided, and the connection point of the resistors R1 and R2 is connected to the base of the transistor Q1.

Next, the operation will be described with reference to the waveform chart of FIG. The induced voltage in the winding L4 is proportional to the resonance voltage V _C. The voltage V _B1 applied to the base of the transistor Q1 is proportional to the induced voltage of the winding L4, that is, the induced voltage of the winding L4 is a combined resistance due to the parallel connection of the resistor R2 and the resistors R4 and R5. , And the resistance R1. Therefore, the voltage V _{B1 is} also the resonance voltage V
Proportional to _C.

When the power source E is at the voltage E ₀₁ , the peak value of the current I _D is I _D1 . In this case, when the voltage of the power source E increases from E ₀₁ to E ₀₂ , the resonance voltage V _C increases and the voltage V _B1
Also increases. Therefore, the transistor Q1 turns on with a smaller current I _D , and the peak value of the current I _D decreases from I _D1 to I _D2 . When the peak value of the current I _D decreases,
Since the energy stored in the primary winding L1 decreases,
Resonance voltage V _C is suppressed, the level of the resonant voltage V _C is held substantially constant.

Further, in this state, when the level of the load E2 decreases from the rated load to no load, the energy transferred from the resonance circuit to the load E2 via the transformer decreases, so that the attenuation of the resonance voltage V _C also decreases. , The resonance voltage V _C increases more than before. When the resonance voltage V _C increases, the voltage V _B1
Also increases, the peak value of the current I _D decreases to I _D3 ,
The energy stored in the primary winding L1 is reduced, the increase of the resonance voltage V _C is suppressed, and the level of the resonance voltage V _C is kept substantially constant.

As described above, according to the third embodiment, the power source E
The resonant voltage V _C can be kept constant even if the voltage level of V fluctuates. Therefore, when the level of the load E2 is constant, the level of the output current I ₀ can be stabilized.

Further, even when the level of the load E2 fluctuates, the resonance voltage V _C can be kept constant.
It is possible to prevent the breakdown voltage level such as T1 from being exceeded. Therefore, it is possible to prevent the breakdown of the voltage of the component and the deterioration of the life.

Since the resonance voltage V _C can be made constant,
For example, the constants of circuit parts may change due to temperature changes,
The output can be made substantially constant even if the component constants vary widely.

Further, as in the present embodiment, when the primary side and the secondary side of the transformer are detachable and electric power is transmitted in a contactless manner by electromagnetic induction in a mounted state, the resonance circuit of the resonance circuit is changed depending on the mounted state. Even if the inductance changes, the resonance voltage V _C can be made constant, so that the transmitted power can be made almost constant.

As shown in FIG. 9, the resistor R2 may be removed and the induced voltage of the rectified and smoothed winding L4 may be connected to the base of the transistor Q1 via the zener diode Z20 and the resistor R1. Good. Also in this case, the same operation as in the case of FIG. 7 can be performed and the same effect can be obtained.

Further, as shown in FIG. 10, a resistor R71 may be connected in parallel with the capacitor C71 of FIG. The operation of the circuit of FIG. 10 when a load is suddenly connected from the no-load state will be described with reference to the voltage waveform diagram of FIG.

Since the energy is transmitted to the secondary side of the transformer at the moment when the load is connected, the resonance voltage is attenuated and the voltage induced in the winding L4 is also reduced. At this time,
If the resistor R71 is not connected, the capacitor C
Since 71 gradually discharged, the voltage V _X of the induced voltage is rectified and smoothed windings L4, as indicated by a broken line in FIG. 11, gradually decreases. For lowering speed of the voltage V _X is slow, then when the FET1 is turned on, is still higher voltage applied to the base of transistor Q1, it is impossible to flow a sufficient current I _D. Therefore, the resonance voltage may be further attenuated, resulting in intermittent oscillation.

However, if the resistor R71 is connected in parallel with the capacitor C71, a discharge loop is formed. Therefore, when the resonance voltage is attenuated and the induced voltage in the winding L4 is lowered, the voltage V _X is discharged through the resistor R71. Therefore, even if the load is suddenly connected, as shown by the solid line in FIG.
_{Since X} decreases rapidly, when the FET1 is turned on next time, the voltage applied to the base of the transistor Q1 decreases, so that the current I _D can sufficiently flow and stable oscillation can be continued. it can.

Further, as shown in FIG. 12, in FIG. 9, the polarity of the winding L4 may be set so that the output is taken out from the ON side. The operation when the power is turned on will be described with reference to the voltage waveform diagrams of FIGS. 13 and 14.

First, a case where the polarity is set so that the output of the winding L4 is taken out from the OFF side, contrary to the case of FIG. 12, will be described with reference to FIG.

After the power is turned on, the voltage V _G rises and the FET1
When the threshold voltage is exceeded, FET1 turns on. When the FET1 is turned on, the voltage V _F drops to GND, but the winding L
FET4 outputs the output from the OFF side, so FET1
The voltage V _X is not generated during the ON period. FET1 is
When the current I _D increases and the voltage across the resistor R4 exceeds the threshold value of the transistor Q1 to turn on the transistor Q1, it turns off. The voltage V _X generated in the off period of the FET1.

When the power is turned on, since the voltage applied from the winding L4 to the base of the transistor Q1 is not generated, the ON period becomes long, so that the energy stored in the primary winding L1 is large. And the voltage V _F jumps greatly. Therefore, the voltage V _X due to the induced voltage of the winding L4, which is proportional to the voltage V _F in the off period, becomes large, and the voltage applied to the base of the transistor Q1 becomes too large. I _D cannot be flowed, and there is a risk of intermittent oscillation.

Next, a case where the polarity is set so that the output of the winding L4 is taken out from the ON side will be described with reference to FIG.

After the power is turned on, the voltage V _G rises and the FET1
When the threshold voltage is exceeded, FET1 turns on. When FET1 is turned on, the windings L4 in order is taken out of the output from the on-side, the voltage V _X may occur at the same time. Therefore, the resistance R
Transistor Q1 due to the voltage generated at 4 and the voltage V _X
Depending on the voltage applied to the base of FET1, FET1 is turned off earlier than in the case of FIG.

Since the voltage V _X is generated at the ratio of the potential difference between the power source E and the GND level, the voltage level is higher than that in the case of the peak voltage of the voltage V _F and the potential difference between the GND level as shown in FIG. Get lower.

Therefore, the voltage applied to the base of the transistor Q1 is also at an appropriate level, and the current I _D of a sufficient level can flow when the FET1 is turned on next time, so that the oscillation can be stably maintained. .

Further, as shown in FIG. 15, FIG. 9 and FIG. 7 may be combined. FIG. 15 shows that a resistor R2 is provided between the power supply terminal B and the base of the transistor Q1 in FIG.
Is connected. In this case, the change amount of the voltage V _B1 can be increased with respect to the change of the voltage V _X. Therefore, the voltage V _C can be held substantially constant in response to the larger level fluctuation of the voltage V _E.

Further, as shown in FIG. 16, FIG. 9 and FIG. 3 may be combined. In FIG. 16, resistors R19 and R20 are connected in series between the power supply terminals A and B in FIG. 9, and the connection point of the resistors R19 and R20 is connected to the base of the transistor Q1. In this case, it is possible to directly respond to the level change of the power source E, and to keep the resonance voltage V _C constant even when the level of the load E2 changes.
Therefore, when the level of the load E2 is constant, the level of the output current I ₀ can be stabilized. Further, it is possible to prevent the breakdown voltage level of the FET 1 from being exceeded.

Next, a fourth embodiment of the power supply device according to the present invention will be described with reference to FIG. FIG. 17 is a circuit diagram showing a fourth embodiment of the power supply device.

[0139] The fourth embodiment is also intended to maintain the voltage V _C of the primary side of the transformer, i.e., the resonant voltage constant. This circuit is the same as the first embodiment except that the primary winding L1 of the transformer is
A winding L5 formed on the same magnetic circuit, a diode D5 and a capacitor C3 that rectify and smooth the induced voltage of the winding L5, and resistors R6 and R7 that divide the rectified and smoothed DC voltage. Is equipped with.

Further, the operational amplifier OP1 and the resistor R81,
The operational amplifier OP1 is equipped with a non-inverting amplifier circuit composed of R9.
The reference voltage V _Y generated by the reference voltage generation circuit including the resistor R10 and the Zener diode Z30 is input to the − input terminal of the resistor via the resistor R9, and the + input terminal is divided by the resistors R6 and R7. Voltage has been input. The output terminal of the operational amplifier OP1 is connected to an integrating circuit composed of a resistor R11 and a capacitor C4, and the output of this integrating circuit is connected to the base of the transistor Q1 via the resistor R12.

Next, the operation will be described. During operation,
The operational amplifier OP1 divides the voltage V _X proportional to the resonance voltage V _C by the resistors R6 and R7 into a voltage V _X1 and
The voltage V _X2 input from the reference voltage V _Y via the resistor R9
Are compared with each other, the difference is amplified, and the voltage V _B1 is applied to the base of the transistor Q1 via the integrating circuit.

When the voltage V _C increases from the stable state due to a decrease in load level or the like, the voltage V _X also increases,
A voltage difference occurs between the input voltage V _X1 of the operational amplifier OP1 and the input voltage V _X2, and the operational amplifier OP1 outputs the output voltage V _X3 that rises so that the input difference becomes zero.

The output voltage V _X4 of the integrating circuit increases slowly compared to the increase of the voltage V _X3 due to the integrating action of the resistor R11 and the capacitor C4. Therefore, the level of the voltage V _B1 gradually increases, so that the current I _D gradually decreases and the voltage V _C decreases.

As described above, according to the fourth embodiment,
By the integration control, the resonance voltage V _C is automatically controlled so that the resonance voltage V _C always becomes a voltage having the same magnification as the reference voltage V _Y , so that the voltage V _C can be held constant.

As shown in FIG. 18, the winding L5 is removed and the anode of the diode D5 is changed to FET1.
The voltage V _C1 = V _DS + I _D · R ₄ may be used as a substitute for the voltage proportional to the resonance voltage V _C. However, V _DS is the drain-source voltage of FET1. Also in this case, the same effect can be obtained by the same operation as in FIG.

Next, a fifth embodiment of the power supply device according to the present invention will be described with reference to FIG. FIG. 19 is a circuit diagram showing a fifth embodiment of the power supply device.

In the fifth embodiment, the output current I _O on the secondary side of the transformer is kept constant. In addition to the first embodiment, this circuit includes a winding L6 formed on the same magnetic path as the secondary winding L2 of the transformer, and a primary winding L7 connected to this winding L6 via a resistor R15. And a secondary winding L8 and a transformer TL7 for transmitting the induced voltage of the winding L6, and
Diode D7 that rectifies and smoothes the induced voltage in the secondary winding L8
The rectified and smoothed DC voltage is provided to the base of the transistor Q1 via the Zener diode Z20 and the resistor R1. A resistor R1 is connected in series between the power supply terminals A and B.
9 and R20 are connected and resistors R19 and R20 are connected.
Is connected to the base of the transistor Q1.

Next, the operation will be described. During operation,
A voltage is induced in the winding L6 by the induced voltage in the secondary winding L2 of the transformer, and this voltage is transferred to the winding L6 via the transformer TL7.
This induced voltage is induced in the secondary winding L8 and this induced voltage is applied to the diode D7.
And the capacitor C5 for rectification and smoothing. Then, this DC voltage is applied to the base of the transistor Q1 via the Zener diode Z20 and the resistor R1.
The power source E is connected to the resistor R at the base of the transistor Q1.
The voltage divided by 19 and R20 is applied.

From the stable state, the diodes D1 and D2 on the secondary side of the transformer are changed due to a decrease in the load level.
When the currents I _L2 and I _L3 flowing in the transformer TL7 increase, the voltage induced in the winding L6 also increases, and the voltage induced in the secondary winding L8 of the transformer TL7 also increases. Therefore, since the voltage V _B1 applied to the base of the transistor Q1 increases and the transistor Q1 turns on earlier, the FET1 turns off earlier and the current I _D decreases, so the voltage V _C decreases, The secondary currents I _L2 and I _L3 are kept constant.

When the voltage level of the power source E rises,
The voltage divided by the resistors R19 and R20 increases. Therefore, since the voltage V _B1 applied to the base of the transistor Q1 increases and the transistor Q1 turns on earlier, the FET1 turns off earlier and the current I _D decreases, so the voltage V _C decreases, Secondary current I _L2 , I
_L3 is kept constant.

As described above, according to the fifth embodiment, the fluctuations of the secondary currents I _L2 and I _L3 are detected, and the resonance voltage V _C is held constant according to the fluctuations. Even if the voltage of E or the level of the load E2 changes, the output current I _O can be stabilized.

Further, by keeping the resonance voltage V _C constant, it is possible to prevent the breakdown voltage level of the FET 1 and the like from being exceeded, so that it is possible to prevent the breakdown of the voltage of the parts and the deterioration of the life.

Since the resonance voltage V _C can be made constant,
For example, the constants of circuit parts may change due to temperature changes,
The output can be made substantially constant even if the component constants vary widely.

Further, as in the present embodiment, when the primary side and the secondary side of the transformer are detachable and electric power is transmitted in a contactless manner by electromagnetic induction in the mounted state, the resonance circuit of the resonant circuit is changed depending on the mounted state. Even if the inductance changes, the secondary current I
_Since the resonance voltage V _C is made constant by detecting the variation of _L2 and I _L3 , the output current I _O can be stabilized.

Next, a sixth embodiment of the power supply device according to the present invention will be described with reference to FIG. FIG. 20 is a circuit diagram showing a sixth embodiment of the power supply device.

Similar to the fifth embodiment, the sixth embodiment keeps the output current I _O on the secondary side of the transformer constant. This circuit is the same as the first embodiment except that the secondary winding L of the transformer is used.
Winding L6 formed on the same magnetic path as 2 and this winding L6
Diode D8 and capacitor C6 for rectifying and smoothing the induced voltage of R1, and resistor R1 connected in parallel with capacitor C6
2 and a series circuit of the light emitting element LD. The light emitting element LD is composed of a light emitting diode or the like, and emits a light amount corresponding to the current level when a current is supplied by a rectified and smoothed DC voltage.

Further, the light receiving element P1 and the resistors R19 and R20 are connected in series between the power supply terminals A and B, and
The connection point of the resistors R19 and R20 is connected to the base of the transistor Q1 and the resistor R13 is connected in parallel to the resistor R20. The light receiving element P1 is formed of, for example, a phototransistor, is configured to receive the light of the light emitting element LD, and conducts at a conductivity corresponding to the amount of received light.

Next, the operation will be described. During operation,
A voltage is induced in the winding L6 by the induced voltage in the secondary winding L2 of the transformer, the voltage is rectified and smoothed, a current is supplied, the light emitting element LD emits light, and the light is received by the light receiving element P1.
Is received by. Since the light receiving element P1 conducts at a conductivity corresponding to the amount of received light, the voltage V _B1 applied to the base of the transistor Q1 is a value obtained by subtracting the collector-emitter voltage V _{CE of the} light receiving element P1 from the voltage of the power source E. , And the combined resistance of the resistors R19 and R13, R20.

Then, when the currents I _L2 and I _L3 flowing through the secondary side diodes D1 and D2 increase from the stable state due to a decrease in load level or the like, the voltage induced in the winding L6 rises, so that light emission occurs. The current flowing through the element LD increases and the emission intensity thereof increases. Therefore, the collector-emitter voltage V _{CE of} the light receiving element P1 decreases, and the voltage V _B1 applied to the base of the transistor Q1 increases. Therefore, the transistor Q1 turns on earlier and the FET1
Turn off faster. Therefore, the current I _D is reduced and the resonance voltage V _C is reduced, so that the secondary currents I _L2 and I _L3 are reduced and the output current I _O is stabilized.

As described above, according to the sixth embodiment, the fluctuations of the secondary currents I _L2 and I _L3 are detected, and the resonance voltage V _C is kept constant in accordance with the fluctuations. Even if the voltage of E or the level of the load E2 changes, the output current I _O can be stabilized.

Further, by keeping the resonance voltage V _C constant, it is possible to prevent the withstand voltage level of the FET 1 and the like from being exceeded, so that it is possible to prevent voltage breakdown of components and deterioration of life.

Since the resonance voltage V _C can be made constant,
For example, the constants of circuit parts may change due to temperature changes,
The output can be made substantially constant even if the component constants vary widely.

Further, as in the present embodiment, when the primary side and the secondary side of the transformer are detachable and electric power is transmitted in a contactless manner by electromagnetic induction in a mounted state, the resonance circuit of the resonance circuit is changed depending on the mounted state. Even if the inductance changes, the secondary current I
_Since the resonance voltage V _C is made constant by detecting the variation of _L2 and I _L3 , the output current I _O can be stabilized.

Next, the peripheral protection circuit for the non-contact type power supply device will be described. FIG. 21 is a block diagram of a power supply device showing an outline of a peripheral protection circuit. Explaining the overall configuration, the present apparatus is configured such that a power supply unit and a load unit can be separated and attached, and the power supply unit includes a connectable power supply E, a current control circuit 11 as a self-oscillation circuit unit, and an FET.
1 and the like, a resonance circuit 13, a primary core K1, a primary winding L1 wound around the primary core K1, and a feedback circuit 14, and a signal processing circuit 15 described later. The load unit side includes a secondary core K2, a secondary winding L2 wound around the secondary core K2, a rectifying / smoothing circuit (not shown), and a load E2, and further includes a signal detection circuit 21 and a signal processing circuit 22 described later. .

The block diagram of FIG. 21 shows, for example, the circuit shown in FIG. 3 as a block except for the signal detection circuit 21 and the signal processing circuits 15 and 22. Also, FIG.
1 to 32, the thick line shows the flow of electric power, and the thin line shows the flow of load state signals.

The signal detection circuit 21 is provided on the line of the output current I ₀ and detects the state of the load. The signal processing circuit 22 is interposed between the signal detection circuit 21 and the secondary winding L2, and performs signal processing so as to convert a signal indicating the detected load state into a signal form that can be transmitted to the primary side through a transformer. It is a thing. The signal processing circuit 15 is provided between the primary winding L1 and the feedback circuit 14 and converts the load state signal transmitted from the secondary side into a signal for feedback. By providing such a peripheral protection circuit, the state of the load is detected, the detection signal is transmitted to the primary side, the self-excited oscillation is feedback-controlled, and the output to the load E2 is stabilized.

FIG. 22 shows the signal processing circuit 1 shown in FIG.
In the block diagram of the power supply device showing 5, 22 more specifically,
Power transmission and load state signal transmission are performed simultaneously.

In the figure, the signal processing circuit 22 is composed of a Vf conversion circuit 221 and a waveform shaping circuit 222. The V-f conversion circuit 221 uses voltage information detected as a load state signal on the load side by using a VCO (voltage controlled oscillation) or the like, and determines a required frequency band at least different from the resonance frequency on the power supply side. It is converted into a frequency signal. The waveform shaping circuit 222 is composed of a bandpass filter or the like, and transforms the waveform into a sine wave that can easily pass through the transformer. Then, the obtained sine wave signal is
It is transmitted from the secondary winding L2 to the primary winding L1. In addition, the signal processing circuit 15 includes a limiter circuit 151 and a waveform shaping circuit 15.
2 and the f-V conversion circuit 153. Since the voltage generated in the primary winding L1 is a composite wave of the sine wave power wave and the sine wave signal wave from the secondary side, the limiter circuit 151 extracts only the signal wave from the primary winding voltage. . The waveform shaping circuit 152 shapes the extracted sine wave signal into a rectangular wave. f-V conversion circuit 15
3 is for converting into a voltage signal corresponding to the pulse frequency of the rectangular wave. Then, the obtained voltage signal is guided to the feedback circuit 14, where feedback control is performed so that the output to the secondary side is stabilized according to the change in the state of the load.

FIG. 23 shows another block diagram of the power supply device shown in FIG. 22, in which power transmission and load state signal transmission are switched, and in addition to the configuration shown in FIG. A circuit SW is provided. Switching circuit SW
For a period of time in the power switching element 12 is operated is Ru is transmitted from the primary side to the secondary side is connected to the power line, alternately to the duration as another period is connected to a load state signal line side It can be switched every time.

FIGS. 24 to 26 are block diagrams for each form of the signal detected by the signal detection circuit 21 in FIG. In FIG. 24, the output current I _{0 is transferred} to the secondary winding L
The current is detected by the current detection circuit 21a such as a low resistance connected in series with the second circuit, so that the output current on the load side can be stabilized. In FIG. 25, the voltage across the secondary winding L2 or the voltage across the load E2 is detected by the voltage detection circuit 21b, whereby the output voltage on the load side can be stabilized. In FIG. 26, both the current and the voltage are detected and the product thereof is obtained, or the electric power is detected by the electric power detection circuit 21c based on the principle of a well-known electric power meter or the like, whereby a stable electric power is output to the load. be able to.

FIG. 27 is a more detailed block diagram of the power supply device for stabilizing the output current I ₀ shown in FIG. In the figure, the same components as those in FIG. 22 are designated by the same reference numerals and the description thereof will be omitted.

A parallel circuit including a resistor R21 and a capacitor C10 as a current detection circuit for detecting the output current I ₀ is connected in series to the load E2. Output current I ₀
Is detected as the voltage V1 across the resistor R21, and V-
After being input to the f conversion circuit 221, it passes through a waveform shaping circuit 222 such as a bandpass filter and a capacitor C11,
It is converted into a sine wave of a predetermined level V2 and guided to the secondary winding L2. The capacitor C12 is connected in series with the primary winding L1 and extracts a sine wave signal which is a load state signal transmitted from the secondary side. Amplifier circuit 150
Is for amplifying the extracted sine wave signal to a required level, and the amplified sine wave signal is the limiter circuit 15 described above.
After being guided to 1, the waveform is shaped into a rectangular wave of level V3 by the waveform shaping circuit 152, and further the voltage V4 is shaped by the fV conversion circuit 153.
And is output to the feedback circuit 14.

The feedback circuit 14 is an operational amplifier OP.
2, OP3 has a configuration in which two stages are connected, and an operational amplifier O
The-input terminal of P2 is connected to the output side of the fV conversion circuit 153, and the output terminal is connected to the-input terminal of the operational amplifier OP3. On the other hand, a resistor R19 is provided between the power supply terminals A and B.
A series circuit of a zener diode Z40 and a zener diode Z40 is connected, and a zener voltage as a reference voltage is supplied to an operational amplifier OP.
2, lead to the OP3 + input terminal. The output terminal of the operational amplifier OP3 is connected to the base of the transistor Q1 and outputs the output voltage V5 as a bias for feedback.

Next, the operation will be described with reference to the waveform chart of FIG. Now, assuming that the state of the load E2, for example, the load resistance decreases (time T1), the decrease in the load resistance appears as an increase in the output current I ₀ , and the detection voltage V1 increases accordingly. Therefore, the V-f conversion circuit 221
And a sine wave signal V2 obtained through the waveform shaping circuit 222
Appears as an increase in frequency (see signal V2). Then, after the sine wave signal V2 is transmitted from the secondary side to the primary side and extracted by the capacitor C12, the waveform shaping circuit 1
The signal is shaped into a rectangular wave signal V3 at 52, and is further output as a voltage V4 at the fV conversion circuit 153. Then, the increase in the frequency of the sine wave signal V2 appears as an increase in the voltage V4. When the voltage V4 increases, the output of the operational amplifier OP2 decreases, so that the output voltage V5 of the operational amplifier OP3 increases. When the output voltage V5 rises, the base voltage of the transistor Q1 rises, and the smaller current I _D
Then, the transistor Q1 turns on, so the FET1
_Turns off quickly and the peak value of the current I _{D changes} from I _DMAX2 to I
It will be reduced to _DMAX1 . As a result, the primary winding L1
Since the energy stored in the output current I ₀ decreases,
Will decrease. On the contrary, when the load resistance of the load E2 becomes large, the output voltage V5 decreases according to the amount of change, and the OFF of the FET1 is delayed, so that the primary winding L1
The energy accumulated in the output current I ₀ increases and the output current I ₀ increases. In this way, a constant current function is realized to make the change in the output current I ₀ extremely small with respect to the large change in the load E2.

Next, another peripheral protection circuit for the non-contact detachable power supply device will be described. This peripheral protection circuit determines whether or not the object placed on the secondary side of the transformer in this type of power supply device is a regular load, and when a foreign object is placed or when there is no load, energy is saved. The purpose is to stop or minimize the supply. That is, in the case of the detachable power supply device, the secondary winding L2 of the transformer and the load E2 are often integrated. Therefore, in the case of a regular load, the secondary winding L
It is only necessary to consider the effect of the presence or absence of 2 and the presence or absence of load.

By the way, the inductance of the transformer is expressed as a function of the magnetic resistance and the number of turns of the winding. Further, in this type of power supply device, it is difficult to fix the positions of the secondary core K2 and the secondary winding L2 of the transformer, so that the magnetic resistance changes depending on the relative mounting position with respect to the primary core K1 and the primary winding L1. To do. In addition, the magnetic resistance differs greatly between the case where a regular load is placed on the secondary side and the case where no load or foreign matter such as metal is placed on the secondary side, and the value of the inductance seen from the primary side changes greatly. Becomes By utilizing this inductance change, it is possible to discriminate them with a simple detecting means. Therefore, the above determination is performed using the inductance and the capacitance of the resonant circuit viewed from the primary side. Now, assuming that the inductance is L ₀ and the capacitor capacity is C ₀ , the frequency f is represented by 1 / 2π√ (L ₀ · C ₀ ). Therefore, a method of detecting this frequency change can be considered.

Further, in this type of power supply device, the energy stored in the resonance circuit is transferred to the load E2. However, in the case of no load or a metallic foreign object with respect to the regular load, the amount of energy transfer due to the difference in inductance. Is also significantly different, so it also appears as a change in resonance voltage or resonance current. Hereinafter, various peripheral protection circuits using the characteristics caused by the difference in the inductance will be described with reference to FIGS.
Will be explained. In each figure, the same parts as those in FIG. 21 are designated by the same reference numerals and the description thereof will be omitted.

FIG. 29 is a block diagram of a power supply device showing an outline of a peripheral protection circuit utilizing a change in resonance frequency. The frequency detection circuit 17 is connected to the resonance circuit and detects the resonance frequency using an fV conversion circuit or the like.

FIG. 30 is a block diagram of a power supply device showing an outline of a peripheral protection circuit utilizing a change in resonance current. The current detection circuit 18 has a low resistance interposed in the resonance circuit, and detects the resonance current from the voltage level obtained across the resonance circuit.

FIG. 31 is a block diagram of a power supply device showing an outline of a peripheral protection circuit utilizing a change in resonance voltage. The voltage detection circuit 19 is adapted to extract the voltage across the resonance circuit.

Then, it is judged whether or not the detected inductance, resonance frequency, resonance current, or resonance voltage is within the range of appropriate values when a regular load is connected, and if it is within the range. Assuming that the load is normal, the self-excited oscillation operation is continued, and if it is out of the proper range, it is judged that there is no load or metal foreign matter is placed, and the self-excited oscillation operation is stopped, or the operation is turned on / off. Try to turn off the switch. The appropriate value may be set and stored in advance, or may be generated as an input signal on the reference side to the comparison means.

FIG. 32 is a block diagram of the power supply device showing an outline of the peripheral protection circuit utilizing the information from the load side. Since the detachable detachable power supply device allows electromagnetic coupling by placing the housing of the load section at a proper place on the upper surface of the housing of the power supply section, the load signal detection circuit 20 is provided in the housing of the power supply section. A magnet is disposed in the load unit casing and a magnet switch is disposed in the power unit side casing at a position facing when the casing of the load unit is properly placed on the body. With this configuration, the magnet switch of the load signal detection circuit 20 is turned on only when a proper load is placed, so that it is possible to reliably confirm that the proper load is placed. The magnet switch of the load signal detection circuit 20 is connected to a switch (not shown) for turning on / off the self-excited oscillation operation. When the magnet switch is turned on, the connection with the power supply is cut off to stop the oscillation operation.

FIG. 33 shows a frequency detecting circuit 1 shown in FIG.
6 is a circuit block diagram showing details of a primary side of a power supply circuit using the No. 7 circuit. In the figure, the same parts as those in FIG. 29 are designated by the same reference numerals and the description thereof will be omitted.

Series voltage dividing resistors R22 and R23 are connected in parallel between the negative electrodes of the resonance circuits C1 and L1 and the ground. An f-V conversion circuit 171 is connected to the connection point of the resistors R22 and R23, divides the resonance waveform signal V11 and captures it, and outputs a voltage signal V12 corresponding to the frequency. When comparator circuit 172 which is appropriately placed on a predetermined mounting position of the power unit normal loads, have a range of resonant frequencies of the resonant circuit as set range data V _S of the voltage, and the setting range Data V _S compares the voltage signal V12, and outputs only high-level signal V13 when outside the range data V _S. The pulse generation circuit 173 outputs the pulse signal V14 of a predetermined cycle while the high level signal is input from the comparison circuit 172. Also, the transistor Q
2 is connected in parallel to both ends of the capacitor C2 and is connected to the output side of the pulse generation circuit 173 via the base resistor R24.

Next, the operation will be described with reference to the waveform chart of FIG. If a regular load is connected now, f
The output voltage V12 from the −V conversion circuit 171 is within the range of the set range data V _S , so it is not output from the comparison circuit 172. Therefore, the output from the pulse generation circuit 173 is in the low level state, and as a result, the transistor Q2
Will remain off. When the transistor Q2 is off, the charging voltage of the capacitor C2 and the feedback circuit 1
The oscillating operation is continued by the feedback signal from 4 and the output current I ₀ is supplied to the load E2.

On the other hand, in this state, if the regular load E2 is removed from the power supply section, the inductance seen from the primary side of the transformer decreases, the oscillation frequency increases, and the output from the fV converting circuit 171 increases. the voltage V12 exceeds the range of the set range data V _S. Therefore, the comparison circuit 1
The output signal V13 is sent from 72, and the pulse generation circuit 1
A high-level pulse signal V14 is output from 73.
As a result, the transistor Q2 is turned on only during the pulse output period to discharge the charge stored in the capacitor C2, so that the on / off operation of the FET1 is stopped and the supply of the output current I ₀ to the load is stopped. In addition, the pulse generation circuit 1
While the capacitor C2 is charged during the rest period of the output pulse of 73 and oscillation is restarted during that period, the pulse signal V14 is immediately sent out, and as a result, the intermittent oscillation operation is repeated. Therefore, when there is no load or when the load is removed, the power loss on the primary side can be suppressed as much as possible (power saving).

Next, when a metallic foreign matter is placed and the value of the inductance seen from the primary side of the transformer becomes larger than when a regular load is placed, the oscillation frequency becomes low and the output voltage V12 becomes It falls below the range of the set range data V _S. Therefore, the comparison circuit 172 outputs the output signal V13 and the pulse generation circuit 173 outputs the high-level pulse signal V14. As a result, the transistor Q2
Since the oscillating operation is intermittently driven during the pulse signal transmission period and the rest period, even if a metallic foreign matter is placed, the power loss on the primary side can be suppressed (power saving) as much as possible, and as a result, the metallic foreign matter can be suppressed. Heat generation is prevented.

Next, the switching element is a voltage drive type F
A circuit configuration for adopting a current drive type transistor instead of ET1 will be described with reference to FIGS. In these figures, the same components as those in FIG. 1 are designated by the same reference numerals and the description thereof will be omitted.

FIG. 35 is a circuit diagram in which FET1 is replaced with a current drive type transistor Q3. Voltage driven F
When ET is used, the bias voltage V _G2 is obtained by the charging current I _S1 and the discharging current I _S3 by the resistor R8 and the capacitor C2, and the FET1 is turned on after the transistor Q1 is turned on.
Is turned off so that the on-timing of the transistor Q1 can be linked with the value of the drain current I _D to control the current flowing through the primary winding L1.

On the other hand, when the current-driven transistor Q3 is used, as shown in FIG. 35, the base current I _S2 continues to flow while the transistor Q3 is on, so that the base control function of the transistor Q1 deteriorates. Will come. The input voltage of the transistor Q3 is about 0.7V.
And turn on at a low voltage. On the other hand, the transistor Q1
Also turns on at the same voltage, so transistor Q1
Cannot fully exercise the function of. Therefore, the transistor Q1
In order to satisfy the function of
It is necessary to make the circuit configuration equivalent to the case of 1. This equivalence means increasing the input impedance and raising the operating voltage to about several V as FET1.

Therefore, as a measure for increasing the operating voltage, as shown in FIGS.
A resistor was inserted into the input of 3 to apply a self-bias voltage to the base and the emitter.

FIG. 36 is a circuit diagram showing a first countermeasure example, in which a resistor R25 is inserted between the base of the transistor Q3 and the anode of the diode D4 and the resistor R shown in FIG.
In place of 3, constant voltage diode Z41 as a constant voltage element
(Self-biasing circuit) is provided. In this first countermeasure example, the resistor R25 raises the input impedance of the transistor Q3 to suppress the base current I _S2 to secure the base control function, and also the constant voltage diode Z41.
Thus, the operating voltage of the transistor Q3 is raised to obtain a desired operating voltage to ensure the on / off control of the transistor Q3 by the transistor Q1.

FIG. 37 is a circuit diagram showing a second countermeasure example, in which a resistor R25 is inserted between the base of the transistor Q3 and the anode of the diode D4 and the resistor R shown in FIG.
3, a plurality of diodes D10 as constant voltage elements are connected in series so that the emitter side of the transistor Q3 serves as an anode. In this second countermeasure example, the input impedance of the transistor Q3 is increased by the resistor R25 to suppress the base current I _S2 to secure the base control function, and the operating voltage of the transistor Q3 is increased by each forward voltage of the diode D10. I am trying. A desired operating voltage can be obtained by providing two or a required number of diodes D10.

FIG. 38 is a circuit diagram showing a third countermeasure example, in which a resistor R25 and a diode D1 as a constant voltage element are provided between the base of the transistor Q3 and the anode of the diode D4.
1 is connected in series with a circuit in which a plurality of 1s are connected in series so that the resistor R25 side serves as an anode. In the third countermeasure example, the resistor R25 increases the input impedance of the transistor Q3 to suppress the base current I _S2 to secure the base control function, and the operating voltage of the transistor Q3 is set to the forward voltage of the diode D11. Q
I try to lift it relative to 1. A desired operating voltage can be obtained by providing two or a required number of diodes D11.

FIG. 39 is a circuit diagram showing a fourth countermeasure example, in which a resistor R25 is provided between the base of the transistor Q3 and the anode of the diode D4, and a constant voltage diode as a constant voltage element in which the anode is connected to the resistor R25 side. Z42 is connected in series. In the fourth countermeasure example, the resistance R2
5, the input impedance of the transistor Q3 is increased to suppress the base current I _S2 to secure the base control function, and the operating voltage of the transistor Q3 is increased relative to the transistor Q1 by the constant voltage of the constant voltage diode Z42. I am trying.

By adopting the circuit configuration shown in FIGS. 36 to 39, the same function as that of the device using FET1 can be obtained, and the cost can be reduced. Further, since the IGBT (isolated gate bipolar transistor) which is a MOS input type transistor is equivalent to the FET 1, it can be used as it is. As described above, the switching element is not limited by the type, and a voltage drive type element such as an FET, an IGBT or a transistor and a current drive type element can be used, so that there is an advantage that an optimum element can be selected according to the application.

By the way, when the transistor Q3 is used as the switching element, the base current I _S2 flowing when the transistor Q3 is turned on can be positively used so that the constant output control is performed by self-excited oscillation.

FIG. 40 is a circuit diagram of a power supply device which performs a self-excited oscillation operation using the transistor Q3. This circuit
From the circuit of FIG. 35, the transistor Q1, the diode D4,
The resistor R5 and the resistors R3 and R4 are removed.
In this circuit, the bias voltage V _B to the base of the transistor Q3 is stabilized by the charging current I _S1 and the discharging current I _S2 , so that the self-excited oscillation is continued. Further, self-excited oscillation continues even if the diode D12 for non-regeneration to the power source E is not provided.

By the way, in the circuit of FIG. 40, the charging current I
_The bias voltage V _B increases as _S1 increases, and the ON period of the transistor Q3 increases. When this ON period increases, the energy stored in the primary winding L1 also increases.
Therefore, it is possible to control the output by controlling the charging current I _S1 from the feedback circuit 14 using this phenomenon.

41 to 43 show examples of the current I _L1 flowing through the primary winding L1 by controlling the charging current I _S1 in the circuit of FIG. 40, that is, a circuit example of a power supply device for controlling the output.

In the circuit of FIG. 41, the resistors R8a to R8d connected in series correspond to the resistor R8 of FIG. 40. By making this resistance value selectable by the switch SW1, the charging current I from the power source E is changed. _It is a variation of _S1 . The switch SW1 is composed of fixed contact pieces s1 to s3 connected to respective connection points with adjacent resistors and a movable contact piece s0 connected to the positive electrode side of the capacitor C2, and is switched by a selection signal from the feedback circuit 14a. It is supposed to be changed. When the movable contact piece s0 is connected to the fixed contact piece s1, the charging path is only the resistance R8a, and when it is connected to the fixed contact piece s2, the charging path is the resistance R8.
8a and the resistor R8b, and when connected to the fixed contact piece s3, the charging path becomes the resistors R8a, R8b and R8c.
The resistor R8d is for protection during switching.

In the above structure, the signal detection circuit 21 detects the output current I ₀ which is the load state and feeds it back to the feedback circuit 14a on the primary side. The feedback circuit 14a is configured to output a switch selection signal according to the feedback amount, and when the output current I ₀ increases, the movable contact s0 of the switch SW1 is set to, for example, so as to increase the resistance value of the charging path. By switching from the fixed contact piece s1 to the fixed contact piece s2 so that the resistance changes from R8a only to the resistances R8a and R8b, the discharge current I _S1 is reduced to 1
The increase of the current I _L1 flowing through the next winding L1, that is, the output current I ₀ is suppressed. On the contrary, when the output current I ₀ decreases, the movable contact s0 of the switch SW1 is switched from, for example, the fixed contact s3 to the fixed contact s2 so that the resistance value of the charging path is reduced, and the resistances R8a, R8b and R
By making the resistances R8a and R8b from 8c, the discharge current I _S1 is increased to suppress the decrease of the current I _L1 flowing through the primary winding _L1 , that is, the output current I ₀ .

In the circuit of FIG. 42, the transistor Q4
Is interposed between the resistor R8 and the capacitor C2. The base of the transistor Q4 is connected to the feedback circuit 14b via the base resistor R26 and the variable resistor R27 for adjusting the base current. In the above configuration, the signal detection circuit 21 detects the output current I ₀ , which is the load state, and
It is fed back to the feedback circuit 14b on the next side. The feedback circuit 14b outputs a level signal corresponding to the amount of feedback, which is the level signal of the active region of the transistor Q4. When the output current I ₀ increases, the output level from the feedback circuit 14b decreases. To increase the voltage drop of the transistor Q4 to increase the charging current I
_{If S1} is decreased and vice versa, the feedback circuit 14
The output level from b is increased to reduce the voltage drop of the transistor Q4, the charging current I _S1 is increased, and the current I _L1 flowing through the primary winding L1 by the increase / decrease control of the charging current I _S1 , that is, the output current The increase / decrease variation of I ₀ is suppressed.

In the circuit of FIG. 43, the charging current to the capacitor C2 is different from the charging current I _S1 from a different power source and the current I
The voltage on the primary side of the transformer is stabilized by supplying _S4 .

This circuit forms a winding L10 as an external control power source on the same magnetic path as the primary winding L1 of the transformer, and a diode D12 and a capacitor C13 for rectifying and smoothing the induced voltage of this winding L10. And a feedback circuit 14 in which the induced voltage is guided as a detection value.
with c. The feedback circuit 14c has the following configuration. That is, a series circuit of a constant voltage diode Z43 and a resistor R28 is connected to both ends of the capacitor C13 so that a reference voltage can be obtained at the cathode of the constant voltage diode Z43. This constant voltage is input to the + input terminal of the operational amplifier OP4 as the current limiting element, and the induced voltage of the winding L10 is smoothed to the − input terminal.
The divided voltage, which has been rectified and divided by the resistors R29 and R30, is input to form an inverting amplifier. From the output terminal of the operational amplifier OP4, a voltage corresponding to the input voltage difference between both input terminals is sent as an output V _OP , and is input as a charging current I _S4 to the positive electrode of the capacitor C2 via the resistor R31. The charging current I _S4 is the winding L10.
The induced electromotive force of is used.

Next, the operation of the above structure will be described with reference to FIG. Now, if the voltage of the power supply E rises from E ₀₁ to E ₀₂ at time T10 in the state where the voltage V _C becomes constant after the power is turned on, the charging current I _S1 slightly increases, and accordingly increases. since the the I _Dmax1 electromagnetic energy accumulated in the primary winding L1 increases, further, also increases the electrostatic energy stored in the capacitor C1, the resonance voltage V _C
Rises, the current I _L1 also rises. Therefore, the voltage induced in the winding L10 increases and the operational amplifier OP4
Since the input voltage of the negative input terminal rises, the output V _OP from the output terminal of the inverting amplifier decreases and the charging current I _S4 decreases. As a result, the rise of the bias voltage V _B2 is suppressed.

On the contrary, when the voltage of the power source E drops, the charging current I _S1 decreases to some extent, and the charging current I _S1 decreases accordingly.
_The electromagnetic energy stored in the primary winding L1 is reduced by _DMAX1, and the electrostatic energy stored in the capacitor C1 is also reduced, so that the resonance voltage V _C is reduced and the current I _L1 is also reduced. For this reason, the voltage induced in the winding L10 decreases and the input voltage of the − input terminal of the operational amplifier OP4 decreases, so that the output V _OP from the output terminal of the inverting amplifier increases and the charging current I _S4 increases. As a result, the decrease of the bias voltage V _B2 is suppressed, and thus the voltage V _C is held at a constant value.

Next, a modification of the circuit showing the fourth embodiment of the power supply device of FIG. 17 will be described with reference to FIGS. 45 to 51. FIG. 45 is a circuit diagram showing a first modified example. In the figure, the parts designated by the same reference numerals as those in FIG. 17 are the same parts and the description thereof will be omitted.

This circuit is similar to that of FIG.
Winding L5 formed on the same magnetic circuit as the next winding L1
A diode D5 and a capacitor C3 that rectify and smooth the induced voltage of the winding L5, and resistors R6 and R7 that divide the rectified and smoothed DC voltage. Further, a non-inverting amplifier circuit including an operational amplifier OP1 and resistors R81 and R9 is provided, and a reference voltage V _Y generated by a reference voltage generating circuit including a resistor R10 and a Zener diode Z30 is provided at a negative input terminal of the operational amplifier OP1.
The voltage divided by the resistors R6 and R7 is input to the + input terminal via the resistor R9 ′.
The output terminal of the operational amplifier OP1 is connected to the base of the transistor Q1 via the resistor R12.

Next, the operation will be described. During operation,
The voltage V _X proportional to the resonance voltage V _C is divided by the resistors R6 and R7 by the operational amplifier OP1, and the voltage V _X1 input via the resistor R9 ′ and the reference voltage V _Y to the resistor R9.
Is compared with the voltage V _X2 input via
The difference is amplified and the voltage V is applied to the base of the transistor Q1.
_B1 is applied.

Now, due to a decrease in load level or the like, the voltage V
_{When C} rises, the voltage V _X rises and the input voltage V _X1 of the operational amplifier OP2 rises, so that the output voltage V _{X3 of the} operational amplifier OP1 rises. Therefore, the base voltage of the transistor Q1 rises and the EFET1 is turned off earlier, so that the voltage V _C drops. On the contrary, when the voltage V _C decreases, the voltage V _X decreases and the input voltage V 1 of the operational amplifier OP1 decreases.
_{Since X1} drops, the output voltage V _{X3 of the} operational amplifier OP1 drops. Therefore, the base voltage of the transistor Q1 drops, and the EFET1 is turned off later, so that the voltage V _C rises, and thus the voltage V _C can be held constant.

By the way, in this circuit, when the power is turned on, F
Excessive voltage is generated in ET1. FIG. 46 is a waveform diagram for explaining the occurrence of this excessive voltage. In FIG. 46, when the voltage V _X transiently rises when the power is turned on, the voltage V _X1 obtained by dividing the voltage V _X by the resistors R6 and R7 also rises with a rising curve corresponding to the voltage division ratio of the voltage V _X. , _-The voltage V _X2 applied to the input terminals is obtained at an early point when the voltage V _X reaches the Zener voltage. By the way, the output voltage V _X3 of the operational amplifier OP1 is V _X1 > V
_Since it cannot be obtained until the time point τ1 at which _X2 is reached, the output timing is delayed because the voltage V _X1 has a rising curve, and during this period, the resonance voltage V _C is not controlled and an excessive voltage is generated.

FIG. 47 is a diagram for explaining the function of the winding L5 having different winding directions. (A) and (b) of FIG.
Circuit diagram that induces voltage by turning on and its voltage waveform diagram,
(C) and (d) are a circuit diagram and a voltage waveform diagram thereof for inducing a voltage when the FET 1 is off.

In (a), as shown in the voltage waveform in (b), the voltage V _X is induced in the winding L5 from the time T1 when the FET1 is first turned on after the power is turned on. In contrast,
In (c), as shown in the voltage waveform in (d), the time T2 at which the FET1 changes from the first on-state to the off-state after the power is turned on.
The voltage V _X is induced in the winding L5 from. Therefore, (a)
It can be seen that the circuit in the winding direction shown in (8) has a faster rise of the voltage V _X than the circuit shown in (c). Therefore,
The circuit of (a) can more effectively prevent the generation of an excessive voltage at the time of voltage inrush.

FIG. 48 is a circuit diagram showing a second modification. Note that, in the figure, those denoted by the same reference numerals as those in FIG. 45 are the same and their description will be omitted. This circuit includes a capacitor C14 connected in parallel with the Zener diode Z30 shown in FIG.
Is connected.

Next, the operation will be described with reference to FIG. When the voltage V _X transiently rises when the power is turned on, the voltage V _X1 obtained by dividing the voltage V _X by the resistors R6 and R7 also rises with a rising curve corresponding to the voltage division ratio of the voltage V _{X. -The} voltage V _X2 applied to the input terminal is also resistor 1
It transiently rises by the charging circuit consisting of 0 and the capacitor C14. This charging circuit is set to a time constant such that V _X1 > V _X2 from the time when the voltage V _X2 rises. Therefore, the output voltage V _X3 is generated from the operational amplifier OP2 from the time when the FET1 is first turned on, so that the transistor Q1
This prevents the voltage V _{C from} becoming an overvoltage due to feedback.

FIG. 50 is a circuit diagram showing a third modification. Note that, in the figure, those denoted by the same reference numerals as those in FIG. 45 are the same and their description will be omitted. In this circuit, a Zener diode Z31 is inserted and connected to the resistor R10 side in the series circuit of the resistor R10 and the Zener diode Z30 shown in FIG. As the Zener diode Z31, one having a Zener voltage such that the sum of Zener voltages of the Zener diodes Z30 and Z31 becomes a value close to the power supply voltage of the operational amplifier OP2 is adopted.

Next, the operation will be described with reference to FIG. When the voltage V _X transiently rises when the power is turned on, the voltage V _X1 obtained by dividing the voltage V _X by the resistors R6 and R7 also rises with a rising curve corresponding to the voltage division ratio of the voltage V _X. On the other hand, -the voltage V applied to the input terminal
_The output of _X2 cannot be obtained until time τ1 when the voltage V _X reaches the level of the sum of the Zener voltages of the Zener diodes Z30 and Z31, and the rise of V _X2 is delayed during this period. Therefore, the voltage V _X1 is V _X1 >
Since it becomes V _X2 , the output voltage V _X3 is generated from the operational amplifier OP2 from the time when the FET1 is first turned on.
Overvoltage of _C is prevented.

FIG. 52 is a circuit diagram showing a fourth modification. Note that, in the figure, those denoted by the same reference numerals as those in FIG. 45 are the same and their description will be omitted. This circuit is similar to the operational amplifier OP2 shown in FIG.
In addition to connecting P20, the voltage V _X1 is applied to the-input terminal of the operational amplifier OP20 in the opposite manner to the operational amplifier OP2,
The voltage V _X2 is introduced to the input terminal. Further, the resistor R12 and the transistor Q at the output terminal of the operational amplifier OP2
The diode D13 is inserted between the two so that the cathode is connected to the base of the operational amplifier OP20.
A resistor R120 and a diode D14 are connected in series to the output terminal of, and the cathode of the diode D14 is connected to the base of the transistor Q1.

The operation will be described below. In the circuit shown in FIG. 45, as shown in voltage waveform diagram of FIG. 46, since the output voltage V _X3 from the operational amplifier OP2 is obtained from τ1 time after power ON, the resonance voltage V _C during this rising
There is a problem that becomes too large. Therefore, during the period when the voltage V _X1 to the − input terminal is less than or equal to the input voltage V _X2 to the + input terminal,
By obtaining the output voltage V _X3 ′ from the operational amplifier OP20 and performing feedback control of the rising period from the power ON to the time τ1 with this output voltage V _X3 ′, the voltage V _C during this period becomes excessive. To prevent. The diodes D13 and D14 are for preventing the currents due to the output voltages V _X3 and V _X3 ′ output from one of the operational amplifiers from flowing into the output terminal side of the other operational amplifier.

In the circuit of FIG. 48, the output voltage V _X3 is obtained from the rise of the power source as described above. However, when a momentary power failure (hereinafter referred to as a momentary power failure) occurs, the output voltage V _X3 is generated. However, there is a problem that the voltage V _C becomes an overvoltage.

FIG. 53 is a waveform diagram for explaining the operation. Now, when the power is turned off due to an instantaneous power failure and the resonance voltage V _C decreases, the voltages V _X1 and V _X2 also decrease accordingly. At this time, the voltage V _X1 rapidly increases in proportion to the decrease in the voltage V _X. On the other hand, the voltage V _X2 transiently decreases because the capacitor C14 is provided, and becomes V _X1 <V _X2 at time τ2 after the power is turned off. Then, when the power supply is restored in this state, the voltages V _X1 and V _X2 are transiently increased by the capacitors C3 and C14 from the state of V _X1 <V _X2 at the time of momentary power failure. Therefore, immediately after the power is restored, the state of V _X1 > V _X2 cannot be achieved, and during that time, the output voltage V _X3 cannot be obtained, and the voltage V _C becomes an over voltage, and the over voltage is applied to the FET 1. Inconvenience occurs.

An embodiment of the power supply device for protection against the instantaneous power failure will be described below with reference to FIGS. 54 to 57.

FIG. 54 is a circuit diagram showing the first embodiment. Note that, in the drawing, those denoted by the same reference numerals in FIG. 48 are the same, and description thereof will be omitted.

In the first embodiment, a diode D15 is connected between the resistor R9 and the resistor R9 'shown in the circuit of FIG. 48 so that the anode faces the resistor R9 side. The diode D15 prevents the voltage V _{X2 from} becoming higher than the voltage V _X1 by the forward voltage or more.

The operation will be described below with reference to FIG.

The power is turned off due to the instantaneous power failure, and the resonance voltage V
_{When C} decreases, the voltage V _X1 decreases in proportion to this, and the voltage V _X2 also decreases following the voltage V _X1 because the diode D15 is provided. Then, when the power supply is restored after the instantaneous blackout ends, the voltages V _X1 , V
_{As X2} rises from almost the same level,
Compared with the case of 3, V _X1 > V _{X2 in a} shorter time (τ4
As a result, as a result, the output voltage V
_X3 is obtained, and the voltage V _C is prevented from becoming an overvoltage.

FIG. 56 is a circuit diagram showing the second embodiment. Note that, in the drawing, those denoted by the same reference numerals in FIG. 48 are the same, and description thereof will be omitted.

In the second embodiment, the circuit shown in FIG. 48 is provided with a detection control circuit for detecting a drop in the voltage E of the power supply and detecting this drop in voltage to decrease the voltage of the capacitor C14. In this detection control circuit, resistors R32 and R33 connected in series that divide the voltage E of the power supply are connected, a base is connected to the connection point, and a transistor Q5 and a transistor Q5 whose collectors are connected to the power supply E via a resistor R34.
A collector of which is connected to the base thereof, and a collector of which is connected to the positive electrode of the capacitor C14. The voltage dividing resistors R32 and R33 have resistance values such that the transistor Q5 is turned off when the voltage E of the power supply is regarded as an instantaneous blackout and drops to a certain level.

The operation will be described below with reference to FIG.

The power is turned off due to the instantaneous power failure, and the resonance voltage V
_{When C} begins to drop and drops to a certain level, transistor Q5 turns off. When the transistor Q5 turns off, the transistor Q6 turns on and the capacitor C
Since the electric charge is discharged from 14, the voltage V _X2 is drastically reduced as compared with the decreasing curve of the voltage V _X1 . Therefore, even during the period of the instantaneous power failure, the output voltage V _X3 is continuously output because the state of V _X1 > V _X2 is maintained. Then, even if the power supply is restored after the momentary power failure, the voltages V _X1 and V _X2 are V _X1 >.
Since it rises from the state of V _X2 , the output voltage V _X3 remains unchanged.
Is obtained, and the voltage V _C is prevented from becoming an overvoltage.

In FIGS. 48 and 50, the transistor Q1 is on-timing controlled by the voltage across the resistor R4 due to the current I _D, with the voltage V _X3 being further added. A method of controlling the on-timing of the transistor Q1 based on the voltage across the resistor R4 and subtracting the voltage V _X3 from the voltage can also be adopted. For example, a transistor or the like is interposed between the base and the emitter of the transistor Q1, and when a voltage is not generated in the winding L5 such as when the output voltage V _X3 is turned on, or a large voltage is induced in the winding L5. In this case, a negative feedback operation is performed so as to decrease to zero or less, and the intervening transistor is turned off by the output voltage V _X3 reduced to zero or so that the on-timing of the transistor Q1 is controlled by the resistor R4. The FET1 is turned off earliest by the voltage across both ends. Conversely, winding L
When the voltage induced in 5 decreases, the output voltage V
_{If X3} rises and the above-mentioned intervening transistor changes so as to approach ON in the active region, the base voltage of the transistor Q1 can be lowered so that E
As FET1 is turned off later, the voltage V _C rises, and thus the voltage V _C can be held constant.

FIG. 58 is a circuit diagram for preventing an overvoltage from being applied to the switching element when the power is turned on in the circuit of the power supply device shown in FIG. In addition, in the figure, those denoted by the same reference numerals as those in FIG. 9 are the same and the description thereof is omitted.

This circuit differs from the circuit shown in FIG. 9 in that the resistance R41 and the resistance R41 interposed between the resistance R4 and the ground.
A transistor Q8 connected in parallel with 41 and a capacitor C41 connected between the base of the transistor Q8 and the ground are provided, and the base of the transistor Q8 is connected to a power supply via a resistor R42.

In operation, the transistor Q8 is turned on in response to the voltage from the power source E and the resistor R41 is not functioning during the steady operation. In this state,
The voltage generated in the resistor R4 and the feedback voltage induced in the winding L4 are obtained by the current I _D flowing through the FET1 which is a switching element, and these voltages determine the on-time point of the transistor Q1.
When 1 is turned off, the voltage V _D applied to FET1 is
Is set. However, when the power is on, there is no voltage for feedback, so if the transistor Q8 remains on as described above, the transistor Q1
The ON time of is determined only by the voltage generated by the resistor R4. However, since there is no feedback voltage, the ON time of the transistor Q1 is delayed.
An overvoltage is applied to T1. Since the overvoltage applied to the FET 1 depends on the number of turns of the winding L4 and the resistance values of the resistors R1, R4, R5, etc., it is necessary to be careful in selecting the set value.

Therefore, as shown in the circuit of FIG. 58, a capacitor C41 is provided at the base of the transistor Q8 so that when the power is turned on, the charging time for the capacitor C41 is used to prevent the transistor Q8 from being turned on immediately. ing. That is, since the transistor Q8 is kept off by the capacitor C41 during the rising period of the power supply, the resistor R41 intervenes in series with the resistor R4, and the transistor Q1 is turned on earliest. As a result, the FET Q1 is turned on. Overvoltage is prevented from being applied. Note that the circuit configuration added to FIG.
It can also be applied to the circuit shown in (1), and by doing so, it is possible to prevent the application of an overvoltage to the FET 1 when the power is turned on.

FIG. 59 is a circuit diagram for stopping the oscillation when the load is opened in the circuit of the power supply device shown in FIG. In addition, in the figure, those denoted by the same reference numerals as those in FIG. 9 are the same and the description thereof is omitted.

This circuit is based on the Zener diode Z220.
The Zener diode Z50 is connected in parallel to the feedback path formed by the series circuit of the resistor R1 and the resistor R1.
It has a feedback path.

The operation will be described below.
As described in the operation of the circuit shown in FIG. 58, the transistor Q is driven by the feedback voltage induced in the winding L4.
The ON time of 1 is determined, and then the FET1 is turned off, so that the voltage V _D applied to the FET1 is set. By the way, when the load is opened, in the non-contact state, or when the secondary load is removed (load open), the resonance voltage V _C rises and the voltage level induced in the winding L4 rises. . The rising voltage of the winding L4 supplies the base current through the Zener diode Z220 and the resistor R1 to accelerate the on-timing of the transistor Q1 and
The time when T1 is turned off is made earlier to suppress the level of the voltage V _C. However, in case of the above load open, FET
It is necessary to stop the oscillation of 1.

Therefore, as shown in the circuit of FIG.
A Zener diode Z50 is provided as a feedback path of the above, and a Zener voltage having a level slightly lower than the high voltage induced in the winding L4 when the load is open is adopted. Then, when the load is opened, the Zener diode Z50 is turned on and the high voltage induced in the winding L4 directly causes the resistance R1.
The base current is supplied to the transistor Q1 (without interposing a resistor such as), so that the transistor Q1 is completely turned on, and as a result, the FET1 is maintained in the off state. When the oscillation of the FET1 is stopped, the voltage V _C is attenuated and the induced voltage of the winding L4 is lowered, while the capacitor C2 is gradually charged through the resistor R8, so that the oscillation is restarted. Even if is restarted, if the load open state continues, the voltage V _C becomes the high level again, so that the FET1 is kept off in the same manner as described above, and by repeating this operation, the load open state is obtained. During a certain period, the intermittent oscillation operation is performed, the primary current is reduced, and the loss can be suppressed as much as possible. The circuit configuration added to FIG. 59 is similar to that of FIG.
It is also applicable to the circuit shown in (1), and by doing so, it is possible to suppress power loss when the load is open.

Furthermore, the present invention can employ the following modified examples.

(1) In each of the above embodiments, the transformer is 1
Although the description is made by using the one divided into the secondary side and the secondary side, the same effect can be obtained by using the one having the higher order winding.

(2) In the transformer, the core has an integral type common to the primary side and the secondary side, and has a gap between the primary side and the secondary side. The primary side core and the secondary side core are The same effect can be obtained by using either of the completely separated non-contact electromagnetic induction.

(3) The load E2 on the secondary side of the transformer is 2
Not only the secondary battery, but also other loads such as motors,
The same effect can be obtained. The output of the secondary winding L2 is full-wave rectified by the center tap using the diodes D1 and D2, but may be rectified and smoothed by another rectifying / smoothing means using a coil, a capacitor, or the like. The same effect can be obtained.

(4) In each of the above embodiments, the windings L4 and L5 and the windings L8 and L formed in the same magnetic path of the transformer.
Although 10 performs rectification and smoothing using a half-wave rectification of one diode and a capacitor input type smoothing circuit, the same effect can be obtained by using another rectification and smoothing method such as full-wave rectification. it can.

(5) In each of the above embodiments, the U-U core shape is used as the core separation type transformer. However, if it is used by being separated and attached, a pot core type, CC core type,
The same effect can be obtained even in the coreless type and other shapes.

(6) In each of the above embodiments, each circuit element has been described as having a simplified structure. However, if the same operation is performed, the arrangement or order of these elements may be changed, or other elements may be changed. Even if elements are connected in series or in parallel and combined, the same effect can be obtained.

[0248]

As described above, according to the invention of claim 1, when the current flowing through the switching element reaches a predetermined level, the switching element is turned on to form a discharge loop.
All the discharge current flowing through the switching hand
Since the stage is provided , the bias voltage of the bias voltage generating circuit can be held at a stable voltage.

According to the second aspect of the invention, since the electric charge accumulated in the capacitor is discharged when the switching means is turned on, the bias voltage can be held at a stable voltage.

Further, according to the invention of claim 3, since the switching means is connected to the switching control end of the switching element, when the current flowing through the switching element reaches a predetermined level, the switching element is turned off. Therefore, the current flowing through the switching element can be controlled, and an excessive current can be prevented from flowing through the switching element.

Further, according to the invention of claim 4, since the switching means is controlled based on the feedback signal generated from the electric signal in the circuit of 1,
The output can be kept almost constant.

Further, according to the invention of claim 5, since the feedback signal is generated according to the voltage level of the power source, the output can be kept substantially constant even when the voltage level of the power source is different. .

Further, according to the invention of claim 6, since the feedback signal according to the electric signal generated on the primary side of the transformer is generated, even if the level of the load or the power supply voltage is changed, The output on the primary side of can be kept almost constant. Therefore, since it is possible to prevent application of overvoltage to the switching element, it is possible to prevent voltage breakdown and shortening of life of the element. Also, the output to the load can be kept almost constant.

Further, according to the invention of claim 7, since the feedback signal is generated in accordance with the electric signal generated on the secondary side of the transformer, even if the load or the level of the power supply voltage changes, The output on the secondary side of can be kept almost constant. Therefore, it is possible to always supply a constant power to the load.

According to the ninth aspect of the invention, the on / off of the switching means is controlled based on the feedback signal generated from the electric signal in the one circuit, so that the output is kept substantially constant. be able to.

According to the tenth aspect of the invention, since the resonance voltage of the resonance circuit is made constant, the resonance voltage can be kept substantially constant even if the load or the level of the power supply voltage changes. it can. Therefore, since it is possible to prevent application of overvoltage to the switching element, it is possible to prevent voltage breakdown and shortening of life of the element.

According to the eleventh aspect of the invention, since the resonance voltage of the resonance circuit is made constant by the automatic control circuit combining the proportional control and the integral control, the level of the load, the power supply voltage, etc. fluctuates. However, the resonance voltage can be kept almost constant.

According to the twelfth aspect of the invention, when the output of the primary side of the transformer is lowered due to an increase in load level or the like, the electric charge of the capacitor is discharged through the resistor. Therefore, the feedback circuit is detected. The winding can output a signal according to the output of the primary winding of the transformer without delaying the output reduction. Therefore, the output on the primary side of the transformer can be kept substantially constant in response to the level fluctuations of the load, the power supply voltage and the like.

According to the thirteenth aspect of the invention, since the feedback signal is output from the ON side of the detection winding, the feedback signal can be output to the switching means immediately after the power is turned on. Therefore, the oscillation can be stably continued.

According to the fourteenth aspect of the present invention, since the current control circuit is provided and the current control circuit is provided with the circuit for feeding back a part of information of the device so that the output is kept substantially constant, the load control circuit is provided. The output can be made almost constant regardless of the state.

According to the fifteenth aspect of the present invention, since the detection circuit for detecting the load-side state signal and the signal processing circuit for performing signal processing on the detected signal and guiding the signal to the feedback circuit are provided. By directly detecting the state of the load on the secondary side, the output can be made substantially constant more accurately.

Further, according to the sixteenth aspect of the present invention, since the detected signal is transmitted to the primary side via the transformer, it is not necessary to separately provide a transmission circuit, and the size and cost of the device can be reduced. Can be down.

Further, according to the invention of claims 17 to 19,
As the detection circuit, one for detecting the output current to the secondary side load, one for detecting the output voltage to the secondary side load, and one for detecting the output to the secondary side load are provided. A circuit can be configured according to the object to be made constant and the application.

According to the twentieth aspect of the present invention, the transformer is provided with the circuit for voltage-frequency converting the detection signal and the circuit for frequency-voltage converting the signal transmitted via the transformer. It can be used for both power transmission and detection signal transmission.

According to the twenty-first aspect of the invention, the load signal state can be detected promptly and accurately by adopting a configuration capable of transmitting the detection signal and power transmission in parallel.

Further, according to the invention of claim 22, since the switching circuit for alternately switching the transmission period of the detection signal and the power transmission period is provided, the detection signal can be obtained more reliably on the primary side. You can

According to the twenty-third aspect of the present invention, the transformer is detachable between the primary side and the secondary side having a load, and the power is transferred in a contactless manner. It can be applied to a wide range of applied equipment.

According to the twenty-fourth aspect of the present invention, the power supply to the secondary side is provided only when the regular load is placed, which is provided with the discrimination circuit for detecting whether the placed load is regular or not. Since the power is supplied, it is possible to prevent power loss when there is no load or a foreign object is placed, and it is possible to reliably prevent unexpected heat generation of the foreign object.

According to the twenty-fifth aspect of the present invention, since the load is discriminated based on the information on the primary side, there is no need to additionally provide a configuration for exchanging information with the load side, and the configuration is simple. Be converted.

According to the twenty-sixth aspect of the present invention, since the load is discriminated based on the information from the secondary side, it is possible to surely discriminate.

According to the inventions of claims 27 to 30,
As the information on the primary side, the information on the inductance change, that is, the change of the resonance frequency, the change of the current, or the change of the voltage can be used. It can be adopted selectively depending on the conditions.

According to the inventions of claims 31 to 33,
By providing a self-bias circuit that applies a self-bias between the base of the transistor and the ground, it is possible to use a transistor instead of the switching element, so that the selectivity of the switching element can be expanded and the cost can be reduced.

According to the inventions of claims 34 to 36,
The self-bias circuit can be realized by a simple element.

According to the thirty-seventh aspect of the present invention, a transistor can be adopted instead of the switching element, and the ON period of this transistor is controlled by controlling the charging current of the bias circuit of the self-excited oscillation circuit. Therefore, the output can be kept substantially constant even when the load changes.

According to the thirty-eighth aspect of the invention, since the charging current of the bias circuit is controlled by switching the resistance value, the charging current can be controlled with a simple structure.

According to the thirty-ninth aspect of the invention, since the charging current of the bias circuit is controlled by the current control element inserted in series with the bias circuit, the charging current can be controlled with a simple structure. Becomes

According to the forty-third aspect of the invention, since the control of the charging current by the current control element is performed by the external control power supply, reliable control is possible.

According to the forty-first aspect of the invention, since the external control power source is generated by the induced electromotive force of the transformer, it is not necessary to attach a separate power source, and the size and cost of the device can be reduced. .

According to the invention of claim 42, the feedback circuit generates a feedback signal for preventing an excessive current from flowing into the switching means when the power is turned on. Can be protected.

According to the forty-third aspect of the invention, the feedback circuit generates the feedback signal for intermittently oscillating the oscillation of the switching element while the load is open, so that power loss can be prevented during this period. .

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a first embodiment of a power supply device according to the present invention.

FIG. 2 is a waveform diagram showing the operation of the first embodiment.

FIG. 3 is a circuit diagram showing a second embodiment of the power supply device according to the present invention.

FIG. 4 is a waveform chart showing the operation of the second embodiment.

FIG. 5 is a circuit diagram showing a modification of the second embodiment.

FIG. 6 is a circuit diagram showing another modification of the second embodiment.

FIG. 7 is a circuit diagram showing a third embodiment of the power supply device according to the present invention.

FIG. 8 is a waveform chart showing the operation of the third embodiment.

FIG. 9 is a circuit diagram showing a modification of the third embodiment.

FIG. 10 is a circuit diagram showing another modification of the third embodiment.

11 is a voltage waveform chart showing the operation of the circuit of FIG.

FIG. 12 is a circuit diagram showing a modified example of FIG.

13 is a waveform diagram illustrating the operation of the circuit of FIG.

14 is a waveform diagram illustrating the operation of the circuit of FIG.

FIG. 15 is a circuit diagram showing another modification of the third embodiment.

FIG. 16 is a circuit diagram showing another modification of the third embodiment.

FIG. 17 is a circuit diagram showing a fourth embodiment of the power supply device according to the present invention.

FIG. 18 is a circuit diagram showing a modification of the fourth embodiment.

FIG. 19 is a circuit diagram showing a fifth embodiment of the power supply device according to the present invention.

FIG. 20 is a circuit diagram showing a sixth embodiment of the power supply device according to the present invention.

FIG. 21 is a block diagram of a power supply device showing an outline of a peripheral protection circuit.

22 is a block diagram of a power supply device more specifically showing the signal processing circuit shown in FIG. 21, in which power transmission and load state signal transmission are performed simultaneously.

23 is another block diagram of the power supply device shown in FIG.
The power transmission and the load state signal transmission are switched.

FIG. 24 is a block diagram for each form of a signal detected by a signal detection circuit, which is for detecting an output current.

FIG. 25 is a block diagram for each form of a signal detected by a signal detection circuit, which detects a voltage.

FIG. 26 is a block diagram for each form of a signal detected by a signal detection circuit for detecting electric power.

27 is a more detailed block diagram of the power supply device shown in FIG. 24 for stabilizing the output current.

FIG. 28 is a waveform chart for explaining the operation.

FIG. 29 is a block diagram of a power supply device showing an outline of a peripheral protection circuit using a change in resonance frequency.

FIG. 30 is a block diagram of a power supply device showing an outline of a peripheral protection circuit using a change in resonance current.

FIG. 31 is a block diagram of a power supply device showing an outline of a peripheral protection circuit using a change in resonance voltage.

FIG. 32 is a block diagram of a power supply device showing an outline of a peripheral protection circuit using information from the load side.

33 is a circuit block diagram showing details of the primary side of a power supply circuit using the frequency detection circuit shown in FIG. 29.

FIG. 34 is a waveform chart for explaining the operation.

FIG. 35 is a circuit diagram in which the FET is replaced with a current drive type transistor.

FIG. 36 is a circuit diagram showing a first countermeasure example.

FIG. 37 is a circuit diagram showing a second countermeasure example.

FIG. 38 is a circuit diagram showing a third countermeasure example.

FIG. 39 is a circuit diagram showing a fourth countermeasure example.

FIG. 40 is a circuit diagram of a power supply device that performs self-oscillation using a transistor.

41 is a circuit diagram of a power supply device that controls the current flowing through the primary winding, that is, the output by controlling the charging current I _S1 in the circuit of FIG. 40.

42 is another circuit diagram of the power supply device that controls the current flowing through the primary winding by controlling the charging current I _S1 in the circuit of FIG. 40, that is, the output control.

43 is still another circuit diagram example of the power supply device that controls the current flowing through the primary winding by controlling the charging current I _S1 in the circuit of FIG. 40, that is, the output control.

FIG. 44 is a waveform chart for explaining the operation.

45 is a circuit diagram showing a first modified example of the circuit showing the fourth example of the power supply device of FIG. 17. FIG.

FIG. 46 is a waveform diagram for explaining generation of an excessive voltage.

47 (a) and 47 (b) are circuit diagrams for inducing a voltage when the FET1 is turned on and a voltage waveform diagram thereof, and FIG. ), (D)
FIG. 3 is a circuit diagram for inducing a voltage when the FET 1 is off and a voltage waveform diagram thereof.

48 is a circuit diagram showing a second modified example of the circuit showing the fourth example of the power supply device of FIG. 17. FIG.

FIG. 49 is a waveform chart for explaining the operation.

50 is a circuit diagram showing a third modified example of the circuit showing the fourth example of the power supply device of FIG. 17. FIG.

FIG. 51 is a waveform chart for explaining the operation.

52 is a circuit diagram showing a fourth modified example of the circuit showing the fourth example of the power supply device of FIG. 17. FIG.

FIG. 53 is a waveform diagram for explaining an operation in which the voltage V _C becomes an overvoltage.

FIG. 54 is a circuit diagram showing a first embodiment of a power supply device for protection against momentary power failure.

FIG. 55 is a waveform chart for explaining the operation.

FIG. 56 is a circuit diagram showing a second embodiment of the power supply device for protection against momentary power failure.

FIG. 57 is a waveform chart for explaining the operation.

58 is a circuit diagram for preventing an overvoltage from being applied to the switching element when the power is turned on in the circuit of the power supply device shown in FIG. 9.

59 is a circuit diagram for stopping oscillation when the load is opened in the circuit of the power supply device shown in FIG. 9.

FIG. 60 is a circuit diagram showing an example of a conventional power supply device.

FIG. 61 is a circuit diagram showing an example of a conventional power supply device.

FIG. 62 is a waveform diagram showing the operation of the conventional example.

[Explanation of symbols]

1 FET (switching element) OP1 to OP4, OP20 operational amplifier A, B power supply terminals C1 to C6, C10 to C14, C41, C71 capacitors D1, D2, D4, D5, D7 to D15 diode E power supply E2 load L1 primary winding L2 secondary winding L3 feedback winding L4, L5, L6 winding (detection winding) L7, L8, L10 winding K1 primary core K2 secondary core LD light emitting element P1 light receiving element Q1, Q2 transistor (switching means) Q3 transistor (switching element) Q4 to Q6, Q8 transistors R1 to R14, R19, R20 to R34, R41, R4
2, R71, R81, R120, R8a to R8d Resistors Z12, Z20, Z30, Z31, Z40 to Z42, Z
50, Z220 Zener diode 11 Current control circuit 12 Switching element 13 Resonance circuit 14, 14a to 14c Feedback circuit 15 Signal processing circuit 150 Amplifier circuit 151 Limiter circuit 152 Waveform shaping circuit 153 f-V conversion circuit SW switching circuit 17 Frequency detection circuit 171 f-V conversion circuit 172 comparison circuit 173 pulse generation circuit 18 current detection circuit 19 voltage detection circuit 20 load signal detection circuit 21 signal detection circuit 21a current detection circuit 21b voltage detection circuit 21c output detection circuit 22 signal processing circuit 221 V-f conversion Circuit 222 Wave shaping circuit SW1 switch

Front page continuation (72) Inventor Hideki Tamura 1048, Kadoma, Kadoma, Osaka Prefecture, Matsushita Electric Works, Ltd. (72) Inventor, Satoru Katakata, 1048, Kadoma, Kadoma, Osaka Prefecture, Matsushita Electric Works, Ltd. (72) Inventor Hiroyasu Kitamura 1048, Kadoma, Kadoma City, Osaka Prefecture, Matsushita Electric Works, Ltd. (56) Reference JP-A-6-46573 (JP, A) JP-A-6-70461 (JP, A) JP-A-6- 105542 (JP, A) European Patent Application Publication 86521 (EP, A1) (58) Fields searched (Int.Cl. ⁷ , DB name) H02M 3/338 H02M 3/28

Claims (43)

(57) [Claims] 1. A and a switching element and a feedback winding,
Connected to a switching control terminal of the switching element via the feedback winding, and a self-excited oscillation circuit for self-oscillating a resonance circuit composed of a primary winding of a transformer and a capacitor connected in series to the switching element. And a bias voltage generating circuit for supplying power to a load connected to the secondary side of the transformer when the power source is connected, a discharge loop is formed when the switching means is turned on.
To maintain a stable bias voltage,
All discharge current flowing through the switching means.
Power supply characterized in that it is. 2. The power supply device according to claim 1, wherein the bias voltage generating circuit comprises a capacitor, and the switching means discharges the electric charge accumulated in the capacitor when turned on. 3. The power supply device according to claim 1, wherein the switching means is connected to a switching control end of the switching element. 4. The power supply device according to claim 3, further comprising a feedback circuit that generates a feedback signal from an electric signal in one circuit and outputs the feedback signal to the switching means. 5. The power supply device according to claim 4, wherein the feedback circuit generates a feedback signal according to a voltage level of the power supply. 6. The power supply device according to claim 4, wherein the feedback circuit is for generating a feedback signal according to an electric signal generated on the primary side of the transformer. 7. The power supply device according to claim 4, wherein the feedback circuit generates a feedback signal according to an electric signal generated on the secondary side of the transformer. 8. The power supply device according to claim 5, wherein the feedback circuit comprises at least one of a resistor and a constant voltage element. 9. The power supply device according to claim 4, wherein a current detection resistor for detecting a current flowing through the switching element, and a resistor interposed between the current detection resistor and a switching control end of the switching means. The power supply device is characterized in that the feedback circuit is configured so that the feedback signal is output between the switching control end of the switching means and the resistor. 10. The power supply device according to claim 6, wherein the feedback circuit keeps the resonance voltage of the resonance circuit constant. 11. The power supply device according to claim 10, wherein the feedback circuit is an automatic control circuit that combines proportional control and integral control. 12. The feedback circuit comprises a detection winding magnetically coupled to the primary winding, a capacitor for smoothing an induced voltage in the detection winding, and a resistor connected in parallel with the capacitor. The power supply device according to claim 6, which is a power supply device. 13. The feedback circuit includes a detection winding magnetically coupled to the primary winding, and the feedback signal is output from an ON side of the detection winding. 6. The power supply device according to 6. 14. The power supply device according to claim 1, further comprising a current control circuit, wherein the current control circuit is provided with a circuit for feeding back a part of information of the device so as to make an output substantially constant. Characteristic power supply device. 15. The power supply device according to claim 14,
A power supply device comprising: a detection circuit that detects a load-side state signal; and a signal processing circuit that performs signal processing on the detected signal and guides the signal to the feedback circuit. 16. The power supply device according to claim 15, wherein the signal processing circuit transmits the detected signal to the primary side via the transformer. 17. The detection circuit detects an output current to a secondary side load.
The power supply described. 18. The detection circuit detects an output voltage to a secondary side load.
The power supply described. 19. The power supply device according to claim 15, wherein the detection circuit detects an output to the secondary load. 20. The signal processing circuit includes a circuit for voltage-frequency converting the detection signal detected by the detection circuit, and a circuit for frequency-voltage converting the signal transmitted via the transformer. 17. The method according to claim 16, wherein
The power supply described. 21. The power supply device according to claim 16, wherein the signal transmission of the detection signal through the transformer is performed in parallel with the power transmission performed through the transformer. 22. The power supply device according to claim 16,
A power supply device comprising: a switching circuit that alternately switches a period of signal transmission performed through the transformer for the detection signal and a period of power transmission performed through the transformer. 23. The power supply device according to claim 1, wherein the transformer is attachable / detachable between a primary side and a secondary side having a load, and transmits power in a contactless manner. . 24. The power supply device according to claim 23,
A power supply characterized by being provided with a discrimination circuit for detecting whether or not the placed load is regular, and supplying power to the secondary side only when a regular load is placed. apparatus. 25. The power supply device according to claim 24, wherein the discriminating circuit discriminates the load based on information on the primary side. 26. The discriminating circuit discriminates a load based on information from the secondary side.
4. The power supply device according to 4. 27. The power supply device according to claim 25, wherein the information on the primary side is a change in inductance. 28. The power supply device according to claim 27, wherein the information on the primary side is a change in resonance frequency. 29. The power supply device according to claim 27, wherein the information on the primary side is a change in current. 30. The power supply device according to claim 27, wherein the information on the primary side is a change in voltage. 31. The power supply device according to claim 1, wherein the switching element is composed of a transistor, and a self-bias circuit for applying a self-bias is provided between the base of the transistor and the ground. 32. The self-biasing circuit is provided between the emitter of the transistor and a current detection resistor connected to the emitter.
The power supply described. 33. The power supply device according to claim 31, wherein the self-bias circuit is connected in series to the base of the transistor. 34. The power supply device according to claim 32, wherein the self-bias circuit is a constant voltage element. 35. The power supply device according to claim 34, wherein the constant voltage element is a constant voltage diode. 36. The power supply device according to claim 34, wherein the constant voltage element is formed by connecting a plurality of diodes in series. 37. The power supply device according to claim 1, wherein the switching element is composed of a transistor, and the ON period of the transistor is controlled by controlling a charging current of a bias circuit of the self-excited oscillation circuit. . 38. The power supply device according to claim 37, wherein the charging current of the bias circuit is controlled by switching a resistance value. 39. The power supply device according to claim 37, wherein the charging current of the bias circuit is controlled by a current control element inserted in series with the bias circuit. 40. The power supply device according to claim 39, wherein the current control element is configured to control a charging current by an external control power supply. 41. The power supply device according to claim 40, wherein the external control power supply is generated by an induced electromotive force of the transformer. 42. The power supply device according to claim 6, wherein the feedback circuit generates a feedback signal for preventing an excessive current from flowing into the switching means when the power is turned on. 43. The power supply device according to claim 6, wherein the feedback circuit generates a feedback signal for intermittently oscillating the switching element while the load is open.