JP3494109B2 - Bandpass filter using TEM mode dielectric resonator - Google Patents

Description

Detailed Description of the Invention

[0001]

TECHNICAL FIELD The present invention has a high no-load Q value and is 1/4 of a low-profile TEM mode (main mode).
The present invention relates to a two-stage bandpass filter using a wavelength plane type dielectric resonator. The coupling between the two resonators is an evanescent E
This is done in a mode waveguide, by coupling the open faces opposite the shorted faces of these resonators. The filter of the present invention is a wideband CDMA (Code Divisi).
on Multiple Access (code division multiplexing) and other cellular phone systems, wireless LAN
In (Local Area Network),
And can be used in other communication systems where filtering is required.

[0002]

2. Description of the Related Art (1) Arun Chandra Kundu and Iku Awai, "Low profile dual mode BPF using rectangular dielectric disk resonator", 1997 China Chapter of Electrical and Information Related Society Union Conference Proceedings,
272, (2) Yoshihiro Konishi, "New Dielectric Waveguide Components-Application of New Ceramic Materials to Microwaves", IEEE Bulletin, Vol. 79, No. 6, 72nd
There are pages 6-740, June 1991.

Further, although not publicly known, there are (3) Japanese Patent Application 2000-012939 and (4) Japanese Patent Application 2000-012940 as patent applications already submitted.

Arun Chandra Kundu, one of the inventors of the present application, discloses in the literature (1) that a TEM dual-mode dielectric disk resonator of a new type having the following structure and a dual using the resonator. We propose a modal bandpass filter.

This dielectric resonator is a dual mode resonator having a square planar shape of 5 mm × 5 mm,
Its upper and lower surfaces are coated with a silver layer. The silver layer on the upper surface side is floating, and the silver layer on the lower surface side is grounded. The inside sandwiched between these two silver layers is filled with a dielectric material having a relative dielectric constant ε _r = 93. All side surfaces of the disk resonator are open surfaces exposed in the air. Therefore, radiation is easily generated through these open surfaces and the disk resonator operates as a half-wave resonator. The electric field has a maximum at the magnetic wall of the resonator and a minimum at the plane of symmetry of the resonator. Therefore, this type of resonator is called a half-wave (λ / 2) dielectric disk resonator.

FIG. 1 is a characteristic diagram showing the results of theoretically and experimentally confirming the relationship between the thickness t and the unloaded Q value of this disk resonator.
It is described in. As is clear from the figure, this disk resonator uses a dielectric material having a relative permittivity ε _r = 93, and when the length and width are 5 mm × 5 mm, the thickness t is 1 mm. The load Q value becomes maximum (≈250 (experimental value)).

When a bandpass filter of 2 GHz is formed by using such a resonator, the dimensions of the resonator and the filter are 8.5 mm × 8.5 mm × 1 mm, and the unloaded Q
The value is about 260.

[0008]

However, recent construction of mobile terminals requires the use of much smaller and higher performance resonators in the filter. In order to achieve this object, the present inventors proposed a very miniaturized dielectric resonator for 2 GHz in the above-mentioned patent application (3). The dimensions of this dielectric resonator are 3 mm x 4.2.
It is 5 mm × 1 mm, and the unloaded Q value is about 240.

In the above-mentioned patent application (4), the present inventors propose to form a two-stage bandpass filter by using two λ / 4 resonators.

That is, in this patent application, the bandpass filter includes first and second dielectric resonators each including a dielectric block having an upper plane, a lower plane and four side surfaces, and an evanescent mode. And a waveguide coupling part of. Each of the first and second dielectric resonators includes a first and second metal layer coated on the upper plane and the lower plane, respectively, and a third metal layer coated on one of the four side surfaces. A third metal layer, the side surface coated with the third metal layer serves as a short-circuit surface, and the remaining three side surfaces out of the four side surfaces serve as open surfaces, thereby operating as a λ / 4 dielectric resonator and electromagnetic field. It is configured to maintain a field independent TEM mode. The waveguide coupling section of the evanescent mode connects the short-circuited surfaces of the first and second dielectric resonators to each other and TE of the first and second dielectric resonators is connected to each other.
It provides M-mode coupling and is configured to operate in an evanescent mode having a cutoff frequency higher than the resonant frequency of each of the first and second dielectric resonators.

The present invention, like this patent application (4),
It is intended to realize a small bandpass filter by using the extremely miniaturized dielectric resonator described above.
An object of the present invention is to provide a bandpass filter using a TEM mode dielectric resonator, which can achieve further miniaturization while maintaining good filter performance.

[0012]

According to the present invention, a TEM
A bandpass filter using a mode dielectric resonator includes first and second dielectric resonators each including a dielectric block having an upper plane, a lower plane and four side surfaces, and an evanescent E mode. And a waveguide coupling part of.
Each of the first and second dielectric resonators includes a first and second metal layer coated on the upper plane and the lower plane, respectively, and a third metal layer coated on one of the four side surfaces. A third metal layer, the side surface coated with the third metal layer serves as a short-circuit surface, and the remaining three side surfaces out of the four side surfaces serve as open surfaces, thereby operating as a λ / 4 dielectric resonator and electromagnetic field. It is configured to maintain a field independent TEM mode. The evanescent E-mode waveguide coupling section connects the open surfaces of the first and second dielectric resonators facing the short-circuited surfaces to each other and couples the TEM mode of the first and second dielectric resonators. Offers the first and second
Is configured to operate in an evanescent E-mode having a cutoff frequency higher than the resonance frequency of each of the dielectric resonators.

As previously mentioned, using a TEM dual mode half-wave structure to form a dual mode filter, the filter size for 2 GHz is 8.5 mm ×
It will be 8.5 mm × 1 mm. In the above-mentioned patent application (3), by making this resonator a TEM mode λ / 4 dielectric resonator, its dimensions are 3 mm × 4.25 mm × 1 m.
m and the no-load Q value are optimized to 240.

The T in the above-mentioned patent application (3) will be described below.
The EM mode λ / 4 dielectric resonator will be described.

FIG. 2 shows a structure of a general λ / 2 dielectric resonator, and FIG. 3 shows λ in the above-mentioned patent application.
It is a perspective view which shows the basic composition of a / 4 dielectric resonator.

In FIG. 2, 20 is a dielectric block having a rectangular planar shape, 21 is a metal layer coated on the upper surface of the dielectric block 20, and 22 is a metal layer coated on the lower surface of the dielectric block 20. Are shown respectively. The upper metal layer 21 is floating, and the lower metal layer 22 is grounded. All four side surfaces of the dielectric block 20 are open surfaces. Figure 2
, The λ / 2 dielectric resonator has a length of a and a thickness of t.
Indicated by.

In this λ / 2 dielectric resonator, assuming a TEM mode in the z-axis direction, the maximum negative electric field is in the plane of z = 0 and the maximum positive electric field is as shown by an arrow 23 in FIG. Is in the plane of z = a. The minimum (zero) electric field is clearly in the plane 24 of z = a / 2, which is the plane of symmetry of the resonator.

If such a λ / 2 dielectric resonator is divided and considered along the plane of symmetry 24, two λ / 4 dielectric resonators can be obtained, and each λ / 4 dielectric resonator is obtained. In the resonator, the z = a / 2 plane acts as a perfect electrical conductor (PEC).

FIG. 3 shows a λ / 4 dielectric resonator thus obtained. In FIG. 3, 30 is a dielectric block having a rectangular plane shape, and 31 is a coating on the upper surface of the dielectric block 30. The metal layers 32 are metal layers coated on the lower surface of the dielectric block 30, respectively. The metal layer 32 on the lower surface is grounded. The metal layer 34 on one side corresponds to a complete electric conductor (PEC) of the λ / 2 resonator, short-circuits the metal layer 31 on the upper surface and the metal layer 32 on the lower surface, and the other three side surfaces are It is an open surface. In the figure, an arrow 33 indicates an electric field, and an arrow 3 indicates
5 has shown the electric current, respectively.

The λ / 4 dielectric resonator of FIG. 3 and the λ / of FIG.
The two dielectric resonators have the same resonance frequency in principle. Since the dielectric has a relatively high relative permittivity of ε _r = 93, the electromagnetic field confinement characteristic is sufficiently strong,
Moreover, the electromagnetic field distribution of the λ / 4 dielectric resonator is divided into λ / 2 along the symmetry plane 24 of the λ / 2 dielectric resonator.
It is almost the same as the electromagnetic field distribution of the dielectric resonator. As shown in FIGS. 2 and 3, the volume of the λ / 4 dielectric resonator is λ / 2.
It is half that of a dielectric resonator. As a result, the total energy amount of the λ / 4 dielectric resonator is also half that of the λ / 2 dielectric resonator. Nevertheless, since the energy loss is reduced to about 50% of the λ / 2 dielectric resonator, the unloaded Q value of the λ / 4 dielectric resonator is half that of the λ / 2 dielectric resonator. Become. That is, the λ / 4 dielectric resonator can be significantly reduced in size without changing the resonance frequency and the unloaded Q value.

When a filter of 2 GHz is formed by using the dual mode dielectric resonator according to the known document (1), the size of the filter becomes 8.5 mm × 8.5 mm × 1 mm. However, the dimensions are 3 mm x 4.25 mm x 1 m
When a two-stage bandpass filter is constructed by using two λ / 4 dielectric resonators according to the above-mentioned patent application (3) having an m and an unloaded Q value of about 240, the size of the filter is 3 m.
It will be m × 8.8 mm × 1 mm.

That is, the volume of the filter according to the present invention is 1/3 that of the conventional filter. Moreover, the performance of the filter of the present invention is sufficiently good. The coupling between the two resonators is done by providing an evanescent E-mode waveguide between the open front faces.

Filters using dielectric resonators are well known because of their versatility in the fields of electronics and communications. In particular, waveguide filters are used in base stations and mobile stations.

The two-stage and multi-stage filters using the λ / 4 waveguide resonator have been studied by Mr. Konishi as described in the above-mentioned document (2). This filter is a TE mode dielectric waveguide resonator type filter, and is superior in performance, size and volume to a direct coupling cavity filter.

However, recent mobile terminals require a filter that is extremely small and has higher performance in order to be a small and lightweight mobile terminal. Therefore, in the present invention, the TEM mode λ / 4 dielectric is used. The body resonator is used to form a two-stage bandpass filter. While the resonance frequency of the TE mode (lowest order mode) resonator changes depending on its length and thickness or width, TEM
The resonant frequency of a mode resonator depends almost exclusively on the length of the resonator. Therefore, the unloaded Q-factor can be optimized as a function of resonator thickness and width without taking the resonance frequency into account. By this method, it is possible to provide a bandpass filter which is smaller and has higher performance than those of the prior art.

The evanescent E-mode waveguide coupling portion preferably has an open upper surface and four side surfaces, and a metal-coated lower surface.

It is highly preferred to have an attenuation pole on each side of the pass band. The bandpass filter of the present invention has unexpected attenuation poles on both sides of its pass band, which improves the characteristics of the band outside the pass band. That is, the above-mentioned patent application (4)
Although the bandpass filter described in No. 1 did not have attenuation poles on both sides of the pass band, this band pass filter of the present invention has attenuation poles on both sides of the pass band, which allows the skirt of the filter to pass. The characteristics of the part can be further improved. Specifically, the direct coupling and the internal coupling between the first and second dielectric resonators through the evanescent E-mode waveguide coupling section are different couplings between the capacitive coupling and the inductive coupling. Is configured.

It is preferable that the first and second dielectric resonators are made of the same dielectric material. More preferably, these first and second dielectric resonators are made of a ceramic dielectric material having a high dielectric constant. It is more preferable that the evanescent E-mode waveguide coupling portion also uses the same dielectric material as the first and second dielectric resonators.

It is preferable that the first and second dielectric resonators have substantially the same dimensions.

It is preferable that the evanescent E-mode waveguide coupling portion has a length and a cross section shorter than the length and cross section of each of the first and second dielectric resonators. More preferably, the dimensions of this evanescent E-mode waveguide coupling are selected such that the coupling between the first and second dielectric resonators is at the desired value.

It is also preferred that the evanescent E-mode waveguide coupling has a rectangular cross section.

Preferably, the evanescent E-mode waveguide coupling is configured to provide a series capacitance and a pair of shunt capacitances between the first and second dielectric resonators.

The lower second metal layer of each of the first and second dielectric resonators is preferably used as a ground plane. The metal coated surface of the evanescent E-mode waveguide coupling is also preferably used as the ground plane.

It is preferable that one side surface of each of the first and second dielectric resonators, which is orthogonal to the short-circuit surface, is used for capacitive excitation. This capacitive excitation is
This will be done with electrical input / output ports on this side.

This electrical input / output port is preferably formed by providing a rectangular, square, trapezoidal or circular metal pattern on this side surface.

The metal pattern is preferably separated from the first metal layer on the upper plane and the second metal layer on the lower plane.

More preferably, the dimensions of the metal pattern are selected so that the strength of the connection with the external circuit has a desired value.

A narrow slit for frequency adjustment is preferably provided in the upper first metal layer on at least one of the first and second dielectric resonators.

[0039]

FIG. 4 is a perspective view schematically showing the structure of a high frequency dielectric resonator type bandpass filter having two dielectric resonators as an embodiment of the present invention, and FIG. FIG. 6 is an exploded perspective view of the bandpass filter, and FIG. 6 is a perspective view schematically showing the configuration of each dielectric resonator.

In these figures, 40 is the first λ / 4
A dielectric resonator 41 is a second λ / 4 dielectric resonator, respectively. These two λ / 4 dielectric resonators 40
And 41 are connected to each other via an evanescent E-mode dielectric waveguide (blocking waveguide) 42.

Each of the λ / 4 dielectric resonators 40 and 41 has a rectangular planar shape of a dielectric block 400 (410) and a dielectric block 400 (410), as clearly shown in FIG. ) And metal block 401 (411) coated on the upper surface of the dielectric block 400.
Metal layer 402 coated on the underside of (410)
(412) and. This lower metal layer 402
(412) is grounded.

Although not shown in FIG. 6, the metal layer 403 (413) on one side corresponds to a perfect electric conductor (PEC) that minimizes the electric field of the λ / 2 resonator, and the metal layer on the top surface. The layer 401 (411) is short-circuited with the lower metal layer 402 (412), and the other three side surfaces are open surfaces. Metal layer 403 of the dielectric block 400 (410)
On one side surface orthogonal to (413), an excitation electrode 404 (414) made of a rectangular metal pattern and for giving a capacitive excitation to the resonator is formed. In addition, the excitation electrode 404 (414) and the metal layer 402 (4) are formed on a part of the lower-side metal layer 402 (412) which is grounded.
12) a notch 402a (412) for separating
a) is provided.

In the present embodiment, the evanescent E-mode dielectric waveguide 42 is composed of a dielectric block having a rectangular plane shape, and only the lower plane thereof is coated with the metal layer 421 to be grounded. Has been done. The upper plane and the four side surfaces of the dielectric waveguide 42 are all open surfaces.

Evanescent E-mode dielectric waveguide 4
The two open side surfaces of 2 are two λ / 4 dielectric resonators 40.
And 41 are connected between the open side surfaces facing the short-circuit surface. In the λ / 4 dielectric resonators 40 and 41, the electric field becomes maximum on the open surface facing the short-circuited surface. Therefore, on this open side, capacitive coupling is most effective.

As described above, the metal layers 402 and 412 formed on the lower surfaces of the dielectric blocks 400 and 410 are grounded.

In the present embodiment, the excitation electrodes 404 and 414 are formed on the side surfaces of the dielectric blocks 400 and 410, which are orthogonal to the short-circuit surface and face the same direction, respectively.

The dielectric blocks 400 and 410 and the dielectric waveguide 42 are made of a dielectric material having a relatively high relative permittivity of ε _r = 93, and the metal layers 401,
411, 402, 412, 403, 413 and 421 and the excitation electrodes 404 and 414 are made of silver.

The use of TEM mode λ / 4 dielectric resonators is an important part of the present invention. This is because the volume of the filter can be essentially reduced.

Moreover, by using a dielectric material having a high dielectric constant, the thickness of the λ / 4 dielectric resonator in this embodiment should be optimized to 1 mm as described in Document (1). And succeeded in forming a filter with a thickness of 1 mm that can easily respond to the latest technological innovation.

The numerical value for evaluating the performance or quality of the resonator is the Q value. The unloaded Q value Q ₀ is defined as follows.

Q ₀ = ω ₀ × (energy stored in the resonance circuit) / (power loss in the resonance circuit) where ω ₀ is the resonance angular frequency.

The λ / 2 dielectric resonator shown in FIG. 2 has three loss coefficients. They are conductor loss due to metal coating, dielectric loss due to dielectric material, and radiation loss due to the dielectric material being open to the air.

The unloaded Q value Q ₀ of the λ / 2 dielectric resonator can be calculated using the following equation.

1 / Q₀= 1 / Q_c+ 1 / Q_d+ 1 / Q_r Where Q_cIs the Q value based on conductor loss, Q_dIs the dielectric loss
Q value based on Q_rIs a Q value based on radiation loss.

Since the Q value is inversely proportional to the loss, the larger the Q value, the smaller the power loss.

Q value Q _d based on dielectric loss × resonance frequency (GHz) = A (constant). However, A is the loss coefficient of the dielectric material and does not depend on the frequency in a certain frequency band. According to the measurement by the applicant, A = 7500 (GHz) in a dielectric material having a relative dielectric constant of ε _r = 93 in a frequency band of 2 GHz to 10 GHz.
Is.

Since the resonance frequency of the λ / 4 resonator is slightly lower than the resonance frequency of the λ / 2 dielectric resonator, the Q value Q _d based on the dielectric loss becomes slightly large.

As is apparent from FIGS. 3 and 2, λ /
Since the area of the open surface of the 4-dielectric resonator is half that of the λ / 2 dielectric resonator, the radiation loss is also halved.

In the λ / 4 dielectric resonator, the area of the portion coated with metal (excluding the plane of the complete electric conductor) is also halved, and the conductor loss is halved.

The only additional loss source in the λ / 4 dielectric resonator is the complete electrical conductor plane. However,
This plane is small and partially compensated by dielectric losses.

Since the volume of the λ / 4 dielectric resonator is half that of the λ / 2 dielectric resonator and the overall loss coefficient is almost half, the unloaded Q value is equal to the λ / 4 dielectric resonance. In both the resonator and the λ / 2 dielectric resonator.

Λ / 2 of 8.5 mm × 8.5 mm × 1 mm
The experimentally obtained unloaded Q value of the dielectric resonator is 260, and is 250 in the case of the 8.5 mm × 4.25 mm × 1 mm λ / 4 dielectric resonator. The cause of this deterioration is due to the conductor loss on the short-circuited surface.

As described above, the volume of the λ / 4 dielectric resonator is half that of the λ / 2 dielectric resonator, but the two important parameters of the resonator, the unloaded Q value, are almost the same. It can be said that they are the same. Also, the resonance frequency is almost the same.

The lowest mode of the λ / 4 dielectric resonator (T
The resonance frequency in the (EM mode) substantially depends on the length of the resonator (a <λg / 2, where λg is the guide wavelength), and hardly depends on its width a. For a resonant frequency of 1.945 GHz, the length of the λ / 4 dielectric resonator is 4.25.
mm, which is constant. Λ in this embodiment
The thickness of the / 4 dielectric resonator is optimized to 1 mm as described in the document (1).

Therefore, one remaining parameter for optimizing the dimensions of the λ / 4 dielectric resonator is the width a of this resonator.

FIG. 7 is a characteristic diagram of the unloaded Q value Q ₀ with respect to the width a of the λ / 4 dielectric resonator.

From the figure, it can be seen that the no-load Q value sharply increases to a = 3 mm and then becomes a substantially constant value. Therefore, a = 3 mm, that is, 3 mm × 4.25
mm × 1 mm has an unloaded Q value Q ₀ ≈240 λ /
The dimensions are optimized for the 4TEM mode dielectric resonator. a>
In the case of 3 mm, the internal energy of the resonator is almost proportional to the loss of this resonator, and the unloaded Q value does not increase.

The reduction of the width of the resonator allows the electrical conductor (PE
The area of C) is reduced and the added magnetic field leakage is reduced. Therefore, the series inductor decreases and the resonance frequency increases.

From the experimental results, when the width of the λ / 4 dielectric resonator was decreased from a = 8.5 mm to 3 mm (the resonator length and thickness were maintained at 4.25 mm and 1 mm, respectively), the TEM mode was used. The resonant frequency of the above has increased from 1.945 GHz to 2.133 GHz.

The external Q value Q _e , which represents the strength of the coupling of the resonator to the external circuit, is the reciprocal of this strength of the coupling. The external Q value Q _e can be controlled by changing the dimensions of the excitation electrodes 404 and 414, that is, the height and width. FIG. 8 shows the result of measuring the external Q value Q _e when the height of the excitation electrode was kept constant at 0.8 mm and the width b thereof was changed. From the figure, the width of the excitation electrodes 404 and 414 can be changed from 1 mm to
It can be seen that increasing to mm reduces the external Q factor Q _e from 35 to 22.

The strength of the coupling between the two λ / 4 dielectric resonators 40 and 41 depends on the dimensions of the evanescent E-mode waveguide 42 made of the same dielectric material as those dielectric resonators.
For example, it can be controlled by changing the thickness h.

FIG. 9 is a characteristic diagram showing the coupling coefficient k with respect to a change in the thickness h of the evanescent E-mode waveguide 42 while keeping the width of the waveguide 42 constant at 0.3 mm. From the figure, it can be seen that the coupling coefficient increases curvilinearly as the waveguide thickness h increases. For example, as the thickness h increases from 0.4 mm to 0.9 mm, the coupling coefficient k increases from 0.007 to 0.106.

To obtain a properly coupled two-stage bandpass filter, the external Q-factor Q _e should be the reciprocal of the coupling strength between the two resonators. From FIG. 8, when the width b of the excitation electrode is 3 mm, the external Q value Q _e is about 22. Therefore, the required internal coupling coefficient is about 0.045.
It can be inferred from FIG. 9 that this can be obtained by setting the thickness h of the E-mode waveguide 42 to 0.7 mm.

As a result of the above, a bandpass filter having the configuration shown in FIG. 4 can be obtained. That is, the height x width of the excitation electrodes 404 and 414 is 0.8 mm x 3 mm, and the length x width of the evanescent E-mode waveguide 42 x width x
The thickness is 0.3 mm × 3 mm × 0.7 mm.

FIG. 10 is a characteristic diagram in which the frequency characteristics of the reflection loss and the passage loss of this bandpass filter are actually measured.

From this figure, this filter is
It can be seen that it is a high-performance and low-loss bandpass filter that can be used for A. In addition, this filter has two unexpected attenuation poles on both sides of the pass band, and the presence of these attenuation poles gives a very sharp dip in the tail band of the pass band. The insertion loss of this filter is 1.3 dB and the reflection loss is 19 dB.
And the 3 dB pass bandwidth was 128 MHz and the filter frequency was 2015 MHz.

The designed filter is of the maximally flat type. The coupling coefficient k of this filter is obtained from the following equation: k = (1 / √ (g ₁ g ₂ )) (B / f ₀ ), where B is the 3 dB pass bandwidth, f ₀ is the filter frequency, and g ₁ And g ₂ are 1.414 for the maximally flat filter. The coupling coefficient k obtained from the above equation is k =
It was 0.0449. This is almost the same as the design value.

The evanescent E-mode waveguide 42, which is primarily capacitive in energy, provides a capacitive coupling and a pair of shunt capacitances to ground. FIG. 11 shows an equivalent circuit of the bandpass filter in this embodiment.

In the figure, two λ / 4 dielectric resonators 40 and 41 are connected to two LC parallel circuits 110 and 11 respectively.
Each is represented by 1. G is due to the loss factor. The electrical input / output port is represented by two capacitors C _e . L _d represents the direct coupling inductance between the electrical input / output ports. The evanescent E-mode waveguide 42 has a series coupling capacitance (internal coupling capacitance) C between the two dielectric resonators 40 and 41.
₁₂ and also a pair of shunt capacitances C ₁₁ that are grounded in the electrical schematic.

Two λ / 4 dielectric resonators 40 and 41
Should have the same dimensions to have the same resonant frequency. If there is a slight difference between the two resonant frequencies, this is a very narrow slit 4 in the upper plane of the resonator.
It can be adjusted by providing 3. Excitation is done on the lateral sides of the resonator, but the main TEM mode current flows along the length of the resonator. Therefore, the slit 43 is provided so as to prevent the current. This slit 43 adds an inductance component to the inductance component of the resonator, and reduces its resonance frequency.

The frequency adjusting slit may be provided on both the dielectric resonators 40 and 41, or may be provided on only one of them as in the present embodiment. Further, it may be located at any position including the central portion and the edge portion on the metal layer on the upper surface, and the extending direction thereof may be any direction as long as it is a direction that disturbs the current in the main TEM mode. Further, the number of slits is not limited to one and may be plural.

In the bandpass filter of this embodiment, since the excitation electrodes 404 and 414 which are the input / output ports are very close to each other, direct coupling occurs between them. In general, the nature of direct coupling (capacitive or inductive) follows that of excitation (capacitive or inductive). As described above, the measured characteristic of the bandpass filter of this embodiment has two finite attenuation poles on both sides of its passband. As described above, since the bandpass filter has two finite attenuation poles on both sides of the passband, the direct coupling and the internal coupling have different properties (that is, one is capacitive and the other is inductive). ). (For example, Yoshihiro Konishi et al., "Design of Communication Filter Circuit and Its Application", Sogo Denshi Publishing Co., Ltd., pages 31 to 41, February 1, 1994, hereinafter known. Reference (5)). In the bandpass filter of the present embodiment, the internal coupling between the two resonators is obtained through the open surface where the electric field is maximum, and the evanescent E-mode waveguide mainly retains the capacitive energy. Therefore, inductive coupling is never possible. That is, the inner join is capacitive.

In the case of capacitive internal coupling, the even mode resonance frequency f _even is equal to the odd mode resonance frequency f.
higher than _odd , and, as shown in FIG. 12, the capacitance C in parallel with the internal coupling capacitance C _{12.
When d} is connected, the odd-mode resonance frequency f _odd will decrease and the even-mode resonance frequency f _even will not change. This is because in the case of even mode resonance, the symmetry plane of the filter operates as an open circuit as shown in FIG. 13, so the even mode resonance frequency f _even is given by f _even = 1 / 2π√ (L (C + C ₁₁ )). In the case of odd mode resonance, the plane of symmetry is short-circuited as shown in FIG.
_odd is f _odd = 1 / 2π√ (L (C + C ₁₁ + 2C ₁₂ +2
This is because it is given by C _d )).

In order to prove this theoretical concept, as shown in FIG. 15, an evanescent E-mode dielectric waveguide 1 is used.
Metal layer 1 formed on the upper plane of two λ / 4 dielectric resonators 150 and 151 connected to each other via 52
The even and odd mode resonance frequencies were actually measured with the input and output of the filter loosely coupled with and without the capacitance C _d connected between 501 and 1511. FIG. 16 shows the measurement result when C _d = 0, and FIG. 17 shows C
The measurement results when _d = 1 pF are shown. Comparing FIG. 16 and FIG. 17, it can be seen that as the capacitance C _d increases, the odd-mode resonance frequency f _odd decreases but the even-mode resonance frequency f _even hardly changes.

Therefore, it was demonstrated that the internal bond has a capacitive property.

On the other hand, if the direct coupling between the input and output ports is of a capacitive nature, the frequency of the attenuation pole will approach the center frequency of the filter as this capacitance increases,
On the contrary, if the direct coupling is inductive, the frequency of the attenuation pole moves away from the center frequency of the filter when this inductance increases. Further, it is known that when direct coupling combines capacitive and inductive in parallel, the frequency of the attenuation pole moves away from or approaches the center frequency of the filter as the capacitance increases and decreases (known document (5)).

[0087] To demonstrate the direct binding properties between the input and output ports, the case where the case and connected not connected to the capacitor capacitance C _p between the input and output ports, actually measure the frequency characteristics of the filter. FIG. 18 shows C _p =
19 shows the measurement result when 0, and FIG. 19 shows the measurement result when C _p = 0.5 pF. Comparing FIG. 18 and FIG. 19, the direct coupling capacitance C between the input and output ports
It can be seen that as _p increases, the frequency of the attenuation pole moves away from the center frequency of the filter.

Therefore, it was demonstrated that the direct bond has an inductive property.

The capacitance C _p added between the input and output ports works in series with the excitation capacitance, thus reducing the equivalent external circuit capacitance. As shown in FIG. 19 as compared with FIG. 18, a coupling imbalance occurs in the filter. The added capacitance C _p and the capacitance of the external circuit are opposite in nature and act to partially cancel each other. As the effective excitation capacitance decreases, the frequency of the attenuation pole approaches the center frequency of the filter.

The very thin dielectric filter in this embodiment is replaced with a conventional dielectric waveguide type filter,
It is possible to significantly reduce the size while maintaining its performance well. This TEM mode dielectric resonator filter has the ability to be applied to mobile terminals in wideband CDMA systems, wireless LANs and various other applications where signal processing is required.

In the filter of this embodiment, both the excitation and the internal coupling between the two resonators are capacitive, which makes it possible to reduce the resonance frequency of the filter and further reduce the size of the filter itself.

FIG. 20 shows another embodiment of the present invention.
1 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators.

In the figure, reference numeral 200 is a first λ / 4 dielectric resonator, and 201 is a second λ / 4 dielectric resonator. These two λ / 4 dielectric resonators 200
And 201 are connected to each other via an evanescent E-mode dielectric waveguide (blocking waveguide) 202.

In the present embodiment, the excitation electrode 2004 of the λ / 4 dielectric resonator 200 is short-circuited with the excitation electrode 2004 of the λ / 4 dielectric resonator 201 on the side surface on the right side as viewed in the resonator direction from the short-circuit surface. It is formed on each of the left side surfaces when the resonator direction is viewed from the surface.

Other constitutions and operational effects of this embodiment are exactly the same as those of the embodiment of FIG.

FIG. 21 schematically shows the structure of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as still another embodiment of the present invention.

In the figure, 210 is a first λ / 4 dielectric resonator, and 211 is a second λ / 4 dielectric resonator. These two λ / 4 dielectric resonators 210
And 211 are connected to each other via an evanescent E-mode dielectric waveguide (blocking waveguide) 212.

In the present embodiment, the excitation electrode of the λ / 4 dielectric resonator 210 is provided on the left side surface of the excitation electrode 2 of the λ / 4 dielectric resonator 211 when viewed in the resonator direction from the short-circuited surface.
114 are formed on the right side surface of the short-circuited surface as viewed in the resonator direction.

The other constitutions and effects of this embodiment are exactly the same as those of the embodiment of FIG.

FIG. 22 schematically shows the structure of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators, as still another embodiment of the present invention.

In the figure, 220 indicates a first λ / 4 dielectric resonator and 221 indicates a second λ / 4 dielectric resonator. These two λ / 4 dielectric resonators 220
And 221 are connected to each other via an evanescent E-mode dielectric waveguide (blocking waveguide) 222.

In this embodiment, the evanescent E-mode dielectric waveguide 222 is composed of a dielectric block having a rectangular cross-sectional shape, and only two side surfaces that are not coupled to the dielectric resonator are provided. It is coated with a metal layer 2221. The metal layer 2221 is grounded via conductors 2215 and 2205 (hidden in the figure) provided on the side surface of the λ / 4 dielectric resonators 220 and 221 opposite to the short-circuit surface. λ / 4 dielectric resonator 22
0 and 221, which are hidden, also have the same structure on the opposite side from the short-circuit surface when viewing the resonator direction. The upper and lower planes of the dielectric waveguide 222 and the two side surfaces connected to each other are all open surfaces.

The λ / 4 dielectric resonators 220 and 22
The excitation electrodes 2204 and 2214 of No. 1 are formed so as to be shifted so as not to contact the metal layer 2221 of the waveguide 222.

Other constitutions and effects of this embodiment are exactly the same as those of the embodiment of FIG.

FIG. 23 shows another embodiment of the present invention.
1 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators.

In the figure, reference numeral 230 indicates a first λ / 4 dielectric resonator, and 231 indicates a second λ / 4 dielectric resonator. These two λ / 4 dielectric resonators 230
And 231 are connected to each other via an evanescent E-mode dielectric waveguide (cutoff waveguide) 232.

In the present embodiment, the excitation electrode 2304 of the λ / 4 dielectric resonator 230 is short-circuited to the excitation electrode 2304 of the λ / 4 dielectric resonator 231 on the side surface on the right side as viewed in the resonator direction from the short-circuit surface. It is formed on each of the left side surfaces when the resonator direction is viewed from the surface.

The other structure, function and effect of this embodiment are exactly the same as those of the embodiment of FIG.

FIG. 24 schematically shows the structure of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators, as still another embodiment of the present invention.

In the figure, reference numeral 240 denotes a first λ / 4 dielectric resonator, and 241 denotes a second λ / 4 dielectric resonator. These two λ / 4 dielectric resonators 240
And 241 are connected to each other via an evanescent E-mode dielectric waveguide (blocking waveguide) 242.

In the present embodiment, the excitation electrode of the λ / 4 dielectric resonator 240 is provided on the left side surface of the excitation electrode 2 of the λ / 4 dielectric resonator 241 when viewed in the resonator direction from the short circuit surface.
414 are respectively formed on the right side surfaces of the short-circuited surface as viewed in the resonator direction.

Other configurations, functions and effects of this embodiment are exactly the same as those of the embodiment of FIG.

It is obvious that the constituent materials of the dielectric block, the evanescent E-mode waveguide, and the metal layer in each of the above-described embodiments are merely examples, and the present invention is not limited thereto. Further, it is apparent that the shape of the excitation electrode is not limited to the rectangular shape, and may be any pattern shape such as a square, a trapezoid or a circle.

The embodiments described above are merely illustrative of the present invention and are not restrictive, and the present invention can be implemented in various other modified modes and modified modes. Therefore, the scope of the present invention is defined only by the claims and their equivalents.

[0115]

As described in detail above, according to the present invention, since the two-stage bandpass filter is formed by using the TEM mode λ / 4 dielectric resonator, further miniaturization and higher performance are achieved. Can provide a band pass filter.

It is configured to have attenuation poles on both sides of the pass band, that is, direct coupling and internal coupling between the first and second dielectric resonators via the evanescent E-mode waveguide coupling section. However, it is possible to further improve the characteristics of the band outside the pass band by configuring the capacitive coupling and the inductive coupling to be mutually different couplings.

[Brief description of drawings]

FIG. 1 is a characteristic diagram showing the results of theoretically and experimentally confirming the relationship between the thickness t and the unloaded Q value of a disk resonator described in a known document.

FIG. 2 is a perspective view showing a configuration of a general λ / 2 dielectric resonator.

FIG. 3 shows λ / used in the bandpass filter of the present invention.
It is a perspective view which shows the basic composition of a 4-dielectric resonator.

FIG. 4 is a perspective view schematically showing the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as an embodiment of the present invention.

5 is an exploded perspective view of the bandpass filter in the embodiment of FIG.

FIG. 6 is a perspective view schematically showing the configuration of a λ / 4 dielectric resonator in the embodiment of FIG.

FIG. 7 is a characteristic diagram of the unloaded Q value Q ₀ with respect to the width a of the resonator.

FIG. 8 is a characteristic diagram showing changes in the external Q value Q _e with respect to the width b of the excitation electrode.

FIG. 9 is a characteristic diagram showing changes in the coupling coefficient k with respect to the thickness h of an evanescent mode waveguide.

FIG. 10 is a characteristic diagram in which frequency characteristics of reflection loss and passage loss of the bandpass filter according to the present embodiment are actually measured.

FIG. 11 is a circuit diagram showing an equivalent circuit of the bandpass filter in the present embodiment.

FIG. 12 is a circuit diagram showing an equivalent circuit for explaining internal coupling of a bandpass filter when a capacitance C _d is connected in parallel.

FIG. 13 is a circuit diagram showing an equivalent circuit in the case of even mode resonance.

FIG. 14 is a circuit diagram showing an equivalent circuit in the case of odd mode resonance.

FIG. 15 is a diagram for explaining a configuration for demonstrating capacitive inner coupling.

FIG. 16 is a characteristic diagram showing measurement results for demonstrating capacitive internal coupling.

FIG. 17 is a characteristic diagram showing a measurement result for demonstrating capacitive internal coupling.

FIG. 18 is a characteristic diagram showing measurement results for demonstrating inductive direct coupling.

FIG. 19 is a characteristic diagram showing measurement results for demonstrating inductive direct coupling.

FIG. 20 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as another embodiment of the present invention.

FIG. 21 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as still another embodiment of the present invention.

FIG. 22 shows still another embodiment of the present invention, 2
1 schematically shows the structure of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators.

FIG. 23 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as another embodiment of the present invention.

FIG. 24 schematically shows the configuration of a high-frequency dielectric resonator type bandpass filter having two dielectric resonators as still another embodiment of the present invention.

[Explanation of symbols]

20, 30, 400, 410 Dielectric block 21, 22, 31, 32, 34, 401, 402, 40
3, 411, 412, 413, 421, 2221 Metal layer 23, 33 Electric field 24 Symmetry plane 35 Current 40, 41, 150, 151, 200, 201, 21
0, 211, 220, 221, 230, 231, 24
0, 241 λ / 4 dielectric resonator 42, 152, 202, 212, 222, 232, 24
2 Evanescent E-mode dielectric waveguide 43 Slits 402a, 412a Notches 404, 414, 2004, 2104, 2114, 22
14, 2304, 2414 Excitation electrodes 2205, 2215 Conductor

Continued front page       (56) References JP-A-53-99848 (JP, A)                 JP-A-63-232602 (JP, A)                 JP-A-7-288404 (JP, A)                 JP-A-9-246808 (JP, A)                 JP-A-7-297604 (JP, A)                 JP-A-6-204706 (JP, A)                 Japanese Patent Laid-Open No. 5-211403 (JP, A)                 Actual Development Sho 56-165401 (JP, U)

Claims (19)

(57) [Claims] 1. A first and a second dielectric resonator each including a dielectric block having an upper plane, a lower plane and four sides, and an evanescent E-mode waveguide coupling. Each of the first and second dielectric resonators includes a first and a second metal layer coated on the upper plane and the lower plane, respectively, and one of the four side surfaces. A side surface coated with a third metal layer, the side surface coated with the third metal layer is a short-circuit surface, and the remaining three side surfaces of the four side surfaces are open surfaces.
/ 4 dielectric resonator and electromagnetic field independent T
The evanescent E-mode waveguide coupling portion is configured to maintain an EM mode, and connects the open surfaces of the first and second dielectric resonators facing the short-circuit surface to each other. For providing TEM mode coupling of the first and second dielectric resonators, and
A bandpass filter using a TEM mode dielectric resonator, wherein the bandpass filter is configured to operate in an evanescent E mode having a cutoff frequency higher than each resonance frequency of the dielectric resonators. 2. The evanescent E-mode waveguide coupling portion has an open top surface and four side surfaces, and a metal-coated bottom surface. 1. The bandpass filter according to 1. 3. The direct coupling and the internal coupling between the first and second dielectric resonators via the evanescent E-mode waveguide coupling section are different from each other among capacitive coupling and inductive coupling. The bandpass filter according to claim 1 or 2, wherein the bandpass filter is configured as follows. 4. The bandpass filter according to claim 1, wherein the first and second dielectric resonators are made of the same dielectric material. 5. The band according to claim 1, wherein the first and second dielectric resonators are made of a ceramic dielectric material having a high dielectric constant. Pass filter. 6. The bandpass filter according to claim 1, wherein the first and second dielectric resonators have substantially the same dimensions. 7. The evanescent E-mode waveguide coupling portion has a length and a cross section shorter than the length and cross section of each of the first and second dielectric resonators. The bandpass filter according to any one of claims 1 to 6, which is characterized. 8. The dimension of the evanescent E-mode waveguide coupling portion is selected so that the coupling between the first and second dielectric resonators has a desired value. The bandpass filter according to claim 7. 9. The bandpass filter according to claim 1, wherein the evanescent E-mode waveguide coupling portion has a rectangular cross section. 10. The evanescent E-mode waveguide coupling portion uses the same dielectric material as that of the first and second dielectric resonators. The bandpass filter according to item 1. 11. The evanescent E-mode waveguide coupling is configured to provide a series capacitance and a pair of shunt capacitances between the first and second dielectric resonators. Claim 1 characterized by
11. The bandpass filter according to any one of items 1 to 10. 12. The method of claim 1, wherein the second metal layer on the lower plane of each of the first and second dielectric resonators is used as a ground plane. A bandpass filter according to item. 13. The coated surface of the metal layer of the evanescent E-mode waveguide coupling portion is used as a ground plane.
The bandpass filter according to any one of 2 above. 14. The one side surface orthogonal to the short-circuit surface of each of the first and second dielectric resonators is used for capacitive excitation.
The bandpass filter according to any one of 1. 15. The electrical input / output port is provided on one side surface orthogonal to the short-circuit surface of each of the first and second dielectric resonators.
4. The bandpass filter according to any one of 4 above. 16. The bandpass filter according to claim 15, wherein the electrical input / output port is formed by providing a rectangular, square, trapezoidal or circular metal pattern on the side surface. 17. The bandpass filter according to claim 16, wherein the metal pattern is separated from the first metal layer on the upper plane and the second metal layer on the lower plane. . 18. The bandpass filter according to claim 16, wherein the size of the metal pattern is selected so that the strength of coupling with an external circuit has a desired value. 19. The narrow slit for frequency adjustment is provided in the first metal layer on the upper plane of at least one of the first and second dielectric resonators. 19. The bandpass filter according to any one of items 1 to 18.