JP2903609B2 - Power supply for arc processing - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an improvement in a power supply for arc processing used for arc welding, cutting, plasma arc processing, and the like, and in particular, converts a DC power supply into a high-frequency AC by a switching element, and converts the power. The present invention relates to an improvement of a power supply of a system in which a converted AC is converted into a voltage suitable for processing by a transformer and then rectified again by a rectifier to obtain a DC.

<Prior Art> A circuit as shown in FIG. 6 is conventionally used as a power source for arc processing in which a DC power source is converted into AC by a switching element, passed through a transformer, rectified and converted back to DC. In FIG. 1, reference numeral 1 denotes a DC power source, which is usually obtained by obtaining power from a commercial AC power source, rectifying the power, and smoothing the rectified power. Reference numerals 2 and 3 denote capacitors connected in series, and reference numerals 4 and 5 denote switching elements connected in series. Usually, switching transistors are used. The switching elements 4 and 5 are bridge-connected to the capacitors 2 and 3 and are supplied with power from the DC power supply 1, and their respective intermediate points are connected to the primary winding of the transformer 6 as shown to form an inverter circuit. ing. The secondary winding of the transformer 6 has a center tap, is rectified by a double-wave rectifier composed of diodes 7 and 8, is again converted to DC, and is connected to output terminals (a) and (b) via a DC reactor 9. Electrode 10 and workpiece
11 is supplied to the arc processing load. Reference numeral 12 denotes an inverter control circuit for alternately turning on and off the switching elements 4 and 5 at a predetermined time rate. 13a to 13d
Is a snubber circuit connected in parallel to each of the switching elements 4 and 5 and diodes 7 and 8 in order to absorb a surge voltage applied thereto, and is usually configured by a series circuit of a capacitor and a resistor.

<Problems to be Solved by the Invention> In the conventional device described above, a large surge voltage is generated when the switching elements 4 and 5 are cut off due to the stray inductance of the circuit. Since this surge voltage is proportional to the square of the current flowing through the switching element whose energy is to be cut off, a large-capacity power supply requires the use of a large-capacity snubber circuit to suppress this. Becomes When the power capacity of the snubber circuit increases, a large-sized cooling mechanism is required to cool the snubber circuit. For this reason, the whole apparatus becomes large and once converted into high-frequency AC, not only does the effect of downsizing the transformer and the smoothing circuit be offset, but also the cost of the apparatus increases.

By the way, among these arc machining power supplies, there are some which set the output voltage relatively high depending on the machining object. In this case, the current supplied to the primary side of the transformer has a much larger value than the current used for processing. For example, in a plasma arc processing power supply or arc cutting power supply with an output voltage of 100 V to 200 V or more, the current flowing on the primary side of the transformer reaches several times the processing current to obtain the required output current. There is.

<Means for Solving the Problems> The present invention is directed to an arc processing power supply in which the primary current of the transformer is larger than the output current. Rectified DC output by a chopper circuit using a switching element, and synchronizing the OFF period of the switching element for the chopper with the ON / OFF timing of the switching element for supplying AC to the transformer primary side. By doing so, the switching element on the primary side is turned on and off with no load.

<Embodiment> FIG. 1 shows a connection diagram of an embodiment of the present invention. 6, reference numerals 1 to 11 denote components having the same functions as those of the conventional device shown in FIG. 6, and reference numeral 14 denotes a chopper connected in series between the common connection point of the diodes 7, 8 and the DC reactor 9. The switching element 15 is a flywheel diode, and constitutes a smoothing circuit for smoothing the output of the switching element 14 together with the DC reactor 9. Snubber circuits 13a to 13f for absorbing surge voltage are connected to each switching element and diode. 16 is the output current detector, 17 is the output current setter, 18 is the output of the output current setter 17
An error amplifier that compares Vr with the output current detector Vf to obtain a difference signal ΔV = Vr−Vf, and 19 is a pulse width for generating a two-phase pulse train having a duty corresponding to the output of the error amplifier 18 to generate s1 and s2 Modulation circuits (hereinafter referred to as PWM circuits) 20a and 20b receive the signals s1 and s2 of the PWM circuit 19 as input and rise at the rise of each input signal, and fall at a certain time delay after the fall of each input signal. The signal delay circuit 21 generates the signals s3 and s4 for conducting the signal, the signal 21 receives the signals s1 and s2 of the PWM circuit 19 as input, and is turned off for a certain time from the fall of both input signals,
During the other periods, the signal s5 that is turned on is output and the switching element 14
Is a chopper control circuit for conducting the current. Of these, as the signal delay circuits 20a and 20b, input signals s1 and s2 are instantaneously operated, a time limit is started by disappearance of the input signal, and a time delay circuit of an instantaneous operation time reset type of a method of returning after a fixed time is input to each input. It is sufficient to provide one set for each signal, and the chopper control circuit 21 becomes a low level from the falling time of either the input signal s1 or s2 for a fixed time, that is, the delay recovery time td1 of the signal delay circuits 20a and 20b. Long td2 ((td1 <td2
A timed circuit that generates an OFF signal (interruption command signal) that continues only during> and outputs an ON signal (conduction command signal) at other times (for example, an OR gate that receives the signals s1 and s2 and an output signal of the OR gate) A circuit configured using a mono-multivibrator that outputs a low-level signal for a certain period of time by falling)
May be used. FIG. 2 shows waveforms at various parts for explaining the operation of the embodiment of FIG. 1 in such a case. In FIG. 2, (a) shows the output signal s1 of the PWM circuit 19,
(B) shows the output signal s2, respectively, and (c) shows the signal s3 obtained by converting the signal s1 by the signal delay circuit 20a.
(D) similarly shows a signal s4 obtained by converting the signal s2 by the signal delay circuit 20b. (E) is the output signal s5 of the chopper control circuit 21, (f) is the waveform of the current Icl flowing through the switching element 4, (g) is the terminal voltage V1 across the switching element 4, (h) and (i) Waveform of current Ic2 flowing through switching element 5 and terminal voltage V2
(J) and (k) show the waveform of the current Ic3 flowing through the switching element 14 and the waveform of the terminal voltage V3, respectively.

The operation of the embodiment of FIG. 1 will be described with reference to the waveform diagram of FIG. PWM with a time width T ₀ determined by the difference ΔV between the output Vr of the output current setting unit 17 and the output Vf of the output current detector, which is determined by ΔV = Vr−Vf
The circuit 19 outputs signals s1 and s2. At the time t = t1 at the end of the signal s1, the output signal s5 of the chopper control circuit 21 falls, and the switching element 14 is cut off.
Ic3 decreases rapidly and goes to zero. The signal delay circuit 20a starts the time period from this time t = t1, and the time t after the set time period td1.
= The signal s3 is turned off at t2. As a result, the switching element 4 is cut off, but by this time the current Ic1
Is also zero according to the current Ic3, so that the switching element 4 is cut off almost without load.
Next, from time t = t1, the set time td2 of the chopper control circuit 21 is set.
At time t = t3, which corresponds to the time after, the signal s5 turns on the switching element 14 again. Time t = t4 after this
Then, when the signal s2 of the PWM circuit 19 is turned on, the signal delay circuit 20b generates the signal s4, the switching element 5 conducts, and the current Ic2 starts flowing. Thereafter, the operation according to the operation after time t = t1 is performed. After that, the same operation is repeated.

In the embodiment shown in FIG. 1, since the switching elements 4 and 5 are always cut off under no load, the surge voltage generated by cutting off the primary current of the transformer is extremely small.

If the chopper control circuit 21 of FIG. 1 is used to perform a delay operation in which the output signal rises with a certain time delay from the rise of the input signal and falls at the same time as the fall of the input signal, the switching elements 4 and 5 can be turned on at the start of conduction. Can also be in a no-load condition. FIG. 3 is a diagram for explaining the operation when a chopper control circuit that operates as described above is used. In FIG. 3, (a) shows the output s1 of the PWM circuit 19, and (b) shows the output of the PWM circuit 19. Outputs s2 and (c) are signals s3 and (d), respectively, a signal s4 delayed by td1 in the signal delay circuit 20a, and (e) is a chopper control circuit.
21 shows how the signal s5 'whose rise is delayed by td3 is changed.

In FIG. 3, when the signal s1 output until then falls at time t = t1, the signal s5 'falls immediately, and the switching element 14 is cut off. However, the signal s3 falls with a delay of time td1 by the signal delay circuit 20a, and the switching element 4 shuts off at time t2 in a no-load state. Next, at time t =
When the output s2 of the PWM circuit 19 rises at t3, the signal s4 rises immediately and the switching element 5 is turned on. The set time of the chopper control circuit 21 from the time t = t3
At time t = t4 delayed by td3, the signal s5 'rises and the switching element 14 becomes conductive. Next, at time t = t5, the signal s2
As soon as the signal falls, the signal s5 'falls and the switching element
14 is cut off, and the signal s is delayed from this time by the time td1.
4 falls, and the switching element 5 is cut off. Thereafter, the same operation is repeated. Where PWM circuit 1
9, the minimum value of the gap tg between the output signals s1 and s2 is set so as not to be shorter than the delay time td1, and the time td1 is set longer than the cutoff delay time of the switching element 14, and the time td3
May be longer than the time required for the switching elements 4 and 5 to conduct. Since each of these delay times td1 and td3 may be about several μs, a sufficiently stable operation can be obtained even if the operating frequency of each switching element, that is, the operating frequency of the PWM circuit is 50 kHz or more. When using the above-described delayed operation as the chopper control circuit 21, the switching elements 4 and 5 are operated in a no-load state both at the time of starting conduction and at the time of shutting off. No surge voltage occurs. Therefore, it is not necessary to prepare a snubber circuit to be connected in parallel with them, or it is possible to use a snubber circuit of very small capacity. Since no current flows through these switching elements 4 and 5 during switching, no switching loss occurs. Therefore, it is only necessary to have a current capacity in a conductive state, so that an inexpensive and small one can be used.

Instead of the embodiment shown in FIG. 1, the switching elements 4 and 5 are controlled to open and close by the output of the pulse width modulation circuit having a fixed ON duty, and the ON duty is determined by the difference between the output detection value and the output set value. The present invention can also be implemented by controlling the switching element 14 to open and close based on the output of the ordinary pulse width modulation circuit determined, and synchronizing both the pulse width modulation circuits with the output of the common oscillator.

FIG. 4 is a connection diagram showing another embodiment of the present invention as described above. Signals s3, s4, and s5 for driving the switching elements 4, 5, and 14 of the embodiment shown in FIG. Only the circuit portions that occur are shown, and the others are omitted or abbreviated. In the figure, 31 is a voltage of a triangular wave having a constant frequency.
An oscillator that generates s0, 32 is a DC power supply that outputs a constant voltage E0 slightly lower than the peak voltage sp of the signal s0 for keeping the pulse width of the signals s3 and s4 constant, and 33 and 34 are comparators and oscillators. The output s0 of the DC power supply 32 is compared with the output E0 of the DC power supply 32.
A high level signal is output during <E0 and s0 <ΔV. 35
Is a two-phase separator using a flip-flop circuit,
This is a circuit that alternately distributes signals s3 and s4 each time the output of the comparator 33 becomes high level. 4, 5, 14 are switching elements, 16 is an output current detector, 17 is a reference signal setter, 18 is an error amplifier, all of which have the same functions as those of the embodiment shown in FIG. Things. FIG. 5 is a diagram for explaining the operation of the embodiment of FIG. 4,
(A) shows the relationship between the output s0 of the oscillator 31, the output E0 of the DC power supply 32, and the output ΔV of the error amplifier 18, and (b) and (c) show the signals s3, s4,
(D) shows the state of the signal s5 that goes high during s0 <ΔV. As shown in the figure, the signals s0 and E0
Is constant, the pulse widths of the signals s3 and s4 are also constant, while the pulse width of the signal s5 is determined in proportion to the error signal ΔV. Since also the signal s3, s4, s5 its phase by the output signal of the oscillator 31 is determined, than the maximum value of ΔV by setting the output E0 of the DC power source 32 to slightly higher, i.e. s _p> E0 > Delta] Vmax (however, s _p is the peak value of the output signal of the oscillator 31) if in the signal rising signal s5 is always the signal s3 or s4 after I rise, signal s5 (or s6) is after me falling s3 Alternatively, s4 falls, and the switching elements 4 and 5 driven by these are always controlled to open and close under no load.

The switching elements 4, 5 as in the embodiment of FIG.
In order to make a no-load state only when the power supply is cut off, the output waveform of the oscillator 31 in FIG.

In each of the above embodiments, the DC power supply is connected to the capacitor 2,
Although a method of converting into high-frequency AC by a half-bridge type inverter circuit composed of 3 and switching elements 4 and 5 has been described, the circuit for converting DC into AC is not limited to these embodiments. Instead, a conversion circuit of another system, for example, a full-bridge inverter circuit, a DC / DC conversion circuit that obtains a direct current of a different voltage through a high frequency stage by a forward converter, or an inverter circuit using a single switching element can be used. Further, the output side of the transformer may be of a double-wave rectification type in addition to the method of converting the rectified output into a direct current by using the center tap type half-wave rectification circuit shown in the embodiment. One may be provided every time to reduce the current burden.

Further, the DC output thus obtained may be switched between positive and negative using a switching element and supplied to the arc load as an AC output.

As described above, in a device in which the output voltage is set to be relatively high and the current flowing on the input side and the output side of the transformer is larger on the input side, the switching element having the larger current is used in the present invention. Since the shut-off or turning-on and shut-off are performed under no load, generation of a surge voltage due to switching can be suppressed to a minimum value. For this reason, the switching element that opens and closes the input side of the transformer does not require a snubber circuit for absorbing surge voltage, or it can be of a very small capacity for safety. An inexpensive device can be obtained.

[Brief description of the drawings]

FIG. 1 is a connection diagram showing an embodiment of an arc machining power supply according to the present invention, FIG. 2 is a diagram for explaining the operation of the embodiment of FIG. 1, and FIG. 3 is a diagram of the embodiment of FIG. FIG. 4 is a diagram for explaining an operation when the chopper control circuit is partially changed; FIG. 4 is a connection diagram showing only a circuit portion for generating a switching element driving signal according to still another embodiment of the present invention; FIG. 5 is a diagram for explaining the operation of the embodiment of FIG. 4, and FIG. 6 is a connection diagram showing an embodiment of a conventional arc machining power supply. 1: DC power supply, 2,3: Capacitor, 4,5,14: Switching element, 6: Transformer, 7,8,15: Diode, 13a to 13f: Snubber circuit, 16: Output current detector, 17: Output Current setter, 18… Error amplifier, 19… PWM circuit, 20a, 20b… Signal delay circuit, 21… Chopper control circuit, 31… Oscillator, 32… DC power supply, 33,34… Comparator, 35… 2 phase separator

Claims (1)

(57) [Claims] An output of a DC power supply is converted into a high-frequency AC by a DC / AC conversion circuit comprising a first switching element, and then converted into a voltage suitable for processing by a transformer, and an output of the transformer is again rectified by a rectifier circuit. In a power source for arc machining of a system of rectifying to DC, a chopper switching element for opening and closing the output of the rectifier circuit and a smoothing circuit for smoothing the output of the chopper switching element are provided on the output side of the rectifier circuit. An arc machining power supply, further comprising a switching element control circuit for controlling switching of the switching element of the DC / AC conversion circuit in synchronization with an OFF period of the chopper switching element.