JP2971033B2 - Apparatus and method for restoring digital carrier in television signal receiver - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a television signal receiving system, and more particularly to a high definition television (Hig).
The present invention relates to a carrier recovery apparatus that compensates for the offset of the frequency and phase of a carrier signal using a pilot signal so that only a frequency band of the carrier signal can be accurately passed through a receiver of a h Definition Television.

[0002]

2. Description of the Related Art Communication systems for transmitting a pilot signal to a carrier include VSB (Vestigial Sideband), DSB (Double Sideband), and SSB (Single Sideband). At this time, the pilot signal is transmitted to the carrier at the time of transmission in order to accurately recover the carrier. FIG. 1 shows an example of a communication system for transmitting a pilot signal to a carrier wave.

FIG. 1 shows a US 8-level VSB HDT.
It is a figure which is a part of V receiver and shows a tuning part, an IF part, a carrier recovery device, etc. The figure shows the united St
ates Advanced Television
April 1, 1995 at the System Committee
Title “Guide to the Us”
e of the Digital Televisi
on For HDTV Transmission ”
Page 100.

Referring to FIG. 1, the operation of the receiver in the receiving process will be described as follows. The RF signal received through the antenna is applied to the double conversion tuner 2. The dual conversion tuner 2 tunes the received RF signal with a fixed local oscillation signal LO1 generated by the frequency synthesizer 8 and converts it into a 920 MHz band primary IF signal. At this time, the double conversion tuner 2 also performs the gain control of the output signal in response to the automatic gain control AGC of the receiver. The IF signal output from the double conversion tuner 2 is filtered through a SAW (surface acoustic wave) filter 4 into an IF signal having a predetermined bandwidth. This SAW filter 4
Referring to the frequency spectrum shown in FIG. 3, the filter has a characteristic of filtering an IF signal into, for example, a 6 MHz bandwidth (41 to 47 MHz). At this time, the pilot signal is
As an example, it is located at 3 dB point of the upper cutoff frequency of the IF signal belonging to the frequency band of the SAW filter 4.

[0005] 6 MHz passed through the SAW filter 4
The IF signal of the bandwidth is amplified by the IF amplifier 6 and then applied to the carrier recovery device 10. The amplification gain at this time is determined by the automatic gain control AGC provided from the receiver. The carrier recovery device 10 at the receiver is usually referred to as a frequency and phase locked loop device (FPLL).
The carrier recovery device 10 extracts a frequency error and a phase error from the demodulated baseband signal, and applies the local oscillation signal LO2 to the double conversion tuner 2. Then, the double conversion tuner 2 performs more precise tuning by precisely correcting the tuning by the local oscillation signal LO1 by the local oscillation signal LO2.

The specific details of the carrier recovery device 10 are as follows.
IEEE Trans. on C
on Electronics Electronics Vol C
E-23 NO3. On pages 358-365. Hereinafter, the operation of the carrier recovery apparatus 10 will be described in more detail. The local oscillator 12 in the carrier recovery device 10 generates a local oscillation signal LO3 of a predetermined MHz for converting the IF signal output from the IF amplifier 6 to a base band. The local oscillation signal LO3 is phase-shifted by 90 ° by the phase shifter 14, applied to the first mixer 16, and directly applied to the second mixer 18.

Therefore, after the amplified IF signal output from the IF amplifier 6 is multiplied by the local oscillation signal LO3 whose phase has been shifted by 90 ° by the first mixer 16 (after tuning), the first low frequency band The frequency is converted to a signal of a base band through a pass filter LPF1 20. Further, the amplified IF signal is multiplied by the local oscillation signal LO3 by the second mixer 18 (after tuning), and then the second low-pass filter LPF2 22
Is converted to a signal of a base band through the frequency conversion. here,
The output signal of the first mixer 16 is an I signal, and the output signal of the second mixer 18 is a Q signal. The first low-pass filter LPF 120 removes the second harmonic component (video frequency component) of the I signal, passes only the baseband signal of the I signal, and the second low-pass filter LPF 222 also has the same function. Similarly, it removes the second harmonic component (video frequency component) of the Q signal and performs the function of passing only the baseband signal of the Q signal.

At this time, the first low-pass filter LPF1
The pilot signal is located at 0 (zero) Hz when the tuning is accurately performed by the double conversion tuner 2, and when the tuning is not accurately performed, the pilot signal Is present in the positive base band or in the negative base band. Where SA
If the W filter 4 does not filter the pilot signal, the carrier recovery device 10 cannot perform carrier recovery. Therefore, when the pilot signal is in the positive base band or in the negative base band, a frequency offset (that is, a difference between the demodulated frequencies of the carrier frequencies) occurs. The frequency offset is generated because there is a difference between the frequency to be tuned by the double conversion tuner 2 and the frequency actually tuned.

When the frequency offset exists, the output signal (I signal) of the first low-pass filter LPF 120 is a cosine wave, and the second low-pass filter LPF 2 2
The output signal (Q signal) 2 is a sine wave. At this time, the output signal (I signal) of the first low-pass filter LPF 120 is automatically controlled by an automatic frequency control low-pass filter (hereinafter, “A”).
FC LPF "24) and a sine wave through a limiter 26. The AFC LPF 24 can follow the frequency detected by the pilot signal in the output signal (I signal) of the first low-pass filter LPF1 20. The fact that the +1 or -1 signal output from the limiter 26 changes with time is as follows.
It means that there is a frequency error.

The output of the limiter 26 is supplied to a mixer 30
Is multiplied by the output signal (Q signal; sine wave) of the second low-pass filter LPF 222 and converted into a DC signal. This DC signal is supplied to an automatic phase control low-pass filter (hereinafter referred to as a
A VCO (Voltage Controlled Oscillator) 34 is controlled in a direction to remove a frequency error through an “APC LPF” 32. The VCO 34 outputs the local oscillation signal LO2 to the double conversion tuner 2 in response to the frequency error removal control of the APC LPF 32.

If all the frequency errors are removed by this control, the output of the limiter 26 is fixed to 1 or -1 without conversion according to time. This means that there is no frequency error. At this time, since only the output of the second low-pass filter LPF2 22 operates in the mixer 30, the carrier recovery device 10 functions as one PLL. At this time, the output of the mixer 30 is
V passes through the LPF 32 to remove the residual phase error.
Controls CO34.

In summary, the operation of the carrier recovery apparatus 10 extracts a carrier signal, a demodulated frequency signal, and a frequency offset, operates first as a loop for correcting a frequency error, and then automatically tracks a phase error. Act as a loop. In this carrier recovery device 10, the frequency pull-in region is a relatively narrow range of about ± 100 kHz.
Then, the pilot signal transmitted for carrier recovery must be extracted from the 6 MHz frequency bandwidth according to the filtering characteristics of the SAW filter 4. So, AFC
From the viewpoint of the pilot signal, the signal input to the LPF 24 has a considerable noise component within the 6 MHz frequency bandwidth excluding the pilot signal.

Further, the SAW filter 4 has a precision of 6 MHz.
Since only the signal of the bandwidth is passed, if the offset of the carrier wave is considerably large, the pilot signal becomes the SAW.
It will be outside the bandwidth of the filter 4. In this case, the carrier recovery apparatus 10 cannot search for the pilot signal at all, thereby completely disabling the carrier recovery.

[0014]

SUMMARY OF THE INVENTION Accordingly, it is an object of the present invention to provide an improved carrier recovery apparatus and method which enables carrier recovery to be performed smoothly even when the frequency offset is considerably large. Another object of the present invention is to provide a digital carrier recovery apparatus and method for digitally processing carrier recovery.

It is still another object of the present invention to provide a digital carrier recovery apparatus and method for tuning so as to be pre-shifted in a direction for correcting a frequency offset.
It is still another object of the present invention to provide an apparatus and method for restoring a digital carrier wave in which a frequency pull-in region for performing carrier restoration can be arbitrarily varied.

[0016]

SUMMARY OF THE INVENTION In order to achieve such an object, the present invention provides a pilot signal having a bandwidth corresponding to a base band of a SAW filter bandwidth and a low band in preparation for a case where carrier tuning is shifted. The frequency conversion is performed by shifting the frequency in the opposite direction by a predetermined frequency in advance so that it can sufficiently exist within the bandwidth of the pass filter.

[0017]

Preferred embodiments of the present invention will be described below in detail with reference to the accompanying drawings. In the drawings, the same constituent elements and the same portions are denoted by the same reference numerals and symbols as much as possible. First, in order to facilitate a clearer understanding of the present invention, a description will be given with reference to the frequency spectra of FIGS. 3, 4, and 5.

In the present invention, the 3 dB frequency band of the IF signal passing through the dual conversion tuner 2 is set to 41.31 as shown in FIG.
~ 46.69 MHz and the pilot signal is 46.69 MH
Assume that it is at z. This is the United States 8-level VSB
HDTV standard. Therefore, this IF signal is SA
After passing through W filter 4, 41.31 to 46.69 in the IF signal
Frequency components other than MHz are completely removed. But,
If the IF signal tuned and frequency-converted by the double conversion tuner 2 at the beginning is shifted by 0.3 MHz (the center frequency of the IF signal is changed from 44 MHz to 44.3 MHz) as shown in FIG. Cannot exist within 3 dB of the filtering bandwidth of Referring to FIG. 4, the pilot signal at this time becomes 46.99 MHz and cannot exist within 3 dB of the filtering bandwidth of the SAW filter 4.

In the present invention, when the IF signal output from the double conversion tuner 2 is shifted by a frequency of 0.3 MHz,
A double conversion tuner 2 and an NCO (Numerically Controlled Oscillator) so that it can look like the IF signal shown in FIG.
Adjust 58. An embodiment for adjusting the dual conversion tuner 2 will be described with reference to FIGS. 2 and 8, and an embodiment for adjusting the NCO 58 will be described with reference to FIGS. 9 and 10. At this time, like the IF signal shown in FIG.
In order to correct the IF signal which is the output of the double conversion tuner 2, the double conversion tuner 2 is tuned so that the frequency is shifted in the opposite direction only by the shift.

Tuning so as to shift in the opposite direction, which is only a shift, is achieved when the system is turned on (V).
This is performed by adjusting the output frequency of CO78. The adjustment is performed by adjusting the local oscillation signal of the NCO 58. If the double conversion tuner 2 is tuned to be shifted in the opposite direction by the local oscillation signal LO2, I
The pilot signal of the F signal is 46.39 M as shown in FIG.
Hz (in a normal case, 46.6
9 MHz). It can be seen that the 46.39 MHz pilot signal sufficiently exists in the filtering bandwidth of the SAW filter 4.

In order to generate the IF signal shown in FIG. 2 output from the double conversion tuner 2 as shown in FIG. 4, an embodiment of the configuration of the present invention is as shown in FIG. FIG. 2 shows a tuning section, an IF section, and a carrier recovery apparatus 5 of a receiver according to the present invention.
0, that is, a diagram showing the FPLL unit, wherein the configurations of the double conversion tuner 2, the SAW filter 4, and the IF amplifier 6 are the same as the components shown in FIG.

The IF signal amplified from the IF amplifier 6 is
The mixer 44 multiplies (tunes) the local oscillation signal LO3 generated from the third local oscillator 42 and frequency-converts the signal into a signal close to the base band. In the present invention, an example will be described in which when a 46.69 MHz pilot signal as shown in FIG. 3 is output from the mixer 44, the frequency is converted to 2.69 MHz as shown in FIG. That is, the frequency of the local oscillation signal LO3 of the third local oscillator 42 at this time is 46.69 MHz + 2.69 MHz. This mixer 4
The second harmonic component (video frequency component) is removed from the output of the first through the first low-pass filter LPF1 46, and the output signal of the first low-pass filter LPF1 46 is A /
The signal is converted into a digital signal through the D converter 48 and applied to the carrier recovery device 50. The third local oscillator 42 and the mixer 44 convert the frequency bandwidth of the signal to a lower frequency band so that the signal can be digitally processed by the A / D converter 48.

The carrier recovery device 50 according to the present invention has the configuration shown in FIG.
, The first phase separator 54, the low-pass filter LPF and the second phase separator 56, the NCO 58, the mixer 6
0 and 62, frequency error quick response detection unit 64, frequency error precision detection unit 66, lock detector 70, multiplexer 7
2, a mixer 74, an APC LPF 76 and a VCO 78.

The digital signal converted by the A / D converter 48 passes through a first phase
₁ and Q ₁ , and the complex signals I ₁ and Q ₁ are
The signal is multiplied by the complex signals I ₀ and Q ₀ of the oscillation frequency of the NCO 58 fixed by the mixer 60 and frequency-converted to a base band. At this time, the first phase separator 54 performs the Hilbert transform to perform the complex signal I _1.
And to output the Q _1.

The digital signal converted by the A / D converter 48 is separated into complex signals I ₂ and Q ₂ through a low-pass filter LPF and a second phase separator 56, and the complex signal I ₂ And Q ₂ are complex signals I ₀ and Q ₀ of the oscillation frequency of the NCO 58 fixed by the mixer 60.
And frequency-converted to a baseband. This NCO
The fixed oscillation frequency 58 is a frequency value for converting the pilot frequency of the input signal of the A / D converter 48 into a base band, and is 2.69 MHz in this embodiment.
In the complex signal component having an oscillation frequency of 2.69 MHz provided from the NCO 58, I ₀ is a real number signal and Q ₀ is an imaginary number signal.

The low-pass filter LPF and the second phase separator in the second phase separator 56 perform a Hilbert transform, and the low-pass filter LPF passes a predetermined low-frequency signal. At this time, it is desirable that the tap length of the delay element of the low-pass filter be set to a time that is delayed by the first phase separator 54 in the upper signal path. This is so that the two signals multiplied by the mixers 60 and 62 are on the same point in time. FIG. 6 shows an example of the low-pass filter LPF according to the present invention.
As shown in FIG. This filtering bandwidth can be determined to vary according to the required frequency pull-in region.

The complex signals I ₂ and Q ₂ output from the low-pass filter LPF and the second phase separator 56 are multiplied by the complex signals I ₀ and Q _{0 of} the NCO 58 fixed by the mixer 62 to obtain a base band. Is frequency-converted. At this time, the signal is converted by the mixer 62 into a baseband frequency generated in a positive frequency sideband. If the carrier tuning is ideal as shown in FIG. 3, the frequency band (frequency pull-in region) of the signal output from the mixer 62 is 2.69 to 3.29 MHz as shown in FIG. The pilot signal at this time is 2.69 MHz, which is located at the edge of this frequency pull-in area.

The complex signal I output from the mixer 62
_{During 3} and Q ₃ , the in-phase component signal I ₃ is simultaneously applied to the frequency error quick response detection unit 64 and the frequency error precision detection unit 66. The frequency error quick response detection unit 64 includes a differentiator 80
And a first limiter 82. This differentiator 8
0, the in-phase component signal I ₃ output from the mixer 62
Is differentiated, and the first limiter 82 limits the signal differentiated from the differentiator 80 and outputs the signal as +1 or −1. The differentiator 80 exhibits excellent performance when the frequency offset is large, and in this case, ensures fast convergence of the carrier recovery device 50. Therefore, when the frequency error is large (the in-phase component signal I ₃ is located at a higher frequency as the frequency error is large). The first limiter 82 repeatedly outputs +1 and -1.

The frequency error precision detection unit 66 is an AFC L
It comprises a PF 84 and a second limiter 86. This AFC
The LPF 84 performs low-pass filtering (integration) on the in-phase component signal I ₃ output from the mixer 62, and a second limiter 86
In this example, the signal that has been low-pass filtered by the AFC LPF 84 is limited and output as +1 or −1. This A
The FC LPF 84 exhibits excellent performance (eliminates a noise component that causes a malfunction) when the frequency offset is small, and in this case, ensures accurate convergence of the carrier recovery device 50. Therefore, when the frequency error is small (the in-phase component signal I ₃ is located at the lower frequency as the frequency error is small). The second limiter 86 has +1 and -1
Is output repeatedly.

The output of the first limiter 82 is applied to the input 1 of the multiplexer 72. Second limiter 8
The output of 6 is applied to the input 0 of the multiplexer 72 and also to the lock detector 70. The lock detector 70 changes the output of the second limiter 86 from +1 to-
The period of transition to 1 (or from -1 to +1) is examined, and the input terminal of the multiplexer 72 is selected. If the transition period is smaller than a predetermined critical period (ie, if the frequency error is large), the lock detector 70 selects the input 1 of the multiplexer 72.
However, if the frequency error gradually decreases, and the output of the second limiter 86 has a longer transition period from +1 to -1 and eventually becomes longer than a predetermined critical period, the lock detector 70 switches the multiplexer 72. Selects input terminal 0. The selection of the input terminal 0 is performed by the mixer 6
The output signal I ₄ 0 which means that we are frequency locked.

Accordingly, the multiplexer 72 selects the input terminal 1 when the frequency error is large, and selects the input terminal 0 when the frequency error is gradually reduced. After all, multiplexer 7
The output values of 2 differ according to the degree of frequency error detection. If no or all frequency errors are eliminated, the output of multiplexer 72 is fixed at 1 or -1. The output value of the multiplexer 72 is applied to a mixer 74 and a low-pass filter LPF2 88.

The mixer 74 mixes the Q ₃ signal output from the mixer 62 with the output value of the multiplexer 72 and outputs the result. As the Q ₃ signal output from the mixer 62 is a value indicating the phase error, the phase error is performed from being performed to correct the frequency error. Therefore, the Q ₃ signal has no mean anything when there is a frequency error. First,
It is assumed that the output value of the multiplexer 72, that is, the degree of frequency error detection is large. The output signal Q ₃ of the mixer 62 and the output of the multiplexer 72 are mixed in a mixer 74, and then the VC signal is applied to an APC LPF 76.
Applied to O78.

Therefore, the VCO 78 not only controls the frequency of the pilot signal and the frequency of the demodulated signal to match, but also controls the carrier wave tuning frequency to shift. That is, for a shifted carrier tuning frequency, it is 46.39 MHz as shown in FIG. 5 (for normal carrier tuning, 46.69 MH as shown in FIG. 3).
z).

The value controlled by the VCO 78 is provided to the double conversion tuner 2 as a local oscillation signal LO2.
In the case of the embodiment shown in FIGS. 3 and 5, the local oscillation signal LO2 has a value obtained by adding 0.3 MHz to a frequency adjustment value for making the frequencies coincide. Therefore, the double conversion tuner 2 performs tuning by converting the frequency by the local oscillation signal LO2.

In FIG. 7, the local oscillation signal LO2 is
By shifting by a frequency value that is tuned so as to be shifted in the opposite direction (or more) just by shifting the frequency of the F signal, it can be seen that the pilot signal is at the center of the frequency pull-in region. By repeating such an operation, the carrier recovery device 50 subsequently performs correction of the frequency error.

Next, if the frequency of the pilot signal matches the frequency of the demodulated signal, the output of the multiplexer 72 in the carrier recovery device 50 is fixed at 1 or -1. At this time, only the output of the mixer 62 is
4 so that the carrier recovery device 50 can
Plays the role of LL. Thus, the output of the mixer 74 controls the VCO 78 in a direction that passes through the APC LPF 76 and removes the residual phase error. Then, VCO
78 provides the local oscillation signal LO2 to the dual conversion tuner 2 with a value corresponding thereto.

When the carrier recovery device 50 completes the carrier recovery, it is ideal that the output of the multiplexer 72 is maintained at +1 or -1. However, when the signal-to-noise ratio (S / N ratio) characteristic is poor, that is, when the signal has much noise, the output of the multiplexer 72 is inverted. Also in this case, the signal output from the output of the multiplexer 72 is low-pass filtered by the second low-pass filter LPF288 to maintain the lock, and is limited by the third limiter 90. Then multiplier 6
8 is applied. If the signal-to-noise ratio (S / N ratio) characteristics are always good, the second low-pass filter LP
The F2 88 and the third limiter 90 may not be provided.

The multiplier 68 multiplies the in-phase signal I ₄ passed through the mixer 60 by the signal output from the third limiter 90 and outputs the result. When the frequency is completely locked to the pilot signal, the output of this third limiter 90 is +1 if the phase does not shift, and 180
It is -1 when shifted. Therefore, this multiplier 6
8 indicates that when the phase is shifted by 180 °, the in-phase signal I ₄
Is again shifted by 180 °. The output of the multiplier 68 is transmitted to the next signal processing section of the receiver.

As described above, according to the present invention, the frequency pull-in region can be determined by adjusting the degree of frequency shift of the double conversion tuner 2 and the transfer characteristics of the low-pass filter and the second phase separator 56 at the beginning. . 8 to 10 show various embodiments of the carrier recovery apparatus 50 according to the present invention.

In the embodiment shown in FIG. 8, the frequency error response detector 64 (that is, the differentiator 80, the first limiter 82), the lock detector 70, and the multiplexer 72 shown in FIG. 2 are omitted. The embodiment shown in FIG. 8 has a disadvantage that the recovery time of the carrier is slightly longer than the embodiment shown in FIG. 2, but has an advantage that a simple logic configuration is possible.

The embodiment shown in FIG. 9 is characterized by the use of a secondary local oscillator 92 for generating a second local oscillation signal LO2. At this time, the frequency of the secondary local oscillator 92 is initially set so as to shift in the opposite direction only by the shift of the frequency of the double conversion tuner 2. That is, the case where the frequency shift of the double conversion tuner 2 becomes a problem at the initial stage is as shown in FIG. 4, so that it is sufficient to set the second local oscillation signal LO2 to shift in the opposite direction. Can be expected. The embodiment shown in FIG. 9 is characterized in that the oscillation frequency value of the NCO 58 in the carrier recovery device 50 varies according to the output value of the APC LPF 76. At this time, the carrier recovery device 50 is realized as in the embodiment shown in FIG. In the embodiment as shown in FIG. 9, since the entire carrier recovery device 50 can be digitized, it has a structure advantageous for manufacturing with one IC.

In the embodiment shown in FIG. 10, similarly to the embodiment shown in FIG. 9, a second local oscillation signal LO2 is generated.
It is characterized in that the next local oscillator 92 is used as a fixed oscillator. Then, the NCO 58 in the carrier recovery device 50
Is characterized in that the oscillation frequency value varies according to the output value of the APC LPF 76. At this time, the carrier recovery device 50 is realized as in the embodiment shown in FIG. Therefore, in the embodiment shown in FIG. 10, since the entire carrier recovery device 50 can be digitized, not only has an advantageous structure for manufacturing with one IC, but also the recovery time of the carrier wave is shortened. is there.

Further, the present invention is not limited to such an embodiment, and various modifications can be made without departing from the scope of the present invention.

[0044]

As described above, according to the present invention, the pilot signal is set to the SA
Since the frequency conversion is performed by shifting the frequency in the opposite direction by a predetermined frequency in advance so that the bandwidth corresponding to the base band of the W filter bandwidth and the bandwidth of the low-pass filter can be sufficiently present, An advantage is that only the frequency band of the carrier signal can be passed accurately.

[Brief description of the drawings]

FIG. 1 is a tuning, IF, and carrier recovery device (ie, FPL) of a US 8-level VSB HDTV receiver.
FIG.

FIG. 2 is a diagram illustrating a tuning unit, an IF unit, and a carrier recovery device (ie, FPLL) of a receiver according to the present invention.

FIG. 3 is a frequency spectrum diagram showing a pilot signal of an IF signal existing within a SAW filter bandwidth when carrier tuning is normal.

FIG. 4 is a frequency spectrum diagram showing a pilot signal of an IF signal existing at the end of the SAW filter bandwidth when carrier tuning is shifted.

FIG. 5 performs carrier tuning compensation in case the carrier tuning shifts so that the pilot signal is SA
FIG. 4 shows that the signal is sufficiently present in the W filter bandwidth.

FIG. 6 is a frequency spectrum diagram showing a marginal section in which a carrier recovery device can detect a pilot signal based on a SAW filter bandwidth and a low-pass filter bandwidth when the operation is normal as in FIG. 3;

FIG. 7 is a frequency spectrum diagram showing that a carrier recovery device sufficiently secures a margin section in which a pilot signal can be detected by compensating as in FIG.

FIG. 8 is a diagram illustrating various examples of a carrier recovery apparatus according to the present invention.

FIG. 9 is a diagram showing another embodiment of the carrier recovery apparatus of the present invention.

FIG. 10 is a diagram showing still another embodiment of the carrier recovery apparatus of the present invention.

[Explanation of symbols]

 2 Double conversion tuner 4 SAW filter 48 A / D converter 50 Carrier recovery device 64 Frequency error quick response detector 66 Frequency error precision detector

Claims (14)

(57) [Claims] 1. A receiving system using a communication system for transmitting a pilot signal, comprising: receiving a radio signal including the pilot signal; and tuning the radio signal to a first tuning frequency corresponding to a channel selection by a user. Tuning means for outputting the intermediate frequency signal as an intermediate frequency signal; filtering means for filtering the intermediate frequency signal with a specific frequency bandwidth capable of passing the pilot signal; and frequency conversion of the output of the filtering means to a frequency band capable of digital signal processing A first frequency conversion means for converting the output of the first frequency conversion means into a digital signal; a first I signal and a first
First phase separation means for outputting as a Q signal; low-pass filtering and second phase separation means for performing low-pass filtering and phase separation of the digital signal to output as a second I signal and a second Q signal; A second frequency conversion means for mixing the signal and the first Q signal with a fixed local oscillation frequency and outputting them as a first I signal and a first Q signal in a base band; and a local oscillation means fixing the second I signal and the second Q signal. A third frequency conversion means for mixing the frequency with the frequency and outputting as a second I signal and a second Q signal of a base band; a frequency error is detected by a pilot signal in the second I signal of the base band; Detect the phase error from and create an error correction value to correct this error,
A second tuning frequency, which is obtained by adding a preliminary correction frequency value for the initial shift situation of the first tuning frequency to the error correction value, is provided to the tuning means, and the tuning means corrects the first tuning frequency. And a tuning correction means for controlling the digital carrier recovery device. 2. A tuning error correcting means for detecting a frequency error in a large range by a pilot signal in the second I signal of the base band and outputting a first error detection value for performing a fast correction of the frequency error. Quick response detection means; frequency error precision detection means for outputting a second error detection value for detecting a frequency error in a smaller range in the pilot signal in the second I signal of the base band and performing precise correction of the frequency error; , Having a predetermined frequency error threshold value, wherein the second error detection value is greater than the threshold value in response to the first frequency error threshold value.
An error detection selection control means for outputting a selection control signal and outputting a second selection control signal in response to a value smaller than the threshold value; and selecting the first error detection value in response to the first selection control signal An error detection selecting means for selecting the second error detection value in response to the second selection control signal; a first error detection value or a second error detection value selected from the error detection selection means; Mixing means for mixing the frequency band and the second Q signal to output a frequency error and a phase error detection value, and an error correction value generated from the frequency error and the phase error detection value. A second tuning frequency generating means for adding the provided pre-correction frequency value to the error correction value and outputting the second tuning frequency; and outputting the second tuning frequency when there is no frequency error and no phase error. 2. The digital carrier restoration apparatus according to claim 1, further comprising: a phase state correction unit that corrects a phase state of the first I signal based on an output value of the error detection selection unit. 3. The frequency error response detecting means includes a differentiator for differentiating the second I signal of the base band and outputting the same, and a limiter for limiting and outputting the output of the differentiator. The digital carrier restoration device according to claim 2. 4. A frequency error precise detection means, comprising: a low-pass filter that low-pass filters and outputs the second I signal of the base band; and a limiter that limits and outputs the output of the low-pass filter. The digital carrier recovery device according to claim 2, comprising: 5. A tuning error correction means comprising: a frequency error detection means for detecting a frequency error based on a pilot signal in a second I signal of the base band and outputting an error detection value; Mixing means for mixing the second Q signal to output a frequency error and a phase error detection value; generating an error correction value from the frequency error and the phase error detection value to prepare for an initial shift state of the first tuning frequency. A second tuning frequency generating means for adding the correction frequency value to the error correction value and outputting the second tuning frequency; and a first I output from the second frequency conversion means in the base band when there is no frequency error and no phase error. 2. The digital carrier restoration apparatus according to claim 1, further comprising: a phase state correction unit that corrects a phase state of the signal based on an error detection value. 6. A low-pass filter that low-pass filters and outputs the second I signal of the base band, a limiter that limits and outputs an output of the low-pass filter, The digital carrier restoration device according to claim 5, comprising: 7. A receiving system using a communication system for transmitting a pilot signal, comprising: receiving a radio signal including the pilot signal, and selecting a first tuning frequency corresponding to a channel selection by a user and the first tuning frequency. Tuning means for adding a pre-correction frequency for an initial shift situation, tuning the radio signal to a second tuning frequency and outputting it as an intermediate frequency signal, and a specific frequency band through which the pilot signal can pass the intermediate frequency signal Filtering means for filtering to a width, first frequency converting means for converting the output of the filtering means into a frequency band capable of digital signal processing, signal converting means for converting the output of the first frequency converting means to a digital signal, The digital signal is phase-separated to obtain a first I signal and a first I signal.
First phase separation means for outputting as a Q signal; low-pass filtering and second phase separation means for performing low-pass filtering and phase separation of the digital signal to output as a second I signal and a second Q signal; A second frequency converting means for mixing the signal and the first Q signal with the local oscillation frequency and outputting them as a first I signal and a first Q signal in a base band; and mixing the second I signal and the second Q signal with the local oscillation frequency. Third frequency conversion means for outputting as a second I signal and a second Q signal of a base band, a frequency error is detected by a pilot signal in the second I signal of the base band, and a phase error is detected from the second Q signal of the base band. And a tuning correction means for generating an error correction value for correcting the error and correcting the local oscillation frequency. 8. A tuning error correction means for detecting a frequency error in a larger range of a pilot signal in the second I signal of the base band and outputting a first error detection value for causing a quick correction of the frequency error. Quick response detection means; frequency error precision detection means for outputting a second error detection value for detecting a frequency error in a smaller range in the pilot signal in the second I signal of the base band and performing precise correction of the frequency error; Having a predetermined threshold value of the frequency error, outputting a first selection control signal in response to the second error detection value being greater than the threshold value, and responding to the response when the second error detection value is smaller than the threshold value. Error detection selection control means for outputting a second selection control signal in response to the first selection control signal, selecting the first error detection value in response to the first selection control signal, and selecting the second error detection value in response to the second selection control signal. Error detection selection means for selecting a difference detection value; frequency error and phase error by mixing the first error detection value or the second error detection value selected from the error detection selection means with the second Q signal of the base band Mixing means for providing the detected value as the local oscillation frequency; and when there is no frequency error and no phase error, the phase state of the first I signal output from the second frequency conversion means in the base band is determined by the output value of the error detection selection means. 8. The digital carrier restoring device according to claim 7, comprising: 9. The frequency error quick response detecting means comprises a differentiator for differentiating the second I signal of the base band and outputting the same, and a limiter for limiting and outputting the output of the differentiator. 9. The digital carrier restoration device according to item 8. 10. A frequency error precise detection means, comprising: a low-pass filter that low-pass filters and outputs the second I signal of the base band; and a limiter that limits and outputs the output of the low-pass filter. 9. The digital carrier restoration device according to claim 8, comprising: 11. A tuning error correction means comprising: frequency error detection means for detecting a frequency error based on a pilot signal in the second I signal of the base band and outputting an error detection value; Mixing means for mixing the second Q signal to provide a frequency error and phase error detection value as the local oscillation signal; and a first I output from the second frequency conversion means in the baseband when there is no frequency error and no phase error. 8. The digital carrier restoration apparatus according to claim 7, further comprising: a phase state correction unit that corrects a phase state of the signal based on an error detection value. 12. A low-pass filter that low-pass filters and outputs the second I signal of the base band, a limiter that limits and outputs an output of the low-pass filter, and The digital carrier restoration device according to claim 11, comprising: 13. A method for recovering a carrier wave in a television receiver using a communication system for transmitting a pilot signal to a carrier wave, comprising: converting a received radio signal including the pilot signal to a predetermined first tuning frequency. A first tuning process for tuning to the frequency signal; a filtering process for filtering the intermediate frequency signal only in a specific frequency band through which the pilot signal can pass; and a frequency capable of digitally processing the signal filtered in the specific frequency band. A first frequency conversion step of converting a first frequency into a band, a signal conversion step of converting the first frequency-converted output into a digital signal, a phase separation of the digital signal, a first I signal and a first I signal,
A low-pass filtering and phase separation process of outputting the digital signal as a low-pass filter and phase-separating the digital signal as a second I signal and a second Q signal; and fixing the first I signal and the first Q signal. The first I signal and the first Q signal of the baseband are output after being mixed with the local oscillation frequency, and the second I signal and the second Q signal of the baseband are mixed by mixing the second I signal and the second Q signal with the fixed local oscillation frequency. A second frequency conversion step of outputting as a 2Q signal, an error for detecting a frequency error based on a pilot signal in the second I signal of the base band, detecting a phase error from the second Q signal of the base band, and correcting the error. Make a correction value,
A carrier correction control step of adding a preliminary correction frequency value for an initial shift state of the first tuning frequency to the error correction value and performing control to correct the first tuning frequency. Recovery method. 14. A method for recovering a carrier in a television receiver using a communication system for transmitting a pilot signal to a carrier, comprising: receiving a radio signal including the pilot signal and responding to a user's channel selection. A tuning step of adding a first tuning frequency and a preliminary correction frequency for an initial shift situation of the first tuning frequency, tuning the radio signal to a second tuning frequency, and outputting the same as an intermediate frequency signal; A first frequency conversion step of filtering a signal filtered in the filtering step into a frequency band capable of digital signal processing, and a first frequency conversion step of filtering the signal filtered in the filtering step into a specific frequency bandwidth through which the pilot signal can pass. A signal conversion step of converting a signal whose frequency has been converted in the conversion step into a digital signal; Signal and the first
A low-pass filtering and phase separating step of outputting the digital signal as a Q signal, and performing low-pass filtering and phase separation on the digital signal as a second I signal and a second Q signal, and local oscillation of the first I signal and the first Q signal. The second I signal and the second Q signal of the baseband are output as the first I signal and the first Q signal of the baseband by mixing with the frequency, and the second I signal and the second Q signal of the baseband are mixed and output as the second I signal and the second Q signal of the baseband. A two-frequency conversion process, a frequency error is detected by a pilot signal in a second I signal of the base band, a phase error is detected from a second Q signal of the base band, and an error correction value for correcting the error is generated. A tuning correction process for correcting a local oscillation frequency.