JP2022152779A - Motor controller, motor control method, and program - Google Patents

Abstract

To achieve motor control that enables avoidance of voltage saturation and reduction of its influence considering a dead time.SOLUTION: Provided is a motor controller including: calculation means for calculating an armature voltage limit value based on a voltage error resulting from a dead time in an AC motor and a power supply voltage of an inverter; and derivation means for deriving a dq current command value for controlling the AC motor using a torque command value and the armature voltage limit value.SELECTED DRAWING: Figure 4

Description

The present invention relates to a motor control device, a motor control method, and a program.

Conventionally, electric power is supplied from a power supply capable of controlling frequency, voltage, current, phase, etc. to an AC motor such as an AC motor provided in an electric power steering device, and the operation thereof is controlled. . When performing drive control of a synchronous motor, the voltage output from the control device to the motor to be controlled generally passes through an inverter. In a typical inverter configuration, a set of six switching elements is used to output voltage to a three-phase motor. By these switching, any three-phase voltage can be output within the DC power supply voltage range.

From the controller's point of view, the above configuration limits the output voltage to the motor. In other words, the motor control system is a nonlinear system with output saturation. If the motor is driven without considering the limitation of the power supply voltage of the inverter (hereafter referred to as voltage limitation), especially when the current command is large or when the motor is rotating at high speed, the voltage command increases and the voltage limitation is imposed. Exceeded voltage saturation state. This is because the voltage command is derived each time based on the rotational speed, the current command value, and the current detection value, so the voltage command value also increases when the control is performed as described above. When voltage saturation occurs, various problems arise, such as a sudden drop in output torque and a drop in responsiveness. Therefore, processing is required to prevent voltage saturation or to reduce the influence of voltage saturation.

For example, the induced voltage increases during high-speed rotation. Therefore, it is common to perform flux weakening (FW) control to keep the induced voltage within the voltage limit value by reducing the interlinking magnetic flux due to the permanent magnet due to the demagnetization effect due to the d-axis armature reaction in the motor. (For example, Patent Document 1).

JP 2019-134612 A

In general flux-weakening control as disclosed in Patent Document 1, only the power supply voltage of the inverter is used as the voltage limit value. However, since dead time insertion is unavoidable in actual control, a difference occurs in the output phase voltages. Accurate flux-weakening control requires an accurate grasp of the usable range of voltage. Since the dead time is inserted, the voltage that can be used is reduced, so the torque that can be output by the flux-weakening control during high-speed rotation is reduced. In other words, if the dead time is not taken into consideration, voltage saturation occurs when trying to output the intended torque, resulting in a decrease in the output torque.

SUMMARY OF THE INVENTION In view of the above problems, it is an object of the present invention to realize motor control that enables avoidance of voltage saturation and reduction of its influence in consideration of dead time.

In order to solve the above problems, the present invention has the following configuration. That is, the motor control device is
calculating means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
deriving means for deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.

Moreover, another form of this invention has the following structures. That is, a motor control method comprising:
a calculating step of calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
and a deriving step of deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.

Moreover, another form of this invention has the following structures. That is, the program
the computer,
calculation means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
Using the torque command value and the armature voltage limit value, it functions as derivation means for deriving the dq current command value for controlling the AC motor.

According to the invention of the present application, it is possible to realize motor control that can avoid voltage saturation and reduce the influence of dead time in consideration of dead time.

1 is a configuration diagram showing an example of a schematic configuration of an electric power steering device using a motor control device according to an embodiment of the present invention; FIG. The figure which shows the structural example of the motor control system which concerns on one Embodiment of this invention. FIG. 4 is a diagram for explaining the influence of dead time according to the present invention; 4 is a flowchart showing control signal value calculation processing in consideration of dead time according to an embodiment of the present invention; FIG. 4 is a diagram for explaining a voltage error due to dead time according to one embodiment of the present invention;

EMBODIMENT OF THE INVENTION Hereafter, the form for implementing this invention is demonstrated with reference to drawings. In addition, the embodiment described below is one embodiment for describing the invention of the present application, and is not intended to be construed as limiting the invention of the present application. Not all configurations are essential configurations for solving the problems of the present invention. Moreover, in each drawing, the same component is indicated by the same reference number to indicate the correspondence.

<First Embodiment>
A first embodiment of the present invention will be described below. The configuration of the electric power steering device shown below is an example of application of the present application, and the present invention can be applied to electric motors in general including AC motors (e.g., electric brake booster, etc.) in addition to the electric power steering device. . Moreover, the control method according to the present invention can also be applied to a megatorque motor as an AC motor.

[Configuration overview]
FIG. 1 shows a configuration example of an electric power steering device as an example of a device to which the motor control method according to this embodiment can be applied. A steering wheel 1 is a steered wheel for a driver to perform a steering operation. A steering shaft 2 of the steering wheel 1 is steered through a reduction gear (worm gear) 3, universal joints 4a and 4b, a pinion rack mechanism 5, tie rods 6a and 6b, and hub units 7a and 7b. It is connected to wheels 8L and 8R.

The steering shaft 2 is configured by connecting an input shaft on the side of the steering wheel 1 and an output shaft on the side of the pinion rack mechanism 5 via a torsion bar 9 . The pinion rack mechanism 5 has a pinion 5a connected to a pinion shaft (not shown) to which steering force is transmitted from the universal joint 4b, and a rack 5b that meshes with the pinion 5a. The rotational motion transmitted to the pinion 5a is converted into a rectilinear motion in the vehicle width direction by the rack 5b.

The steering shaft 2 is provided with a torque sensor 10 for detecting steering torque _Tdct applied to the torsion bar 9 . Further, the steering shaft 2 is provided with a steering angle sensor 14 for detecting a steering angle _θh indicating the rotation angle of the steering shaft 2 around the steering wheel 1 side (input shaft side). Further, the steering shaft 2 is provided with an output shaft angle sensor 15 for detecting an output shaft angle _θc indicating the rotation angle of the steering shaft 2 around the pinion rack mechanism 5 side (output shaft side). That is, the steering angle sensor 14 detects the rotation angle of the input shaft with respect to the torsion bar 9 as the steering angle _θh , and the output shaft angle sensor 15 detects the rotation angle of the output shaft with respect to the torsion bar 9 as the output shaft angle _θc . do. The torque sensor 10 detects the steering torque _Tdct based on the twist of the torsion bar 9 caused by the difference between the steering angle _θh and the output shaft angle _θc .

The steering angle sensor 14 and the output shaft angle sensor 15 may be integrated sensors. Further, in FIG. 1, the steering shaft 2 and the torque sensor 10 are shown separately for ease of explanation, but they may be integrated together. The configuration of the torque sensor 10 is not particularly limited, and for example, a sleeve type or ring type that detects torque from the torsion of the torsion bar 9 may be used. In the above configuration, the steering torque _Tdct is detected based on the torsion of the torsion bar 9 caused by the difference between the steering angle _θh and the output shaft angle _θc , but it is not limited to this. For example, the torque value may be detected using the difference between the angle signal of the torsion bar 9 on the steering wheel 1 side and the angle signal on the pinion rack mechanism 5 side. In the following description, the steering wheel 1 side of the steering shaft 2 is also referred to as the upstream side, and the pinion rack mechanism 5 side is also referred to as the downstream side.

The steering torque T _dct detected by the torque sensor 10 includes the driver torque based on the driver's operation on the steering wheel 1 as well as the torque generated by the input (disturbance, etc.) from the downstream side. The command value based on the steering torque T _dct is corrected so as to suppress the vibration caused by the downstream input. The suppressing method here is not particularly limited, and any method may be used.

A steering assist motor 20 that assists the steering force to the steering wheel 1 is connected to the steering shaft 2 via the reduction gear 3 . An EPS-ECU (Electronic Control Unit) 30, which is a controller for controlling an electric power steering (EPS) device, is supplied with electric power from a battery 13 and receives an ignition key signal via an ignition (IGN) key 11. is entered.

A steering assist motor 20 (hereinafter also simply referred to as motor 20) according to the present embodiment is, for example, a three-phase AC motor, which is a type of synchronous motor, and has a permanent magnet field or a winding field. The motor 20 rotates by supplying three-phase AC currents with phases different by 120° to u-phase, v-phase, and w-phase coils (not shown). A rotation angle detection unit 215 including a resolver, a rotary encoder, and the like is installed on the shaft of the rotor (not shown) of the motor 20 . Rotation angle θ of motor 20 detected by rotation angle detection unit 215 is output to EPS-ECU 30 . The use of the rotation angle θ detected here will be described later.

The EPS-ECU 30 calculates a voltage command value as an assist command value based on the steering torque T _dct detected by the torque sensor 10 and the vehicle speed V _h detected by the vehicle speed sensor 12 . Further, the EPS-ECU 30 controls the electric power (voltage value V _ref ) supplied to the steering assist motor 20 according to the voltage command value based on the steering torque T _dct and the voltage command value based on the driving support function. The steering assist motor 20 operates the reduction gear 3 based on the voltage value V _ref input from the EPS-ECU 30 to perform assist control for the steering wheel 1 .

The EPS-ECU 30 may comprise, for example, a computer including a processor and peripheral components such as a storage device. The processor may be, for example, a CPU (Central Processing Unit) or an MPU (Micro-Processing Unit). The storage device may comprise any one of a semiconductor storage device, a magnetic storage device and an optical storage device. The storage device may include memory such as a register, a cache memory, a ROM (Read Only Memory) used as a main memory, and a RAM (Random Access Memory). The functions of the EPS-ECU 30 to be described below are realized, for example, by the processor of the EPS-ECU 30 executing a computer program stored in a storage device.

Note that the EPS-ECU 30 may be formed of dedicated hardware for executing each information process described below. For example, the EPS-ECU 30 may comprise a functional logic circuit set in a general-purpose semiconductor integrated circuit. For example, the EPS-ECU 30 may have a programmable logic device (PLD: Programmable Logic Device) such as a Field-Programmable Gate Array (FPGA).

[Motor control functional configuration]
FIG. 2 is a diagram showing a configuration example of a control system for controlling the motor 20 according to this embodiment. The control system shown in FIG. 2 is configured inside the EPS-ECU 30, for example.

In the control system according to the present embodiment, first, torque command value τ ^* is input to current command generator 201 . In the following description, the symbol " ^* " indicates a command value. Based on the angular velocity ω of the motor 20 input from the angular velocity calculator 217, the current command generator 201 adjusts the d-axis and q-axis components so that the torque generated by the motor 20 and the torque command value τ ^* match. Calculate the current command value for each. Here, the d-axis and the q-axis are axes in the dq rotating coordinate system. The d-axis indicates the direction of magnetic flux of a rotor (not shown) included in the motor 20 . Also, the q-axis indicates a direction perpendicular to the d-axis.

A current command generator 201 generates a dq current command value corresponding to the rotation speed. In this embodiment, MTPA (Maximum Torque Per Ampere) control and FW (Flux Weaking) control are used. Details of the processing by the current command generation unit 201 will be described later with reference to the drawings. The current command value i _d ^* of the d-axis component generated by the current command generator 201 is output to the subtractor 202 . Also, the q-axis component current command value i _q ^* generated by the current command generation unit 201 is output to the subtractor 202 .

_Subtractor 202 subtracts actual current value id output from 3-phase/2-phase converter 216 from current command value _id ^* , and outputs the result to PI control unit 204 as deviation _Δid ^* . Subtractor 203 subtracts actual current value i _q output from 3-phase/2-phase converter 216 from current command value i _q ^* , and outputs the result to PI controller 204 as deviation Δi _q ^* .

A PI control unit 204 receives the deviations Δi _d ^* and Δi _q ^* from the subtractors 202 and 203 and performs PI (Proportional-Integral) control. Here, PI control section 204 calculates voltage command values v _d ^* and v _q ^* using a predetermined transfer function, and outputs them to anti-windup section 205 . At this time, the PI control unit 204 receives from the anti-windup unit 205 the excesses Δv _d ^* and Δv _q ^* from the limit values for the voltage command value as negative feedback to the integrator (not shown), and reflects them in the PI control. Let This prevents saturation in the integrator (not shown).

The anti-windup unit 205 limits the voltage command values v _d ^* and v _q ^* from the PI control unit 204 so that they do not exceed preset limit values, and the excess amounts are Δv _d ^* and Δv _q ^* . Negative feedback is provided to the PI control unit 204 . This prevents the integral value in the integrator (not shown) from continuing to increase when the control output is saturated, and improves followability to command changes. Anti-windup section 205 also outputs voltage command values v _d ^** and v _q ^** after the limitation to decoupling section 206 .

The decoupling unit 206 applies an induced voltage to the voltage command values v _d ^** and v _q ^** input from the anti-windup unit 205 according to the angular velocity ω of the motor 20 input from the angular velocity calculation unit 217. are removed, and decoupling between the q-axis and the d-axis is performed. Decoupling section 206 then outputs the processed voltage command values v _d ^*** and v _q ^*** to 2-phase/3-phase conversion section 207 .

The two-phase/three-phase conversion unit 207 converts the voltage command values v _d ^*** and v _q ^*** input from the decoupling unit 206 into the rotation angle θ of the motor 20 input from the rotation angle detection unit 215 . Based on this, the three-phase voltage command values v _u ^* , v _v ^* , v _w ^* corresponding to the motor 20 are converted. Two-phase/three-phase converter 207 then outputs three-phase voltage command values v _u ^* , v _v ^* , and v _w ^* to dead time compensator 208 .

The dead time compensator 208 compensates for the three-phase voltage command values v _u ^* , v _v ^* , v _w ^* input from the two-phase/three-phase converter 207 to remove the effects of dead time. . The dead time compensator 208 cancels the voltage error by superimposing the voltage error caused by the dead time on each phase voltage command value in advance. The voltage superimposed here has the same magnitude as the voltage error caused by the dead time. Dead time compensation section 208 then outputs the processed three-phase voltage command values v _u ^** , v _v ^** , and v _w ^** to SVM section 209 .

The SVM unit 209 performs SVM (Space Vector Modulation) processing on the three-phase voltage command values v _u ^** , v _v ^** , and v _w ^** input from the dead time compensation unit 208, Duty _u , Duty _v , and Duty _w , which are the duties of each phase, are generated. Then, SVM section 209 outputs the generated 3-phase duty to PWM generation section 210 .

The PWM generation unit 210 generates PWM (Pulse Width Modulation) signals PWM _u , PWM _v , and PWM _w , which are control signals for the inverter 211 , from each of the three-phase duties input from the SVM unit 209 . PWM generator 210 then outputs PWM signals PWM _u , PWM _v , and PWM _w to inverter 211 .

Inverter 211 converts AC power supplied from battery 13 into three-phase AC voltages v _u , v _v , and v _w based on an input from PWM generator 210 , and supplies the three-phase AC voltages v u , v v , and v w to motor 20 . Note that FIG. 1 shows the voltage value V _ref as the command value output from the EPS-ECU 30 to the motor 20, which corresponds to the three-phase AC voltages v _u , v _v , and v _w .

Current detection circuits 212, 213, and 214 for detecting current values corresponding to the three-phase AC voltages _vu , _vv , and _vw are provided on the connection line between the inverter 211 and the motor 20. The values of alternating currents i _u , i _v , i _w are detected. The current detection circuits 212 , 213 and 214 output the detected current values to the 3-phase/2-phase converter 216 . Note that the current detection circuits 212 , 213 and 214 may be provided inside the inverter 211 .

Rotation angle detection unit 215 detects rotation angle θ of motor 20 and outputs it to 2-phase/3-phase conversion unit 207 , 3-phase/2-phase conversion unit 216 , and angular velocity calculation unit 217 .

3-phase/2-phase converter 216 converts 3-phase current values (3-phase AC currents i _u , _iv , and i _w supplied from inverter 211 to motor 20 ) and rotation of motor 20 from rotation angle detector 215 . With the angle θ as an input, the three-phase current values are converted into two-phase actual current values i _d and i _q respectively corresponding to the d-axis and q-axis of the dq rotating coordinate system. Three-phase/two-phase converter 216 then outputs actual current value _{id to subtractor 202 and outputs actual current value iq to} subtractor 203 .

Angular velocity calculator 217 differentiates rotation angle θ from rotation angle detector 215 to calculate angular velocity ω of motor 20 , and outputs it to current command generator 201 and decoupling section 206 .

[Influence of dead time on torque]
FIG. 3 is a conceptual diagram for explaining the influence of dead time in flux-weakening control. In FIG. 3, the horizontal axis indicates the number of revolutions [rpm] of the motor, and the vertical axis indicates the torque [Nm]. A line 301 indicates the change without considering the dead time, and a line 302 indicates the change when the dead time is taken into consideration. Since the dead time is inserted, the voltage that can be used is reduced, so the torque that can be output by the flux-weakening control during high-speed rotation is reduced. That is, as shown in FIG. 3, the torque (line 302) with the dead time taken into account is a lower value than the torque (line 301) calculated without taking the dead time into account. Therefore, as a result of performing flux-weakening control so as to output torque as indicated by line 301, voltage saturation occurs, resulting in a decrease in output torque.

In this embodiment, as shown in FIG. 2, a dead time compensator 208 is provided to remove the influence of dead time. However, in this method, the voltage error due to the dead time is preliminarily superimposed on the phase voltage command value. The voltage superimposed here has a size equivalent to the voltage error of the dead time, and is a voltage that cannot be used for control. Therefore, in this embodiment, in addition to the configuration provided with the dead time compensator 208, the current command generator 201 generates a dq current command value considering the influence of the dead time.

[Current vector control method]
Here, the current vector control method used in this embodiment will be described. In this embodiment, maximum torque/current (MTPA: Maximum Torque Per Ampere) control and flux-weakening (FW: Flux-Weaking) control are used as methods for controlling the current vector in the motor 20 .

Here, definitions of parameters used in various formulas used in the following description are shown. Further, parameters used other than the parameters shown here will be explained as appropriate.

i _d : Current value of d-axis component in dq coordinate system i _q : Current value of q-axis component in dq coordinate system _ia : Current vector or scalar value (norm) of current vector
R _a : Winding resistance in the motor ψ _a : Magnetic flux linkage by permanent magnet in dq coordinate system L _d : Inductance of d axis in dq coordinate system L _q : Inductance of q axis in dq coordinate system β : Current from q axis Vector phase v _d : voltage value of d-axis component in dq coordinate system v _q : voltage value of q-axis component in dq coordinate system v _a : armature voltage vector or scalar value (norm) of armature voltage vector
v _o : induced voltage vector or scalar value (norm) of the induced voltage vector
V _dc : Inverter power supply voltage (DC link voltage)

(MTPA control)
MTPA control is a method of controlling a current vector so that the generated torque is maximized with respect to the current. In the MTPA control, among the current vectors that generate the same torque, the current phase with the smallest amplitude (that is, the most efficient one) is specified as the current command value. In this embodiment, MTPA control is used in a low speed region where the voltage (induced voltage) does not reach a predetermined limit value and the rotation of the motor 20 is low. When the magnitude of the current vector is determined as _ia , the current vector phase β is defined by the following equation (1).

Then, the dq-axis current command values can be calculated using the magnitude _ia of the current vector and the current vector phase β. The dq-axis current command values can be calculated by the following equations (2) and (3).

(FW control)
The FW control is a method of controlling the current vector so that the magnetic flux is reduced by the demagnetization effect due to the d-axis armature reaction due to the negative d-axis current and the induced voltage is kept within the limit value. In the present embodiment, FW control is performed in a high-speed region where the rotation of the motor 20 is high such that the voltage (induced voltage) reaches a predetermined limit value (that is, the voltage is saturated) when MTPA control is applied. use. The FW control suppresses the induced voltage generated by the rotation to enable even higher speed rotation. When the induced voltage limit value is _{vo_lim} and the _q -axis current iq is determined, the d-axis current is defined by the following equation (4).

[Processing flow]
FIG. 4 is a flowchart of control processing by the current command generator 201 according to this embodiment.

In S401, current command generator 201 calculates a limit value and a threshold. Before explaining the calculation of the limit value and the threshold value according to this embodiment, the basic flow of calculation in this process will be explained first. Assuming that the phase current amplitude is I _p , the system maximum phase current amplitude is I _{p_max} , the armature voltage limit value is _{va_lim} , and the current limit value is _{ia_lim} , the relational expression (limiting expression) relating to current and voltage limits is, for example, It can be represented by the following formulas (4) and (5).

Here, the maximum amplitude of the phase voltage due to the power supply voltage V _dc of the inverter 211 is normally V _dc /2, but the fundamental wave amplitude of the phase voltage output by the SVM processing by the SVM unit 209 as shown in FIG. becomes V _dc /√3. Therefore, the armature voltage limit value v _{a_lim} can be calculated by the following equation (7) in consideration of the SVM processing.

Also, the induced voltage limit value v _{o_lim} can be calculated by the following equation (8) using the power factor angle φ.

Here, for the sake of simplification, the minimum value among the values derived from Equation (8) is treated as the induced voltage limit value _{vo_lim} . As a result, the induced voltage limit value v _{o_lim} can be expressed by the following equation (9).

Therefore, the induced voltage limiting formula can be expressed by the following formula (10). The relational expression shown in Expression (10) is treated as a voltage limiting expression.

The threshold used for switching from MTPA control to FW control is determined based on the angular velocity ω of the motor 20, and the angular velocity corresponding to this threshold is called base velocity ω _base . Assuming that the dq-axis currents when i _a = _{ia_lim} are i _d1 and i _q1 , respectively, as a condition of the MTPA control, the base speed ω _base is expressed by the following equation (11).

As shown so far, the current limit formula, the voltage limit formula, and the base velocity ω _base are obtained by this process (formula (5), formula (10), formula (11)). In this embodiment, part of the calculation formula is changed in order to obtain the armature voltage limit value v _{a_lim} that takes into consideration the effect of dead time. Details of the calculation formula according to the present embodiment will be described later with reference to FIG. 5 and the like.

In S402, the current command generator 201 calculates the q-axis current limit value _{iq_lim} corresponding to the number of rotations (that is, the angular velocity ω). When the angular velocity ω of the motor 20 is equal to or lower than the base speed ω _base (ω≦ω _base ), the q-axis current limit value i _{q_lim} is set to i _q1 as shown in the following equation (12) in order to apply the MTPA control. .

On the other hand, when the angular velocity ω of the motor 20 is greater than the base speed ω _base (ω>ω _base ), the q-axis current limit value i _{q_lim} is set to Ask.

In S403, current command generation unit 201 calculates phase current command value I _{p_CMD} based on input torque command value τ ^* . For the phase current command value _{Ip_CMD} , for example, a table defining the correspondence between the output torque (that is, the torque indicated by the torque command value τ ^* ) and the phase current command value is defined in advance and held. may be derived by referring to Then, the current command generator 201 calculates the temporary q-axis current command value _{i′q_CMD} under the MTPA conditions (that is, formula (1) and formula (3)) based on the calculated phase current command value _{Ip_CMD} . . Under the MTPA conditions, the provisional q-axis current command value i′ _{q_CMD} can be calculated, for example, by the following equation (15).

In S404, the current command generation unit 201 limits the temporary q-axis current command values i′ _{q_CMD} calculated in S403 by the q-axis current limit values i _{q_lim} according to the rotation speed, Let the current command value _{be iq_CMD} . That is, the upper limit of the q-axis current command value is the q-axis current limit value i _{q_lim} calculated in S402.

In S405, the current command generator 201 determines whether or not the induced voltage v _o is equal to or less than the induced voltage limit value v _{o_lim} based on the above equation (10). If induced voltage v _o is equal to or lower than induced voltage limit value v _{o_lim} (YES in S405), the process of current command generation unit 201 proceeds to S406. On the other hand, if induced voltage v _o is greater than induced voltage limit value _{vo_lim} (NO in S405), the process of current command generation unit 201 proceeds to S407.

In S406, the current command generator 201 performs MTPA control and calculates the dq current command value. The current command generation unit 201 sets the q-axis current command value _{iq_CMD} calculated in S404 as the _q -axis component current command value iq ^* . Further, the current command generation unit 201 calculates the current command value i _d ^* of the d-axis component using the value of the current command value i _q ^* of the q-axis component and the above equations (1) and (2). do. Then, the current command generation unit 201 outputs the calculated current command values _{id* and iq} ^* _, ^and the processing flow ends. Note that this processing flow is also repeatedly executed while the drive control of the motor 20 continues.

In S407, the current command generator 201 performs FW control and calculates a dq current command value. The current command generation unit 201 sets the q-axis current command value _{iq_CMD} calculated in S404 as the _q -axis component current command value iq ^* . Further, the current command generation unit 201 calculates the current command value id ^* of the _d -axis component using the value of the current command value iq ^* of the _q -axis component and the above equation (4). Then, the current command generation unit 201 outputs the calculated current command values _{id* and iq} ^* _, ^and the processing flow ends. Note that this processing flow is also repeatedly executed while the drive control of the motor 20 continues.

[Limit value considering the effect of dead time]
Details of the calculation process in S401 of FIG. 4 according to the present embodiment will be described. First, the voltage error that appears due to dead time will be described. A voltage error V _dt that appears on average due to the dead time is expressed by the following equation (16) when the DC link voltage V _dc of the inverter 211, the PWM frequency f _pwm , and the dead time T _d .

This becomes a voltage error on a substantially rectangular wave with respect to the phase voltage, depending on the sign of the phase current. FIG. 5 is a conceptual diagram for explaining an overview of the superimposition of voltage errors due to dead time. In FIG. 5, the horizontal axis indicates electrical angle [deg], the left vertical axis indicates voltage [V], and the right horizontal axis indicates current [A]. A line 501 (chain line) indicates the ideal phase voltage, and a line 502 (solid line) indicates the phase voltage. Line 503 (thick line) indicates the voltage error, which is the difference between the voltages indicated by lines 501 and 502 . Line 504 (dashed line) indicates the phase current.

Since the value of the voltage error indicated by the line 503 is added in advance by the dead time compensator 208, this amount of voltage cannot be used for control. As a result, it is assumed that the phase voltage maximum value of the primary electrical angle is reduced. A certain primary phase voltage V _p1 is expressed by the following equation (17), where θ is the phase voltage phase and φ is the phase current phase difference with respect to the phase voltage phase θ.

When the phase current phase difference φ=0, the amplitude represented by Equation (17) becomes the minimum value, which is the maximum phase current amplitude in this embodiment. The maximum phase voltage _{Vp1_max} at this time is represented by the following equation (18).

When the armature voltage limit value v _{a_lim} is calculated using this maximum phase voltage V _{p1_max} , it is obtained by the following equation (19). That is, in this embodiment, the following equation (19) is derived by using equation (18) instead of equation (7).

Then, by using the armature voltage limit value v _{a_lim} obtained by calculating using the formula (19) and the formulas (8) to (11), the induced voltage limit value v _{o_lim} and the base speed ω _base are calculated as calculate. Furthermore, by calculating a current command value corresponding to MTPA control or FW control, it is possible to generate a current command value considering the influence of dead time. In this embodiment, the DC link voltage V _dc , PWM frequency f _pwm , and dead time T _d of the inverter 211 are held and managed in advance by the current command generation unit 201, and used when generating the current command value. .

As described above, according to the present embodiment, it is possible to realize motor control that can avoid voltage saturation and reduce the influence of dead time in consideration of dead time.

<Other embodiments>
In the present invention, a program or application for realizing the functions of one or more embodiments described above is supplied to a system or device using a network or a storage medium, and one or more processors in the computer of the system or device can also be implemented by reading and executing the program.

As described above, the present invention is not limited to the above-described embodiments, and those skilled in the art can make modifications and applications by combining each configuration of the embodiments with each other, based on the description of the specification and well-known techniques. It is also contemplated by the present invention that it falls within the scope of protection sought.

As described above, this specification discloses the following matters.
(1) calculating means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
and deriving means for deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.
According to this configuration, it is possible to realize motor control that enables the avoidance of voltage saturation and the reduction of its influence, taking dead time into account.

(2) When the armature voltage limit value v _{a_lim} , the power supply voltage V _dc of the inverter, the PWM (Pulse Width Modulation) frequency f _pwm , and the dead time T _d are used, the calculation means uses the following equation to obtain: The motor control device according to (1), wherein the armature voltage limit value is calculated.

According to this configuration, the armature voltage limit value can be calculated in consideration of the voltage error obtained from the DC link voltage of the inverter, the PWM frequency, and the dead time.

(3) the motor control device is capable of switching between MTPA (Maximum Torque Per Ampere) control and FW (Flux-Weaking) control;
The derivation means is
calculating a q-axis current limit value based on the armature voltage limit value and the rotation speed of the AC motor;
determining whether to perform the MTPA control or the FW control based on the rotation speed of the AC motor;
The d-axis current command value corresponding to the determined control is calculated using the command value obtained by limiting the q-current command value specified from the torque command value by the q-axis current limit value ( 1) or a motor control device according to (2).
According to this configuration, in a configuration capable of switching between MTPA control and FW control, it is possible to realize motor control that enables avoidance of voltage saturation and reduction of its influence in consideration of dead time.

(4) The derivation means uses the armature voltage limit value to obtain an induced voltage limit value, which is a threshold value for determining whether to perform the MTPA control or the FW control, and a base velocity. The motor control device according to (3), wherein the calculation is performed.
According to this configuration, it is possible to calculate a threshold value for determining whether to perform MTPA control or FW control using the armature voltage limit value that takes dead time into consideration.

(5) first converting means for converting the dq current command values derived by the deriving means into three-phase voltage command values;
second conversion means for generating a three-phase duty signal based on the three-phase voltage command values;
(1) to (4), further comprising second generation means for generating a control signal for supplying a power supply voltage to the AC motor by the inverter based on the three-phase duty signal. A motor control device according to any one of the preceding claims.
According to this configuration, it is possible to realize motor control that enables the avoidance of voltage saturation and the reduction of its influence, taking dead time into account.

(6) a calculation step of calculating an armature voltage limit value based on the voltage error caused by the dead time in the AC motor and the power supply voltage of the inverter;
and a derivation step of deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.
According to this configuration, it is possible to realize motor control that enables the avoidance of voltage saturation and the reduction of its influence, taking dead time into account.

(7) a computer;
calculation means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
A program for functioning as derivation means for deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.
According to this configuration, it is possible to realize motor control that enables the avoidance of voltage saturation and the reduction of its influence, taking dead time into account.

1 Steering Wheel 2 Steering Shaft 3 Reduction Gears 4a, 4b Universal Joint 5 Pinion Rack Mechanisms 6a, 6b Tie Rods 7a, 7b Hub Units 8L, 8R Steering Wheels 9 Torsion Bar 10 Torque Sensor 11 Ignition (ING) Key 12 Vehicle Speed Sensor 13 Battery 14 steering angle sensor 20 steering assist motor 30 EPS (Electric Power Steering)-ECU (Electronic Control Unit))
201 current command generation units 202, 203 subtractor 204 PI control unit 205 anti-windup unit 206 decoupling unit 207 2-phase/3-phase conversion unit 208 dead time compensation unit 209 SVM unit 210 PWM generation unit 211 inverters 212, 213, 214 Current detection circuit 215 Rotation angle detection unit 216 Three-phase/two-phase conversion unit 217 Angular velocity calculation unit

Claims (7)

calculating means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
and deriving means for deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value. The _{calculation means calculates} the _armature 2. The motor control device according to claim 1, wherein the voltage limit value is calculated.

The motor control device is capable of switching between MTPA (Maximum Torque Per Ampere) control and FW (Flux-Weaking) control,
The derivation means is
calculating a q-axis current limit value based on the armature voltage limit value and the rotation speed of the AC motor;
determining whether to perform the MTPA control or the FW control based on the rotation speed of the AC motor;
The d-axis current command value corresponding to the determined control is calculated using a command value obtained by limiting the q-current command value specified from the torque command value by the q-axis current limit value. Item 3. A motor control device according to Item 1 or 2. The derivation means uses the armature voltage limit value to calculate an induced voltage limit value, which is a threshold value for determining whether to perform the MTPA control or the FW control, and a base speed. 4. The motor control device according to claim 3, characterized by: a first converting means for converting the dq current command value derived by the deriving means into a three-phase voltage command value;
second conversion means for generating a three-phase duty signal based on the three-phase voltage command values;
5. The apparatus according to any one of claims 1 to 4, further comprising second generation means for generating a control signal for supplying power supply voltage to said AC motor by said inverter based on said three-phase duty signal. 1. The motor control device according to claim 1. a calculating step of calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
and a derivation step of deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value. the computer,
calculation means for calculating an armature voltage limit value based on a voltage error caused by dead time in the AC motor and the power supply voltage of the inverter;
A program for functioning as derivation means for deriving a dq current command value for controlling the AC motor using the torque command value and the armature voltage limit value.

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