JP2020031508A - Control device of ac motor and control method thereof - Google Patents

Abstract

【Task】
In an AC motor control device, when the AC motor is driven from a low speed range by position sensorless control, a harmonic voltage is applied to the AC motor, and in order to accurately estimate a rotor position by a harmonic current generated by the harmonic current, a harmonic wave is required. It is necessary to increase the current to improve the detection sensitivity, but there is a trade-off relationship with problems such as electromagnetic noise, torque pulsation, and reduced response. Further, when driving an AC motor having a low salient pole ratio, a minute harmonic current flows, so that the estimation accuracy of the rotor position is reduced.
[Solution]
A method for controlling an AC motor that applies a harmonic component to the AC motor using a power conversion device and performs vector control of the AC motor without a position sensor, wherein different processes are performed for the vector control process and the current detection process for the AC motor, respectively. Starting with a signal, vector control without a position sensor is performed using a harmonic component included in the current detected by the current detection processing.
[Selection diagram] Fig. 1

Description

  The present invention relates to a vector control device and an AC motor control system that control an AC motor based on a harmonic current generated by superimposing a harmonic voltage on the AC motor.

  Drive systems for AC motors (hereinafter referred to as "motors") are used in industrial applications such as machine tools, fans, pumps (hydraulic pumps, water pumps), compressors, and air conditioners, as well as AC motors for electric vehicles and railway vehicles. In particular, the motor drive device using a permanent magnet synchronous motor having features of small size and high efficiency has been expanding.

  In order to drive a permanent magnet synchronous motor, it is necessary to accurately detect the rotor position and speed of the motor. 2. Description of the Related Art In recent years, a motor drive device employs sensorless control for estimating a rotor position and a speed from back electromotive force information generated in a motor without using a position sensor or a speed sensor. However, when estimating the rotor position from the back electromotive force information, it is difficult to apply the method in the vicinity of an extremely low speed because the back electromotive force is small. Therefore, as a method of estimating the rotor position at low speed, there is a method using the salient pole ratio of the electric motor.

  As background arts in this technical field, there are Patent Literature 1 and Patent Literature 2. Patent Literature 1 discloses a conventional technique for estimating a rotor position using a salient pole ratio of an electric motor. Here, in order to improve the accuracy of estimating the rotor position, a technique for reducing the influence of the detection error of the harmonic current of the electric motor is disclosed. Accuracy is ensured by setting the cycle of the harmonic current longer than the current sampling cycle and increasing the number of times of detection during one cycle of the harmonic current.

  Further, in the prior art of Patent Document 2, in order to accurately estimate the stop / initial state of the permanent magnet synchronous motor, the harmonic current of the electric motor is set to at least two harmonic currents for each of the positive side and the negative side. Two current values are detected, and the initial position of the rotor is estimated from the amount of change in the harmonic current.

JP 2012-161143 A JP 2005-020918 A

  In Patent Literature 1, by setting the period of the harmonic current to be sufficiently long with respect to the number of times of sampling, the number of times of detection of the harmonic current is increased, and the accuracy of estimating the rotor position is improved. However, since the detection timing of the harmonic current is synchronized with the triangular wave carrier for performing PWM (pulse width modulation) of the controller, the number of detections is restricted by the switching frequency of the power semiconductor element. Therefore, in order to realize this method, it is necessary to set the frequency of the harmonic superimposed on the switching frequency sufficiently low. When harmonics having a low frequency are superimposed, harsh electromagnetic noise or torque pulsation and accompanying mechanical vibration may occur, which is not practical. Also, by lowering the frequency of the harmonic, the responsiveness of the essential sensorless portion is reduced, and it may be difficult to achieve high response as a system.

  Furthermore, in recent years, there has been an increasing tendency to apply low-speed position sensorless control to inexpensive electric motors in the industrial field. Since many inexpensive motors have a low salient pole ratio, there is a problem that the estimation accuracy of the rotor position is reduced. In order to drive a motor having a low salient pole ratio by sensorless driving using the method disclosed in Patent Document 1, it is necessary to further reduce the frequency of the harmonics or increase the amplitude to improve the sensitivity. Problems occur. Also, it is difficult to improve the control response.

  The method of Patent Document 2 is basically a task process of initial position estimation before execution of vector control, and thus does not disclose position estimation in a state where the electric motor is driven, that is, a state at the time of vector control. However, as a method of detecting the amount of change in the harmonic current, a method of setting a peak value and an intermediate value of a triangular wave carrier for performing PWM (pulse width modulation) as detection timing is disclosed. However, the two timings of the peak and the intermediate value of the triangular wave carrier are susceptible to noise and error, and it is difficult to greatly improve the detection accuracy.

  The present invention has been made in view of such problems, and in a control of a rotor position sensorless by harmonic superposition, a control device of an AC motor that can accurately, quietly, and universally estimate a rotor position. The purpose is to provide.

  The present invention has been made in view of the background art and the problems described above, and, for example, a power conversion device for driving an AC motor, and applying a harmonic component to the AC motor using the power conversion device to provide an AC motor without a position sensor. A control method for an AC motor including a controller that performs vector control, wherein the vector control process and the current detection process of the AC motor are started with different trigger signals, and are included in the current detected by the current detection process. Vector control without a position sensor is performed by using the higher harmonic components.

  ADVANTAGE OF THE INVENTION According to this invention, in the rotor position sensorless control by harmonic superposition, the control apparatus and control method of the AC motor which can estimate a rotor position accurately, quietly, and versatilely can be provided.

FIG. 2 is a configuration diagram of a control device of the AC motor according to the first embodiment. FIG. 2 is a configuration diagram of a vector control processor according to the first embodiment. FIG. 2 is a configuration diagram of a current detection processor according to the first embodiment. FIG. 4 is a diagram illustrating a relationship between an actual rotation coordinate axis (d-q axis) of the AC motor, a rotation coordinate axis (dc-qc axis) inside the controller, and an axis error Δθc. FIG. 4 is an explanatory diagram of an operation of the controller according to the first embodiment. FIG. 3 is a configuration diagram of a filter processor in the current detection processor according to the first embodiment. FIG. 3 is a configuration diagram of a ΔIq calculator in a current detection processor according to the first embodiment. FIG. 14 is an explanatory diagram of an operation of the interrupt signal generator according to the second embodiment. FIG. 9 is a configuration diagram of an interrupt signal generator according to a second embodiment. FIG. 14 is a configuration diagram of a ΔIq calculator in a current detection processor according to a third embodiment. FIG. 14 is an explanatory diagram of the operation of the ΔIq calculator according to the third embodiment. FIG. 14 is a configuration diagram of a current detection processor according to a fourth embodiment. FIG. 14 is a configuration diagram of a polarity discriminator in a current detection processor according to a fourth embodiment. FIG. 14 is an explanatory diagram of an operation of a polarity discriminator according to the fourth embodiment. FIG. 14 is a configuration diagram of a control device for an AC motor according to a fifth embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  A control device for an AC motor according to the present embodiment will be described with reference to FIGS.

  FIG. 1 is a configuration diagram of a control device for an AC motor according to the present embodiment. In FIG. 1, a control device 20 includes a power converter 3 and a controller 4. The power supply 1 is a DC power supply that supplies power to a power converter 3 for driving a permanent magnet synchronous motor 2 (hereinafter abbreviated as PM motor). The power converter 3 includes an inverter main circuit section 31 composed of six power semiconductor elements, a gate driver 32 for generating an on / off signal for each power semiconductor element of the inverter main circuit section 31, and a filter capacitor 33. The power converter 3 is controlled based on a PWM signal (pulse width modulation signal) output from the controller 4. The current sensors 8 a and 8 b detect a phase current flowing through the PM motor 2.

  In the controller 4, the vector control processor 6 starts the control process by the trigger signal TRG1 output from the interrupt signal generator 5, and the phase currents Iu and Iw detected by the current sensors 8a and 8b become desired values. Thus, the output voltage command of power converter 3 is calculated. Finally, a PWM signal is generated and output to the gate driver 32.

  On the other hand, the current detection processor 7 starts the arithmetic processing in response to the trigger signal TRG2 output from the interrupt signal generator 5, samples the Iu and Iw detected by the current sensors 8a and 8b, and sends them to the vector control processor 6. The necessary currents Idm and Iqm are calculated, and the axis error Δθc required to drive the PM motor 2 without the rotational position sensor is calculated.

  Next, the operation of the vector control processor 6 will be described with reference to FIG. As shown in FIG. 2, the vector control processor 6 starts the control process by the trigger signal TRG1, and the speed command generator 61 for generating the rotation speed command of the PM motor 2 generates the speed command ωr *, The speed controller 62 operates so that the rotation speed of the PM motor 2 (here, sensorless control is premised and becomes a rotation speed estimation value) ωrc ^, and the current command values id *, dq, Calculate and output iq *. The vector control calculator 63 calculates the voltage commands vdc * and vqc * such that the deviation between the current command values id * and iq * and the currents Idm and Iqm of the PM motor 2 is eliminated. Δvdc * and Δvqc * output from the harmonic voltage generator 67 are added to these values, and converted into three-phase AC voltage commands vu *, vv * and vw * in the dq inverse converter 64. These three-phase AC voltage commands are pulse-width-modulated by the PWM signal generator 66 and output as PWM signals Pu, Pv, Pw. Note that the harmonic voltage generator 67 generates a harmonic voltage required for the position sensorless control of the PM motor 2, and simultaneously generates a sign signal Pvh necessary for performing position estimation. 7 is output.

Since this system is based on the rotor position sensorless control of the PM motor 2, there is an error between the actual rotor position θd and the rotor position θdc assumed inside the controller. It is defined as the axis error Δθc.
Δθc = θdc-θd (1)

  This Δθc is calculated by the current detection processor 7 and output to the vector control processor 6. In the vector control processor 6, the speed estimator 68 corrects the rotation speed ωrc ^ such that Δθc becomes zero, thereby achieving both speed estimation and position control. The phase calculation is performed using ωrc ^, and the phase θdc is used as the phase of the dq inverse transformer 64. The obtained θdc is also used in the current detection processor 7 for coordinate conversion.

  Next, the current detection processor 7, which is a feature of the present invention, will be described in detail with reference to FIGS. FIG. 3 is a configuration diagram of the current detection processor in the present embodiment. As shown in FIG. 3, the current detection processor 7 starts arithmetic processing in response to a trigger signal TRG2, and uses the current detection values Iu and Iw of the PM motor 2 to cause the dq coordinate converter 71 to convert dq with respect to the coordinate of θdc. Coordinate conversion is performed, and the phase currents Iu and Iw are converted into currents Idc and Iqc on the dq axes. Idc and Iqc contain a large amount of harmonic components generated by the harmonic voltage generator 67 of the vector control processor 6, and are filtered by the filter processor 72 in order to cut them. Outputs Idm and Iqm.

  The ΔIq calculator 9 calculates ΔIq, which is a component that changes in accordance with the generation amount of the axis error Δθc, from the harmonic components included in Iqc. The axis error calculator 73 calculates and outputs an axis error Δθc by multiplying ΔIq by a coefficient.

  Here, the relationship between the axis errors Δθc and ΔIq will be described with reference to FIG. This part is a known technique.

  FIGS. 4A to 4C respectively show an actual dq coordinate axis dq axis of the PM motor 2 and a dc-qc axis which is a coordinate axis inside the controller 4, and both have an axis error Δθc. This is an example.

As shown in FIG. 4A, a harmonic voltage Vh is applied to a dc axis which is a coordinate axis on the control side. That is, in the harmonic voltage generator 67 in FIG.
Δvdc * = Vh, Δvqc * = 0 ... (2)
And The harmonic voltage Vh on the dc axis is vector-decomposed on the actual dq coordinate axes of the PM motor 2, and harmonic currents Ihd and Ihq are generated on each axis.

  At this time, as shown in FIG. 4B, when the d-axis inductance Ld is equal to the q-axis inductance Lq, Ihd and Ihq are generated according to the vector ratio of the harmonic voltage. Under such conditions, the axis error Δθc cannot be estimated. However, when the inductance of the dq axes is different, a difference occurs in the amount of generated current. As shown in FIG. 4C, a large amount of current flows on the d-axis having a small inductance, and a small amount of current flows on the q-axis having a large inductance. As a result, the harmonic current Ih is inclined in the direction in which the d axis exists, and a harmonic component Iqhc is generated on the qc axis. This generated amount is related to the axis error Δθc. If the phase is corrected so that this Iqch becomes zero, the position sensorless control is established.

  These operations will be described with reference to waveform diagrams. 5A shows a triangular wave carrier for PWM generation used in the PWM signal generator 66 and a phase voltage command. A PWM signal is generated by comparing the triangular carrier wave with the voltage command. At this time, the interrupt signal TRG1 for the vector control processor 6 is generated at the peak of the peak and the valley of the triangular wave carrier. Vector control processing is performed using this signal TRG1 as a trigger (FIG. 5D). Also, as shown in FIG. 3E, the harmonic voltage is applied as a waveform synchronized with the peaks and valleys of the triangular wave carrier. In the PWM method using a triangular wave carrier, a signal having a frequency higher than the triangular wave carrier frequency cannot be applied. Therefore, the application method as shown in FIG. In addition, in synchronization with the superimposed voltage waveform, a code signal Pvh of the harmonic voltage is also generated and provided to the current detection processing 7.

  By applying the harmonic, the q-axis current Iqc includes the harmonic component Iqhc (FIG. 5 (g)). FIG. 5 (g) shows a conventional current sampling timing. Sampling is performed at the timing of the carrier wave, the peak and the valley, and the sensorless control based on the difference (ΔIq) is performed. However, the timing at which current detection is performed is only at the moment of peaks and valleys of carriers, and is easily affected by noise and detection errors. Considering the S / N ratio of detection, it is necessary to apply a harmonic voltage having a large amplitude in order to obtain ΔIq with sufficient accuracy. In this case, an increase in electromagnetic noise and an increase in loss due to harmonics are required. There is also concern about the development.

  In this embodiment, the current detection processor 7 is operated at a shorter cycle than the vector control processor 6 to solve this problem. FIG. 5H shows the trigger signal TRG2 in the high-speed sampling, and the current waveform is sampled as shown in FIG.

  As shown in FIG. 3, the phase current sampled at high speed is subjected to coordinate processing by the filter processor 72 and the ΔIq calculator 9 after the coordinate conversion.

  FIG. 6 is a configuration diagram of the filter processor 72 in FIG. 3. Here, the filtering by the moving average process is performed using the delay unit 10, the adder 11, and the operation gain 12. “2N−1” delay units 10 are used, and 2N moving average processes are performed. Here, N is the number of samples during a half cycle of the harmonic voltage Vh (here, coincides with T1). By taking 2N moving averages, the influence of harmonic components can be eliminated.

  FIG. 7 is a detailed block diagram of the ΔIq calculator 9 in FIG. In FIG. 7, noise contained in Iqc is removed by a filter processor 91, and then a first-order difference of the waveform is calculated by a difference calculator 92. This difference calculation is a calculation corresponding to “difference calculation at the timing of the peak and valley of the carrier wave” shown in FIG. 5 (g), and the result and the code signal Pvh synchronized with the polarity of the harmonic voltage are calculated. By multiplying by the multiplier 13, ΔIq is calculated. At this time, a delay unit 93 is inserted into the code signal Pvh in consideration of the signal delay in the filter processor 91. Further, since there is a possibility that a variation may remain in ΔIq, the calculation result is smoothed by the low-pass filter (LPF) 14.

  The filter processor 91 in FIG. 7 performs the moving average processing for K pieces. If the value of K is too large, the harmonic component is completely eliminated, so that the moving amount for several times for noise removal is reduced. An average is desirable. Alternatively, there is no problem if a band pass filter is used by digital filter processing.

  In the present embodiment, in the position sensorless control by applying the harmonics, it is possible to detect the current and perform position estimation by introducing a short sample period different from the vector control process. Harmonic voltage can be reduced.

  In this embodiment, the harmonic voltage is applied only to the dc axis. However, there is no problem if the harmonic voltage is applied only to the cc axis or to both axes. Also, a harmonic having a frequency equal to the triangular carrier frequency is applied, but this can also be achieved if the carrier frequency is equal to or lower than the carrier frequency. In this case, it is also possible to apply a sine wave harmonic instead of a rectangular wave. .

  As described above, according to the present embodiment, in the control device for an AC motor, even a minute current can be detected by detecting the current in a cycle shorter than the vector control cycle. Therefore, the amount of superimposed voltage can be reduced, and the electromagnetic noise and torque pulsation generated in the electric motor can be reduced. In addition, by applying the present invention to a motor having a low salient pole ratio, the rotor position can be accurately estimated and the position sensorless control can be performed from a low speed. Further, the response of the control system can be set to the same frequency as the frequency of the harmonic current and the frequency of the PWM triangular wave carrier, so that the response of the control system can be increased.

  A control device for an AC motor according to the present embodiment will be described with reference to FIGS.

  FIG. 8 is an explanatory diagram of the operation of the interrupt signal generator in the present embodiment. 8, TRG1, TRG2, and Iu are the trigger signal and the phase current in FIG. The trigger signal TRG1 is a signal synchronized with the triangular wave carrier as described above. Therefore, T1 which is the cycle of the trigger signal TRG1 is a cycle in which the inverter main circuit performs a switching operation, and at least one ON or OFF operation accompanies the period of T1. As a result, the pulsating component included in the phase current Iu becomes a component having a cycle based on the T1.

  Since the pulsation component due to the harmonic voltage shown in the first embodiment is also added to this pulsation component, it is desired to detect the pulsation component as accurately as possible. For this purpose, it is desirable that the synchronization is performed so that the period T2 for generating the trigger signal TRG2 is always an integer number in the period T1. If the synchronization is not established, "swelling (beat)" due to sampling may occur. Therefore, in order to prevent a detection error due to undulation, the trigger signals TRG1 and TRG2 are generated such that an integral multiple of T2 becomes T1.

  FIG. 9 shows the configuration of the interrupt signal generator 5 that generates a trigger signal under such conditions. In FIG. 9, a transmitter 51 generates a signal of TRG2 at a cycle T2, and counts the signal at a counter 52 to generate a signal of a cycle T1. By setting the counter 52, it is possible to set how many times T1 is larger than T2.

  As described above, according to the present embodiment, it is possible to realize a control device for an AC motor capable of sensing the rotor position with higher accuracy.

  In the present embodiment, a predetermined frequency component is extracted from the detected value of the harmonic current, and the amplitude of the harmonic component is extracted to perform the rotation position sensorless control.

  In the first embodiment, the means has been described in which noise is reduced by filtering by increasing the number of times of sampling of the current, and the position estimation accuracy without the position sensor is improved. However, the position estimation calculation needs to calculate a “difference” with respect to the current value after noise removal, and the resolution at the time of current detection is important. That is, there is a problem that the accuracy cannot be obtained unless a high-resolution AD converter is used as the number of bits of the AD converter used in the controller 4. In order to solve this problem, a method that does not use the difference calculation will be described in the present embodiment.

  A control device for an AC motor according to the present embodiment will be described with reference to FIGS.

  This embodiment can be realized by replacing the ΔIq calculator 9 shown in FIG. 7 in the first embodiment with a ΔIq calculator 9B shown in FIG.

  FIG. 10 shows the configuration of the ΔIq calculator 9B in the present embodiment. The ΔIq calculator 9B includes a sine wave generator 94 that generates a sine wave synchronized with the harmonic voltage code Pvh, a multiplier 13, and a filter processor 91B. Here, the q-axis current Iqc is input, and the amplitude component ΔIq2 of the harmonic current Iqhc included in the q-axis current Iqc is output.

  FIG. 11 shows the operation of the ΔIq calculator 9B of FIG. The sine wave generator 94 generates a sine wave function S (FIG. 11B) equal to the harmonic frequency in synchronization with the harmonic voltage sign Pvh in FIG. 11A, and adds the harmonic current Iqhc to the multiplier 14. Multiply by. Here, the amplitude component of the harmonic current Iqhc can be extracted by setting the magnitude of the sine wave function S to 1 and setting the phase so as to be in phase with the harmonic current Iqhc. The phase may be set so that the peak of the sine wave is obtained at the fall timing of the harmonic voltage code Pvh (for the harmonic voltage, the PM motor 2 can be regarded as an inductance and its current The phase is delayed by 90 degrees).

  The result of multiplication of the sine wave function S and the harmonic current Iqhc is shown in the waveform of FIG. From the waveform shown in FIG. 11D, it can be seen that the amplitude component of the harmonic current Iqhc is extracted. However, since the double frequency component of the harmonic is included, the frequency component doubled by the filter processor 91B. Is removed. The filter processor 91B performs the number of times of moving average that can sufficiently remove the double frequency component, and calculates the amplitude component ΔIq2 of the harmonic current Iqhc corresponding to the axis error Δθc. Note that the moving average period of the filter processor 91B is set to coincide with the half period (the period of T1) of the harmonic voltage Vh.

  As described above, according to the present embodiment, it is possible to extract the frequency component set in advance from the detected value of the harmonic current to obtain the amplitude of the harmonic component, and furthermore, to use the difference calculation. This makes it possible to perform high-precision rotational position sensorless control even when an AD converter with low resolution is used.

  In this embodiment, for the detected value of the harmonic current, the frequency component, which is the main component of the harmonic current, is extracted and the rotation position sensorless control is performed. This is to perform a polarity discrimination process of the rotor position of the PM motor.

  In the estimation calculation of the axis error Δθc in the first embodiment, the position can be estimated only when the axis error Δθc is in a range of ± 90 degrees. However, in the state of the initial position of the rotor, the axis error Δθc is not always in the range of ± 90 degrees, and the axis error Δθc may converge to either 0 degree or 180 degrees. That is, in the case of the coordinate axes shown in FIG. 4, when the axis error Δθc is 0 degree, it means that the dc axis coincides with the d axis. On the other hand, when the axis error Δθc is 180 degrees, it means that the dc axis is inverted by 180 degrees from the d axis. A polarity discrimination process is performed to discriminate these two points. Note that the extraction operation of the frequency component in the present embodiment performs the same processing as in the third embodiment. Hereinafter, the operation of the polarity determination processing will be described.

  A control device for an AC motor according to the present embodiment will be described with reference to FIGS. 12, 13, and 14. FIG.

  This embodiment can be realized by replacing the current detection processor 7 shown in FIG. 3 in the first embodiment with a current detection processor 7C shown in FIG.

  FIG. 12 shows the configuration of the current detection processor 7C including the polarity discrimination processing in this embodiment. The current detection processor 7C includes a dq coordinate converter 71, a filter processor 72, a ΔIq calculator 9, an axis error calculator 73, and a polarity discriminator 15. The current detection processor 7C has the same configuration and operation as the first embodiment except for the polarity discriminator 15.

  FIG. 13 shows the configuration of the polarity discriminator 15. The polarity discriminator 15 includes a second harmonic generator 95, a multiplier 13, and a filter processor 91C for performing a moving average. Here, the d-axis current Idc is input, the polarity is determined using Idc, and a polarity determination signal NS is output.

  FIG. 14 shows the operation of the polarity discriminator 15 of FIG. In synchronization with the harmonic voltage code Pvh in FIG. 14A, the second harmonic generator 95 generates a sine wave function S2 (FIG. 14B) having a frequency twice as high as the harmonic. The peak value of the sine wave function S2 is synchronized with the falling timing of the harmonic voltage code Pvh. The multiplier 14 multiplies the sine wave function S2 by the harmonic current Idhc of the d-axis current to obtain a waveform shown in FIG. The waveform shown in FIG. 14D is subjected to the number of times of moving average capable of sufficiently removing harmonic components by the filter processor 91C, and the average value shown in FIG. 14E is calculated as the discrimination signal NS. The polarity is determined by the sign of. Here, the period of the moving average is one cycle of Vh (twice T1).

  Although the principle of the polarity determination performed here is publicly known, the principle is briefly described as follows. The polarity is determined using the magnetic saturation characteristic of the magnet flux of the PM motor. When the harmonic voltage Vh is applied to the dc axis, which is the coordinate axis on the control side, the harmonic current Idhc of the d-axis current Idc is affected by the magnetic saturation of the magnet magnetic flux, and the positive and negative asymmetric current shown in FIG. It becomes a waveform. Polarity discrimination is performed using a second harmonic component included in the asymmetric waveform. In order to extract the second harmonic component of Idhc, the waveform is multiplied by a sine wave function S2 having a frequency twice as high as the harmonic to obtain a waveform shown in FIG. The moving frequency of the waveform in FIG. 14D allows the average value of the required frequency to be extracted, and becomes the determination signal NS shown in FIG. 14E. Here, if the discrimination signal NS is positive, the axis error Δθc is 0 degrees (the dc axis coincides with the d axis), and if the discrimination signal NS is negative, the axis error Δθc is 180 degrees (the dc axis is 180 degrees with respect to the d axis).

  As described above, according to the present embodiment, for the detected value of the harmonic current, the extraction operation of the frequency component, which is the main component of the harmonic current, and the extraction operation of the frequency component of the second harmonic of the harmonic current are further performed. Accordingly, it is possible to realize a control device for an AC motor capable of performing rotor position sensorless control and polarity determination.

  In the present embodiment, a function of freely setting the cycle T1 for performing vector control and the processing cycle T2 for current detection in the first embodiment is applied.

  FIG. 15 is a configuration diagram of a control device for an AC motor in the present embodiment. This embodiment is applicable to the first to fourth embodiments, and its configuration is realized by replacing the controller 4 in FIG. 1 with the controller 4b shown in FIG. 15 and using the same other components. it can.

  15 differs from FIG. 1 in that the controller 4b adds a communication unit 16 so that it can communicate with other devices connected to the controller 4 of the first to fourth embodiments. Here, the processing period T1 of the vector control and the processing period T2 of the current detection in the present embodiment can be set via the communication unit 16.

  The communication unit 16 performs wired communication by connecting to, for example, a local area network on the personal computer 17 or a field bus of another connected device, and freely sets the processing period T1 and the processing period T2. May be set. Alternatively, for example, communication may be performed with another device wirelessly connected, and the processing cycle T1 and the processing cycle T2 may be set.

  As described above, according to the present embodiment, the processing cycle T1 of the vector control and the processing cycle T2 of the current detection can be freely set by wire communication or wireless communication.

  Although the embodiments have been described above, the present invention is not limited to the above-described embodiments, and includes various modifications. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described above. Further, a part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of one embodiment can be added to the configuration of another embodiment. Further, for a part of the configuration of each embodiment, it is also possible to add, delete, or replace another configuration. Further, the controller may be realized by hardware, for example, by designing a part or all of the controller with an integrated circuit, or by a processor interpreting and executing a program for realizing each function. May be realized by software according to.

1: power supply, 2: permanent magnet synchronous motor (PM motor), 3: power converter, 4, 4b: controller, 5: interrupt signal generator, 6: vector control processor, 7, 7c: current detection processor , 8a, 8b: current sensor, 9: ΔIq calculator, 10: delayer, 11: adder, 12: calculation gain, 13: multiplier, 14: low-pass filter, 15: polarity discriminator, 16: communication means, 17: personal computer, 20: control device, 31: inverter main circuit section, 32: gate driver, 33: filter capacitor, 51: oscillator, 52: counter, 71: dq coordinate converter, 72: filter processor , 73: axis error calculator, 721a: filter processor for dc axis current Idc, 721b: filter processor for qc axis current Iqc, 9B: ΔIq calculator, 91B: filter processor (implementation) Mode 3), 91C: filter processor (within polarity discriminator), 94: sine wave generator, 95: secondary harmonic generator

Claims (10)

A control device for an AC motor comprising: a power conversion device that drives an AC motor; and a controller that applies a harmonic component to the AC motor using the power conversion device and performs vector control of the AC motor without a position sensor. So,
The controller is a vector control processor that performs the vector control process, a current detection processor that performs a current detection process of the AC motor, and starts the processes of the vector control processor and the current detection processor. It has an interrupt signal generator that outputs different trigger signals,
A control device for an AC motor, wherein the vector control without the position sensor is performed using the higher harmonic components included in the current detected by the processing of the current detection processor. The control device for an AC motor according to claim 1,
In the processing cycle T1 of the vector control and the processing cycle T2 of current detection of the AC motor, the current detection processing of the processing cycle T2 is performed twice or more within the processing cycle T1. Control device. The control device for an AC motor according to claim 2, wherein
The control apparatus for an AC motor, wherein the processing cycle T1 of the vector control is a cycle synchronized with a cycle of outputting a switching command of a power semiconductor element of the power converter. The control device for an AC motor according to claim 2 or 3,
The control apparatus for an AC motor, wherein the processing cycle T1 of the vector control is an integral multiple of the processing cycle T2 of current detection of the AC motor. The control device for an AC motor according to claim 1,
The control device for an AC motor, wherein the current detection processor performs a process including a difference between a detected value of the harmonic component from a past detected value and filtering. The control device for an AC motor according to claim 1,
The control device for an AC motor, wherein the current detection processor performs a peak value calculation or a preset frequency component extraction calculation process on the detected value of the harmonic component. The control device for an AC motor according to claim 1,
The current detection processor performs, on the detected value of the harmonic component, an extraction operation of a frequency component which is a main component of the harmonic component and an extraction operation process of a frequency component twice as high as the frequency component. AC motor control device. The control device for an AC motor according to claim 2, wherein
A control device for an AC motor, comprising a function of freely setting the processing cycle T1 and the processing cycle T2. A control device for an AC motor according to claim 8,
A control device for an AC motor, wherein the setting function of the processing period T1 and the processing period T2 is set by another device connected to the AC motor control device by wire communication or wireless communication. An AC motor control method comprising: a power converter that drives an AC motor; and a controller that applies a harmonic component to the AC motor using the power converter and performs vector control of the AC motor without a position sensor. So,
The vector control process and the current detection process of the AC motor are started with different trigger signals, respectively, and using the harmonic components included in the current detected by the current detection process, the vector control without the position sensor is performed. A method for controlling an AC motor.

Images