JP2018110455A - Analog-to-digital converter and diagnosis probe - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an analog-to-digital converter capable of suppressing increase in occupied area.SOLUTION: An analog-to-digital converter has a multiplication type digital-to-analog conversion circuit. The multiplication type digital-to-analog conversion circuit has: a capacitance circuit that samples and amplifies an input signal; a quantizer that quantizes the input signal; and a control circuit that determines a voltage to be supplied to the capacitance circuit according to an output of the quantizer. The capacitance circuit includes a first capacitive element and a second capacitive element each having a first electrode supplied with a positive-phase signal corresponding to the input signal when the input signal is sampled and a second electrode supplied with a reverse-phase signal. When the input signal is amplified, each second electrode is supplied with an output of the control circuit, and a signal at each first electrode becomes an amplified residual amplification signal.SELECTED DRAWING: Figure 1

Description

  The present invention relates to an analog-digital converter, and more particularly to a cyclic analog-digital converter useful for a diagnostic probe and a medical diagnostic system including the diagnostic probe.

  Examples of the diagnostic apparatus constituting the medical diagnostic system include an ultrasonic diagnostic apparatus and an X-ray CT scanner apparatus. Many of these diagnostic apparatuses are used together with a diagnostic probe (including a movable detection unit in the case of an X-ray CT scanner or the like) that comes into contact with a human body to constitute a medical diagnostic system. In medical diagnosis systems, it is required to increase the resolution of in-vivo images of the human body so that diagnosis can be performed more accurately. The signal measured by the diagnostic probe is an analog signal, and processing is performed with a digital signal in the diagnostic apparatus. Therefore, an analog / digital converter is provided in the medical diagnosis system.

  With the demand for higher resolution of in-vivo images, medical diagnostic systems are required to be provided with a higher number of analog-digital converters with higher performance. For example, a medical diagnosis system is required to have many high-performance analog-digital converters having a high conversion rate of several tens of Msps or more and a high resolution of ten or more bits. When such a high-performance analog-digital converter is provided in a diagnostic probe, from the viewpoint of miniaturization of the diagnostic probe, the high-performance analog-digital converter has the occupied area and power consumption of the conventional one. It is required to be an order of magnitude smaller than That is, as an analog-digital converter mounted on a diagnostic probe, a high-performance analog-digital converter having a small area and power consumption, a high conversion rate of several tens Msps or more, and a high resolution of ten or more bits. Will be required.

  A cyclic analog-digital converter is known as an analog-digital converter having a small occupation area, in other words, suitable for mounting in a small area. A cyclic analog-digital converter is disclosed in Non-Patent Document 1, for example.

M.M. Kim. P. Hanumulu and U. Moon, “A10 MS / s 11-bit 0.19 mm 2 algorithmic ADC with improved clocking scheme,” IEEE Journal of Solid-State Circuits, Vol. 44, pp. 2348-2355, Sep. 2009. Imran Ahmed, Jan Mulder, David A, Johns, “A 50MS / s 9.9mW Pipelined ADC With 58dB SNDR in 0.18um CMOS Using Captive Emission Cump 9”. 164-165, Feb. 2009.

  Here, the configuration and operation of the cyclic analog-digital converter will be described. FIGS. 9A to 9D and FIG. 10 are diagrams created by the present inventor in order to study the cyclic analog-digital converter 900 prior to the present invention.

  FIG. 9A is a block diagram showing a configuration of a cyclic analog-to-digital converter 900 configured using a single-stage multiplying digital-to-analog converter circuit (Multiple DAC: hereinafter also referred to as MDAC) 901. FIG. 9B is a timing chart showing the conversion operation by the MDAC 901 shown in FIG. FIG. 9C is a diagram showing a configuration of a cyclic analog-digital converter 900 configured by connecting two MDACs 901a and 901b in series and configured by two stages of MDACs. FIG. 9D is a timing chart showing the conversion operation by the two-stage MDACs 901a and 901b shown in FIG. 9C. Although not particularly limited, the MDACs 901, 901a and 901b have the same configuration, and an example of the configuration is shown in FIG.

9A, the cyclic analog-digital converter 900 includes an MDAC 901 and a switch 904. First, with the switch 904 on the lower side, the node 904a and the node 904c of the switch 904 are brought into conduction. As a result, an input signal 902 that is an analog signal is supplied to the MDAC 901. When the switch 904 is in the lower, the supplied input signal is in this example is converted from 4-bit digital signals D ₁ to D _4. In this 4-bit digital conversion, the MDAC 901 performs time-series conversion of the input signal bit by bit from the upper bit side (eg, D ₁ ) to the lower bit side (eg, D ₄ ) of the digital signal. .

That is, the MDAC 901 samples the supplied input signal, obtains a 1-bit (for example, D ₁ ) digital value corresponding to the sampled voltage value, and outputs the digital value. Next, a residual (difference) between the voltage corresponding to the obtained digital value of 1 bit (for example, D ₁ ) and the input voltage obtained by sampling is obtained, amplified, and the residual obtained by amplification. The voltage is supplied to the node 904 b of the switch 904 through the path 903. The switch 904 is on the upper side so that the node 904b and the node 904c are electrically connected during the analog-digital conversion operation. As a result, the amplified residual voltage is input again to the MDAC 901, and sampling and amplification operations are performed. During this sampling, the next bit (eg, D ₂ ) is determined and output. In this way, the analog signal is converted into a 4-bit digital signal. Here, each of D _{1 to} D ₄ takes, for example, two values of ± 1, or three values of ± 1, 0.

  In FIG. 9B, each of 1S, 2S, 3S, and 4S indicates a period during which sampling is performed by the MDAC 901 (sampling period), and each of 1A, 2A, 3A, and 4A indicates a residual. An amplifying period (residual amplification period) is shown. Although not particularly limited, the sampling period and the residual amplification period are substantially the same time. In this example, since an analog signal is converted into a 4-bit digital signal, one conversion cycle is from a sampling period 1S to a residual amplification period 4A as shown in FIG. 9B. When converting an analog signal into a digital signal in time series, the conversion cycle is repeated.

FIG. 9C shows the configuration of another cyclic analog-digital converter 900. In the figure, each of 901a and 901b is an MDAC having the same configuration as the MDAC 901 shown in FIG. However, unlike the MDAC 901, the MDAC 901a (901b) converts the input signal into 2-bit digital signals D ₁ and D ₃ (D ₂ and D ₄ ). The output of the MDAC 901a is connected to the input of the MDAC 901b via the path 906, and the output of the MDAC 901b is connected to the node 904b of the switch 904 via the path 905. That is, the MDACs 901a and 901b are connected in series and have a two-stage configuration.

The MDAC 901a and the MDAC 901b operate so as to overlap each other. That is, as shown in FIG. 9D, the sampling period (1S, 3S) of MDAC 901a and the residual amplification period (4A, 2A) of MDAC 901b overlap, and the residual amplification period (1A, 3A) of MDAC 901a. And the sampling period (2S, 4S) of MDAC 901b overlap. Thereby, the conversion period in each MDAC 901a and 901b can be made half of the conversion period shown in FIG. 9B. In this case, 2-bit digital signals (D ₁ , D ₃ ) are output from the MDAC 901a in the sampling periods 1S and 3S, and 2-bit digital signals (D ₂ , D ₄ ) are output from the MDAC 901b in the sampling period 2S. 4S. As a result, the number of conversion bits is the same as that of the cyclic analog-digital converter shown in FIG. 9A, but the conversion cycle can be shortened. That is, the conversion rate can be doubled. However, since a two-stage MDAC is used, it is conceivable that the mounting area (occupied area) increases.

  In the cyclic analog-digital converter shown in FIGS. 9A and 9C, in order to realize a higher conversion rate, the conversion processing time of each bit performed in the MDACs 901, 901a, and 901b is shortened. Is needed.

  Next, a multiplying digital-to-analog converter circuit (MDAC) 901 examined by the present inventors will be described with reference to FIG.

  The MDAC 901 is basically configured by an analog circuit, and includes a coarse quantizer 1000 that roughly quantizes an input signal Vin, a digital-analog converter (DAC) 1001, a difference unit 1002, and an amplifier 1003. . Here, the input signal Vin is a signal supplied to the MDAC 901 via the switch 904 in FIG. 9A or 9C. Further, the output Vout of the amplifying unit 1003 is a signal supplied to the path 903, 905, or 906 in FIG. 9A or 9C.

The input voltage Vin is roughly quantized by the coarse quantizer 1000, and the result is a digital signal D _i (i-th bit) that is the output of the MDAC 900. The digital signal D _i is converted again into an analog voltage corresponding to the digital signal D _i by the digital-analog converter 1001. The difference between the converted analog voltage and the input voltage Vin is obtained by the differentiator 1002. This difference is because the input voltage Vin, a voltage obtained by subtracting a voltage corresponding to the digital signal D _i, the residual voltage. This residual voltage is amplified by an amplifying unit 1003 having a gain G and becomes an output Vout of the MDAC. This output Vout becomes the next input voltage Vin of the MDAC 901, and a conversion process for obtaining the next bit is performed. Such an operation (MDAC operation) is repeated N times to finally obtain N-bit digital signals D _{1 to DN} . In the example shown in FIG. 9A, the MDAC operation (conversion process) is executed four times.

At this time, the relationship between the analog voltage Vin, which is an input signal, and the digital signals D _{1 to DN} is expressed by Expression (1). Here, Q is a quantization error generated in the coarse quantizer 1000, G _i is a gain of the amplification unit 1003 in the i-th stage (i = 1 to N−1) MDAC, and Vref is a reference Voltage. As shown in FIG. 9C, when a plurality of MDACs 901a and 901b are connected in series, the gain G of the amplification unit 1003 in each MDAC is assumed to vary between MDACs. In 1), the gain in each MDAC is represented separately. Of course, as shown in FIG. 9A, when the cyclic analog-digital converter 900 is configured with one stage of MDAC 901, each of the gains G _{1 to} G _N−1 in Expression (1) The gain G of the amplifying unit 1003 may be used.

Here, the term excluding the final term (Q / G ₁ · G ₂ ... _GN−2 · G _N-1 ) in the equation (1) corresponds to the analog-digital conversion result. That is, the term portion excluding the final term represents the relationship between the input voltage Vin and the digital signals D _{1 to DN} . In this case, the last term corresponds to a conversion error. Therefore, in the cyclic analog-digital converter, the conversion error is increased by increasing the gain G (in other words, residual amplification factor) of the amplification unit 1003 provided in the MDAC to more than 1 and increasing the number of conversions N. Can be reduced.

  Non-Patent Document 1 discloses a cyclic analog-digital converter using MDAC. In the MDAC described in Non-Patent Document 1, residual amplification is realized by a feedback operation of an operational amplifier. In order to increase the conversion rate of the cyclic analog-digital converter, it is necessary to shorten the conversion processing time of each bit as described above. In Non-Patent Document 1, in order to shorten the conversion processing time, it is required to use a wide-band operational amplifier as the operational amplifier, and it is considered that the power consumption of the operational amplifier increases. That is, when the conversion rate is increased, the power consumption of the cyclic analog-digital converter may be increased.

  Non-Patent Document 2 discloses a pipeline type analog-digital converter. Non-Patent Document 2 discloses a technique for realizing residual amplification without using an operational amplifier. In order to examine the technique disclosed in Non-Patent Document 2, the inventor of the present application created a diagram of the circuit under consideration based on the disclosure of Non-Patent Document 2. FIG. 11 is an explanatory diagram showing a configuration of a study circuit created by the inventor of the present application. Next, this examination circuit will be described.

  The study circuit shown in FIG. 11 includes capacitive elements C11 and C12, a switch 1101 and a coarse quantizer 1100. The operation of the study circuit is similar to the MDAC 901 shown in FIGS. 9A and 9B during the sampling period (left side of the arrow in FIG. 11) and the residual amplification period (right side of the arrow in FIG. 11). I know. That is, it operates in the sampling period and then operates in the residual amplification period. As the input signal Vin, a positive phase input signal + Vin corresponding to the input signal and a negative phase input signal −Vin having a reverse (inverted) phase with respect to the positive phase input signal Vin are supplied.

  In the sampling period, the positive phase input signal + Vin is supplied to one electrode of the capacitive element C11, and the negative phase input signal −Vin is supplied to one electrode of the capacitive element C12. In the figure, in order to clearly indicate that the positive phase input signal + Vin and the negative phase input signal −Vin are in the inverted phase, the positive phase input signal + Vin and the negative phase input signal −Vin are respectively inverted with the sine wave. Although drawn as a sine wave, the input signal supplied to one electrode of each of the capacitive elements C11 and C12 in the sampling period is, for example, a signal (voltage) at time t1 in FIG. At the time t1, the other electrode of each of the capacitive elements C11 and C12 is connected to an AC ground (ground voltage Vs). That is, a so-called pseudo differential is configured in the sampling period. Thus, during the sampling period, the positive phase input signal + Vin is applied to the AC ground (Vs) between the pair of electrodes of the capacitive element C11, and charge is charged. Similarly, a negative-phase input signal −Vin is applied to the AC ground (Vs) between the pair of electrodes of the capacitive element C12, and charge is charged.

In the sampling period, the coarse quantizer 1100 quantizes the input signal Vin. In FIG. 11, the voltage range of the input signal is divided into three stages, and quantization is performed by determining in which range the voltage of the input signal exists by the coarse quantizer 1100. Here, the voltage value of the input signal, 0, + 1, the digital signal D _i represent either a -1 is output from the coarse quantizer 1100.

Next, in the residual amplification period, one electrode of the capacitive element C12 is connected to the other electrode of the capacitive element C11, and the other electrode of the capacitive element C12 is connected to the switch 1101. Further, the output signal Vout is extracted from one electrode of the capacitive element C11. Switch 1101, one of three switches 1102 to 1104 are turned on in accordance with the three includes a switch 1102 to 1104, the value of the digital signal _{D i.} Thus, the other electrode of the capacitor C12 in accordance with the value of the digital signal _{D i,} the reference voltage Vref, either 0V and -Vref are applied.

In the residual amplification period, the capacitive elements C11 and C12 that are charged in the sampling period are connected in series. Therefore, according to the law of conservation of charge, the voltage value of the output signal Vout with respect to the other electrode of the capacitive element C12 is, for example, twice the input signal Vin. Further, the other electrode of the capacitor C12, a voltage in accordance with the value of the digital signal _{D i (+ Vref, 0V,} -Vref) because is applied, the output signal Vout is a value of the converted digital signal _{D i} The reflected value (residual). By performing such a series of operations (operation in the sampling period and operation in the residual amplification period), it is possible to provide the same function as the residual amplification in MDAC. When the capacitance values of the capacitive elements C11 and C12 are the same capacitance value C, the residual gain G is almost doubled.

  According to this examination circuit, an MDAC can be configured without using an operational amplifier. However, a capacitive element having a total capacitance value of 2C is necessary, and the occupied area may be increased. In addition, since it is necessary to generate three types of reference voltages, + Vref, 0V, and −Vref, the power consumption increases in the reference voltage generation circuit that generates these reference voltages, and the overall power consumption may increase. Conceivable.

  An object of the present invention is to provide an analog-digital converter capable of suppressing an increase in occupied area.

  Of the inventions disclosed in the present application, the outline of typical ones will be briefly described as follows.

  That is, the analog-to-digital converter includes one or more multiplying-type digital-to-analog conversion circuits, and the multiplying-type digital-to-analog conversion circuit includes a capacitor circuit that samples and amplifies the input signal, and a quantizer that quantizes the input signal. And a control circuit that determines a voltage supplied to the capacitor circuit according to the output of the quantizer. Here, when sampling the input signal, the capacitor circuit is supplied with a first electrode to which a positive phase signal corresponding to the input signal is supplied, and a second electrode to which a negative phase signal having an opposite phase to the positive phase signal is supplied. And a second capacitor element having a first electrode to which a positive phase signal is supplied and a second electrode to which a negative phase signal is supplied. When amplifying the input signal, a voltage according to the output of the quantizer is supplied from the control circuit to the second electrodes of the first capacitor element and the second capacitor element, and the first capacitor element and the second capacitor element The signal at the first electrode is an amplified signal.

  When sampling the input signal, a voltage difference between the positive phase signal and the negative phase signal is applied between the first electrode and the second electrode of each of the first capacitive element and the second capacitive element. Therefore, it is possible to increase the amount of charge accumulated in each of the first capacitor element and the second capacitor element with respect to the same capacitance value. Thereby, since the capacitance values of the first capacitor element and the second capacitor element included in the capacitor circuit can be reduced, it is possible to reduce the size of these capacitor elements and suppress an increase in the area occupied by the analog-digital converter. It becomes possible.

  Further, when the reference voltage is equivalently set by the capacitance ratio of the capacitive element included in the capacitive circuit and the input signal is amplified, the first voltage is determined according to the output of the quantizer obtained by the quantization based on the reference voltage. A voltage supplied to the second electrode of the capacitive element and the second capacitive element is determined. Thereby, the control circuit supplies the power supply voltage or the ground voltage to the second electrode of the first capacitor element and the second capacitor element according to the output of the quantizer, for example, thereby outputting the quantizer output based on the reference voltage. Can be reflected in the amplified signal. As a result, it is possible to suppress an increase in power consumption in a circuit that forms a voltage supplied to the second electrode of each of the first capacitor element and the second capacitor element. It is possible to suppress an increase in power.

  In one embodiment, the capacitor circuit included in the multiplying digital-to-analog converter circuit includes a first capacitor bank including first and second capacitor elements each having a first electrode and a second electrode; And a second capacitor bank including third and fourth capacitors each having a first electrode and a second electrode. Here, when sampling the input signal, a positive phase signal is supplied to each of the first electrodes of the first, second, third and fourth capacitive elements, and the first, second, third and fourth are supplied. A negative phase signal is supplied to each second electrode of the capacitive element. When the input signal is amplified, the first electrodes of the first and second capacitors are used as output nodes of the multiplying digital-analog converter circuit, and the first electrodes of the third and fourth capacitors are The second electrodes of the first and second capacitive elements are coupled to the second electrodes, and the voltage from the control circuit is supplied to the second electrodes of the third and fourth capacitive elements.

  According to this embodiment, when the first capacitor element and the second capacitor element in the first capacitor bank and the third capacitor element and the fourth capacitor element in the second capacitor bank amplify the input signal. Connected in series. This makes it possible to increase the amplification gain of the multiplying digital-to-analog converter circuit by about four times.

  Furthermore, in one embodiment, a diagnostic probe is disclosed. The diagnostic probe includes a plurality of analog-digital converters each receiving a signal under measurement as an input signal. Each of the plurality of analog-digital converters has a multiplication type digital-analog conversion circuit. Here, each of the multiplying digital-to-analog conversion circuits includes a quantizer that quantizes an input signal based on a reference voltage, a passive circuit that samples and amplifies the input signal, and a buffer circuit that receives an output of the passive circuit. And a control circuit for generating a voltage to be supplied to the passive circuit according to the output of the quantizer. The passive circuit includes the first capacitive element and the second capacitive element described above, and is composed of passive elements. Thereby, it is possible to suppress an increase in power consumption in the passive circuit.

  Furthermore, in the form relating to the diagnostic probe, a plurality of analog-digital converters are formed in one semiconductor integrated circuit device, and a common voltage is supplied to the control circuit in each of the plurality of analog-digital converters. The The voltage supplied to the control circuit is supplied to the second electrodes of the first capacitor element and the second capacitor element when the input signal is amplified in the passive circuit, and becomes a reference voltage. is there. As in this embodiment, by making the voltage supplied to the control circuit common among the plurality of analog-digital converters, the reference voltage varies among the plurality of analog-digital converters. It becomes possible to suppress the measurement variation among the signals under measurement, and it is possible to provide an accurate diagnostic probe.

  Furthermore, in one embodiment, a diagnostic system including the above-described diagnostic probe and diagnostic apparatus is disclosed. In the diagnostic probe, the signal under measurement is converted into a digital signal and supplied to the diagnostic apparatus as a radio signal. Therefore, it is possible to provide a diagnostic system that is easy to handle.

  Of the inventions disclosed in the present application, effects obtained by typical ones will be briefly described as follows.

  That is, it is possible to provide an analog-digital converter that can suppress an increase in occupied area.

1 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit according to Embodiment 1. FIG. 6 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit according to Embodiment 2. FIG. (A) And (b) is the block diagram and timing diagram which show the structure of the cyclic type analog-digital converter concerning Embodiment 1. FIG. (A) And (b) is the block diagram and timing diagram which show the structure of the cyclic type analog-digital converter concerning Embodiment 3. FIG. FIG. 10 is a circuit diagram illustrating a configuration of a multiplication type digital-analog conversion circuit according to a fourth embodiment. (A) And (b) is the block diagram and timing diagram which show the structure of the cyclic type analog-digital converter concerning Embodiment 5. FIG. FIG. 10 is a block diagram illustrating a configuration of an ultrasonic diagnostic system according to an eighth embodiment. It is explanatory drawing for demonstrating a basic concept. (A)-(d) is a block diagram and timing diagram which show the structure of the cyclic type analog-digital converter which this inventor examined. It is a block diagram which shows the structure of a multiplication type digital analog conversion circuit. It is explanatory drawing for demonstrating examination by this inventor. It is a wave form diagram for demonstrating operation | movement of a multiplication type digital analog conversion circuit. FIG. 10 is a circuit diagram illustrating a configuration of a multiplication type digital-analog conversion circuit according to a sixth embodiment. FIG. 10 is a circuit diagram illustrating a configuration of a multiplication type digital-analog conversion circuit according to a seventh embodiment.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. Note that components having the same function are denoted by the same reference symbols throughout the drawings for describing the embodiment, and the repetitive description thereof will be omitted.

  In the following embodiments, when it is necessary for the sake of convenience, the description will be divided into a plurality of sections or embodiments. However, unless otherwise specified, they are not irrelevant to each other. There are some or all of the modifications, details, supplementary explanations, and the like. Further, in the following embodiments, when referring to the number of elements (including the number, numerical value, quantity, range, etc.), especially when clearly indicated and when clearly limited to a specific number in principle, etc. Except, it is not limited to the specific number, and may be more or less than the specific number. Further, in the following embodiments, the constituent elements (including element steps) are not necessarily indispensable unless otherwise specified and clearly considered essential in principle. Needless to say.

  Similarly, in the following embodiments, when referring to the shapes, positional relationships, etc. of the components, etc., the shapes are substantially the same unless otherwise specified, or otherwise apparent in principle. And the like are included. The same applies to the above numerical values and ranges.

(Basic concept)
A plurality of embodiments will be described below. First, a basic concept will be described with reference to FIG. Here, a multiplication type digital-analog conversion circuit will be described. The multiplying digital-to-analog converter circuit (MDAC) described here constitutes a cyclic analog-to-digital converter by using, for example, as described in FIG.

  As will be described later in Embodiment 1, the MDAC includes a coarse quantizer 114 that roughly quantizes the input signal Vin, a control circuit including the control unit 115, and a capacitive circuit including a plurality of capacitive elements. doing. Of these elements, only the coarse quantizer 114 and the capacitive circuit are shown in FIG. 8 to explain the basic concept. Note that the capacitor circuit includes a plurality of capacitor elements each being a passive element, and thus can be regarded as a passive circuit or a passive circuit. Therefore, in this specification, the capacitor circuit may be referred to as a passive circuit.

  FIG. 8 is an explanatory diagram for explaining the basic concept. In FIG. 8, the MDAC operates in a sampling period for sampling an input signal and an operation in a residual amplification period for amplifying the sampled input signal. Is schematically shown. The sampling period is shown on the left side of the arrow in FIG. 8, and the residual amplification period is shown on the right side of the arrow in FIG.

In the sampling period, the input signal Vin is roughly quantized by the coarse quantizer 114. That is, coarse quantizer 114 receives reference voltages + Vref and -Vref, and sets three voltage ranges using reference voltages + Vref and -Vref. As the three voltage ranges, for example, (a) a voltage range larger than 1/4 of the reference voltage + Vref, and (b) a voltage between 1/4 of the reference voltage + Vref and 1/4 of the reference voltage −Vref. A range and a voltage range smaller than 1/4 of (c) the reference voltage-Vfre are set. Crude quantizer 114, the voltage value of the input signal Vin, depending present in any of these voltage ranges (a) ~ (c), performs quantization, and outputs a digital signal D _i. Here, for ease of explanation, the digital signal _Di includes “1”, “1”, “0”, and “1” according to which of the voltage ranges (a) to (c) the voltage value of the input signal Vin is present. It shall be 0 ”and“ −1 ”.

  The capacitive circuit includes capacitive elements C1, C2, C3a, and C3b. In the sampling period, the capacitive elements C1, C2, C3a, and C3b are connected in parallel to each other. That is, the first electrodes P1 of the capacitive elements C1, C2, C3a, and C3b are connected to each other, and the second electrodes P2 of the capacitive elements C1, C2, C3a, and C3b are connected to each other. In the sampling period, the positive phase input signal + Vin corresponding to the input signal Vin is supplied to the first electrodes P1 of the capacitive elements C1, C2, C3a, and C3b, and is in reverse phase with respect to the positive phase input signal + Vin. The negative-phase input signal −Vin is supplied to the second electrodes P2 of the capacitive elements C1, C2, C3a, and C3b. In the same figure, similarly to FIG. 11, in order to clearly show that the positive phase input signal + Vin and the negative phase input signal −Vin are inverted from each other, the waveforms of the respective sine waves are drawn. However, in the sampling period, the positive phase input signal + Vin and the negative phase input signal −Vin supplied to the first electrode P1 and the second electrode P2 of each capacitive element are, for example, values at time t1.

That is, in the sampling period, sampling is performed in a state where the positive-phase input signal voltage and the negative-phase input signal voltage of the input differential signal are connected to the electrodes P1 and P2 at both ends of each capacitive element, respectively. In other words, as shown in FIG. 11, a pseudo-phase structure in which a positive-phase or reverse-phase input signal is supplied to one electrode of a capacitive element and a pseudo AC ground is supplied to the other electrode. This is not complete differential sampling (pseudo differential sampling), but complete differential sampling (fully differential sampling) in which a positive phase signal and a negative phase signal are supplied to both electrodes P1, P2 of the capacitive element. Thereby, in the sampling period, when the charge amount that can be charged in the capacitive elements C11 and C12 in the study circuit shown in FIG. 11 and the charge amount that can be charged in the capacitive circuit shown in FIG. The total capacitance value of the capacitive elements C1, C2, C3a, and C3b is half of the capacitance value C of the capacitive element shown in FIG. 11 (when the capacitive element C11 and the capacitive element C12 have the same capacitance value C). . Also, the differential thermal noise voltage (k _B T / C noise) generated by the resistance of the sampling switch is logically the same in FIGS.

  That is, by performing fully differential sampling instead of pseudo-differential sampling, even if the total capacity of the capacitive elements required for sampling is reduced to ¼ compared to the study circuit (FIG. 11), the signal pair The noise ratio can be maintained at the same level as the study circuit. As a result, it is possible to suppress an increase in the area occupied by the capacitor circuit. Furthermore, since the capacitive circuit is constituted by a plurality of capacitive elements C1, C2, C3a, and C3b, it is possible to arrange each capacitive element so that the occupied area is small.

As described above, in the sampling period, the input signal Vin is quantized into three values (1, 0, −1) by the coarse quantizer 114. In the residual amplification period (right side in FIG. 8), the first electrodes P1 of the capacitive elements C1, C2, C3a, and C3b are connected to each other, and this common connection node corresponds to the output node Nout of the MDAC, and the output Output signal Vout is taken out from node Nout. In the residual amplification period, the second electrode P2 of the capacitive element C1 is connected to the power supply voltage Vdd, and the second electrode P2 of the capacitive element C2 is connected to the ground voltage (ground) Vs. Meanwhile, the voltage supplied to the second electrode P2 of the respective capacitance elements C3a and C3b will vary according to the value of the digital signal D _i is the output of the coarse quantizer 114. That is, according to the value of the digital signal _{D i,} second electrode P2 of the respective capacitance elements C3a and C3b are connected to the power supply voltage Vdd or the ground voltage Vs. 8, the value of the digital signal _{D i} is the case of "1", state (a), the value of the digital signal _{D i,} if "0", state (b), the digital signal _{D i} The case where the value is “−1” is shown as the state (c).

Typically, for example, the capacitive element C3a and the capacitive element C3b have the same capacitance value, and as will be understood from the states (a) to (c) illustrated in FIG. If the value of the digital signal D _i is “−1” (state (c)), the second electrodes P2 of the two capacitive elements C3a and C3b are both connected to the power supply voltage Vdd, and the digital signal D _i Is “1” (state (a)), the second electrodes P2 of the two capacitive elements C3a and C3b are both connected to the ground voltage Vs. The value of the digital signal _{D i} is "0", (condition (b)), 2 two capacitive elements C3a, second electrode P2 of the capacitor C3a is one of C3b is connected to the power supply voltage Vdd, The second electrode P2 of another capacitive element C3b is connected to the ground voltage Vs. In FIG. 8, the ● marks indicate the same places in the sampling period and the residual amplification period.

  Here, the relationship between the output signal Vout of the MDAC and the input signal Vin follows equation (2). In Expression (2), C1 and C2 are the capacitance values of the capacitive elements C1 and C2, C3 is the capacitance value of the capacitive element C3a (or C3b), and Vref is the reference voltage Vref. Here, the reference voltage Vref is expressed by Expression (3). Also in Formula (3), C1, C2, and C3 are the same as Formula (2). However, in the derivation of these equations, it is used that the actual circuit is realized by two differential circuits that operate in a complementary manner, as will be seen in each of the following embodiments.

In the cyclic analog-digital converter, the MDAC output signal Vout becomes the MDAC input signal Vin in the next sampling period, as described above with reference to FIG. That is, the MDAC output signal is fed back as an MDAC input signal a predetermined number of times, and the input signal Vin, which is an analog signal, is converted into a digital signal D _i (multiple bits).

As understood from the equation (3), the reference voltage Vref can be equivalently determined by a ratio of capacitance values (capacitance ratio) of the capacitor elements included in the capacitor circuit. In the capacitive circuit illustrated in FIG. 8, the capacitive circuit includes capacitive elements C1, C2, C3a, and C3b, and the capacitive element C3a and the capacitive element C3b have the same capacitance value C3. Therefore, the reference voltage Vref is equivalently determined by the capacitance ratio of these capacitive elements (formula (3)). Further, as understood from the equation (2), in the residual amplification period, the output voltage Vout of the MDAC is the product of the digital value D _i obtained by the coarse quantizer 114 and the reference voltage Vref. It is obtained by subtracting from Vin. In this embodiment, the connections of the capacitive elements C3a and C3b are changed in accordance with the output of the coarse quantizer 114 during the residual amplification period. As a result, the output of the coarse quantizer 114 and the reference voltage Vref are reflected in the amplified signal (amplified voltage).

  Here, the reference voltage Vref is a reference voltage appearing in an equation (Equation (3)) and is an equivalent reference voltage. The reference voltages + Vref and −Vref corresponding to the equivalent reference voltage values are used in the coarse quantizer 114 when quantizing the positive phase input signal + Vin and the negative phase input signal −Vin. Although the reference voltages + Vref and −Vref used in the coarse quantizer 114 are voltage values corresponding to the equivalent reference voltage Vref, the configuration of the generation circuit thereof is not limited. The accuracy of the reference voltage generated by this generation circuit is not particularly required to be high because it is used for coarse quantization. Therefore, for example, a voltage difference between the power supply voltage Vdd and the ground voltage Vs may be generated and used as reference voltages + Vref and −Vref corresponding to the equivalent reference voltage Vref by resistance voltage division or capacitance voltage division.

Also, the digital value _{D i,} by connecting the second electrode of the capacitor C3a and C3b to the power supply voltage Vdd or the ground voltage Vs, is reflected in the amplifying operation. Therefore, a highly accurate reference voltage generation circuit is not required. In general, since the reference voltage of an analog-digital converter requires high accuracy, it is known that the power consumption of the reference voltage generation circuit is a bottleneck in reducing the power consumption of the analog-digital converter. Yes. Since a highly accurate reference voltage generating circuit is not required, it is possible to reduce power consumption.

In the configuration of the basic concept shown in FIG. 8, the residual amplification period, according to the value of the digital signal D _i, capacitance elements C3a and destination is the supply voltage of the second electrode P2 of C3b Vdd or the ground voltage Vs It becomes. When viewed in this light, the residual amplification period, the voltage division ratio of the power supply voltage Vdd by configured capacitive division by the capacitance elements C1, C2, C3a and C3b, vary according to the value of the digital signal D _i, further, It can also be seen that a reference voltage is equivalently generated by using a differential circuit configuration that performs complementary operations.

  According to the configuration of the basic concept shown in FIG. 8, since sampling is performed by fully differential sampling, it is possible to prevent an increase in the area occupied by the capacitor circuit. In addition, since a highly accurate reference voltage generation circuit is not required, an increase in power consumption can be suppressed.

  Hereinafter, a plurality of embodiments will be described. Here, a cyclic analog-digital converter will be described as an example of an analog-digital converter using a multiplying digital-analog conversion circuit.

(Embodiment 1)
FIG. 3A is a block diagram showing the configuration of the cyclic analog-digital converter according to the first embodiment, and FIG. 3B is a timing diagram showing the operation of this cyclic analog-digital converter. It is.

In FIG. 3A, reference numeral 300 denotes a multiplication type digital-analog conversion circuit (MDAC). The MDAC 300 will be described in detail later with reference to FIG. 1 or FIG. 2, but the input signal supplied to the input node Nin of the MDAC 300 is sequentially received in the same manner as the MDAC 901 described in FIG. The corresponding digital signal D _i (i = 1 to N) is converted. Also outputs residual amplified signal was obtained by amplifying the residual between the voltage of the voltage and the input signal corresponding to the digital signal D _i obtained in the conversion to the output node Nout. The residual amplified signal transmitted to the output node Nout of the MDAC 300 is supplied to the node 303b of the switch 303 via the buffer circuit 302. The node 303 c of the switch 303 is connected to the input node Nin of the MDAC 300, and the node 303 a of the switch 303 is connected to the output of the analog circuit 301.

  In the first embodiment, the analog circuit 301 includes a positive-phase input signal + Vin (hereinafter also referred to as VinP) corresponding to the input signal Vin and a negative-phase input signal −Vin having a reverse phase with respect to the positive-phase input signal + Vin. (Hereinafter also referred to as VinN) is supplied to the switch 303. In order to clearly show that the analog circuit 301 outputs the positive phase input signal + Vin and the negative phase input signal −Vin, the analog circuit 301 schematically shows an inverter circuit 304. Of course, the normal phase input signal + Vin and the negative phase input signal −Vin are not limited to being generated by an inverter circuit, and a normal inverting amplifier may be used. The differential outputs of the differential output type amplifier may be + Vin and −Vin, respectively.

  In the switch 303, when converting the positive phase input signal + Vin and the negative phase input signal −Vin corresponding to the input signal Vin (analog signal) into digital signals, that is, when performing analog-digital conversion, the node 303c is connected to the node 303a. Is done. Thereby, the voltage value of the input signal Vin (normal phase input signal + Vin, reverse phase input signal −Vin) at the time of analog-digital conversion is supplied to the input node Nin of the MDAC 300. On the other hand, in a period during which analog-digital conversion is performed, the node 303c of the switch 303 is connected to the node 303b. As a result, during the analog-to-digital conversion period, the residual amplified signal that is the output of the MDAC 300 is supplied to the input node Nin of the MDAC 300 via the buffer circuit 302 and the switch 303.

When the switch 303 shown in FIG. 3 (a) is moved downward, that is, when the node 303a and the node 303c are connected, the MDAC 300 converts as shown in FIG. 3 (b). Perform the action. By switch 303 is moved to the lower side, the input signal Vin (+ Vin, -Vin) supplied to the input node Nin is the sampling period 1S, sampled digital value _{D 1} is produced. Further, the residue amplification period 1A, the input signal Vin (+ Vin, -Vin) is the residual between the voltage corresponding to the digital value _{D 1} is amplified. Digital values D ₁ is output as a digital signal obtained by the conversion operation. On the other hand, the amplified residual amplified signal via a buffer circuit 302 and the switch 303 is supplied to the input node Nin, performed sampling in the next sampling period 2S, digital value D ₂ is output, residue amplification An amplification operation is performed in the period 2A.

  The above operation is repeated for the required number of bits of the digital signal (1S, 1A to NS, NA). Thereafter, the switch 303 is moved downward, an input signal having a new voltage value is transmitted to the input node Nin, and an analog-digital conversion operation is performed on the new input signal (1S, 1N˜). In FIG. 3B, analog-digital conversion is performed by a one-stage multiplication type digital-analog conversion circuit 300. Therefore, the conversion cycle of the multiplication type digital-analog conversion circuit 300 is from 1S to NA.

  Next, the multiplication type digital-analog conversion circuit 300 will be described with reference to FIG. FIG. 1 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit (MDAC) 300.

  The MDAC 300 includes capacitive circuits 100P and 100N, a coarse quantizer 114, a control unit 115, and voltage supply units 101P and 101N. Capacitance circuits 100P and 100N have the same configuration, and voltage supply units 101P and 101N also have the same configuration. Here, the capacity circuit 100P and the voltage supply circuit 101P are used to amplify the residual signal related to the positive phase input signal + Vin, and the capacity circuit 100N and the voltage supply circuit 101N are used to a residual signal related to the negative phase input signal −Vin. Is used to amplify. The residual amplified signal amplified by the capacitance circuit 100P and the voltage supply unit 101P is transmitted to the output node NoutP and transmitted to the input of the buffer circuit 17P. Similarly, the residual amplified signal amplified by the capacitance circuit 100N and the voltage supply unit 101N is transmitted to the output node NoutN and transmitted to the input of the buffer circuit 17N.

  For convenience of explanation, FIG. 1 shows buffer circuits 17P and 17N, but these buffer circuits 17P and 17N are collectively shown as a buffer circuit 302 in FIG. Further, the output nodes NoutP and NoutN shown in FIG. 1 are shown as an output node Nout in FIG. 3 by combining these two output nodes. Similarly, the input nodes NinP and NinN shown in FIG. 1 are shown as input nodes Nin in FIG.

  In FIG. 3, the switch 303 is shown as one switch, but the first switch (FIG. 3) supplies the positive phase input signal + Vin (VinP) from the analog circuit 304 or the buffer circuit 17P to the input node NinP. And a second switch (not shown) for supplying an anti-phase input signal −Vin (VinN) from the analog circuit 304 or the buffer circuit 17N to the input node NinN.

A coarse quantizer 114 is connected to input nodes NinP and NinN of the MDAC 300. The coarse quantizer 114 roughly quantizes the difference voltage between the positive phase input signal VinP and the negative phase input signal VinN supplied via the switch 303 (FIG. 3). In the first embodiment, quantized into three values, and outputs the data obtained by the quantization as a digital value D _i. The digital value D _i output from the coarse quantizer 114 is supplied to the control unit 115 and output as a result of analog-digital conversion. The MDAC 300 according to the first embodiment is a so-called 1.5-bit converter. Therefore, by performing arithmetic processing on a plurality of digital values D _i (i = 1 to N) obtained in one conversion cycle, the voltage of the input signal Vin that is an analog signal is a plurality of bits (binary expression). Converted to digital signal.

The control unit 115 receives the output (digital value D _i ) from the coarse quantizer 114, generates control signals p00, n00, p10, and n10 according to the output and supplies them to the voltage supply units 101P and 101N. . The voltage supply units 101P and 101N supply the voltage according to the control signal from the control unit 115 to the capacitance circuits 100P and 100N in the residual amplification period. Therefore, the control unit 115 and the voltage supply units 101P and 101N can be considered as a control circuit that controls the voltage supplied to the capacitor circuit.

The capacitance circuit 100P samples the positive phase input signal VinP and the negative phase input signal VinN in the sampling period, and calculates a voltage difference between the sampled positive phase input signal VinP and the negative phase input signal VinN as a residual amplification period. Amplify. Further, in the residual amplification period, the voltage obtained by the amplification reflects the output (digital value D _i ) of the coarse quantizer 114 and the equivalent reference voltage Vref. The amplified output reflecting the output of the coarse quantizer 114 and the equivalent reference voltage Vref appears at the output node NoutP and is transmitted to the input of the buffer circuit 17P. Similarly, the capacitor circuit 100N samples the negative-phase input signal VinN and the positive-phase input signal VinN in the sampling period, and stores a voltage difference between the sampled negative-phase input signal VinN and the positive-phase input signal VinP. Amplify during the difference amplification period. Further, in the residual amplification period, the voltage obtained by the amplification reflects the output (digital value D _i ) of the coarse quantizer 114 and the equivalent reference voltage Vref. The amplified output reflecting the output of the coarse quantizer 114 and the equivalent reference voltage Vref appears at the output node NoutN and is transmitted to the input of the buffer circuit 17N.

  The capacitive circuit 100P includes a switch 11P connected between the input node NinP and the output node NoutP, and capacitive elements 12P, 13P, 14P and 15P each having a first electrode P1 connected to the output node NoutP. Yes. The capacitive circuit 100P includes a switch row 16P connected between the second electrode P2 of each of the capacitive elements 12P to 15P and the input node NinN, and between the second electrode P2 of the capacitive element 12P and the power supply voltage Vdd. And a switch 19P connected between the second electrode P2 of the capacitive element 13P and the ground voltage Vs. Further, the capacitive circuit 100P includes a switch 111P connected between the second electrode P2 of the capacitive element 14P and the output node Niv1 of the voltage supply unit 101P, and an output of the second electrode P2 of the capacitive element 15P and the voltage supply unit 101P. And a switch 113P connected between the node Niv2.

  Here, the switch row 16P is a switch group including switches a to d. The switch a is connected between the input node NinN and the second electrode P2 of the capacitive element 12P, and the switch b is connected to the input node NinN. The capacitor 13P is connected between the second electrode P2. Similarly, the switch c is connected between the input node NinN and the second electrode P2 of the capacitive element 14P, and the switch d is connected between the input node NinN and the second electrode P2 of the capacitive element 15P. .

  The switch 11P and the switch row 16P are turned on in a sampling period (for example, 1S: FIG. 3), and are turned off in a residual amplification period (for example, 1A: FIG. 3). Thus, in the sampling period (1S), the first electrodes P1 of the capacitive elements 12P to 15P are connected to the input node NinP via the switch 11P, and the second electrodes P2 are connected to the switch row 16P (switch It is connected to the input node NinN via a to d). As a result, in the sampling period (1S), the positive phase input signal VinP and the negative phase input signal VinN are applied to each of the capacitive elements 12P to 15P, and the fully differential sampling is performed. Note that in the sampling period (1S), each of the switches 18P, 19P, 111P, and 113P is in an OFF state.

  On the other hand, in the residual amplification period (1A) following the sampling period (1S), each of the switch 11P and the switch row 16P is turned off, and each of the switches 18P, 19P, 111P, and 113P is turned on. Thereby, in the residual amplification period (1A), the power supply voltage Vdd is supplied to the second electrode P2 of the capacitive element 12P via the switch 18P, and the second electrode P2 of the capacitive element 13P is supplied to the second electrode P2 via the switch 19P. A ground voltage Vs is supplied. At this time, the voltages according to the output of the coarse quantizer 114 from the output nodes Niv1 and Niv2 of the voltage supply unit 101P are applied to the respective second electrodes P2 of the capacitive elements 14P and 15P by the switches 111P and 113P. Supplied through.

  The voltage supply unit 101P is not particularly limited, but includes inverter circuits 110P and 112P. The inverter circuit 110P receives the control signal p10 from the control unit 115, and supplies a voltage obtained by inverting the voltage of the control signal p10 to the output node Niv1 of the voltage supply unit 101P. The inverter circuit 112P receives the control signal n00 from the control unit 115 and supplies a voltage obtained by inverting the voltage of the control signal n00 to the output node Niv2 of the voltage supply unit 101P. Here, the power supply voltage Vdd and the ground voltage Vs are supplied to the voltage supply unit 101P as operating voltages. The power supply voltage Vdd is supplied via the power supply node Nvd, and the ground voltage Vs is supplied via the power supply node Nvs. That is, power supply voltage Vdd and ground voltage Vs are supplied to inverter circuits 110P and 112P as operating voltages via power supply nodes Nvd and Nvs, respectively.

  By receiving power supply voltage Vdd and ground voltage Vs as operating voltages, inverter circuits 110P and 112P are connected to respective second electrodes P2 of capacitive elements 14P and 15P and to power supply node Nvd according to the voltages of control signals p10 and n00. The supplied power supply voltage Vdd or the ground voltage Vs supplied to the power supply node Nvs is supplied. That is, in the first embodiment, the voltages supplied to the capacitor circuit 100P are the power supply voltage Vdd and the ground voltage Vs, and the high-precision reference voltage is not supplied. The first electrodes P1 of the capacitive elements 12P to 15P are connected to the output node NoutP in both the sampling period (1S) and the residual amplification period (1A).

  The capacity circuit 100N is configured similarly to the capacity circuit 100P. That is, the capacitance circuit 100N includes a switch 11N connected between the input node NinN and the output node NoutN to which the reverse-phase input signal VinN is supplied, and a capacitance in which the first electrode P1 is connected to the output node NoutN. Elements 12N to 15N, and a switch row 16N (switches a to d) connected between the input node NinP to which the positive phase input signal VinP is supplied and the second electrodes P2 of the capacitive elements 12N to 15N. It has. The capacitive circuit 100N includes a switch 18N connected between the second electrode P2 of the capacitive element 12N and the power supply voltage Vdd, and a switch connected between the second electrode P2 of the capacitive element 13N and the ground voltage Vs. 19N, a switch 111N connected between the second electrode P2 of the capacitive element 14N and the output node Niv1 of the voltage supply unit 101N, and a second electrode P2 of the capacitive element 15N and the output node Niv2 of the voltage supply unit 101N And a switch 113N connected therebetween.

  The switches 11N, 18N, 19N, 111N, 113N and the switch row 16N in the capacitive circuit 100N correspond to the switches 11P, 18P, 19P, 111P, 113P and the switch row 16P in the capacitive circuit 100P, respectively. That is, the switch 11N and the switch row 16N are turned on in the sampling period (1S) and turned off in the subsequent residual amplification period (1A). Accordingly, in the sampling period (1S), the voltage of the negative phase input signal VinN and the voltage of the positive phase input signal VinP are applied to the first electrode P1 and the second electrode P2 of each of the capacitive elements 12N to 15N. As a result, fully differential sampling is performed. On the other hand, in the residual amplification period (1A), each of the switches 18N, 19N, 111N, and 113N is turned on. Accordingly, in the residual amplification period (1A), the power supply voltage Vdd is supplied to the second electrode P2 of the capacitive element 12N, and the ground voltage Vs is supplied to the second electrode P2 of the capacitive element 13N.

  Further, in the residual amplification period (1A), the second electrodes P2 of the capacitive elements 14N and 15N follow the output of the coarse quantizer 114 from the output nodes Niv1 and Niv2 of the voltage supply unit 101N, respectively. A voltage will be supplied. Similarly to the voltage supply unit 101P described above, the voltage supply unit 101N is configured by inverter circuits 110N and 112N in the first embodiment, and each inverter circuit 110N and 112N includes a power supply node Nvd. The power supply voltage Vdd is supplied via the power supply, and the ground voltage Vs is supplied via the power supply node Nvs. Thereby, the inverter circuit 110N supplies the power supply voltage Vdd or the ground voltage Vs obtained by inverting the voltage of the control signal p00 to the output node Niv1 in the residual amplification period (1A). Similarly, inverter circuit 112N supplies power supply voltage Vdd or ground voltage Vs obtained by inverting the voltage of control signal n10 to output node Niv2. As can be understood from the drawing, also in the capacitive circuit 100N, the second electrodes P2 of the capacitive elements 12N to 15N are connected to the output node NoutN.

  The voltages at the output nodes NoutP and NoutN of the MDAC 300 are buffered by the buffer circuits 17P and 17N, and supplied to the switch 303 (FIG. 3) as the output signals VoutP and VoutN of the buffer circuits 17P and 17N, respectively. During the conversion period, that is, during a period in which the node 303b and the node 303c of the switch 303 are connected, the output signals VoutP and VoutN from the buffer circuits 17P and 17N are input to the MDAC 300 for the next conversion operation. The node Nin (NinP, NinN) is supplied as the normal phase input signal VinP and the negative phase input signal VinN.

  In FIG. 1, reference numeral 103 denotes a reference voltage generation circuit. The reference voltage generation circuit 103 includes, but is not limited to, a voltage-dividing resistor element or capacitor element, receives the power supply voltage Vdd and the ground voltage Vs, and receives a difference voltage between the power supply voltage Vdd and the ground voltage Vs. Is supplied to the coarse quantizer 114 as a reference voltage Vref.

  In the first embodiment, each of capacitive elements 12P and 12N corresponds to capacitive element C1 described in FIG. 8, and each of capacitive elements 13P and 13N corresponds to capacitive element C2 in FIG. Furthermore, each of capacitive elements 14P and 14N in the first embodiment corresponds to capacitive element C3a in FIG. 8, and each of capacitive elements 15P and 15N corresponds to capacitive element C3b in FIG. Therefore, the first embodiment can be regarded as applying the basic concept described above with reference to FIG. 8 to each conversion of the normal phase input signal VinP and the negative phase input signal VinN. The differential voltage (voltage at output node NoutP−voltage at output node NoutN) output from MDAC 300 corresponds to the differential voltage (VoutP−VoutN) between output signal VoutP of buffer circuit 17P and output signal VouN of buffer circuit 17N.

  Next, the operation of the MDAC 300 shown in FIG. 1 will be described. The input signal Vin is an analog signal, and the voltage of the positive phase input signal VinP changes corresponding to the voltage change of the input signal Vin. The voltage waveform of the negative phase input signal VinN has a negative phase with respect to the positive phase input signal VinP. Here, it is assumed that the negative phase input signal VinN has a voltage waveform having a phase opposite to that of the positive phase input signal VinP with the common voltage as the center.

  First, the normal phase input signal VinP and the negative phase input signal VinN corresponding to the input signal Vin are supplied to the input node Nin (NinP and NinN in FIG. 1) of the MDAC 300 via the switch 303 shown in FIG. . In the sampling period (1S: FIG. 3), the switches 11P and 11N and the switch trains 16P and 16N are in the on state. Thereby, the voltage of the positive phase input signal VinP transmitted via the switch 303 is applied to the first electrodes P1 of the capacitive elements 12P to 15P and the second electrodes P2 of the capacitive elements 12N to 15N. At this time, the voltage of the negative phase input signal VinN transmitted through the switch 303 is applied to the second electrodes P2 of the capacitive elements 12P to 15P and the first electrodes P1 of the capacitive elements 12N to 15N. Applied. Thereby, each of the capacitive elements 12P to 15P and 12N to 15N is charged by the voltage applied to the first electrode P1 and the voltage applied to the second electrode P2. Since the voltage of the differential input signal (VinP, VinN) is applied to the first electrode P1 and the second electrode P2 of each capacitive element, complete differential sampling is performed.

  On the other hand, in the sampling period (1S), the normal phase input signal VinP and the negative phase input signal VinN are supplied to the coarse quantizer 114 via the switch 303 (FIG. 3). The coarse quantizer 114 roughly quantizes the input signal. In the first embodiment, the coarse quantizer 114 is a voltage difference between the voltage of the supplied positive phase input signal VinP and the voltage of the negative phase input signal VinN, that is, the voltage of the positive phase input signal VinP. -The difference voltage (VinP-VinN) which is the voltage of the negative phase input signal is quantized based on the comparison with the reference voltage Vref. FIG. 12 is a waveform diagram showing the relationship between the reference voltage Vref and the difference voltage (VinP−VinN).

  In FIG. 12, the horizontal axis indicates time, and the vertical axis indicates voltage. The reference voltage Vref has a positive reference voltage + Vref and a negative reference voltage −Vref centered on 0. In this case, the absolute value voltage of the reference voltage + Vref is the same as the absolute value voltage of the reference voltage −Vref. Such reference voltages + Vref and -Vref can be easily generated by dividing the voltage between the ground voltage Vs and the power supply voltage Vdd as described above. Further, the reference voltages + Vref and −Vref are reference voltages for performing rough quantization, and high accuracy is not required. In FIG. 1, a reference voltage generating circuit 103 is shown as a circuit for generating the reference voltages + Vref and −Vref. In FIG. 1, these reference voltages + Vref and −Vref are collectively shown as the reference voltage Vref. is there.

  The voltage waveform of the normal phase input signal VinP and the voltage waveform of the negative phase input signal VinN change symmetrically around the common voltage. Therefore, the difference voltage (VinP−VinN) between the input differential signals VinP and VinN increases as the voltage difference between these input voltages and the common voltage increases. FIG. 12 schematically shows a voltage waveform of the differential voltage (VinP−VinN) as an example. In the example of the differential voltage shown in FIG. 12, the normal phase input signal VinP has a positive voltage with respect to the common voltage, and the negative phase input signal VinN has a negative voltage with respect to the common voltage. The case where the normal phase input signal VinP changes to a negative voltage with respect to the common voltage and the negative phase input signal VinN changes to a positive voltage with respect to the common voltage is shown. Due to such voltage changes of the positive phase input signal VinP and the negative phase input signal VinN, the voltage value of the differential voltage (VinP−VinN) changes from the plus side to the minus side with the passage of time. In the figure, the plus side indicates the reference voltage + Vref side, and the minus side indicates the reference voltage -Vref side.

In the first embodiment, although not particularly limited, the input differential signal is quantized using a voltage that is 1/4 of the reference voltage Vref (+ Vref, −Vref) as a threshold voltage. That is, the differential voltage (VinP−VinN) of the input differential signal is (a) greater than the reference voltage Vref / 4, or (b) exists between the reference voltage Vref / 4 and the reference voltage −Vref / 4. Or (c) whether the reference voltage is smaller than Vref / 4, and the input differential signal is quantized. By making this determination, the coarse quantizer 114, the difference between the voltage of the input differential signals, for example when the (a), outputs "1" as digital signals D _i, the difference between the voltage of the input differential signals when for example the (b), outputs "0" as a digital signal _{D i,} the difference between the voltage of the input differential signals, for example, when the (c), outputs "-1" as the digital signal _{D i.}

Referring to FIG. 12 as an example, when the analog-to-digital converter captures the input signals VinP and VinN at time t0, the coarse quantizer 114 determines that the difference voltage of the input differential signal is the reference voltage Vref / 4. Therefore, it is determined as (a), and the digital signal D _i = 1 is output. When the input signals VinP and VinN at time t1 are captured, the difference voltage of the input differential signal exists between the reference voltage + Vref / 4 and −Vref / 4. The signal D _i = 0 is output. Similarly, when the input signals VinP and VinN at time t2 are captured, since the difference voltage is smaller than the reference voltage −Vref / 4, the coarse quantizer 114 outputs the digital signal D _i = −1. .

Although not particularly limited, when the digital signal D _i of the coarse quantizer 114 is converted into a binary digital value, the digital signal D _i = 1 corresponds to the binary digital value “10” The signal D _i = 0 corresponds to “01”, and the digital signal D _i = −1 corresponds to “00”. Thereby, so-called 1.5-bit conversion is performed.

Control unit 115 receives the digital signal _{D i} is the output from the coarse quantizer 114, a control signal in accordance with the value of the digital value signal _{D i} p10, n10, p00, to generate the n00. In this example, the control signal p10, when the digital signal D _i is "1", the high level (power supply voltage Vdd), and the when otherwise, so that the low level (ground voltage Vs), are produced. Further, the control signal n10, when the digital signal _{D i} is "1", a low level (Vs), and the at other times, such that a high level (Vdd), is generated. Control signal p00, when the digital signal _{D i} is "-1", a high level (Vdd) becomes, when otherwise, so that a low level (Vs), are produced. Control signal n00, when the digital signal _{D i} is "-1", a low level (Vs), and the at other times, such that a high level (Vdd), is generated.

  In the residual amplification period (1A) following the sampling period (1S), the switches 11P and 11N are turned off. Further, the switch trains 16P and 16N are also turned off. On the other hand, in the residual amplification period, each of the switches 18P, 19P, 111P, 113P, 18N, 19N, 111N, and 113N is changed from the off state to the on state. Thereby, in the residual amplification period, the second electrodes P2 of the capacitive elements 12P and 12N are connected to the power supply voltage Vdd, and the second electrodes P2 of the capacitive elements 13P and 13N are connected to the ground voltage Vs. .

In the residual amplification period, the voltages of the second electrodes P2 of the capacitive elements 14P, 15P, 14N, and 15N are determined by the voltage supply units 101P and 101N. Since the voltages output from the voltage supply units 101P and 101N are determined by the control signal from the control unit 115, the voltages of the second electrodes P2 of the capacitive elements 14P, 15P, 14N, and 15N are determined by the coarse quantizer 114. Is determined by the output from the digital signal (digital signal D _i ).

That is, when the digital signal _{D i} is "1", the control signal p10, n00 becomes high level, from each of the inverter circuits 110P and 112P are ground voltage corresponding to a ground voltage Vs which is fed to the power supply node Nvs Are supplied to the second electrodes P2 of the capacitive elements 14P and 15P. At this time, since the control signals p00 and n10 are at the low level, the power supply voltage corresponding to the power supply voltage Vdd supplied to the power supply node Nvd is supplied from each of the inverter circuits 110N and 112N to each of the capacitive elements 14N and 15N. It is supplied to the second electrode P2. Thereby, the connection state of the capacitive elements 12P to 15P is the same as the state (a) shown in FIG. 8, and the connection state of the capacitive elements 12N to 15N is the same as the state (c) shown in FIG. . As described above, the capacitive elements 12P and 12N correspond to the capacitive element C1 in FIG. 8, and the capacitive elements 13P and 13N correspond to the capacitive element C2 in FIG. The capacitive elements 14P and 14N correspond to the capacitive element C3a in FIG. 8, and the capacitive elements 15P and 15N correspond to the capacitive element C3b in FIG.

Further, the power supply when the digital signal _{D i} is "-1", since the control signals P10, n00 is low, from each of the inverter circuits 110P and 112P, in response to the power supply voltage Vdd is powered to power supply node Nvd The voltage is supplied to the second electrodes P2 of the capacitive elements 14P and 15P. At this time, since the control signals p00 and n10 are at a high level, the ground voltages corresponding to the ground voltage Vs fed to the power supply node Nvs are respectively supplied from the inverter circuits 110N and 112N to the capacitive elements 14N and 15N. To the second electrode P2. Thereby, the connection state of the capacitive elements 12P to 15P is the same as the state (c) shown in FIG. 8, and the connection state of the capacitive elements 12N to 15N is the same as the state (a) shown in FIG. .

Further, when the digital signal _{D i} is "0", the control signal P10 is low, since the control signal n00 becomes high level, the inverter circuit 110P, power supply voltage according to the power supply voltage Vdd, a capacitor element 14P The ground voltage corresponding to the ground voltage Vs is supplied from the inverter circuit 112P to the second electrode P2 of the capacitive element 15P. At this time, since the control signal p00 is at the low level and the control signal n10 is at the high level, the power supply voltage corresponding to the power supply voltage Vdd is supplied from the inverter circuit 110N to the second electrode P2 of the capacitive element 14N, and the inverter circuit 112N The ground voltage corresponding to the ground voltage Vs is supplied to the second electrode P2 of the capacitive element 15N. Thereby, the connection state of the capacitive elements 12P to 15P is the same as the state (b) shown in FIG. 8, and the connection state of the capacitive elements 12N to 15N is the same as the state (b) shown in FIG. .

In the residual amplification period, the connection state of the capacitor 12P~15P (12N~15N) in each of the capacitor circuits 100P and 100N, as described above, in accordance with the output of the digital signal _{D i,} similarly to the state shown in FIG. 8 Changes to. Therefore, the output voltage VoutP corresponding to the voltage at the output node NoutP (corresponding to the output node of the MDAC 300) of the capacitor circuit 100P and the output voltage VoutN corresponding to the voltage at the output node NoutN (corresponding to the output node of the MDAC 300) of the capacitor circuit 100N. The difference voltage Vout (VoutP−VoutN) is in accordance with the above equation (2). In this case, the output voltage Vout in the expression (2) is read as a difference voltage between VoutP and VoutN, the input voltage Vin is read as a difference voltage between VinP and VinN, and the reference voltage Vref is read as + Vref or −Vref.

  In the first embodiment, the output nodes NoutP and NoutN are connected to the inputs of the buffer circuits 17P and 17N. Since buffering is performed by the buffer circuits 17P and 17N, the output nodes of the buffer circuits 17P and 17N and the output nodes NoutP and NoutN are electrically separated. As a result, when the output voltage at each of the output nodes NoutP and NoutN is transmitted to the output nodes of the buffer circuits 17P and 17N, the capacitance existing at the output nodes of the buffer circuits 17P and 17N (the sampling capacitance and the parasitic capacitance of the next stage). It is possible to prevent charge dispersion from being performed between the sum of the capacitance) and the capacitive elements 12P to 15P and 12N to 15N included in the capacitive circuits 100P and 100N. In addition, since the input impedance of the buffer circuits 17P and 17N is high, it is possible to prevent the output voltage values at the output nodes NoutN and NoutP from being changed (destroyed), and the difference given to the inputs of the buffer circuits 17P and 17N. The voltage value can be maintained.

  Capacitance elements 12P to 15P and 12N to 15N included in the capacitance circuits 100P and 100N hold the charge corresponding to the supplied input signal Vin (+ Vin and −Vin) in the sampling period, and thus the switch 303 (FIG. 3). And the MDAC 300 may not include a hold circuit for holding an input signal. Of course, it is needless to say that a hold circuit may be provided in order to allow a deviation between the sampling timings of the capacitance circuits 100P and 100N and the determination timing of the coarse quantizer 114.

  According to the first embodiment, the output signal (voltage) at the output node Nout (NoutP, NoutN) of the MDAC 300 illustrated in FIG. 1 is returned to FIG. 1 via the buffer circuit 302 (17P, 17N) and the switch 303. It is supplied to the input node Nin (NinP, NinN) of the MDAC 300 shown. The supplied output signal is subjected to the above-described coarse quantization and complete differential sampling as the input signal Vin (VinP, VinN) in the sampling period (2S), and further in the residual amplification period (2A) The residual is amplified. In this way, by repeating the sampling period and the subsequent residual amplification period, the input signal Vin (VinP, VinN) is converted into a digital signal having a predetermined number of bits. Note that the process of converting quantized data obtained by so-called 1.5-bit conversion into a binary digital signal is well known, and is omitted here.

  Although not shown in the figure, the cyclic analog-digital converter includes the switches 11P, 18P, 19P, 111P, 113P, 11N, 18N, 19N, 111N, 113N, the switch 303 (FIG. 3), and the switch. A control circuit for controlling the columns 16P and 16N is provided. By this control circuit, a predetermined switch and a switch string are controlled to be turned on / off during the sampling period and the residual amplification period. The switch 303 is controlled so that an input signal is supplied to the MDAC 300 at a predetermined timing. Further, the coarse quantizer 114 and the control unit 115 are also controlled by this control circuit. The coarse quantizer 114 is controlled to perform quantization in the sampling period, and the control unit 115 performs residual amplification on the control signals p10, n00, p00, and n10 according to the output of the coarse quantizer 114. Control is performed so that the voltage supply units 101P and 101N are supplied during the period.

  Next, the power supply voltage Vdd, the reference voltage Vref (+ Vref, −Vref), the input common (average) voltage Vcm of the buffer circuits 17P and 17N, and the capacitive elements 12P to 15P (12N to 15N) included in the capacitive circuit 100P (100N) ) Shows the relationship with each capacitance value. Note that the capacitive element 12P (12N) is a capacitive element C1, the capacitive element 13P (13N) is a capacitive element C2, and the capacitive elements 14P (14N) and 15P (15N) are each shown as a capacitive element C3.

  First, the relationship between the reference voltage Vref and the capacitive elements C1 to C3 (12P to 15P, 12N to 15N) has already been shown in the equation (3). The relationship between the input common voltage Vcm of the buffer circuits 17P and 17N and the capacitive elements C1 to C3 is as shown in Expression (4). The relationship with the voltage Vcm is as shown in Equation (5). Further, the relationship among the capacitive elements C2, C3, the reference voltage Vref, and the input common voltage Vcm of the buffer circuits 17P, 17N is as shown in Expression (6).

  For example, by determining the common voltage Vcm, the reference voltage Vref, and the power supply voltage Vdd at the inputs of the buffer circuits 17P and 17N, the capacitance ratio of the capacitive elements C1 to C3 included in the capacitive circuit is expressed by the above equations (5) and (6 ).

  According to the first embodiment, complete differential sampling is performed on the capacitive element included in the capacitive circuit during the sampling period. As a result, the capacity element can be reduced in size while maintaining the amount of charge accumulated by sampling, and an increase in the area occupied by the multiplying digital-to-analog converter circuit can be suppressed. As a result, it is possible to suppress an increase in the area occupied by the cyclic analog-digital converter using the multiplication digital-analog converter circuit. Further, since the power supply voltage Vdd and the ground voltage Vs can be used as the voltage supplied to the capacitor circuit in the residual amplification period without using the high-precision reference voltage, the high-precision reference voltage is used. It is not necessary to provide a reference voltage generation circuit for generation, and it is possible to suppress an increase in power consumption. In addition, since the multiplication type digital-analog conversion circuit is configured by a capacitive circuit (passive circuit) that is a passive circuit, it is possible to suppress an increase in power consumption in the multiplication type digital-analog conversion circuit.

  Furthermore, according to the first embodiment, the output of the capacitor circuit is buffered by the buffer circuit, so that the charge accumulated in the capacitor circuit can be prevented from being destroyed, and cyclic analog-digital conversion is performed. It is possible to further suppress an increase in the area occupied by the vessel.

(Embodiment 2)
FIG. 2 is a circuit diagram showing the configuration of the multiplication type digital-analog conversion circuit 300 and the buffer circuit according to the second embodiment. The MDAC 300 shown in FIG. 2 has a configuration similar to that of the MDAC 300 shown in FIG. Therefore, here, only the parts different from the first embodiment will be mainly described, and description of the configuration and operation of the MDAC 300 will be omitted.

  The main difference between the first embodiment and the second embodiment is that the buffer circuit connected to the output node Nout (NoutP, NoutN) of the MDAC 300 differs between the first embodiment and the second embodiment. ing. In the second embodiment, each of the buffer circuits 17P and 17N used in the first embodiment is configured by a source follower circuit. That is, the source follower circuit 200P is used instead of the buffer circuit 17P, and the source follower circuit 200N is used instead of the buffer circuit 17N.

  The source follower circuit 200P connected to the output node NoutP of the capacitive circuit 100P that performs sampling and residual amplification of the positive phase input signal VinP is composed of an input field effect transistor (hereinafter referred to as MOSFET) 21P and a current source. MOSFET 22P. Similarly, the source follower circuit 200N connected to the output node NoutN of the capacitance circuit 100N that performs sampling and residual amplification of the negative phase input signal VinN includes an input MOSFET 21N and a current source MOSFET 22N. These MOSFETs 21P, 22P, 21N and 22N are N-channel type MOSFETs in this embodiment. The MDAC 300 shown in FIG. 2 is not particularly limited, but is formed in one semiconductor device by a known semiconductor manufacturing process.

  The input MOSFET 21P in the source follower circuit 200P has its gate electrode connected to the corresponding output node NoutP, its drain connected to the power supply voltage Vdd, and its source S connected to the drain of the current source MOSFET 22P. The source of the current source MOSFET 22P is connected to the ground voltage Vs, and a predetermined bias voltage Vb is supplied to its gate. The output signal VoutP of the source follower circuit 200P is taken out from the source S of the input MOSFET 21P. The back gates B of the input MOSFET 21P and the current source MOSFET 22P are connected to the sources of the corresponding MOSFETs.

  Similarly to the source follower circuit 200P, the input MOSFET 21N in the source follower circuit 200N has its gate electrode connected to the corresponding output node NoutN, its drain connected to the power supply voltage Vdd, and its source S connected to the current source MOSFET 22N. Connected to the drain. The source of the current source MOSFET 22N is connected to the ground voltage Vs, and a predetermined bias voltage Vb is supplied to its gate. The output signal VoutN of the source follower circuit 200N is also taken out from the source S of the input MOSFET 21N. The back gates B of the input MOSFET 21N and the current source MOSFET 22N are directly connected to the sources of the corresponding MOSFETs.

  The gate widths of the input MOSFETs 21P and 21N and the currents of the current source MOSFETs 22P and 22N so that the transient response waveforms of the output signals VoutP and VoutN of the source follower circuits 200P and 200N converge within the residual amplification period. The value is adjusted. In this case, the current values of the current source MOSFETs 22P and 22N can be adjusted by adjusting the value of the bias voltage Vb. In this way, by making the waveforms of the respective output voltages converge within the residual amplification period, it is possible to accelerate the timing for starting the next sampling.

  Further, the substrate bias effect is reduced in the input MOSFET 21P (21N) by directly connecting the back gate B of the input MOSFET 21P (21N) to the source S of the MOSFET 21P (21N). Thereby, the voltage gain of the source follower circuit 200P (200N) can be brought close to 1.

  As described above, the N-channel MOSFET is used as the input MOSFET 21P (21N), and in order to connect the back gate to the source, the P-channel well for forming the input MOSFET 21P (21N) is used as the semiconductor of the semiconductor device. It may be required to form on a substrate. Therefore, it is conceivable to connect the back gate B to the ground voltage Vs instead of the source S. In this case, a substrate bias effect is generated in the input MOSFET 21P (21N), and the voltage gain of the source follower circuit 200P (200N) is smaller than 1. However, the influence is compensated for by a generally known digital correction technique. Of course, this may be realized.

  It is also conceivable to configure the source follower circuits 200P and 200N using P-channel MOSFETs instead of N-channel MOSFETs. The P-channel MOSFET has a low noise characteristic although the response speed is inferior to that of the N-channel MOSFET. Further, when the P-channel type MOSFET is an input MOSFET, it can be formed on the N-channel type well using a well-known semiconductor manufacturing process, so that the back gate and the source can be directly connected. .

  As shown in the second embodiment, by using the source follower circuits 200P and 200N as a buffer circuit, the low power consumption and high-speed transient response of the source follower circuit can be used. As a result, the cyclic type It is possible to reduce the power consumption and improve the speed of the analog-digital converter. Also in the second embodiment, since the gate and the source of the input MOSFET 21P (21N) are electrically separated, the charge held in the capacitor circuit is destroyed by the charge dispersion. It is possible to prevent.

  Note that the power supply nodes Nvd and Nvs are omitted in FIG. 2 in order to avoid the complexity of the figure.

(Embodiment 3)
FIG. 4A is a block diagram showing a configuration of the cyclic analog-digital converter 402 according to the third embodiment. FIG. 4B is a timing chart showing the operation of the cyclic analog-digital converter 402.

  In FIG. 4A, reference numerals 400a and 400b denote multiplication type digital-analog conversion circuits. The MDAC 300 shown in FIG. 1 or 2 is used for each of the multiplication type digital-analog conversion circuits 400a and 400b. In FIG. 3, reference numeral 301 denotes an analog circuit having the same configuration as the analog circuit schematically shown in FIG. Reference numeral 303 denotes a switch having a configuration similar to that of the switch shown in FIG.

  In the first embodiment described with reference to FIGS. 1 and 3, the cyclic analog-digital converter is configured by one stage of MDAC 300. On the other hand, in the third embodiment, MDACs 400a and 400b that perform the same configuration and the same operation as the MDAC 300 shown in FIG. 1 or FIG. 402 is configured.

  Since the configurations and operations of the MDACs 400a and 400b are the same as those of the MDAC 300 described in the first embodiment or the second embodiment, detailed description thereof is omitted here.

  The input node Nin of the MDAC 400a is connected to the switch 303. When the input signals VinP and VinN, which are analog signals output from the analog circuit 301, are converted into digital signals, the switch 303 takes the input signals VinP and VinN into the input node Nin of the MDAC 400a. During the period in which the input signals VinP and VinN thus captured are converted into digital signals, the node 303C and the node 303b of the switch 303 are connected. As a result, during the period of conversion to the digital signal, signals relating to the input signals VinP and VinN circulate in a closed loop including the MDAC 400a, the buffer circuit 401a, the MDAC 400b, the buffer circuit 401b, and the switch 303. That is, from the viewpoint of processing signals related to the input signals VinP and VinN, these circuits and the switch 303 are connected in series. Here, the buffer circuits 401a and 401b are the buffer circuit or the source follower circuit shown in FIGS.

  FIG. 4B shows the operation timing of the MDACs 400a and 400b. In FIG. 4B, the operation timing of the MDAC 400a is shown on the upper side, and the operation timing of the MDAC 400b is shown on the lower side. Each of the MDACs 400a and 400b operates in the sampling period and the subsequent residual amplification period as described in FIGS. That is, coarse quantization and fully differential sampling are performed during the sampling period, and residual amplification is performed during the residual amplification period.

  First, in the sampling period 1S, the MDAC 400a performs fully differential sampling of the input signals VinP and VinN via the switch 303, and performs coarse quantization in this period. In the residual amplification period 1A following the sampling period 1S, the MDAC 400a performs the residual amplification operation. The amplified residual is transmitted from the output node Nout to the input node Nin of the MDAC 400b through the buffer 401a. Even in the residual amplification period 1A, the output of the MDAC 400a is transmitted to the input node Nin of the MDAC 400b via the buffer circuit 401a. To start. The MDAC 400b performs full differential sampling and coarse quantization in the sampling period 2S, and performs residual amplification in the subsequent residual amplification period 2A. From the time of the residual amplification period 2A, the output of the MDAC 400b is transmitted to the input node Nin of the MDAC 400a through the buffer circuit 401b and the switch 303. As a result, the residual amplification period 2A of the MDAC 400b and the sampling period 3S of the MDAC 400a overlap. Thereafter, processing is performed so that the sampling period and the residual period overlap until an N-bit digital signal is obtained. That is, processing is executed in a pipeline manner. Thereby, in the converted digital signal, odd-numbered bits counted from the most significant bit are converted by MDAC 400a, and even-numbered bits are converted by MDAC 400b. In this embodiment, there is an advantage that the conversion rate can be doubled by operating the multiplication type digital-analog conversion circuit in two stages.

  In addition, the buffer circuits 401a and 401b electrically isolate the input node Nin and the output node Nout between the two stages of MDACs 400a and 400b. For example, the output node Nout of the MDAC 400a and the input node Nin of the MDAC 400b are electrically separated by the buffer circuit 401a. Thus, even when the residual amplification period and the sampling period overlap, it is possible to prevent charge dispersion from occurring between the capacitor circuit of the MDAC 400a and the capacitor circuit of the MDAC 400b. For example, in the residual amplification period 1A, even if the charge held in the capacitor circuit of the MDAC 400a overlaps with the sampling period 2S, charge sharing with the capacitor circuit of the MDAC 400b is not performed. As a result, the MDAC 400a can maintain an accurate voltage during the residual amplification period 1A, thereby suppressing a decrease in accuracy and improving the conversion rate.

(Embodiment 4)
FIG. 5 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit 500 according to the fourth embodiment. In the fourth embodiment, a multiplying digital-to-analog converter circuit having a higher residual amplification factor G is provided as compared with the MDAC 300 described in the first embodiment or the second embodiment. In the first or second embodiment, the residual amplification factor G is almost double, but according to the fourth embodiment, a multiplication type digital-analog conversion circuit having a residual amplification factor G of about four times is provided. Provided. The MDAC 500 shown in FIG. 5 is used for the cyclic analog-digital converter shown in FIG. 3 or FIG. That is, the MDAC 500 is used as the MDAC 300 in the case of FIG. 3, and is used as the MDACs 401a and 401b in the case of FIG.

  In FIG. 5, MDAC 500 generates control signals PC1 to PC6 and NC1 to NC6 according to the coarse quantizer 510 that roughly quantizes the normal phase input signal VinP and the negative phase input signal VinN, and the output of the coarse quantizer 510. And a control unit 511. Further, the MDAC 500 samples the positive phase input signal VinP and the negative phase input signal VinN and performs amplification based on the capacitance circuits 501P and 501N that perform amplification, and the control signals PC1 to PC6 and NC1 to NC6 from the control unit 511. Voltage supply units 502P and 502N that determine voltages supplied to 501P and 501N are provided. Although not particularly limited, also in the fourth embodiment, the output nodes NoutP and NoutN of the MDAC 500 are connected to the inputs of the buffer circuits 17P and 17N. In FIG. 5, these buffer circuits 17P and 17N are also shown together with the MDAC 500.

  Since the capacitance circuits 501P and 501N have the same configuration, the capacitance circuit 501P will be described in detail first. The capacitor circuit 501P has two capacitor banks in order to obtain a residual amplification factor G of 2 times or more (in this embodiment, a residual amplification factor G of about 4 times). In order to avoid complication of the drawing in the figure, one of the two capacity banks is designated as a first capacity bank BK1, and the other capacity bank is designated as a second capacity bank. This will be described as BK2.

  The first capacitor bank BK1 includes a plurality of capacitor elements each having the first electrode P1 connected to the output node NoutP, and the second capacitor bank BK2 includes the first capacitor P1 via the switch. It includes a plurality of capacitive elements connected to the second electrode P2 of the corresponding capacitive element in the bank BK1. Referring to FIG. 5, the first capacitor bank BK1 includes capacitive elements 52P, 61P, 62P, 63P, 64P, 65P, and 66P in which the first electrode P1 is connected to the output node NoutP. In addition, the second capacitor bank BK2 is connected to each of the second capacitor elements 52P, 61P, 62P, 63P, 64P, 65P and 66P via the switches 41P, 42P, 43P, 44P, 45P, 46P and 47P, respectively. The electrode P2 includes capacitive elements 53P, 71P, 72P, 73P, 74P, 75P and 76P to which the first electrode P1 is connected.

  The first electrodes P1 of the capacitive elements 52P, 61P, 62P, 63P, 64P, 65P and 66P included in the first capacitive bank BK1 are connected to the input node NinP via the switch 31P, and the capacitive elements 52P, 61P, Each of the second electrodes P2 of 62P, 63P, 64P, 65P and 66P is connected to the input node NinN via a switch row 39P configured by a plurality of switch groups. The second electrodes P2 of the capacitive elements 53P, 71P, 72P, 73P, 74P, 75P, and 76P included in the second capacitive bank BK2 are connected to the switch row 39P formed by the plurality of switch groups described above. ing. As can be seen from the figure, the second electrodes P2 of the capacitive elements included in the first capacitor bank BK1 and the second electrodes of the capacitive elements included in the second capacitor bank BK2 are included in the switch row 39. In the switch group, the switches are connected to the input node NinN via different switches.

  The first electrodes of the capacitive elements 53P, 71P, 72P, 73P, 74P, 75P and 76P included in the second capacitive bank BK2 are connected via the switches 32P, 33P, 34P, 35P, 36P, 37P and 38P. It is connected to the input node NinP. Further, the second electrode P2 of the capacitive element 53P in the second capacitive bank BK2 is connected to the power supply voltage Vdd via the switch 82P, and the second electrodes P2 of the capacitive elements 71P, 72P, 73P, 74P, 75P and 76P, respectively. Are connected to the output of the voltage supply unit 502P via switches 83P, 84P, 85P, 86P, 87P and 88P, respectively.

  Further, the capacitance circuit 501P is connected to the input node NinN via the first electrode P1 connected to the output node NoutP, the switch in the switch row 39P, and to the power supply voltage Vdd via the switch 81P. A capacitive element 51P having an electrode P2 is provided.

  In this fourth embodiment, voltage supply unit 501P has a plurality of inverter circuits 91P, 92P, 93P, 94P, 95P, and 96P, and each inverter circuit is connected to power supply nodes Nvd and Nvs, The power supply nodes Nvd and Nvs operate with the power supply voltage Vdd and the ground voltage Vs supplied from the power supply nodes Nvd and Nvs as operating voltages. That is, each of the inverter circuits 91P to 96P operates with a common power supply voltage.

  Control signals PC1 to PC6 from the control unit 511 are input to the inverter circuits 91P to 96P. Thereby, the inverter circuit 91P supplies a voltage inverted with respect to the voltage of the control signal PC1 to the switch 83P. In this case, the inverter circuit 91P supplies the power supply voltage Vdd or Vs supplied to the power supply node Nvd or Nvs to the switch 83P. Similarly, the inverter circuit 92P supplies the voltage of the signal obtained by inverting the control signal PC2 to the switch 84P, and the inverter circuit 93P supplies the voltage of the signal obtained by inverting the control signal PC3 to the switch 85P. The inverter circuit 94P supplies the voltage of the signal obtained by inverting the control signal PC4 to the switch 86P, the inverter circuit 95P supplies the voltage of the signal obtained by inverting the control signal PC5 to the switch 87P, and the inverter circuit 96P The voltage of the signal obtained by inverting the signal PC6 is supplied to the switch 88P.

  As in the first or second embodiment, the input node NinP is supplied with the normal phase input signal VinP, and the input node NinN is supplied with the negative phase input signal VinN. In the fourth embodiment, input nodes NinP and NinN are input nodes of MDAC 500 and are also input nodes of capacitive circuits 501P and 501N. The output node NoutP is an output node of the MDAC 500 and also an output node of the capacitor circuit 501P. The output node NoutN is an output node of the MDAC 500 and an output node of the capacitor circuit 501N.

  Output nodes NoutP and NoutN are connected to the inputs of buffer circuits 17P and 17N. The buffer circuits 17P and 17N correspond to the buffer circuit 302 in FIG. 3, and correspond to the buffer circuit 401a or 401b in FIG. The positive-phase input signal VinP is supplied to the input node NinP, and the negative-phase input signal VinN is supplied to the input node NinN, so that the residual corresponding to the positive-phase input signal VinP is amplified from the output node NoutP. A difference amplified signal is output, and a residual amplified signal corresponding to the negative phase input signal VinN is output from the output node NoutN. As a result, a differential signal of the residual amplified signal is output between the output nodes NoutP and NoutN. This differential signal is supplied to the switch 303 via the buffer circuits 17P and 17N (for example, 302 in FIG. 3). Thereafter, in the example of FIG. 3, the feedback is made to the input nodes NinP and NinN of the MDAC 500.

  Similarly to the first capacitor circuit 501P, the second capacitor circuit 501N also includes a first capacitor bank BK1 and a second capacitor bank BK2, although not shown. The first capacitor bank BK1 includes capacitor elements 52N and 61N to 66N whose first electrodes are connected to the output node NoutN, and the second capacitor bank BK2 includes capacitor elements 52N and 61N to 66N via the switches 41N to 47N. Capacitive elements 53N and 71N to 76N connected to the second electrode P2. The second electrodes P2 of the capacitive elements 52N and 61N to 66N of the first capacitive bank BK1 are connected to the input node NinP via a switch row 39N configured by a plurality of switch groups. The first electrodes P1 of the capacitive elements 53N and 71N to 76N of the second capacitive bank BK2 are connected to the input node NinN via the switches 32N to 38N. Further, the second electrodes P2 of the capacitive elements 53N and 71N to 76N are connected to the input node NinP via the switch row 39N, and the second electrode P2 of the capacitive element 53N is connected to the power supply voltage Vdd via the switch 82N. The second electrodes P2 of the capacitive elements 71N to 76N are connected to the output of the voltage supply unit 501N via the switches 83N to 88N.

  In the capacitor circuit 501N, the first electrode P1 is connected to the output node NoutN, the second electrode P2 is connected to the input node NinP via the switch row 39N, and further to the power supply voltage Vdd via the switch 81N. The capacitive element 51N connected to is included.

  Similarly to the voltage supply unit 501P, the voltage supply unit 501N includes a plurality of inverter circuits 91N to 96N, and the respective operation power supplies are commonly fed from the power supply nodes Nvd and Nvs. These inverter circuits 91N to 96N are supplied with control signals NC1 to NC6 from the control unit 511, and the inverter circuits 91N to 96N supply inverted voltages with respect to the control signals NC1 to NC6 to the switches 83N to 88N. . It should be noted that the voltages supplied to the switches 83N to 88N by the inverter circuits 91N to 96N are voltages corresponding to power supply voltages or ground voltages supplied to the power supply nodes Nvd and Nvs, respectively.

  In the fourth embodiment, the capacitance values of the capacitive elements 51P and 51N are C1, and the capacitance values of the capacitive elements 52P, 52N, 53P and 53N are C2. The capacitance values of the capacitive elements 61P to 66P and 61N to 66N included in the first capacitance bank BK1 are C3, and the capacitance values of the capacitive elements 71P to 76P and 71N to 76N included in the second capacitance bank BK2. Is also C3.

  Next, the operation of the MDAC 500 according to the fourth embodiment will be described.

  This MDAC 500 also operates in a sampling period and a residual amplification period as in the first or second embodiment.

  In the sampling period, the switches 31P to 38P and 31N to 38N and the switch trains 39P and 39N are turned on, and the remaining switches 41P to 47P, 41N to 47N, 81P to 88P, and 81N to 88N are turned off. Is done. As a result, the positive electrodes of the capacitive elements 52P and 61P to 66P in the first capacitive bank BK1 in the capacitive circuit 501P and the first electrodes P1 of the capacitive elements 53P and 71P to 76P in the second capacitive bank BK2 in the capacitive circuit 501P. The input signal VinP is supplied, and the negative phase input signal VinN is supplied to each second electrode P2. The positive phase input signal VinP is also supplied to the first electrode P1 of the capacitive element 51P in the capacitive circuit 501P, and the negative phase input signal VinN is supplied to the second electrode P2. That is, the positive phase input signal VinP and the negative phase input signal VinN are applied to the pair of electrodes P1 and P2 of each capacitive element, and sampling is performed with full differential.

  On the other hand, at this time, the negative phase input is applied to the first electrodes P1 of the capacitive elements 52N and 61N to 66N in the first capacitive bank BK1 and the capacitive elements 53N and 71N to 76N in the second capacitive bank BK2 in the capacitive circuit 501N. The signal VinN is supplied, and the positive phase input signal VinP is supplied to each second electrode P2. At this time, the negative-phase input signal VinN is supplied to the first electrode P1 of the capacitive element 51N in the capacitive circuit 501N, and the positive-phase input signal VinP is supplied to the second electrode P2. That is, the negative phase input signal VinN and the positive phase input signal VinP are applied to the pair of electrodes P1 and P2 of each capacitive element, and sampling is performed with full differential. As a result, even if the capacitance value of each capacitive element is reduced, the amount of charge held can be maintained, and an increase in occupied area can be prevented.

  The same applies to the first and second embodiments. However, in the capacitance circuit 501P, the first electrodes P1 of the capacitive elements 51P, 52P and 61P to 66P to which the first electrode P1 is connected to the output node NoutP During the sampling period, the positive phase input signal VinP is supplied. On the other hand, in the capacitance circuit 501N, the negative phase input signal VinN is supplied to the first electrodes P1 of the capacitive elements 51N, 52N and 61N to 66N having the first electrode P1 connected to the output node NoutN during the sampling period. Is done. Thereby, the capacitive circuit 501P outputs the residual amplified signal related to the positive phase input signal VinP to the output node NoutP, and the capacitive circuit 501N outputs the residual amplified signal related to the negative phase input signal VinN to the output node NoutN. Become.

  In this sampling period, coarse quantization is performed by the coarse quantizer 510 on the positive phase input signal VinP and the negative phase input signal VinN supplied to the input nodes NinP and NinN. In FIG. 5, the input nodes VinP and VinN and the input of the coarse quantizer 510 are depicted as being separated, but the input of the coarse quantizer 510 is coupled to the input nodes NinP and NinP. I want to be understood.

  The quantization by the coarse quantizer 510 is similar to the quantization by the coarse quantizer 114 described in the first embodiment, but in this fourth embodiment, seven-value quantization is performed. . That is, in the first embodiment, the difference voltage (VinP−VinN) between the positive phase input signal VinP and the negative phase input signal VinN is larger than (a) the reference voltage Vref / 4, or (b) the reference voltage. The input signal is quantized by determining whether it exists between Vref / 4 and the reference voltage-Vref / 4 or (c) smaller than the reference voltage-Vref / 4.

  On the other hand, in the fourth embodiment, the coarse quantizer 510 determines in which of the following voltage ranges the difference voltage (VinP−VinN) of the input differential signal exists, and performs quantization. Do. That is, the difference voltage (VinP−VinN) is (a1) 5Vref / 8 or higher, (a2) exists between 5Vref / 8 and 3Vref / 8, or (a3) between 3Vref / 8 and Vref / 8. Exists, (b) exists between Vref / 8 and −Vref / 8, (c3) exists between −Vref / 8 and −3 Vref / 8, or (c2) −3 Vref / 8 and − It is determined whether it exists between 5 Vref / 8 and (c1) −5 Vref / 8 or less.

Crude quantizer 510, when judging a difference voltage between (a1), the value "3" of the digital signal D _i to be output, if it is determined differential voltage and (a2), the digital signal D _i to be output When the value is “2” and the difference voltage is determined to be (a1), the value of the output digital signal _Di is set to “1”. Further, the coarse quantizer 510, when judging a difference voltage between (b), and the value "0" of the digital signal D _i to be output, if it is determined differential voltage and (c3), the digital signal D to be output _i values of the "-1", if it is determined differential voltage and (c2), the value of "-2" of the digital signal D _i to be output, if it is determined differential voltage and (c1), it outputs a digital and the value of the signal _{D i} "-3". Digital signal D _i obtained by such quantization is output from the cyclic analog-to-digital converter. Further, by performing a predetermined process for a plurality of digital signals D _i outputted time series from the crude quantizer 510, it is converted into a binary digital signal. Furthermore, the digital signal D _i is the output from the coarse quantizer 510 is supplied to the control unit 511.

  In the residual amplification period following the sampling period, the switches 31P to 38P and 31N to 38N and the switch trains 39P and 39N are turned off. On the other hand, the switches 41P to 47P, 81P to 88P, the switches 41N to 47N, and 81N to 88N are turned on. Thereby, in the capacitive circuit 501P, the first electrodes of the capacitive elements 53P and 71P to 76P of the second capacitive bank BK2 are respectively connected to the second electrodes P2 of the capacitive elements 52P and 61P to 66P of the first capacitive bank BK1. P1 will be connected. That is, the capacitor element in the first capacitor bank and the capacitor element in the second capacitor bank BK2 are connected in series. Similarly, in the capacitive circuit 501N, the first electrodes of the capacitive elements 53N and 71N to 76N of the second capacitive bank BK2 are connected to the second electrodes P2 of the capacitive elements 52N and 61N to 66N of the first capacitive bank BK1, respectively. P1 will be connected. Thereby, also in the capacitive circuit 501N, the capacitive element in the first capacitive bank BK1 and the capacitive element in the second capacitive bank BK2 are connected in series.

  If an example of a capacitive element connected in series in the residual amplification period is described, in the capacitive circuit 501P, the second electrode P2 of the capacitive element 52P is connected to the first electrode P1 of the capacitive element 53P, and the capacitive element 66P The second electrode P2 is connected to the first electrode P1 of the capacitive element 76P. Similarly, in the capacitive circuit 501N, the second electrode P2 of the capacitive element 52N is connected to the first electrode P1 of the capacitive element 53N, and the second electrode P2 of the capacitive element 66N is connected to the first electrode P1 of the capacitive element 76N. Is done.

  In the residual amplification period, the switches 81P to 82P and 81N to 82N are turned on, so that the capacitive element 51P in the capacitive circuit 501P is connected between the power supply voltage Vdd and the output node NoutP. Capacitance elements 52P and 53P in capacitance circuit 501P are connected in series between power supply voltage Vdd and output node NoutP. Similarly, the capacitive element 51N in the capacitive circuit 501N is connected between the power supply voltage Vdd and the output node NoutN. Capacitance elements 52N and 53N in capacitance circuit 501N are connected in series between power supply voltage Vdd and output node NoutN.

  These capacitive elements 51P to 53P (51N to 53N) are used to appropriately set the common voltage Vcm at the output node NoutP (NoutN). The common voltage Vcm is desirably determined in consideration of the operating point of the buffer circuit 17P (17N). The capacitive elements 12P, 13P, 12N, and 13N described in the first and second embodiments are also used for appropriately setting the common voltage Vcm. It should be noted that the common voltage Vcm can be adjusted in various ways other than the configuration and method shown in FIG. 5, and of course, the present invention is effective in any of them.

  Since the switches 83P to 88P and 83N to 88N are turned on in the residual amplification period, the second electrodes P2 of the capacitive elements 71P to 76P in the second capacitive bank BK2 are connected via the switches 83P to 88P. Thus, the voltage from the voltage supply unit 502P is supplied. Similarly, the voltage from the voltage supply unit 502N is supplied to the second electrodes P2 of the capacitive elements 71N to 76N in the second capacitive bank BK2 via the switches 83N to 88N. In the fourth embodiment, output voltages of inverter circuits 91P to 96P are applied to second electrodes P2 of corresponding capacitive elements 71P to 76P, and output voltages of inverter circuits 91N to 96N are applied to corresponding capacitive elements 71N to 71N. The voltage is applied to the 76N second electrode P2. Here, each inverter circuit uses, as an output voltage, the power supply voltage Vdd supplied to the power supply node Nvd or the ground voltage Vs supplied to the power supply node Nvs according to the supplied control signals PC1 to PC6 and PN1 to PN6. The voltage is applied to the second electrode P2 of the corresponding capacitive element.

In residue amplification period, the control unit 511, in accordance with the digital signal _{D i} outputted from the coarse quantizer 510, generates a control signal PC1~PC6 and NC1～NC6. In the fourth embodiment, the control unit 511, when the digital signal _{D i} is "3" (a1), the six capacitor element 71P~76P the second capacitor bank BK2 in normal phase of the capacitance circuit 501P Control signals PC1 to PC6 are generated such that an output voltage of the ground voltage Vs (ground) is applied to each second electrode P2 from the inverter circuits 91P to 96P. At this time, the output voltage of the power supply voltage Vdd is applied from the inverter circuits 91N to 96N to the second electrodes P2 of the six capacitors 71N to 76N in the second capacitor bank BK2 in the capacitor circuit 501N on the opposite phase side. Such control signals NC1 to NC6 are generated.

Further, when the digital signal _{D i} is "2" (a2), the control unit 511, out of the capacitive element 71P~76P in the second capacitor bank BK2 in capacitance circuit 501P, the second electrode of the five capacitive element P2 In addition, the control signals PC1 to PC6 are generated so that the output of the ground voltage Vs is applied from the inverter circuit and the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrode P2 of the remaining one capacitive element. . At this time, the control unit 511 applies the output of the power supply voltage Vdd from the inverter circuit to the second electrodes P2 of the five capacitive elements among the capacitive elements 71N to 76N in the second capacitive bank BK2 of the capacitive circuit 501N. Control signals NC1 to NC6 are generated so that the output of the ground voltage Vs is applied from the inverter circuit to the second electrode P2 of the remaining one capacitive element.

When the digital signal _{D i} is "1" (a3), the control unit 511, out of the capacitive element 71P~76P in the second capacitor bank BK2 in capacitance circuit 501P, the second electrode P2 of the four capacitive elements, The output of the ground voltage Vs is applied from the inverter circuit, and the control signals PC1 to PC6 are generated so that the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the remaining two capacitive elements. At this time, the output of the power supply voltage Vdd is applied to the second electrodes P2 of the four capacitive elements among the capacitive elements 71N to 76N in the second capacitive bank BK2 of the capacitive circuit 501N, and the remaining two capacitive elements Control signals NC1 to NC6 are generated so that the output of the ground voltage Vs is applied to the second electrode P2.

When the digital signal D _i = 0 (b), the control unit 511 applies the ground voltage to the second electrodes P2 of the three capacitive elements among the capacitive elements 71P to 76P in the second capacitive bank BK2 in the capacitive circuit 501P. The control signals PC1 to PC6 are generated such that the output of Vs is applied from the inverter circuit and the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the remaining three capacitors. At this time, the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the three capacitors among the capacitors 71N to 76N in the second capacitor bank BK2 in the capacitor circuit 501N, and the remaining three Control signals NC1 to NC6 are generated such that the output of the ground voltage Vs is applied from the inverter circuit to the second electrode P2 of the capacitive element.

When the digital signal _{D i} is "-1" (c3), the control unit 511, out of the capacitive element 71P~76P in the second capacitor bank BK2 in capacitance circuit 501P, the second electrode P2 of the two capacitive elements The control signals PC1 to PC6 are generated so that the output of the ground voltage Vs is applied from the inverter circuit, and the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the remaining four capacitive elements. At this time, the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the two capacitive elements among the capacitive elements 71N to 76N in the second capacitive bank BK2 in the capacitive circuit 501N, and the remaining Control signals NC1 to NC6 are generated such that the output of the ground voltage Vs is applied from the inverter circuit to the second electrodes P2 of the four capacitive elements.

When the digital signal _{D i} is "-2" (c2), the control unit 511, out of the capacitive element 71P~76P in the second capacitor bank BK2 in capacitance circuit 501P, the second electrode P2 of one capacitor element The control signals PC1 to PC6 are generated such that the output of the ground voltage Vs is applied from the inverter circuit, and the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrodes P2 of the remaining five capacitive elements. At this time, the output of the power supply voltage Vdd is applied from the inverter circuit to the second electrode P2 of one capacitor among the capacitors 71N to 76N in the second capacitor bank BK2 in the capacitor circuit 501N, and the remaining five Control signals NC1 to NC6 are generated such that the output of the ground voltage Vs is applied from the inverter circuit to the second electrode P2 of the capacitive element.

Finally, when the digital signal _{D i} is "-3" (c1), the control unit 511, the second electrode P2 of all of the capacitor 71P~76P in the second capacitor bank BK2 in capacitance circuit 501P, supply voltage Control signals PC1 to PC6 are generated such that the output of Vdd is applied from the inverter circuit. At this time, the control unit 511 controls the control signal NC1 so that the output of the ground voltage Vs is applied from the inverter circuit to the second electrodes P2 of all the capacitive elements 71N to 76N in the second capacitive bank BK2 in the capacitive circuit 501N. ~ NC6 is generated.

  According to the fourth embodiment, in the sampling period, the capacitive elements 61P to 66P (61N to 66N) in the first capacitive bank BK1 and the capacitive elements 71P to 76P in the second capacitive bank BK2 in the capacitive circuit 501P (501N). (71N to 76N) are charged in parallel by fully differential sampling. In the residual amplification period following the sampling period, the charged capacitive elements 61P to 66P (61N to 66N) in the first capacitive bank BK1 and the capacitive elements 71P to 76P (71N to 76N) in the second capacitive bank BK2 are respectively charged. Connected in series. As a result, the voltage at the output node NoutP (NoutN) can be doubled or more. The output of the coarse quantizer is given to the capacitor circuit by the voltage supply circuit, and is reflected in the voltage at the output node NoutP (NoutN).

  With the above-described operation, the difference voltage Vout between the output voltage VoutP of the buffer circuit 17P and the output voltage VoutN of the buffer circuit 17N, that is, the output voltage Vout of the MDAC 500 follows the equation (7). Here, the gain G of the MDAC is a value close to 4. Vin is a difference voltage between the positive phase input signal VinP and the negative phase input signal VinN, and Vref is expressed by Vdd and each capacitance ratio as in the equation (3).

  In the fourth embodiment, similarly to the first embodiment, the capacitor used for sampling is configured by a plurality of capacitive elements, and the power supply voltage Vdd or the ground voltage Vs is applied to the second electrode P2 of the capacitive element during the residual amplification period. Is applied, the output of the coarse quantizer 510 is reflected. At this time, the reference voltage Vref is set equivalently by the capacitance ratio of a plurality of capacitive elements used for sampling. In other words, the voltage difference between the power supply voltage Vdd and the ground voltage Vs is divided according to the capacitance ratio, and the differential circuit configuration that performs complementary operation is equivalent. Thus, the reference voltage Vref is set. Thus, the voltage supply units 502P and 502N may apply the power supply voltage Vdd or the ground voltage Vs to a plurality of capacitance elements used for sampling in the residual amplification period, and separately have a highly accurate reference voltage generation circuit. This eliminates the need for power consumption. Further, since sampling is performed by fully differential sampling, the capacitance value of the capacitive element used for sampling can be reduced to ¼, and an increase in occupied area can be suppressed.

  In the fourth embodiment, since the first capacitor bank BK1 and the second capacitor bank BK2 are connected in series in the residual amplification period, the voltage amplification factor G of the MDAC 500 is almost quadrupled. Can be. On the other hand, in the first embodiment, an amplification factor of approximately twice can be obtained by the basic concept described above. In addition to this, in the fourth embodiment, a voltage amplification factor of about twice can be obtained by connecting the first capacitor bank BK1 and the second capacitor bank BK2 in series. As a result, it is possible to obtain a residual amplification factor of approximately 4 times (G≈4) in total. By increasing the amplification factor in the residual amplification period, the number of conversions can be reduced as can be understood from the equation (1). For example, in the first embodiment, the amplification factor G is approximately 2. On the other hand, in the fourth embodiment, the amplification factor G is approximately 4. Therefore, the fractional denominator value in the final term of Equation (1) is the same value in the first and fourth embodiments at half the number of conversions N. Therefore, compared with the case of the first embodiment (G = 2), the same conversion error can be realized with half the number of conversions N.

  Since the number of conversions N is half, the conversion processing time for each bit can be doubled at the same conversion rate. As a result, the transient response times of the buffer circuits 17P and 17N can be relaxed by a factor of 2, so that the power consumption of the buffer circuit can be reduced and the cyclic analog-digital converter can be further reduced in power. On the other hand, the first embodiment has an effect that the occupied area can be further reduced because the number of elements such as switches and capacitors is small and the structure is simple as compared with the first embodiment.

(Embodiment 5)
FIG. 6A is a block diagram showing the configuration of the cyclic analog-digital converter 600 according to the fifth embodiment, and FIG. 6B is a timing showing the operation of the cyclic analog-digital converter 600. FIG. Since the cyclic analog-digital converter 600 according to the fifth embodiment is similar to the cyclic analog-digital converter 402 shown in the third embodiment, the differences will be mainly described.

  In FIG. 6A, each of 601a and 601b is a multiplication type digital-analog conversion circuit, and the MDAC described in the first, second or fourth embodiment is used. In the figure, 17A and 17B are buffer circuits, and each buffer circuit 17A and 17B includes buffer circuits 17P and 17N (Embodiment 1 or 4) or 200P and 200N (Embodiment 2). . Reference numerals 63 and 64 denote switches, which functionally correspond to the switch 303 described in the third embodiment.

  In this embodiment, the buffer circuit 17B associated with the MDAC 601b is used as an input buffer circuit that receives the input signal Vin in the cyclic analog-digital converter 600. In general, in an analog-digital converter, a backflow of a signal to a pre-stage circuit called kickback occurs with a sampling operation. Therefore, in order to prevent kickback, an input buffer circuit is provided immediately before the analog-digital converter.

  In this embodiment, the multiplying digital-to-analog converter circuit described in the first, second, or fourth embodiment is used as MDACs 601a and 601b and connected in two stages in series via a buffer circuit 17A. The basic operation of the cyclic analog-digital converter 600 is the same as the operation described in the third embodiment. When the switch 63 is turned on, the MDACs 601a and 601b are fed back via the switch 603 and the buffer circuits 17A and 17B so that the residual amplified signal draws a loop. Here, for example, when the MDAC 601a is viewed as the first stage MDAC, the output of the capacitor circuit in the first stage MDAC 601a is supplied from the output node Nout to the buffer circuit 17A, and the output of the capacitor circuit in the next stage MDAC 601b is output from the output node Nout. To the buffer circuit 17B.

  In this embodiment, a switch 63 is connected between the buffer circuit 17B associated with the next-stage MDAC 601b and the output node Nout of the next-stage MDAC 601b, and the input signal Vin serving as the input of the cyclic analog-to-digital converter 600. And a switch 64 is connected to the input of the buffer circuit 17B. The switch 63 and the switch 64 constitute a multiplexer (selection circuit). When the input signal Vin is input to the input of the buffer circuit 17B, the switch 64 is turned on and the switch 63 is turned off. On the other hand, at the time of analog-digital conversion, the switch 64 is turned off and the switch 63 is turned on. As a result, either the residual amplified signal from the MDAC 601b or the input signal (analog input voltage) Vin to the cyclic analog-digital converter 600 is selected by the switches 63 and 64 and input to the buffer circuit 17B. .

  That is, in FIG. 6B, in the sampling period (1S) for capturing the input signal Vin, the switch 64 is turned on and the switch 63 is turned off, so that the analog input to the analog-digital converter 600 is performed. The voltage is input to the buffer circuit 17B, and is input to the first-stage MDAC 601a via the buffer circuit 17B. In this way, the buffer circuit 17B can function as an input buffer circuit provided in the preceding stage of the analog-digital converter 600.

  This is possible because the MDAC 601b does not need to perform residual amplification during the sampling period (1S). In the timing chart of FIG. 6B, the N-th bit residual amplification period (NA) overlaps with the sampling period (1S) of the next input signal Vin. Residual amplification is not required (only the coarse quantization function is required). In FIG. 6B, in a period other than the sampling period (1S), the switch 63 is turned on and the switch 64 is turned off. Thus, the buffer circuit 17B can function as the buffer circuit (equivalent to 17P, 17N, 200P, and 200N) described in the first, second, or fourth embodiment.

  In the fifth embodiment, since the input buffer for preventing kickback that is separately required can be used in the buffer circuit attached to the MDAC 601b, the occupied area and the power consumption can be further reduced.

(Embodiment 6)
FIG. 13 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit 1300 according to the sixth embodiment. The MDAC 1300 is similar to the MDAC 300 according to the first embodiment. In the following description, reference numerals of corresponding parts in the MDAC 300 are shown in parentheses, and detailed description of the corresponding parts is omitted here in principle.

In FIG. 13, a multiplying digital-to-analog converter circuit 1300 includes a coarse quantizer 810 (114) that receives a normal phase input signal VinP and a negative phase input signal VinN, and a digital signal D that is an output from the coarse quantizer 810. receiving a _i, it is provided with a control unit 811 for generating a control signal PC1, NC1 in accordance with the digital signal _{D i} (115). Further, the MDAC 1300 includes a capacitance circuit 1301P (100P) corresponding to the positive phase input signal, a capacitance circuit 1301N (100N) corresponding to the negative phase input signal, and a voltage supply unit 1302P (101P) and 1302N (101N). Yes. Also in this figure, buffer circuits 17P (17P) and 17N (17N) associated with the MDAC 1300 are shown, and the input of the buffer circuit 17P is connected to the output node NoutP (NoutP) of the capacitor circuit 1301P. The input of 17N is connected to the output node NoutN (NoutN) of the capacitor circuit 1301N.

  The capacitive circuit 1301P includes capacitive elements 802P (12P), 803P (13P), and 804P each having the first electrode P1 connected to the output node NoutP. The capacitive circuit 1301N is connected to each output node NoutN. Capacitance elements 802N (12N), 803N (13N), and 804N to which one electrode P1 is connected are provided.

  The capacitor circuit 1301P includes a switch 801P (11P) connected between an input node NinP (NinP) to which the positive phase input signal VinP is supplied and an output node NoutP (also corresponding to an output node of the MDAC 1300), and an input It has a switch row 805P (16P) made up of a plurality of switches provided between the node NinN and the second electrodes P2 of the capacitive elements 802P to 804P. Further, the capacitor circuit 1301P is connected between the switch 806P (18P) connected between the second electrode P2 of the capacitor 802P and the power supply voltage Vdd, and between the second electrode P2 of the capacitor 803P and the ground voltage Vs. Switch 807P (19P), and a switch 809P connected between the second electrode P2 of the capacitor 804P and the output node of the voltage supply unit 1302P.

  The capacitor circuit 1301N includes a switch 801N (11N) connected between the input node NinP to which the reverse-phase input signal VinN is supplied and the output node NoutN of the capacitor circuit 1301N, and the input node NinP and the capacitor elements 802N to 804N. And a switch row 805N (16N) composed of a plurality of switches respectively provided between the second electrode P2. Further, the capacitor circuit 1301N is connected between the switch 806N (18N) connected between the second electrode P2 of the capacitor 802N and the power supply voltage Vdd, and between the second electrode P2 of the capacitor 803N and the ground voltage Vs. Switch 807N (19N) and a switch 809N connected between the second electrode P2 of the capacitor 804N and the output node of the voltage supply unit 1302N.

  Also in this embodiment, although not particularly limited, each of voltage supply units 1302P and 1302N includes inverter circuits 808P and 808N. Inverter circuit 808P in voltage supply unit 1302P operates using power supply voltage Vdd supplied to power supply node Nvd and ground voltage Vs supplied to power supply node Nvs as power supply voltages.

  The inverter circuit 808P receives the control signal PC1 from the control unit 811 and supplies the phase-inverted signal to the second electrode P2 of the capacitor 804P via the switch 809P. That is, the power supply voltage Vdd or the ground voltage Vs corresponding to the signal obtained by inverting the phase of the control signal PC1 is supplied to the second electrode P2 of the capacitive element 804P through the switch 809P. Similarly, inverter circuit 808N in voltage supply unit 1302N operates using power supply voltage Vdd and ground voltage Vs supplied to voltage nodes Nvd and Nvs as operating voltages, and power supply voltage Vdd corresponding to a signal obtained by inverting the phase of control signal NC1. Alternatively, the ground voltage Vs is supplied to the second electrode P2 of the capacitive element 804N via the switch 809N.

  Similar to the first embodiment, the MDAC 1300 operates in a sampling period and a residual amplification period. That is, it operates in the sampling period, and then operates in the residual amplification period.

First, in the sampling period, the normal phase input signal VinP and the negative phase input signal VinN are roughly quantized by the coarse quantizer 810. In this quantization, a difference voltage (VinP−VinN) between the positive phase input signal VinP and the negative phase input signal VinN is obtained. On the other hand, in coarse quantizer 810, a predetermined voltage range is determined in advance based on reference voltage Vref. The obtained voltage difference (VinP-VinN) is the where to the presence of voltage within a predetermined range, the value of the digital signal D _i is determined. In this embodiment, the digital signal _Di is a binary digital signal, and becomes “1” or “−1” of the binary signal depending on whether the differential voltage is positive or negative.

Control unit 811, according to the value of the digital signal _{D i,} determining the voltage of the control signal PC1 and NC1. In this case, the control signals PC1 and NC1 are complementary voltages. That is, when the control signal PC1 is at a high level (binary “1”) corresponding to the power supply voltage Vdd, the control signal NC1 becomes the ground voltage Vs (“0”).

  In the sampling period, the switches 801P and 801N and the switch trains 805 and 805N are turned on, and the switches 806P, 807P, 809P, 806N, 807N, and 809N are turned off. Thus, the positive phase input signal VinP is supplied to the first electrodes P1 of the capacitive elements 802P to 894P in the capacitive circuit 1301P, and the negative phase input signal VinN is supplied to the second electrode P2. As a result, a voltage difference between the positive phase input signal VinP and the negative phase input signal VinN is applied between the respective electrodes of the capacitive element in the capacitive circuit 1301P and charged. That is, fully differential sampling is performed. Similarly, the negative phase input signal VinN is supplied to the first electrodes P1 of the capacitive elements 802N to 894N in the capacitive circuit 1301N, and the positive phase input signal VinP is supplied to the second electrode P2. As a result, a voltage difference between the negative-phase input signal VinN and the positive-phase input signal VinP is applied between the respective electrodes of the capacitive element in the capacitive circuit 1301N and charged. That is, fully differential sampling is performed.

  In the residual amplification period following the sampling period, the switches 806P, 807P, 809P, 806N, 807N and 809N are turned on, and the switches 801P and 801N and the switch trains 805P and 805N are turned off. As a result, the input nodes NinP, NinN and the output nodes NoutP, NoutN are electrically separated, and the second electrodes P2 of the capacitive elements 802P and 802N are connected to the power supply voltage Vdd via the switches 806P and 806N. The second electrodes P2 of 803P and 803N are connected to the ground voltage Vs via the switches 807P and 807N.

Further, during the residual amplification period, the output of the voltage supply unit 1302P (1302N), that is, the output of the inverter circuit 808P (808N), is supplied to the second electrode P2 of the capacitor 804P (804N) via the switch 809P (809N). To be applied. That is, the output of the inverter circuit 808P (808N), which is the output of the voltage supply unit 1301P (1301N), is applied to the second electrode P2 of the capacitor 804P (804N) via the switch 809P (809N). As in the first embodiment, a voltage according to the output (digital signal D _i ) of the coarse quantizer 810 is applied to the voltage supply units 1301P and 1301N. Specifically, when D _i = 1, PC1 is at the power supply voltage Vdd, NC1 is at the ground voltage Vs, and when D _i = −1, PC1 is at the ground voltage Vs and NC1 is at the power supply voltage Vdd. Is set. Thereby, the voltages at the output nodes NoutP and NoutN reflect the output of the coarse quantizer 810 and become an amplified voltage value (residual amplified value).

  Also in this embodiment, since sampling is performed by fully differential sampling, it is possible to reduce the size of the capacitor elements in the capacitor circuits 1301P and 1301N while maintaining the amount of charge to be charged, and increase the occupied area. It becomes possible to prevent. Also in this embodiment, since the voltage applied to the second electrode P2 of each of the capacitive elements 804P and 804N may be the power supply voltage Vdd or the ground voltage Vs, the voltage supply units 1302P and 1302N have high accuracy. It is possible to prevent an increase in the occupied area without requiring a reference voltage generation circuit, and to reduce power consumption.

  In this embodiment, the output of the MDAC 1300 is transmitted via the buffer circuits 17P and 17N, as in the first embodiment. Here, the output of the MDAC 1300 is a difference voltage Vout (= VoutP−VoutN) between the output voltage VoutP of the buffer circuit 17P and the output voltage VoutN of the buffer circuit 17N, and follows the equation (2).

  Note that the reference voltage Vref is set equivalently using the capacitance ratio of the plurality of capacitor elements 802P to 804P (802N to 804N) included in the capacitor circuit, as in the first embodiment. Also in this case, as in the embodiment described so far, the reference voltage Vref supplied to the coarse quantizer 810 is a voltage value along the equivalently determined reference voltage Vref, but with high accuracy. There is no need. As in the first embodiment, FIG. 13 is provided with a controller for controlling the above-described switches and the like, but is omitted in FIG. Similarly to the first embodiment, the voltage supply units 1302P and 1302N and the control unit 811 can be collectively regarded as a control circuit.

(Embodiment 7)
FIG. 14 is a circuit diagram showing a configuration of a multiplication type digital-analog conversion circuit 1400 according to the seventh embodiment. The MDAC 1400 is similar to the MDAC 500 described in the fourth embodiment. In the following description, reference numerals of corresponding parts in the MDAC 500 are shown in parentheses, and detailed description of the corresponding parts is omitted here in principle.

  The MDAC 1400 generates the control signals PC1, PC2, NC1, and NC2 based on the coarse quantizer 910 (510) that receives the normal phase input signal VinP and the negative phase input signal VinN, and the output of the coarse quantizer 910. And a control unit 911 (511). The MDAC 1400 includes a capacitor circuit 4101P (501P) corresponding to the positive phase input signal VinP and a capacitor circuit 1401N (502N) corresponding to the negative phase input signal VinN.

  Here, each of the capacitor circuit 1401P and the capacitor circuit 1401N has a first capacitor bank BK1 and a second capacitor bank BK2, as in the fourth embodiment, though not explicitly shown in the drawing. Unlike the fourth embodiment, the first capacitor bank BK1 and the second capacitor bank BK2 in this embodiment have a smaller number of capacitor elements included in each capacitor bank. That is, the first capacitor bank BK1 included in the capacitor circuit 1401P includes three capacitor elements 907P, 909P, and 911P to which the first electrode P1 is connected at the output node NoutP, and the second capacitor bank BK2 , Three capacitive elements 908P, 910P and 912P each having a first electrode P1 connected to an input node NinP through switches 902P to 904P. The first capacitor bank BK1 included in the capacitor circuit 1401N includes three capacitor elements 907N, 909N, and 911N connected to the first electrode P1 at the output node NoutN. The second capacitor bank BK2 , Three capacitive elements 908N, 910N and 912N each having a first electrode P1 connected to an input node NinN through switches 902N to 904N. Further, the capacitor circuit 1401P includes a capacitor element 906P (51P), and the capacitor circuit 1401N includes a capacitor element 906N (51N). In this embodiment, the capacitance values of the capacitive elements 911P, 911N, 912P, and 912N are selected to be twice the capacitance values of the capacitive elements 909P, 909N, 910P, and 910N.

  Similarly to the fourth embodiment, the first electrode P1 of the capacitor included in the second capacitor bank BK2 in the capacitor circuit 1401P is switched to the second electrode P2 of the capacitor included in the first capacitor bank BK1. 913P to 915P are connected. In addition, the first electrode P1 of the capacitor included in the second capacitor bank BK2 in the capacitor circuit 1401N is also connected to the second electrode P2 of the capacitor included in the first capacitor bank BK1 via the corresponding switches 913N to 915N. It is connected.

  In this embodiment, each of the voltage supply units 1402P and 1402N includes two inverter circuits 920P, 921P, 920N, and 921N. Each inverter circuit operates using power supply voltages Vdd and Vs supplied to power supply nodes Nvd and Nvs as operating voltages.

In the fourth embodiment described above with reference to FIG. 5, the coarse quantizer 510 performs quantization to seven values. In this embodiment, the coarse quantizer 910 performs four-value quantization. Quantize. That is, the coarse quantizer 910 predetermines four voltage ranges using the reference voltage Vref supplied thereto. The coarse quantizer 910 determines whether the difference voltage (VinP−VinN) between the positive phase input signal VinP and the negative phase input signal VinN exists in one of the four voltage ranges. D _i is output. If the difference voltage is Vdd / 2 or _{D i = 3, D i =} 1 if between differential voltage of Vdd / 2 and 0, if between the differential voltage is zero and -Vdd / 2 _D i = −1, if the difference voltage is −Vdd / 2 or less, D _i = −3. Control unit 911, based on the digital signal _{D i} is the output from the coarse quantizer 910, as stated in the fourth embodiment, the control signal PC1, PC2, NC1 and NC2, respectively a high level ( Vdd) or low level (Vs).

  Also in the seventh embodiment, the MDAC 1400 operates separately in the sampling period and the subsequent residual amplification period.

  First, in the sampling period, the switches 901P to 904P and 901N to 904N and the switch trains 905P and 905N are turned on. As a result, the capacitance elements 907P to 912P and 907N to 912N included in the first capacitance bank BK1 and the second capacitance bank BK2 in the capacitance circuit 1401P and the capacitance circuit 1401N, respectively, are connected to the electrodes via the on-state switches. A normal phase input signal VinP and a negative phase input signal VinN are supplied. At this time, the positive-phase input signal VinP and the negative-phase input signal VinN are also supplied to the electrodes of the capacitive elements 906P and 906N via the switch in the on state. Thus, during the sampling period, each capacitive element is charged by fully differential sampling. Note that in the sampling period, the switches 913P to 919P and 913N to 919N are off.

  In the residual amplification period following the sampling period, the switches 913P to 919P and 913N to 919N are turned on, and the switches 901P to 904P and 901N to 904N and the switch trains 905P and 905N are turned off. Accordingly, the power supply voltage Vdd is supplied to the second electrodes P2 of the capacitive elements 906P and 917P in the capacitive circuit 1401P, and the power supply voltage Vdd is supplied to the second electrodes P2 of the capacitive elements 906N and 917N in the capacitive circuit 1401N. The On the other hand, the voltages from the inverter circuits 920P and 921P are applied to the second electrodes P2 of the capacitive elements 910P and 912P included in the second capacitive bank BK2 in the capacitive circuit 1401P. Similarly, the voltages from the inverter circuits 920N and 921N are applied to the second electrodes P2 of the capacitive elements 910N and 912N included in the second capacitive bank BK2 in the capacitive circuit 1401N.

In the residual amplification period, similarly to the fourth embodiment, the capacitors 908P, 910P, and 912P (908N, 910N, and 912N) included in the second capacitor bank BK2 via the switches 913P to 915P (913N to 915N). The first electrode P1 is connected to the second electrode P2 of the capacitive elements 907P, 909P and 911P (907N, 909N and 911N) included in the corresponding first capacitive bank BK1. That is, the capacitor element of the first capacitor bank BK1 and the capacitor element of the second capacitor bank BK2 are connected in series. As a result, as in the fourth embodiment, the voltages at the output nodes NoutP and NoutN are boosted approximately twice. At this time, since the voltage applied to the second capacitor bank BK2 is set according to the output of the coarse quantizer 910, the output of the coarse quantizer 910 is reflected in the voltages at the output nodes NoutP and NoutN. Residual amplification is performed. For example, when D _i = 3, both PC1 and PC2 are set to the power supply voltage Vdd, and both NC1 and NC2 are set to the ground voltage Vs. When D _i = 1, PC1 is the ground voltage Vs, PC2 is the power supply voltage Vdd, NC1 is the power supply voltage Vdd, and NC2 is the ground voltage Vs. When D _i = −1, PC1 is the power supply voltage Vdd, PC2 is the ground voltage Vs, NC1 is the ground voltage Vs, and NC2 is the power supply voltage Vdd. When D _i = −3, both PC1 and PC2 are set to the ground voltage Vs, and both NC1 and NC2 are set to the power supply voltage Vdd.

  According to the seventh embodiment, the output voltage Vout of the MDAC 1400 is a difference voltage (VoutP−VoutN) between the output voltage VoutP of the buffer circuit 17P and the output voltage VoutN of the buffer circuit 17N, and is expressed by Expression (7). The Further, the reference voltage Vref can be determined by the capacitance ratio of a plurality of capacitor elements included in the capacitor circuit.

  In the seventh embodiment, as in the fourth embodiment, it is possible to suppress an increase in the occupied area and to reduce power consumption. In particular, in this embodiment, the number of capacitive elements can be reduced, which is effective in reducing the occupied area.

(Embodiment 8)
FIG. 7 is a block diagram showing a medical diagnosis system according to the eighth embodiment. In the figure, reference numeral 700 denotes an ultrasonic diagnostic apparatus probe, and reference numeral 703 denotes an ultrasonic diagnostic apparatus. An ultrasonic diagnostic apparatus probe (hereinafter referred to as a diagnostic probe) 700 supplies a plurality of probes 71 and 72 and high-voltage pulses to the probes 71 and 72, respectively. And a processing device 701 for processing an analog signal (signal under measurement). The result processed by the processing device 701 is supplied as a digital signal to the ultrasonic diagnostic apparatus 703 via the digital cable 714. In the ultrasonic diagnostic apparatus 703, the processing unit 704 performs necessary processing on the digital signal received via the cable 714.

  When one probe, a transmission system that supplies a high voltage pulse to the probe, and a reception system that processes an analog signal from the probe are combined into one channel, the diagnosis in this embodiment The probe 700 is provided with more than 1000 channels. In the figure, of these channels, channels corresponding to two probes are shown as an example. Since each channel has the same configuration, one channel including the probe 71 will be described as an example here.

  The processing device 701 includes a transmission unit 73 connected to the probe 71 via a switch 75 and a reception unit connected to the probe 71 via a switch 77. Here, the receiving unit includes an analog front end circuit 79 having an amplifier and possibly a filter, and an analog / digital converter 711 connected to the analog front end circuit 79 and supplied with an analog signal. The cyclic analog-digital converter described in the embodiment is used as the analog-digital converter 711. That is, the signal from the analog front end circuit 79 becomes the input signal Vin described above.

  In the diagnosis, the switch 75 is turned on, and the high voltage pulse generated in the transmission unit 73 is sent to the probe 71 via the switch 75. The probe 71 converts the received high-voltage pulse into vibration and sends it as ultrasonic waves into the body of the human body to be diagnosed. The sent ultrasonic wave is reflected by an internal organ or the like and received by the probe 71 again. The received ultrasonic vibration is converted into an electric signal, and the converted electric signal is subjected to processing such as amplification by an analog front end circuit 79, and an input signal Vin is generated. The input signal Vin which is an analog signal is converted into a digital signal by an analog-digital converter 711.

The processing described above is performed in each channel (for example, a channel including the probe 74, the switches 76 and 78, the analog front-end circuit 710, and the analog-digital converter 712). In each channel, the converted digital signal D _i is supplied the digital phasing unit (digital circuit) 713. The digital phasing unit 713 obtains in-vivo information and reduces the amount of data by performing delay addition processing on the analog-digital conversion output of each channel. The output of the digital phasing unit 713 is sent to the ultrasonic diagnostic apparatus 703 via the digital cable 714 and used.

  In this embodiment, in the diagnostic probe 700, the processing device 701 excluding the probes 71 and 72 is constituted by one semiconductor device. That is, a plurality of transmitters 73 and 74, a plurality of analog front-end circuits 79 and 710, a plurality of analog-digital converters 711 and 712, switches 73 to 78, and a digital phasing unit 713 corresponding to a plurality of channels are provided. The semiconductor device is formed. Each of the analog-digital converters uses the above-described cyclic analog-digital converter, and the power supply nodes Nvd and Nvs are commonly connected in each analog-digital converter. In this way, by using a common power supply node, it is possible to prevent the voltage values supplied to the capacitor circuits in the multiplication type digital-analog conversion circuit from differing from each other during the residual amplification period. It is possible to suppress variations such as conversion gain at the time of conversion. Further, by forming the semiconductor device in one semiconductor device, variation in the amplification factor G can be reduced, and also in this respect, variation in conversion can be suppressed.

  According to the embodiment described above, it is possible to suppress an increase in the occupied area, and it is possible to reduce the power consumption. It can be integrated in one semiconductor device. By incorporating analog-to-digital converters for all channels, it is possible to digitize the probe output for diagnosis and further reduce the data. As a result, the weight of the cable required for transmission to the ultrasonic diagnostic apparatus 703 is dramatically increased. Can be reduced. Furthermore, since signal quality deterioration due to transmission using a conventional analog cable can be prevented, it contributes to higher image quality.

  Further, the digital cable 714 may be transmitted from the diagnostic probe 700 to the ultrasonic diagnostic apparatus 703 by wireless transmission instead of wired communication. In this case, handling of the diagnostic probe 700 is facilitated. In FIG. 7, reference numeral 702 denotes a controller that controls the switches 73 to 77 and the like.

  Considering an embodiment (Embodiment 7) in which the MDAC converts an analog input signal into a binary digital signal, it can be understood that the invention to be added next is also described in the present application.

<Appendix>
One or more multiplying digital-to-analog converter circuits having an input node supplied with an input signal, an output node supplying an output signal, and a quantizer that quantizes the input signal based on a reference voltage; An analog signal in which an output signal of the multiplying digital-to-analog conversion circuit is supplied to an input node of the multiplying-type digital-to-analog conversion circuit or an input node of the multiplying-type digital-to-analog conversion circuit via another multiplying-type digital-to-analog conversion circuit A digital converter,
The multiplying digital-to-analog converter circuit samples and amplifies an input signal supplied to the input node, supplies the output signal to the output node, and supplies the capacitor circuit according to the output of the quantizer. A control circuit for determining the voltage,
The capacitance circuit includes a plurality of capacitance elements, and a first capacitance element of the plurality of capacitance elements is coupled to the output node, and when the input signal is sampled, a positive phase signal corresponding to the input signal A first electrode to which is applied, and a second electrode to which a negative-phase signal opposite in phase to the positive-phase signal is applied,
The reference voltage is equivalently set by the capacitance ratio of the capacitive elements included in the capacitive circuit, and is supplied to the second electrode of the first capacitive element by the control circuit when a sampled input signal is amplified. The capacitance circuit supplies an amplified signal reflecting the output of the quantizer to the output node by determining a voltage to be output.

  Although the invention made by the inventor has been specifically described based on the above embodiment, the present invention is not limited to the above embodiment, and various modifications can be made without departing from the scope of the invention.

11P, 18P, 19P, 31P-38P, 41P-47P, 81P-88P, 111P, 113P, 801P-804P, 806P, 807P, 809P, 901P-904P, 913P-919P Switch 11N, 18N, 19N, 31N-38N, 41P to 47P, 81P to 88P, 111N, 113N, 801N to 804N, 806N, 807N, 809N, 901N to 904N, 913N to 919N Switch 16P, 16N Switch group 12P to 15P, 51P to 53P, 61P to 66P, 71P to 76P , 802P to 804P, 906P to 912P Capacitance elements 12N to 15N, 51N to 53N, 61N to 66N, 71N to 76N, 802N to 804N, 906N to 912N Capacitance elements 16P, 39P, 805P, 905P, 16N 39N, 805N, 905N Switch row 17P, 17N, 200P, 200N Buffer circuit 100P, 501P, 1301P, 1402P, 100N, 501N, 1301N, 1402N Capacitance circuit 101P, 101N, 502P, 502N, 1302P, 1302N, 1402P, 1402N Voltage supply Unit 114, 510, 810, 910 Coarse quantizer 115, 511, 811, 911 Control unit 300, 400a, 400b, 500, 601a, 601b, 1300, 1400 Multiplication type digital-analog conversion circuit 402, 711, 712 Cyclic type Analog to digital converter

Claims (4)

And including at least one multiplication type digital-to-analog conversion circuit having an input node to which an input signal is supplied, an output node for supplying an output signal, and a quantizer for quantizing the input signal into a binary value, The output signal of the type digital / analog conversion circuit is supplied to the input node of the multiplication type digital / analog conversion circuit via the input node of the multiplication type digital / analog conversion circuit or another multiplication type digital / analog conversion circuit, An analog to digital converter,
The multiplying digital-to-analog converter circuit samples and amplifies the input signal supplied to the input node, supplies the input signal to the output node, and supplies the capacitor circuit according to the output of the quantizer. A control circuit for determining a voltage to be
The capacitance circuit is
When sampling the input signal coupled to the output node, a first electrode to which a positive phase signal corresponding to the input signal is applied, and a negative phase signal opposite to the positive phase signal are applied. A first capacitive element having a second electrode;
When the reference voltage is equivalently set by the capacitance ratio of the capacitive elements included in the capacitive circuit and amplifies the sampled input signal, the control circuit determines the first voltage based on the output of the quantizer. By determining a voltage supplied to the second electrode of the capacitive element, the capacitive circuit supplies an amplified signal reflecting the output of the quantizer and the reference voltage to the output node. . A plurality of analog-to-digital converters each receiving a signal under measurement as an input signal; and a digital circuit that receives the digital signals converted by the plurality of analog-digital converters and outputs a measurement signal based on the digital signals A diagnostic probe,
Each of the plurality of analog-digital converters is
One or more multiplication type digital-to-analog conversion circuits each having an input node to which the input signal is supplied and an output node to supply an output signal are provided, and the output signal of the multiplication type digital-to-analog conversion circuit is the multiplication type An analog-to-digital converter supplied to the input node of the multiplying digital-to-analog conversion circuit via the input node of the digital-to-analog conversion circuit or another multiplying-type digital-to-analog conversion circuit;
The multiplying digital-to-analog converter circuit is
A quantizer that quantizes the corresponding input signal into binary values;
A passive circuit that samples and amplifies the corresponding input signal;
A buffer circuit for receiving the output of the passive circuit;
A control circuit for forming a voltage to be supplied to the passive circuit according to the output of the quantizer;
Comprising
The passive circuit is
When sampling the corresponding input signal, a first electrode to which a normal phase signal corresponding to the input signal is supplied, and a second electrode to which a negative phase signal opposite to the normal phase signal is supplied A first capacitive element having
The first electrode of the first capacitive element is coupled to the buffer circuit, and when the sampled input signal is amplified, the second electrode of the first capacitive element follows the output of the quantizer. Voltage is supplied from the control circuit,
A reference voltage is set equivalently by the capacitance ratio of the capacitive elements included in the passive circuit, and the passive circuit sends an amplified signal reflecting the output of the quantizer and the reference voltage to the buffer circuit. Diagnostic probe supplied. An analog digital comprising two or more multiplying digital-to-analog conversion circuits each having an input node to which an input signal is supplied, an output node for supplying an output signal, and a quantizer for quantizing the input signal into binary values A converter,
The multiplying digital-to-analog converter circuit samples and amplifies the input signal supplied to the input node, supplies the input signal to the output node, and supplies the capacitor circuit according to the output of the quantizer. A control circuit for determining a voltage to be
The capacitance circuit is
When sampling the input signal coupled to the output node, a first electrode to which a positive phase signal corresponding to the input signal is applied, and a negative phase signal opposite to the positive phase signal are applied. A first capacitive element having a second electrode;
When the reference voltage is equivalently set by the capacitance ratio of the capacitive elements included in the capacitive circuit and amplifies the sampled input signal, the control circuit determines the first voltage based on the output of the quantizer. By determining a voltage supplied to the second electrode of the capacitive element, the capacitive circuit supplies an amplified signal reflecting the output of the quantizer and the reference voltage to the output node. . Two or more multiplying digital-to-analog conversion circuits each having an input node to which an input signal is supplied, an output node for supplying an output signal, and a quantizer for quantizing the input signal based on a reference voltage are provided. An analog to digital converter,
The multiplying digital-to-analog converter circuit samples and amplifies the input signal supplied to the input node, supplies the input signal to the output node, and supplies the capacitor circuit according to the output of the quantizer. A control circuit for determining a voltage to be
The capacitance circuit is
When sampling the input signal coupled to the output node, a first electrode to which a positive phase signal corresponding to the input signal is applied, and a negative phase signal opposite to the positive phase signal are applied. A first capacitive element having a second electrode;
A second capacitive element coupled to the output node and having a first electrode to which the positive phase signal is applied and a second electrode to which the negative phase signal is applied when sampling the input signal;
Including
The reference voltage is set equivalently by the capacitance ratio of the capacitive element included in the capacitive circuit, and when the sampled input signal is amplified, the control circuit is based on the output of the quantizer, By determining the voltage supplied to the second electrode of each of the first capacitive element and the second capacitive element, the capacitive circuit outputs an amplified signal reflecting the output of the quantizer and the reference voltage. An analog-to-digital converter that supplies the output node.

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