JP2011172481A - Motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor in which the cost is reduced, the performance is improved, and the size is reduced. <P>SOLUTION: The motor includes: M pieces of stator magnetic poles in 360 degrees by an electrical angle; back yokes for each stator; windings of each phase wound around each slot between stator magnetic poles at a pitch of approx. 180 degrees by an electrical angle; K pieces of rotor magnetic poles whose number of pieces is determined differently from the M pieces of the rotor in 360 degrees by an electrical angle; back yokes for each rotor; a DC power supply PS1 which supplies DC voltage VM; a DC power supply PS2 which supplies regenerative voltage VH larger than the voltage VM; and a current control means of each phase which increases current in one direction in each of the power supplies PS1 and PS2 and conducts electricity to windings of each phase. One end of a winding of each phase is connected to the DC power supply PS1, the other end is connected to an anode of a diode, and a cathode of the diode is connected to the DC power supply PS2. Conductive paths of current to windings of each phase from the DC power supplies PS1 and PS2 comprise three or more parallel conductive paths, and the current control means supplies power in parallel with two or more paths simultaneously. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a motor that is mounted on an automobile, a truck, or the like and that can be applied to industrial equipment, home appliances, and the like.

  Three-phase AC motors have been widely used conventionally. FIG. 116 is an example of a longitudinal sectional view showing a schematic configuration thereof. 811 is a motor output shaft, 812 is a rotor core, 817 and 818 are N pole permanent magnets and S pole permanent magnets attached to the rotor surface, 813 is a bearing, 814 is a stator core, 815 is a coil end of a winding, and 816 is a motor case. It is.

117 is a transverse sectional view showing a section AA-AA in FIG. This motor has three-phase alternating current, two poles, and twelve slots. The teeth of the stator are 821, 822, 823, 824, 825, 826, 827, 828, 829, 82A, 82B, and 82C in the circumferential direction. In each slot sandwiched between the teeth, a three-phase winding is wound, and the U-phase winding is a winding wound from 82Q to 82K and a winding wound from 82D to 82J.
The V-phase winding is a winding wound from 82G to 82P and a winding wound from 82H to 82N.
The W-phase winding is a winding wound from 82L to 82F and a winding wound around 82M and 82E. The pitch of each winding is an electrical angle of 180 °.

  FIG. 118 is a winding diagram, and the horizontal axis indicates each position in the rotational direction of the stator as an electrical angle. Since this motor is an example of a two-pole motor, the mechanical angle and the electrical angle are the same. U is the terminal of the U-phase winding, the U-phase current Iu is applied, V is the terminal of the V-phase winding, the V-phase current Iv is applied, and W is the W-phase winding, the terminal is supplied with the W-phase current Iw. , N is the neutral point of the three-phase Y connection. 831 is a connecting wire for the U-phase winding, 832 is a connecting wire for the V-phase winding, and 833 is a connecting wire for the W-phase winding.

  FIG. 119 is a connection diagram of a three-phase winding and is a star connection. 834 is a U-phase winding, 835 is a V-phase winding, and 836 is a W-phase winding. The surface magnet type brushless motors shown in FIGS. 116 to 119 are widely used as excellent motors. However, from the viewpoints of higher performance, smaller size, and lower cost, there are the following problems depending on the application. As basic characteristics of the motor, the force F is expressed by Fleming's law, and the torque T is expressed by the following equation.

F = BIL
T = FR
Here, I is the acting current, L is the length of the winding that works effectively, and R is the radius of the rotor.

  In the case of the motor shown in FIG. 117, the maximum magnetic flux density Bmax = 2 [T] of the magnetic steel plate at the tooth portion, the circumferential length Ws of the slot near the air gap portion between the stator and the rotor, and the circumferential direction of the tooth Assuming that the lengths Wt are the same, the average magnetic flux density acting as the motor torque is 1 [T]. Therefore, as a problem of the motor shown in FIG. 117, there is a problem that the maximum magnetic flux density Bmax = 2 [T] obtained as a magnetic material is not sufficiently utilized. The peak torque is an important characteristic from the viewpoint of miniaturization of the motor, but the peak torque is also insufficient from the same viewpoint.

  As inferred from FIG. 117, the problem of the stator is that the opening of the slot is narrow and the winding space factor is lowered due to the difficulty of winding of the winding, so the torque is reduced, Since the length of the coil end in the axial direction of the rotor is long, there is a problem that the motor is enlarged, and the productivity of the winding is also low, so that there is a problem of production cost.

  The problem of the rotor shown in FIG. 117 includes a problem of the rotor strength and a problem of the maximum allowable rotational speed due to the centrifugal force strength. In the case of a structure that can withstand the centrifugal force acting on the permanent magnets 817 and 818, there is a problem of the reinforcement cost. Since the output Pout of the motor is expressed as a product of the torque and the rotational angular velocity ωm, there is a problem that the rotational speed restriction becomes an output restriction.

  Further, in the case of a surface magnet type brushless motor, there is a problem that field weakening control is difficult and constant output control by field control is difficult.

  A so-called rare earth magnet composed of neodymium Nd, iron Fe, and boron B is generally used for a high-performance permanent magnet type brushless motor, but it is an expensive material and has a problem of cost. In particular, in recent years, it has attracted attention and concern as a problem of resource depletion.

  In addition, motors are used under various load conditions, and electric vehicles and hybrid vehicles require a large maximum torque, but they are relatively light loads during normal operation, and the motor efficiency at light loads is the same as that of automobiles. It works strongly as fuel efficiency. From the viewpoint of motor efficiency, there are Joule loss due to current flowing through the windings, iron loss accompanying rotation of field magnetic flux, mechanical loss caused by bearings, etc., and iron loss accompanying permanent magnet field at light loads. There are not a few operating areas where this is a problem. In particular, in an operating region where the motor rotates and the torque does not need to be generated, the iron loss component torque is also referred to as “drag torque”, and the presence of the magnetic flux of the permanent magnet itself may be a problem.

  As shown in the winding diagram of FIG. 118, there are all-pitch windings and distributed windings, the problem is that the arrangement of each winding is complicated and difficult to manufacture, the problem that the space factor of the windings is reduced, and the coil due to the intersection of windings There is a problem that the end portion becomes longer in the axial direction.

  FIG. 119 shows a configuration when the motor is driven by a three-phase inverter. 84D is a DC power source such as a battery, 841, 842, 843, 844, 845, and 846 are power transistors, and 847, 848, 849, 84A, 84B, and 84C are diodes arranged in parallel to the power transistors. 834 is a U-phase winding, 835 is a V-phase winding, and 836 is a W-phase winding. Many power transistors and the like are necessary, and there is a problem of the cost. In addition, since two transistors are connected in series when supplying current to the winding, there is a problem of inverter efficiency due to on-loss of the transistors. As a matter of course, a motor and a control device including an inverter are a set, and competitiveness is required in terms of performance, size, and cost.

  FIG. 120 shows a cross-sectional view of another conventional motor. A motor called a switched reluctance motor, which has six magnetic pole teeth on the stator and four magnetic poles on the rotor. Although much research has been done, there are still few examples of practical application.

  Reference numeral 861 denotes an A-phase stator magnetic pole, and the A-phase windings 867 and 868 are concentratedly wound around 861 in the positive and negative directions indicated by symbols in the drawing as indicated by a broken line 86N. Reference numeral 864 denotes a negative A-phase stator magnetic pole, and the A-phase windings 86E and 86D are concentratedly wound around the 864 in the positive and negative directions indicated by symbols in the drawing as indicated by a broken line 864. The two windings are connected in series with a jumper so that the directions of the currents coincide. When an A-phase current is passed at the rotor rotational position θr shown in FIG. 120, a magnetic flux indicated by an arrow 86M is generated, an attractive force is generated, and torque is generated in the counterclockwise direction CCW.

  Reference numeral 863 denotes a B-phase stator magnetic pole, and B-phase windings 86B and 86C are concentratedly wound around 863 as indicated by broken lines in the positive and negative directions indicated by symbols in the figure. Reference numeral 866 denotes a negative B-phase stator magnetic pole, and the B-phase windings 86J and 86H are concentratedly wound around the 866 in the positive and negative directions indicated by symbols in the figure as indicated by broken lines. The two windings are connected in series with a jumper so that the directions of the currents coincide.

  Reference numeral 865 denotes a C-phase stator magnetic pole, and the C-phase windings 86G and 86F are concentratedly wound around the 865 as indicated by broken lines in the positive and negative directions indicated by symbols in the figure. Reference numeral 862 denotes a negative C-phase stator magnetic pole, and the C-phase windings 869 and 86A are concentratedly wound around the 862 in the positive and negative directions indicated by symbols in the drawing as indicated by broken lines. The two windings are connected in series with a jumper so that the directions of the currents coincide.

  The motor shown in FIG. 120 sequentially energizes the A-phase current, the B-phase current, and the C-phase current according to the rotational position θr of the rotor 86L, and generates a continuous torque as a total torque. The direction of the A-phase current does not change the direction of the torque because the torque is generated by the attractive force of the soft magnetic material even if the current directions of the two windings are simultaneously reversed. The same applies to the B phase current and the C phase current. However, in order to reduce the iron loss of the rotor, the current direction shown in FIG. 120 is generally known in order to reduce the number of times the rotor magnetic poles are alternated.

  The switched reluctance motor shown in FIG. 120 does not use a permanent magnet and is low in cost. The stator windings are concentrated and simple, and the magnetic flux acting on the stator salient poles and the rotor salient poles is as follows. Since it operates at the saturation magnetic flux density of the magnetic steel sheet, it can be used for the torque by the electromagnetic action of high magnetic flux density, and the rotor is robust and can rotate at high speed.

  The problem with the switched reluctance motor shown in FIG. 120 is that the radial force acting between the stator and the rotor changes with the rotation to a position different by 90 ° in the circumferential direction, and the drive current acts like a switch. Therefore, there is a problem that the radial deformation of the stator is particularly large, and vibration and noise are large. With respect to the use efficiency of the windings, the current that is energized to generate torque is energized to four of the twelve windings shown in FIG. 120, and 4/12 = 1/3 windings. There is a problem that the use efficiency is low and the use efficiency is low, and as a result, the Joule loss, which is the heat generation of the windings, becomes large.

  FIG. 3 shows an example of an inverter that drives the motor of FIG. Reference numerals 871, 872, 873, 874, 875, and 876 denote power transistors, and reference numerals 877, 878, 879, 87A, 87B, and 87C denote diodes arranged in antiparallel to the respective power transistors and the respective windings 87D, 87E, and 87F. Each power transistor is turned on to supply electric energy to each winding to the DC power supply 84D. When each power transistor is turned off, the magnetic energy is returned to the DC power supply 84D. A flywheel current can also flow through each transistor and winding. Here, the winding 87D is a winding in which A-phase windings 867, 868 and 86E, 86D are wound in series with a jumper. The winding 87E is a winding in which B-phase windings 86B, 86C and 86J, 86H are wound in series with a jumper. The winding 87F is a winding obtained by winding C-phase windings 86G, 86F and 869, 86A in series.

JP 2005-110431 A JP 2002-272071 A

Consideration on egg-shaped diagram and error factor of switched reluctance machine, IEEJ Transactions, Industrial Application Division D, Vol.123, No.2, 2003, p82-89, Chiba Akira, Fukao Tadashi

As described above, the conventional motor is costly, low-performance and large in mechanical structure such as a stator, rotor, and winding, and is also expensive and large in the control circuit such as an inverter. As a result, there have been problems of high cost, low performance, and large size.

  The present invention has been made in view of such circumstances, and an object of the present invention is to provide a motor that can be reduced in cost, performance, and size.

In order to achieve the above-mentioned object, the invention according to claim 1 is characterized in that an electrical angle is set between an arbitrary M stator poles of the stator and 360 slots between the stator poles. Windings of each phase wound at a pitch of about 180 °, K rotor magnetic poles that are arbitrarily determined with an electrical angle of 360 ° differing from the M number of rotors, and a DC voltage VM A DC power supply PS1 to be supplied; a DC power supply PS2 that supplies a regenerative voltage VH that is a voltage higher than the voltage VM; and a DC power supply PS1 that is used to increase the unidirectional current to the windings of each phase Current control means for each phase that reduces and supplies current in one direction using a DC power source PS2, and one end of each phase winding is connected to the DC power source PS1, and the other end of the winding is connected to the other end of the winding. Connect the anode of the diode and An anode cathode is connected to the DC power supply PS2, and current supply paths from the DC power supplies PS1 and PS2 to the windings of each phase have three or more parallel current supply paths, and the current control means The power is supplied in parallel with two or more at the same time. Here, M is an integer of 6 or more, and K is an integer of 2 or more.

  According to this configuration, the motor is inexpensive because an expensive permanent magnet is not used for the rotor. In addition, when the high-speed running is carried out only by the internal combustion engine by using it as an electric drive motor of a hybrid vehicle, the motor is accompanied by a problem with iron loss of the motor. However, the motor of the present invention uses a permanent magnet. Therefore, there is no problem of iron loss called so-called drag torque. Further, it can operate at a high magnetic flux density of about 2.0 T, which is the saturation magnetic flux density of iron, and can generate a high torque. Moreover, since the rotor is robust, it can be driven up to high-speed rotation and a high power output can be obtained. Each winding can be used to drive both stator poles adjacent to each other in the circumferential direction, and the use efficiency is good. Specifically, the winding resistance can be reduced to ½ compared to a switched reluctance motor. However, there is a problem that the winding length of the coil end portion is slightly longer.

  Since the current passed through each phase winding of the stator is a one-way current and not an alternating current, the inverter as the current control means can be simplified. Each of the stator magnetic poles is also a one-way magnetic flux, and iron loss of the stator can be reduced. Further, as described above, each winding is connected independently, and can be used to drive the stator magnetic poles on both sides of the winding. Since the winding can also be used, a power transistor for passing the current can be shared, the operating rate of each power transistor can be doubled, and the current capacity of the power transistor can be halved. For example, when M = 6 and K = 4, the number of windings is three. When driving with three transistors to three windings and controlling the DC current to be turned on and off independently, the rotor magnetic pole rotates and power is simultaneously supplied to both adjacent windings of the stator magnetic pole to generate torque. can do. The motor can be rotationally driven by three transistors, and power can be supplied to the motor through two power supply paths with two of the three transistors. A normal three-phase AC inverter uses 6 transistors, and compared with the fact that the power supply path is one path on average, (2 path power supply) / (3 transistors) = 2/3 ( 1 power supply path) / (6 transistors) = 1/6, the current capacity of the inverter can be reduced to 1/4, and the inverter can be greatly reduced in size, cost and weight. is there. At this time, it is necessary to pay attention to the fact that the withstand voltage of the transistor is increased in the former driving method.

  In addition, in the case of an inverter that controls three unidirectional currents with six transistors, the current capacity can be halved for the same reason, and the inverter is significantly reduced in size, cost, and weight. Is possible.

  As for the winding connection, in the case of a conventional three-phase AC motor, a star connection is often used, and the current of one phase affects the current of other phases, and the current of each winding cannot be controlled independently. It has a configuration. In addition, since the conventional switched reluctance motor in which the winding is wound around each tooth is wound around two adjacent slots, the currents in the two slots cannot be controlled independently. The motor of the present invention can eliminate the limitations of conventional motors.

  The invention according to claim 2 is arranged such that M stator group magnetic poles arbitrarily defined in the stator within an electrical angle of 360 °, and arranged on the inner diameter side of the stator magnetic poles of the first group, The second group of M stator poles, in which the same-phase stator poles are arranged at the same angle in the circumferential direction, the first group of windings and the second group of windings are the first group The back yoke and the second group of back yokes are so-called toroidal windings, and each slot between the magnetic poles of the stator of the first group is wound at an electrical angle of approximately 180 °. The winding of each phase of the first group that is wound and the winding of each phase of the second group that is wound at an electrical angle of approximately 180 ° in each slot between the magnetic poles of the stator of the second group. Line and said first glue K first group of rotor magnetic poles, which are arranged on the outer diameter side opposite to the stator magnetic poles of the rotor and are arbitrarily determined with a difference of the number of M rotors between 360 ° in electrical angle, It is arranged on the outer diameter side so as to face the second group of stator magnetic poles, and is provided with K second group of rotor magnetic poles of the rotor between 360 ° in electrical angle.

  According to this configuration, in the combined winding, the winding can be a so-called toroidal winding and the winding length can be shortened, so that the winding resistance can be reduced. Since aligned winding is possible, the space factor of the winding can be improved. The windings of the same phase and the windings of the opposite phase are connected in series by a jumper, and voltage and current are supplied by the same transistor. The windings of slots having different phases on the circumference are similarly connected to supply voltage and current. The windings of different phases can supply voltage and current independently. Therefore, since the winding can be efficiently wound, the winding resistance can be reduced, and there are many other features similar to the motor described above.

  According to a third aspect of the present invention, in the magnetic pole shape of each of the stators, the shape of the pair of stator magnetic poles facing the rotor is an uneven shape, and two or more convex portions facing the rotor are provided. In the shape, the pitch of the convex portions of the rotor magnetic pole is substantially the same as the pitch of the concave and convex shapes of the pair of stator magnetic poles.

  According to this configuration, the number of teeth of the stator magnetic pole can be increased to two times or three times, and the torque can be increased in proportion. A proportional constant between current and torque, so-called torque constant can be increased by a factor of two or three, and Joule loss can be reduced. However, in general, the maximum torque does not increase so much and the iron loss increases. Therefore, for example, the motor is suitable for applications that require a large torque at a relatively low speed.

  The invention according to claim 4 is characterized in that a field winding is added to each stator magnetic pole.

  According to this configuration, it is possible to reduce the current load of the winding wound around the slot. In addition, when all the field windings wound around each stator magnetic pole are wound in series and a direct current is applied, the magnetic energy inside the motor automatically moves between the windings as the rotor rotates. Thus, the transfer of field energy between the motor control circuit and the motor is reduced, and the control is facilitated. On the other hand, the winding current that generates torque reduces the burden of moving the field energy, so that controllability is greatly improved and control with good responsiveness can be achieved.

  The invention according to claim 5 is characterized in that a permanent magnet is provided on a surface of each stator magnetic pole facing the rotor.

  According to this configuration, the field magnetic flux of the motor can be induced by arranging the permanent magnet on the surface of the stator magnetic pole of each motor, that is, the surface of the stator magnetic pole facing the rotor. The reluctance motor has a large field burden for generating the magnetomotive force of the field, and the efficiency of torque generation is particularly poor in a small current region. However, the reluctance motor can be improved by arranging a permanent magnet on each stator pole. In the motor of the present invention, the winding current is direct current, and the direction of magnetomotive force generated by the current is the direction of the magnetic flux generated by the magnet. This excitation direction is a direction that does not demagnetize. Accordingly, there are few restrictions on demagnetization of the magnet, excitation can be performed with a relatively thin magnet, and there are few cost problems. There are various magnet arrangements such as a method of arranging magnets on the entire surface facing the rotor of the stator, a method of arranging magnets for each stator magnetic pole, and a structure having both a magnet part and a soft magnetic body part on the surface of the stator magnetic pole. Forms are possible. As the magnetic flux direction, various magnets such as a unidirectional magnetic flux magnet, a radial magnetic flux magnet, a so-called Halbach structure magnet arrangement, and a so-called polar anisotropic magnetic flux arrangement magnet can be used.

  The invention described in claim 6 is characterized in that a permanent magnet is provided inside the soft magnetic body of each stator magnetic pole.

  According to this configuration, the permanent magnet can be arranged inside the soft magnetic body of the stator magnetic pole of each motor. By devising the shape and arrangement of the permanent magnet, various magnetic characteristics can be realized and the magnet can be held. For example, the surface magnetic flux density of the stator magnetic pole can be made higher than the magnetic flux density of the magnet surface by arranging the magnets obliquely. On the contrary, it is possible to achieve magnetic characteristics capable of high-speed rotation by suppressing the magnetic flux density of the stator magnetic pole. Further, the stator magnetic pole can be excited by arranging a magnet in a part of the back yoke portion of the stator.

  According to a seventh aspect of the present invention, the value of M, which is the number of magnetic poles of each stator magnetic pole between electrical angles of 360 °, is 6, and the value of K, which is the number of magnetic poles of each rotor magnetic pole, is 2. Features.

  According to this configuration, the winding configuration is a winding configuration of concentrated winding configuration in all joints, an annular winding configuration, two motors arranged on the inner and outer diameters, and two stator slots with inner and outer diameters A winding configuration such as a so-called toroidal winding is possible by winding a winding between them. A three-phase direct current motor can be configured. In addition, a plurality of power supply paths can be configured at the same time, and the motor control device can be configured at low cost.

  Various motors of the present invention, including this motor, can also be driven by an alternating current by changing the connection of windings, conditions for energizing the winding current, and timing. However, AC current driving complicates the control device including the configuration of the power transistor.

  According to an eighth aspect of the present invention, the value of M, which is the number of magnetic poles of each stator magnetic pole between electrical angles of 360 °, is 6, and the value of K, which is the number of magnetic poles of each rotor magnetic pole, is 4. Features.

  According to this configuration, the winding configuration is a winding configuration of concentrated winding configuration in all joints, an annular winding configuration, two motors arranged on the inner and outer diameters, and two stator slots with inner and outer diameters A winding configuration such as a so-called toroidal winding is possible by winding a winding between them. A three-phase direct current motor can be configured. In addition, a plurality of power supply paths can be configured at the same time, and the motor control device can be configured at low cost.

  According to a ninth aspect of the present invention, in the shape of each stator magnetic pole facing the rotor, one of the rotor axial directions is large in the circumferential direction such that the rotor axial shape of each stator magnetic pole is a trapezoidal shape. The other end of the direction is a small shape DIJ in the circumferential direction, and the shape of the stator magnetic pole adjacent to the circumferential direction in the rotor axial direction is a shape DIK obtained by reversing the shape DIJ in the rotor axial direction. The magnetic pole shapes DIJ and DIK are alternately arranged, and the stator magnetic pole of the shape DIJ and the stator magnetic pole of the shape DIK are formed by relatively separating the rotor axial position between the area center of the shape DIJ and the area center of the shape DIK. The size of the slot opening between the two can be increased.

  The invention according to claim 10 is characterized in that a soft magnetic material and a permanent magnet are added to the end of each stator magnetic pole in the rotor axial direction.

  According to this configuration, the magnetic saturation can be reduced, and the maximum torque of the motor can be improved.

  The invention described in claim 11 is characterized in that a soft magnetic material and an excitation winding for exciting magnetic flux are added to the end of each stator magnetic pole or each rotor magnetic pole in the rotor axial direction.

  According to this configuration, the problem of magnetic saturation can be reduced by the current of the excitation winding, and the maximum torque of the motor can be improved.

  The invention described in claim 12 is characterized in that a permanent magnet is arranged in the vicinity of the rotor side end of the slot of each stator.

  According to this configuration, the magnetic saturation can be reduced, and the maximum torque of the motor can be improved.

  According to a thirteenth aspect of the present invention, there is provided a structure in which intermediate taps of the total number of turns can be taken out on both sides of the winding of each stator, and means for switching the connection destination of the winding is provided.

  According to this configuration, the motor voltage can be lowered by connecting the output of the inverter to the intermediate tap, and the motor can be driven to a higher speed.

  The invention described in claim 14 is characterized by comprising demagnetizing means for reducing the magnetized state of the permanent magnet of the motor and magnetizing means for increasing the magnetized state of the permanent magnet of the motor.

  The invention according to claim 15 is characterized in that in the current control of each winding, a voltage signal is generated by using the flux linkage information of each winding determined by the current flowing through each winding and the rotor rotational position, and the current is controlled. To do.

  According to this configuration, in the current control of each winding, the voltage can be controlled by using the flux linkage information determined by the value of the current flowing through each winding and the rotor rotational position. As a result, it is possible to accurately supply the voltage necessary for energizing each winding even at high speed rotation, and it is possible to achieve highly accurate and highly responsive current control, with good torque output and response. It becomes possible to obtain sex.

A three-phase, full-pitch winding, concentrated winding, unidirectional current motor, which is an embodiment of the present invention, including a cross section of a motor having four rotor magnetic poles between electrical angles of 360 °, winding arrangements of each phase, and It is a figure which shows the example of magnetic flux. It is an example of the control device of the present invention combined with the motor winding of FIG. It is an example of the control device of the present invention combined with the motor winding of FIG. It is a figure which shows the electric current and magnetic flux accompanying a rotor rotation when the motor of FIG. 1 generate | occur | produces the torque of CCW. It is a figure which shows the relationship between the electric current of this invention motor, and a torque. FIG. 5 is a diagram showing the relationship between the rotor rotational position θr of the motor of FIG. 4, the current of each winding, and the torque in the CCW direction. It is a figure which shows the electric current and magnetic flux accompanying a rotor rotation when the motor of FIG. 1 generate | occur | produces a braking torque. It is a figure which shows the relationship between the rotor rotational position (theta) r of the motor of FIG. 7, the electric current of each winding, and the braking torque of a CW direction. It is a figure which shows the cross section of a motor provided with two rotor magnetic poles between electrical angles of 360 degrees, and winding arrangement | positioning of each phase by the three-phase motor which is embodiment of this invention. It is a figure which shows the cross section of the stator core and rotor core of the motor which multi-polarized the motor of FIG. 9 to 8 poles. It is a figure which shows the electric current and magnetic flux accompanying a rotor rotation when the motor of FIG. 9 generate | occur | produces the torque of CCW. It is a figure which shows the relationship between the rotor rotational position (theta) r of the motor of FIG. 11, the electric current of each winding, and the torque of a CCW direction. The stator is indicated by S, the rotor is indicated by R, and the horizontal axis indicates a combination of the number M of stator magnetic poles and the vertical axis indicates the number K of rotor magnetic poles as an SM-RK motor type. FIG. 3 is a diagram showing a cross section of a motor having four rotor magnetic poles at an electrical angle of 360 ° and a winding arrangement of each phase in a three-phase, annular winding, one-way current motor according to an embodiment of the present invention. . Two motors, an outer diameter side motor and an inner diameter side motor, are combined in the radial direction, the stators of both motors are arranged between both motors, and the same-phase windings are mutually connected between the slots of both motors. It is a cross section of the composite motor of 3 phases and 8 poles which is an embodiment of the present invention of the arrangement arranged. 16 is a cross-sectional view of a three-phase, eight-pole composite motor that is an embodiment of the present invention having a configuration in which the circumferential phase of both motors differs by 180 ° in electrical angle with respect to the motor of FIG. 15. FIG. 2 is a diagram illustrating an example of a cross section of a motor including ten rotor magnetic poles, winding arrangements of respective phases, and magnetic fluxes in a stator having tooth tips divided into two as compared with the stator of FIG. 1. FIG. 2 is a diagram showing an example of a cross section of a motor including 16 rotor magnetic poles, a winding arrangement of each phase, and magnetic fluxes in a stator having tooth tips divided into three as compared with the stator of FIG. 1. It is a figure which shows the cross section of the three-phase motor which added the concentrated field winding to the motor of FIG. 1, the winding arrangement | positioning of each phase, and the example of magnetic flux. It is a figure which shows the example of the cross section of the three-phase motor which added the field winding of full-pitch winding to the motor of FIG. 1, the winding arrangement | positioning of each phase, and magnetic flux. It is a figure which shows the current control apparatus of a field winding, the coil | winding in the case of using as a generator, and a rectifier circuit. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated-winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed at the tip of a stator magnetic pole and four rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated-winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed at the tip of a stator magnetic pole and four rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed at the tip of a stator magnetic pole and two rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed at the tip of a stator magnetic pole and two rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. It is a figure which shows the relationship between the magnetic field strength H and the magnetic flux density B of a permanent magnet. A three-phase, full-pitch, concentrated winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed inside a soft iron portion of a stator magnetic pole, and two rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the cross section, the winding arrangement | positioning of each phase, and the example of magnetic flux. It is a figure which shows the example of the form of the permanent magnet arrange | positioned inside the soft iron part of a stator magnetic pole. It is a figure which shows the example of the form of the permanent magnet arrange | positioned inside the soft iron part of a stator magnetic pole. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated-winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed on a back yoke portion of a stator and four rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. 3 is a cross-sectional view of a three-phase, full-pitch, concentrated-winding motor according to an embodiment of the present invention, in which a permanent magnet is disposed in a part of a stator magnetic pole, and two rotor magnetic poles are provided at an electrical angle of 360 °. It is a figure which shows the winding arrangement | positioning of each phase, and the example of magnetic flux. The figure which shows the example of the cross section of a motor which is a motor of 3 phases, concentrated winding, and a unidirectional current which is embodiment of this invention, and has a permanent magnet magnetic pole of 10 poles on the surface of a rotor, and arrangement | positioning of each phase, and magnetic flux It is. The figure which shows the cross section of a motor which is a motor of 4 phases, concentrated winding, and a unidirectional current which is an embodiment of the present invention, and which has a permanent magnet magnetic pole of 12 poles on the surface of a rotor, the winding arrangement of each phase, and an example of magnetic flux It is. The figure which shows the example of the cross section of a motor which is a motor of 5 phases, concentrated winding, and a unidirectional electric current which is embodiment of this invention, and has a permanent magnet magnetic pole of 16 poles on the surface of a rotor, and arrangement | positioning of each phase, and magnetic flux It is. It is a figure showing a longitudinal section and a transverse section of a linear motor having four magnetic poles between sliders with an electrical angle of 360 ° in a three-phase, concentrated winding, one-way current linear motor according to an embodiment of the present invention. . The three-phase, full-pitch winding, concentrated winding, and unidirectional current motor, which is an embodiment of the present invention, includes a cross section of a motor having three rotor magnetic poles between electrical angles of 360 ° and winding arrangements of each phase. It is a figure which shows an example. The three-phase, full-pitch winding, concentrated winding, and unidirectional current motor, which is an embodiment of the present invention, having a cross section of a motor having five rotor magnetic poles between electrical angles of 360 ° and winding arrangement of each phase It is a figure which shows an example. FIG. 2 is a diagram showing an example in which the circumferential width of the rotor magnetic pole is changed, an example in which the shape of the opening of the slot is changed, and an example in which a slit is arranged inside the rotor magnetic pole as compared with the motor shown in FIG. 1. The stator is indicated by S, the rotor is indicated by R, and the horizontal axis shows a cross section of an example in which the motor type SM-RK is a combination of the number M of the stator magnetic poles and the vertical axis shows the number K of the rotor magnetic poles 8S4R. is there. It is a figure which shows the cross section of the motor whose motor type is 8S6R and is an annular winding. It is a figure which shows the cross section of the motor whose motor type is 10S4R. FIG. 42 is a view showing a cross section of the motor in which the slot opening of the motor of FIG. 41 is reduced and the circumferential width of the rotor magnetic pole is increased. It is a figure which shows the cross section of the motor which divided the rotor of the motor of FIG. It is a figure which shows the cross section of the motor which changed the relative positional relationship of the rotor piece divided into 2 of the motor of FIG. It is a figure which shows the cross section of the motor whose motor type is 10S6R. It is a figure which shows the cross section of the motor whose motor type is 10S8R. It is a figure which shows the cross section of the motor whose motor type is 12S10R. It is a figure which shows the cross section of the motor whose motor type is 14S4R. It is a figure which shows the cross section of the motor whose motor type is 14S6R. It is a figure which shows the cross section of the motor whose motor type is 14S8R. It is a figure which shows the cross section of the motor whose motor type is 14S10R. It is a figure which shows the cross section of the motor whose motor type is 14S12R. It is a figure which shows the cross section of a motor with a larger stator magnetic pole width and rotor magnetic pole width with respect to the motor of FIG. It is a figure which shows the cross section of the motor which added the auxiliary salient pole magnetic pole with a small circumferential direction width | variety to the motor of 6S2R of FIG. FIG. 55 is a diagram showing a cross section of a stator core and a rotor core of a motor in which the motor of FIG. 54 is multipolarized to 8 poles. FIG. 54 is a cross-sectional view showing current and magnetic flux that generate CCW torque at each rotor rotational position θr of the motor of FIG. 53; FIG. 57 is a diagram showing the relationship between the rotor rotational position θr of the motor of FIG. 56, the current of each winding, and the torque in the CCW direction. FIG. 3 is a cross-sectional view of the motor showing the relationship among the stator magnetic pole width, slot opening width, rotor main salient pole magnetic pole width, and rotor auxiliary salient pole magnetic pole width of the motor of the present invention. It is a motor cross-sectional view which shows the conditions which do not generate | occur | produce the torque to the direction where the rotor magnetic pole adjacent to the circumferential direction opposes. It is a table | surface which shows the preferable relationship between the main salient pole magnetic pole width Hm of a rotor, the auxiliary salient pole magnetic pole width Hh, and the slot opening part width Hs. It is a figure which shows the example which deform | transforms the shape of a rotor mechanically. It is a figure which shows the example which deform | transforms the shape of a rotor mechanically. FIG. 4 is a cross-sectional view of a motor having a structure in which a circumferential arrangement of a main salient pole magnetic pole and an auxiliary salient pole magnetic pole is asymmetric. It is a figure which shows the cross section of the motor whose motor type is 12S8R. FIG. 2 is a cross-sectional view of a motor in which the motor of FIG. 1 is doubled and the 1/2 rotor magnetic pole is shifted in the circumferential direction. It is a figure which shows the example of the structure which skewed the stator magnetic pole or the rotor magnetic pole, or the shape which deform | transformed. It is a figure which shows the relationship of the torque which can generate | occur | produce between the shape which faces the rotor of a stator magnetic pole, the shape which faces the stator of a rotor magnetic pole. It is a figure which shows the relationship of the torque which can generate | occur | produce between the shape which faces the rotor of a stator magnetic pole, the shape which faces the stator of a rotor magnetic pole. It is a figure of winding arrangement and winding shape in a slot. It is a figure which shows the example of a structure which arrange | positions a conductor to the circumferential direction side surface of a rotor magnetic pole, and a rotor axial direction side surface. It is a figure which shows the example of a structure which arrange | positions a conductor inside the soft magnetic body of the circumferential direction end or rotor axial direction end of a rotor magnetic pole. It is a figure which shows the structure which adds a soft magnetic body to the rotor axial direction end of a stator core. It is a figure which shows the structure which adds a soft magnetic body and a permanent magnet to the rotor axial direction end of a stator core. It is a figure which shows the magnetic characteristic which increases the amount of magnetic flux which can pass by carrying out negative bias of a stator core magnetically. It is a figure which shows the structure which adds the winding which supplies a soft magnetic body and the exciting current for bias to the rotor axial direction end of a stator core. It is a figure which shows the structure which arrange | positions a permanent magnet to the opening part of a slot in the direction opposite to the magnetic flux which passes a stator magnetic pole. It is a figure which shows the structure which adds a soft magnetic body to the rotor axial direction end of a rotor magnetic pole. It is a figure which shows the structure which makes the air gap between a stator magnetic pole and a rotor magnetic pole equivalent small. It is a figure which shows the structure which adds a coil | winding to a rotor magnetic pole and supplies electric power through the slip ring attached to the brush and the rotor. It is a figure which shows the method of implement | achieving the driving | operation of a wide rotation speed range by switching the coil | winding of this invention motor with a contactor. It is a figure which shows the method of selecting the winding wire diameter in a slot, and changing the current density distribution, and the method of reducing the leakage magnetic flux of the slot opening part vicinity by use of a high magnetic flux density material. It is a figure which shows the relationship between the slot and winding arrangement | positioning of the conventional 3 phase alternating current motor. FIG. 83 is a longitudinal sectional view showing a shape in the vicinity of a coil end of the motor of FIG. 82. It is a figure which shows the winding arrangement | positioning of the rotor axial direction side surface and coil end part of the three-phase, four pole motor of this invention. It is a figure which shows the coil | winding arrangement | positioning of the rotor axial direction side surface and coil end part of the 3-phase 6-pole motor of this invention. It is a longitudinal cross-sectional view of the AE-AE cross section of the motor of FIG. It is a longitudinal cross-sectional view of the AF-AF cross section of the motor of FIG. It is a figure which shows the core shape and winding shape of the rotor axial direction vicinity of the slot of this invention motor. FIG. 89 is a side view of the stator shown in FIGS. 86, 87, and 88 viewed from the rotor axial direction. It is a figure which shows this invention motor of the structure which divides a stator core into four. It is an example of the longitudinal cross-sectional view of the motor shown in FIG. FIG. 92 is a longitudinal sectional view of a configuration in which the stator and rotor support structure of the motor of FIG. 91 are supported by bearings on both sides in the axial direction. FIG. 93 is a longitudinal sectional view of a motor having a configuration in which a motor case E1D is added to the motor of FIG. 92. FIG. 53 is a cross-sectional view of a motor having a configuration in which each winding of the motor of FIG. 52 is bifilar wound or two coils arranged in parallel. FIG. 95 is a diagram showing a configuration of a control device that drives the motor of FIG. 94 with three power transistors. FIG. 95 is a diagram showing a configuration of a control device that drives the motor of FIG. 94 with three power transistors. FIG. 95 is a diagram showing a configuration of a control device that drives the motor of FIG. 94 with three power transistors. It is a figure which shows the structure of the control apparatus which drives the three-phase motor of this invention. It is a figure which shows the structure which arrange | positions soft iron and a permanent magnet on the side surface of a stator core, and restrict | limits the quantity of the magnetic flux which passes a rotor and a stator. It is a figure which shows the structure which restrict | limits the quantity of the magnetic flux which passes along a rotor and a stator with the magnetic flux of the permanent magnet arrange | positioned to the opening part of a slot. It is a figure which shows the structure of the control apparatus comprised by eight power transistors, and the coil | winding of a three-phase motor. It is a figure which shows the structure of the control apparatus comprised by 12 power transistors, and the winding of a three-phase motor. It is a figure which shows the method of performing gradually the change accompanying rotation of radial attraction force by setting the rotor magnetic pole width to 75 in the three-phase motor of the present invention. It is a figure which shows the electric current and torque of each phase winding of the motor of FIG. It is a figure which shows the method of making the rotor magnetic pole width of the motor of FIG. 1 into 45 degrees, and performing the change accompanying rotation of radial attraction force gently. It is a figure which shows the electric current and torque of each phase winding of the motor of FIG. It is a figure which shows the structure of the control apparatus which controls the electric current and voltage of the motor of this invention precisely and rapidly using the flux linkage information of each winding. It is a figure which shows expressing a motor with the electric current I, the voltage V, the magnetic flux linkage number (PSI) of each winding, and rotational angular velocity (omega) r. 4 is a table showing, as an image, data on the number of magnetic flux linkages of a motor under conditions of each winding current I of the motor and each rotor rotation angle θr. 4 is a table showing data such as the number of magnetic flux linkages Ψ, current, and voltage of the motor under the conditions of each motor torque T and each rotor rotation angle θr. It is a figure which shows the structure of the control apparatus which added the structure which subtracts the measured value of a torque from the command value of a torque and feeds back to the control apparatus of FIG. It is a figure which shows the structure which generates a torque alternately with a main salient pole magnetic pole and an auxiliary salient pole magnetic pole in low speed rotation, and generates a torque with a main salient pole magnetic pole at high speed. It is a figure which shows the characteristic which produces | generates a torque alternately with a main salient pole magnetic pole and an auxiliary salient pole magnetic pole in low speed rotation, and changes the burden ratio of a torque gradually as it becomes high speed. It is an example of the figure which shows the relationship between rotor rotational position (theta) r, the electric current value of a coil | winding, and a linkage flux. It is a figure which shows the inductance characteristic example of each coil | winding of the three-phase motor of this invention, such as FIG. It is a figure which shows the longitudinal cross-section of the conventional motor. It is a figure which shows the cross section of the conventional motor. 116 is a winding diagram of the motor of FIGS. 116 and 117. FIG. It is a figure which shows the connection of a structure of a three-phase alternating current inverter, and a three-phase alternating current motor. It is a figure which shows the example of the structure of the coil | winding of the conventional switched reluctance motor, and magnetic flux.

  Embodiments of the present invention will be described below with reference to the drawings. A motor according to an embodiment of the present invention is shown in FIG. A01, A02, A03, A04, A05, A06 are stator magnetic poles, and the number M of stator magnetic poles is six. A07, A08, A09, A0A, A0B, and A0C are slots sandwiched between the stator magnetic poles. A0D and A0G are A-phase windings, and A0L indicated by a broken line indicates a winding path of the coil end portion. The winding path A0L is arranged on one side of the circular back yoke, but can also be arranged on both sides in half. Each phase winding is controlled by applying a one-way current in the direction of the symbol, as indicated by a current symbol. That is, it is not direct current but direct current control. These windings are concentrated windings because they are wound with full-pitch winding at an electrical angle of 180 ° pitch, and each winding is concentrated in one slot. A0F and A0J are B-phase windings, and A0M indicated by a broken line indicates a winding path of the coil end portion. A0H and A0E are C-phase windings, and A0N indicated by a broken line indicates a winding path of the coil end portion.

  A0Z is the rotor shaft. The rotor has four salient pole-shaped rotor magnetic poles indicated by A0K arranged at equal intervals on the circumference. The number K of rotor magnetic poles is four. FIG. 1 shows an example in which the circumferential widths of the stator magnetic pole and the rotor magnetic pole are both 30 ° in electrical angle. The angle from the center of the positive winding of the A phase to the end of the rotor magnetic pole in the counterclockwise direction is defined as the rotor rotational position θr.

  Here, the naming method related to the motor model format described in the present invention will be defined. When the number of stator magnetic poles is M and the number of rotor magnetic poles is K, it will be called MSNR. As a specific example, when the number of stator magnetic poles M = 6 and the number of rotor magnetic poles K = 4, the motor model type is referred to as 6S4R.

  When the torque T in the counterclockwise rotation direction CCW is generated at the rotor rotational position θr in FIG. 1, as shown in the winding symbol, the A-phase winding A0D is moved from the front side of the drawing to the back side of the drawing, and the A-phase winding A0G is turned on. By applying a direct current Ia from the back side to the front side of the paper surface, and applying a direct current Ic from the front side of the paper surface to the back side of the paper surface to the C phase winding A0H and from the back side of the paper surface to the front side of the paper surface to the C phase winding A0E. The magnetic flux indicated by the thick arrow A0P is induced, and torque in the counterclockwise direction is generated in the rotor. At this time, no current Ib flows through the B-phase windings A0F and A0J. Further, since the magnetomotive forces of the currents Ia and Ic cancel each other in the direction perpendicular to the magnetic flux indicated by the thick arrow A0P, that is, in the direction of the stator magnetic poles A02, A03 and A05, A06, no magnetic flux is generated. When the magnitudes of the currents Ia and Ic are different, a magnetomotive force proportional to the difference acts, so that the magnetic flux is induced in a direction perpendicular to the magnetic flux indicated by the arrow A0P.

  An example of a control device for energizing the currents Ia, Ib, and Ic of the A-phase, B-phase, and C-phase windings is shown in FIGS. Since the current of each winding is a direct current and a unidirectional current, for example, an inverter (current control means) having a simple configuration as shown in FIG. 2 can be obtained. Reference numerals 561, 562, and 563 denote A-phase, B-phase, and C-phase windings, 564, 565, and 566 denote transistors that conduct current, and 567, 568, and 569 denote diodes. The regenerative current collected by the diodes 567, 568, and 569 is charged into the capacitor 56C, and is converted into a voltage by the transistor 56A that is a DC-DC converter, the choke coil Ldcc, and the diode 56B, and is charged into the DC power supply 53A. This DC-DC converter has a common configuration that is often used. The transistor 56A supplies the current Irc to the choke coil Ldcc by the transistor 56A and turns off the transistor 56A. At this time, the magnetic energy accumulated in the choke coil Ldcc can be charged to the DC power supply 53A in the form of a current Irc using the diode 56B. VM is the voltage of the DC power supply 53A. In this way, the kinetic energy of the motor and the magnetic energy of the motor are regenerated and efficiently regenerated to the DC power source 53A.

  The regenerative voltage VH in FIG. 2 can be set according to the required regenerative characteristics. In particular, when the multi-pole motor of the present invention is rotated and used at a high speed, it is necessary to shorten the time for reducing the current. The decrease in current means regeneration of the magnetic energy of the winding, and the current decrease time can be shortened as the regeneration voltage VH increases. Further, the DC-DC converter or the like can be modified. A snubber circuit that protects each transistor, a diode for blocking reverse current, and the like can be added as necessary.

  As another example, FIG. 3 shows an inverter that controls currents Ia, Ib, and Ic of three windings with six transistors. 84D is a DC voltage power supply. The A-phase winding, B-phase winding, and C-phase winding in FIG. 1 correspond to the windings 87D, 87E, and 87F in FIG. For example, when a current is passed through the winding 87D, it is possible to control the transistors 871 and 872 to apply a DC voltage, regenerate, and control the flywheel. The diodes 877 and 878 can regenerate the energy of the winding 87D to the power source by turning off both the transistors 871 and 872 while the current is supplied to the winding 87D. At this time, if only one of the transistors, for example, the transistor 872 is turned on, a flywheel current can be passed through the loop of the winding 87D, the transistor 872, and the diode 878. The same control can be performed for the B-phase winding 87E, the transistors 873 and 874, and the diodes 879 and 87A. The same control can be performed for the C-phase winding 87F, the transistors 875 and 876, and the diodes 87B and 87C. However, only one-way current can be applied to each three-phase winding. In addition, when there is a possibility that a negative voltage is applied to each transistor due to the convenience of the motor, a protection diode as shown by a diode 56D in FIG. 2 may be added to each transistor. The same applies to the other one-way current driving inverter shown in the present invention.

  Also in the inverter of FIG. 3, in order to shorten the current reduction time, it is possible to increase the regenerative voltage by using a power source connected to each diode as a power source different from the DC power source 84D.

  Next, the rotational position θr of the rotor and the current supplied to each phase winding will be described with reference to FIG. 4 (a) shows the same rotational position θr = 30 ° as in FIG. 1, and the DC currents Ia and Ic are applied to the A-phase windings A0D and A0G and the C-phase windings A0H and A0E as described above. A magnetic flux indicated by an arrow is induced to generate a torque T in the counterclockwise direction CCW. At this time, the B-phase windings A0F and A0J are not energized. Note that the magnetomotive force in the horizontal direction on the paper surface of FIG. 4A, that is, in the direction from the B-phase winding A0F to A0J, cancels because the currents Ia and Ic act in opposite directions. Horizontal magnetic flux is not excited. At this time, the B-phase current Ib is zero, but the voltage Vb of the B-phase winding is such that the magnetic flux φ indicated by the arrow is linked to the B-phase winding, and the voltage Vb = Nw × dφ / dt is appear. Here, Nw is the number of winding turns.

  FIG. 4 (b) shows the rotational position θr = 45 °. The DC currents Ia and Ib are energized to the A-phase windings A0D and A0G and the B-phase windings A0F and A0J to induce the magnetic flux φ indicated by the arrows. A torque T in the counterclockwise rotation direction is generated. The C-phase windings A0H and A0E are not energized.

  (C) of FIG. 4 is the rotational position θr = 60 °, energizes the DC currents Ia and Ib to the A-phase windings A0D and A0G and the B-phase windings A0F and A0J, and induces the magnetic flux indicated by the arrows. Torque T in the counterclockwise direction is generated. The C-phase windings A0H and A0E are not energized.

  (D) of FIG. 4 is the rotational position θr = 30 °, energizes the B-phase windings A0F, A0J and the C-phase windings A0H, A0E with the direct currents Ib and Ic, and induces the magnetic flux indicated by the arrows. Torque T in the counterclockwise direction is generated. The A phase windings A0D and A0G are not energized. Thus, rotational torque can be continuously obtained while changing the winding to be energized according to the rotor rotational position θr. Torque is generated by applying currents Ia, Ib, and Ic in the determined direction to the windings of the slots on both sides in the circumferential direction of the stator magnetic pole that magnetically attracts the rotor magnetic pole, so that Rotational torque T is obtained by generating a magnetic flux φ.

  At this time, what is characteristic is that the currents Ia, Ib, and Ic of each winding are unidirectional currents, and each winding and its current are related to two different electromagnetic actions. The two windings serve as two independent power supply paths, and each winding independently supplies power from the DC power source to the motor at the same time. The fact that each winding can be used for driving two stator magnetic poles also means that a power transistor for driving the current of the winding can also be used. In addition, the reluctance torque also takes advantage of the feature of generating an attractive force in the same direction regardless of the direction of the magnetic flux. As a result, it is possible to reduce the size of the motor and reduce the current capacity of the power transistor of the control device, which will be described later. These features can be applied to other motor types shown later than the motor shown in FIG.

  The motor of the present invention can naturally be controlled by a bidirectional current control means capable of supplying current in both directions, and the present invention does not exclude the bidirectional current control means. As will be described later, by supplying current in both positive and negative directions, it may be possible to improve the average motor output torque, the motor peak output torque, the motor constant output characteristics, and the like.

  Next, the torque T generated by the motor will be considered qualitatively. In motors for automobiles, the torque required for motors during normal operation is ½ or less of the maximum torque, and the frequency of using the maximum torque is low, and the motor efficiency at the maximum torque is not so problematic. There is. In order to reduce the size and cost of the motor in such applications, the maximum torque characteristic is important.

  Now, the torque characteristics are divided into a linear operation region Aa where the soft magnetic material of the portion where the stator teeth and the rotor salient poles face each other is not magnetically saturated, and a nonlinear operation region As where the magnetic saturation is performed. In the motor model of FIG. 1, the rotor rotation position θr is considered near 30 °.

  In the magnetic linear operation area Aa of the soft magnetic material, the magnetic flux density is Bx, the number of winding turns Nw of each phase, the current Ix that is passed through each of the A-phase winding and the C-phase winding, Ix, When voltage Vx, magnetic flux density proportional coefficient Kb, motor rotor axial thickness tc, rotor rotational angular velocity ωr, and rotor radius R, magnetic flux density Bx and interlinkage magnetic flux φ are roughly expressed by the following equation (1). It is given by (5). However, the A-phase current Ia and the C-phase current Ic in FIG. 1 are constant and the B-phase current Ib is zero. Time is indicated by t.

ωr = dθr / dt (1)
Vx = Nw × dφ / dt (2)
= Nw × dφ / dθr × dθr / dt (3)
= Nw × dφ / dθr × ωr (4)
Bx = Kb × Ix × Nw (5)
The rotor angle change rate dφ / dθr of the flux linkage φ expressed in the voltage formula of (4) is expressed by the following formulas (6) and (7).

dφ / dθr≈Δφ / Δθr
= (Tc × ωr × Δt × R × Bx) / (ωr × Δt)
= (Tc * [omega] r * [Delta] t * R * Kb * Ix * Nw) / ([omega] r * [Delta] t) (6)
= Tc * R * Kb * Ix * Nw (7)
At this time, the input power Pin is assumed that the winding resistance Ra is zero, the iron loss Pfe is zero, and the mechanical loss of the motor is zero, and the air gap between the rotor and the stator is sufficiently small. It becomes.

Pin = 2 × Vx × Ix = 2 × Nw × dφ / dt × Ix (8)
Here, the coefficient of 2 is that power is supplied by the A-phase winding and the C-phase winding, so the power of the winding on one side is calculated and doubled.

  The mechanical output Pout is expressed by the following equation (9).

Pout = T × ωr (9)
Therefore, from these equations, the torque T of the motor in the linear operation region Aa is given by the following equation (10).

T = 2 × Nw × dφ / dt × Ix / ωr
= 2 × Nw × dφ / dθr × dθr / dt × Ix / ωr
= 2 × Nw × (tc × R × Kb × Ix × Nw) × ωr × Ix / ωr
= 2 × Nw × tc × R × Kb × Nw × Ix ² (10)
The torque T in the linear operation region Aa is proportional to the proportional coefficient Kb of the magnetic flux density and is a value proportional to the square of the current Ix.

  FIG. 5 shows a characteristic example of the current Ix and torque T of the reluctance motor shown in FIG. The characteristic Trm shown in the bold line is a representative torque characteristic. In this figure, the region from the origin to the operating point Tn is a region where the soft magnetic body of the motor is magnetically almost linear, and the current and torque have characteristics close to a quadratic function. is there. The characteristics of the region from the operating point Tn to the torque saturation point Ts are the magnetic region where the vicinity of the air gap where the stator magnetic pole and the rotor magnetic pole face each other is 2.0 T, and the soft magnetic material in the vicinity thereof is close to magnetic saturation, This is an area to be described next. The region where the current is larger than the operating point Ts is not only in the vicinity of the air gap but also in a region where some part of the motor magnetic path such as the back yoke is magnetically saturated. In the magnetic saturation region of the motor magnetic path, it is also a region where it is difficult to give electric energy to the air gap portion of the motor.

  The characteristic Tgs in FIG. 5 is a characteristic in which the length of the air gap where the stator magnetic pole and the rotor magnetic pole face each other is reduced to about ½ that of the motor having the characteristic Trm. When the air gap length approaches zero, the characteristic approaches the characteristic Tgz. The characteristic Tspm is a torque characteristic of a surface magnet type synchronous motor as shown in FIG. 117, and is an example in which a neodymium, iron, or boron NdFeB based permanent magnet operates at about 1.2 T. The magnetic flux density is about 1.2T, which is smaller than 2.0T of the soft magnetic material, but the magnetic flux on the surface of the permanent magnet is position-dependent, and the rotational change rate dφ / dθr of the interlinkage magnetic flux φ of each winding is doubled. Therefore, it can be designed to have a steeper characteristic than the characteristic Tgz. For example, when the average magnetic flux density of a neodymium, iron, or boron NdFeB-based permanent magnet is 1.0 T, the characteristics Tspm and Tg in FIG. 5 have the same gradient. The gradient of these characteristics has the freedom of motor design.

  On the other hand, the saturation magnetic flux density is obtained as Bsat for the torque T in the nonlinear operation region As where the soft magnetic material in the vicinity of the air gap is magnetically saturated. This is the region from the operating point Tn to Ts in the torque characteristics of FIG. The rotor angle change rate dφ / dθr of the linkage flux φ is expressed by the following equation (11).

dφ / dθr≈Δφ / Δθr
= (Tc × ωr × Δt × R × Bsat) / (ωr × Δt)
= Tc x R x Bsat (11)
The voltage Vx of the winding generating the torque is given by the following equation (12).

Vx = Nw × dφ / dt
= Nw × dφ / dθr × dθr / dt
= Nw × tc × R × Bsat × ωr (12)
The input pin and output torque T of the motor are expressed by the following equations (13) and (14).

Pin = 2 × Vx × Ix = T × ωr (13)
= 2 × Nw × tc × R × Bsat × Ix × ωr (14)
Torque T is given by the following equation (15).

T = 2 × Nw × Vx × Ix / ωr
= 2 × Nw × tc × R × Bsat × ωr × Ix / ωr
= 2 × Nw × tc × R × Bsat × Ix (15)
The torque T in the non-linear operation region As of the soft magnetic material is a value proportional to the saturation magnetic flux density Bsat and the current Ix. In the torque characteristics of FIG. 5, the region is from the operating point Tn to Ts. Naturally, the maximum torque T is also proportional to the number of windings Nw of each phase, the thickness tc of the motor in the rotor axial direction, and the rotor radius R. From this result, it can be qualitatively estimated that the maximum torque characteristic of the motor, which is an important characteristic for reducing the size and cost of the motor, greatly depends on the saturation magnetic flux density Bsat. However, as a condition for satisfying the expression (11), there is no magnetic saturation except for the portion where the stator and the rotor face each other, and the leakage magnetic flux density in the space portion in the circumferential direction of the rotor A0K in FIG. The condition that it is sufficiently small compared with the density is necessary. These conditions are assumed to be equivalent for the motors being compared.

  The magnetic energy Ew related to each winding is expressed by the following equation (16).

Ew = 1/2 × Nw × φ × Ix (16)
For example, in the state of FIG. 4A, when the A-phase current Ia and the C-phase current Ic are the same value of current Ix, the energy of the equation (16) exists in the A-phase winding and the C-phase winding. Since there are two windings, the magnetic energy of the entire motor is twice that of the equation (16). This magnetic energy is needed to calculate the voltage, current and time required to increase and decrease the current.

  The saturation torque density Bsat operating as a motor is important for the maximum torque of the motor, and it can be said that the maximum magnetic flux density is used at the operating point of the reluctance motor of FIG. A magnetic flux density of about 2T (Tesla) can be utilized if it is made of a normal electromagnetic steel sheet. This will be described in comparison with the conventional surface magnet type three-phase AC synchronous motor shown in FIG. Assuming that this permanent magnet is a high-performance magnet having a neodymium, iron, or boron NdFeB-based configuration, and assuming that the average magnetic flux density is 1T, the magnetic flux density of each tooth of the stator in FIG. 117 is the limit of 2T. The magnetic flux density acting on the torque generation of the motor in FIG. 117 acts on the average magnetic flux density of the teeth and slots of the stator. Therefore, the magnetic flux density acts approximately 1T which is ½ of the saturation magnetic flux density Bsat of the teeth. It is a low value in terms of magnetic flux density as compared with the motor. However, the rotational change rate of the interlinkage magnetic flux of each winding of the surface magnet type motor of FIG. 117 is twice that of the reluctance motor of FIG. As is well known, the reason why the magnetic flux is doubled is that the magnetic flux of the permanent magnet is highly position dependent, and the reluctance motor can move freely within the range of the soft magnetic material. As a conclusion, the torque of the reluctance motor shown in FIG. 1 is twice that of the surface magnet type three-phase AC synchronous motor shown in FIG. 117, but is ½ in terms of the position dependency of the magnetic flux. Therefore, it can be said that the total is equivalent. Here, it is assumed that the air gap between the stator and the rotor is sufficiently small, and the excitation load of the reluctance motor of FIG. 1 is small. In the motor model of FIG. 1, when the maximum torque of the motor is output near the rotor rotational position θr = 30 °, it can be said that the soft magnetic material is used up to the limit of the saturation magnetic flux density Bsat.

  Next, the motor driving algorithm shown in FIG. 1 will be described with reference to FIG. 4A is the same position as in FIG. 1, and the currents Ia, Ib, Ic, magnetic flux φ, and torque T are as described above, and torque T in the counterclockwise direction is generated. Next, when rotating to the rotational position θr = 45 ° in FIG. 4B, the torque T in the counterclockwise direction CCW is generated by energizing the A-phase current Ia and the B-phase current Ib. Further, even in the vicinity of the rotational position θr = 60 ° in FIG. 4C, the torque T in the counterclockwise rotation direction is generated by energizing the A-phase current Ia and the B-phase current Ib. And when it rotates to rotation position (theta) r = 75 degree of (d) of FIG. 4, the torque T of a counterclockwise rotation direction will be generate | occur | produced by supplying with electricity B phase current Ib and C phase current Ic. Similarly, as shown in FIG. 6, the rotational torque can be continuously generated and rotated by controlling the currents Ia, Ib, and Ic of each phase according to the rotational position θr of the rotor. Driving in the counterclockwise direction and the clockwise direction is possible, and power running and regeneration are possible.

  FIG. 6 shows the relationship between the motor currents Ia, Ib and Ic shown in FIGS. 1 and 4 and the torques Ta, Tb, Tc and Tm. FIG. 6A shows an A-phase current Ia energized to the windings A0D and A0G in FIG. FIG. 6C shows a B-phase current Ib energized to the windings A0F and A0J of FIG. FIG. 6E shows a C-phase current Ic that is passed through the windings A0H and A0E of FIG. 6B shows the torque Ta applied to the rotor by the stator magnetic poles A01 and A04 of FIG. 6D shows the torque Tb applied to the rotor by the stator magnetic poles A03 and A06 in FIG. FIG. 6F shows the torque Tc applied to the rotor by the stator magnetic poles A05 and A02 of FIG. (G) in FIG. 6 is a value obtained by adding torques Ta, Tb, and Tc, and is the rotational torque Tm of the rotor. The motor model in FIG. 1 is drawn with the circumferential width of the stator magnetic pole and the rotor magnetic pole at 30 ° so that it can be easily understood. Therefore, the torque generated by each stator magnetic pole is changed to the other phase. The torque Tm decreases. By making the circumferential width of the stator magnetic pole and the rotor magnetic pole larger than 30 °, it is possible to reduce the torque drop at the torque transfer portion.

  A basic example of the current supply and control method for each phase current is shown in FIG. However, the present invention is not necessarily limited to this method, and it is possible to drive more effectively by correcting the phase of each current, the magnitude of each current, and the like. For example, in the case of high-speed rotation of a certain degree or more, increase and decrease in response of each current becomes a problem, and it is effective to advance the phase of each phase current. Even in the state where the same current flows through the winding, the magnitude of the magnetic flux φ varies depending on the rotational position, and the magnetic energy expressed by the equation (16) varies. Further, the voltage induced in each winding is generated with rotation. Therefore, the increase in current can be accelerated by applying a voltage in a section where the magnetic energy is small and no positive voltage is induced in the winding. In that sense, advancing the phase of each current is effective for improving the response delay of the current.

  In addition, in order to give an accurate voltage to the winding at an accurate time, it is possible to estimate and calculate the interlinkage magnetic flux of each winding, calculate the winding voltage, and give the accurate voltage by feedforward control. Accurate and fast response current control. This control method will be described later.

  In addition, the magnetic flux of each stator magnetic pole is electromagnetic, and the basic idea is to pass currents in opposite directions to the two slots on both sides in the circumferential direction of the stator magnetic pole to be torque-generated. However, the magnitudes of the currents may be different, and the current may be supplied to all three windings.

  Further, the A-phase current Ia in FIG. 6 flows a constant current from 15 ° to 75 °, the current is zero from 75 ° to 105 °, and the same cycle thereafter. The B-phase current Ib has a relationship delayed by 30 ° with respect to the A-phase current Ia, and the C-phase current Ic has a relationship delayed by 30 ° with respect to the B-phase current Ib. The relationship is such that two phase currents flow at the same time, and when one phase current decreases, the other one phase current increases.

  In addition, as shown by a broken line in FIG. 6, for example, another method in principle is to increase the currents Ia and Ic at a point of 15 °, for example, and to make the currents Ia and Ic zero at a point of 45 °. Immediately after that, the currents Ib and Ia are increased and energized. At the point of 75 °, the currents Ib and Ia are decreased to zero A. Immediately after that, the currents Ic and Ib are increased and energized, and at the point of 105 °. The currents Ic and Ib are decreased to zero A, and immediately after that, the currents Ia and Ic are increased and energized. At a point of 135 °, the currents Ia and Ic are decreased to zero A. It can continue to rotate counterclockwise. With such an algorithm, only the magnetic flux in the intended direction is induced. Note that the temporary decrease in current shown by the broken line in FIG. 6 does not decrease until the current becomes zero, but the effect can be obtained even when the current decreases halfway. Further, at this time, in high-speed rotation over a certain rotation, it is necessary to advance the current phase as shown in FIG. 6 in order to improve the current response.

  Further, as a method different from these, when only one phase current is applied, a state in which magnetic flux is induced in two paths occurs, and the operation becomes complicated. Also, when a current is passed through the three-phase windings, various electromagnetic effects can be created depending on the magnitude of the current. Conversely, torque can be generated by combining these various electromagnetic actions.

  Next, the operation when the brake is applied when the motor shown in FIG. 1 rotates to the CCW, that is, when the regenerative braking is performed will be described with reference to FIGS. The torque depends only on the current, and has no relation in principle with the rotation direction and the rotation speed. If the rotation direction and the torque direction coincide with each other, power running is performed, and if the rotation direction and the torque direction are opposite, regenerative operation is performed.

  The currents Ia, Ib, and Ic, the rotor rotation angle θr, and the torques Ta, Tb, Tc, and Tm of the motors shown in FIGS. 7 and 8 are the same as those shown in FIGS. At θr = 60 ° in FIG. 7A, the rotor is rotating to CCW, and in order to generate CCW torque, the A-phase current Ia is supplied to the A-phase windings A0D and A0G, A C-phase current Ic is applied to the C-phase windings A0H and A0E, and a magnetic flux indicated by an arrow is induced to generate a torque T in the CW direction. The direction of rotation and the direction of torque generation are opposite directions, which is regenerative braking. Next, when rotating to the rotational position θr = 75 ° in FIG. 7B, the A-phase current Ia and the B-phase current Ib are energized to prepare for generating the torque T of the subsequent CW. From the rotation angle θr = 75 ° to 105 °, as represented by the rotation position θr = 90 ° in FIG. 7C, the A-phase current Ia and the B-phase current Ib are energized to pass CW. Torque T is generated. Then, when rotating to the rotational position θr = 105 ° in FIG. 4 (d), the B-phase current Ib and the C-phase current Ic are energized, so that the CW torque is maintained between the rotational position θr and 105 ° to 135 °. Prepare to generate T. Similarly, by controlling the currents Ia, Ib, and Ic of each phase according to the rotational position θr of the rotor, it is possible to continuously generate rotational torque and perform regenerative braking.

  FIG. 8 shows the relationship between the rotor rotational position θr, currents Ia, Ib, Ic and torque T shown in FIG. The rotor rotates to CCW, and currents Ia, Ib, Ic and torques Ta, Tb, Tc of each phase when CCW torque is generated. In FIG. 8, since the torques Ta, Tb, and Tc are regenerative torques, they are shown as negative values. Further, the current and torque in FIG. 8 are in a relationship delayed by 30 ° compared to the timing at the time of powering shown in FIG. This is because the angle at which the CCW torque is generated and the angle at which the CW torque is generated in the motor of FIG. (A) of FIG. 8 is an A-phase current Ia energized to the A-phase windings A0D and A0G of FIG. (C) of FIG. 8 is the B-phase current Ib energized to the B-phase windings A0F and A0J of FIG. (E) in FIG. 8 is a C-phase current Ic energized to the C-phase windings A0H and A0E in FIG. FIG. 8B shows the torque Ta applied to the rotor by the stator magnetic poles A01 and A04 of FIG. 8D shows the torque Tb applied to the rotor by the stator magnetic poles A03 and A06 in FIG. FIG. 8F shows the torque Tc applied to the rotor by the stator magnetic poles A05 and A02 of FIG. (G) in FIG. 8 is a value obtained by adding torques Ta, Tb, and Tc, and is the rotational torque Tm of the entire rotor.

  The motor model in FIG. 7 is drawn with the circumferential width of the stator magnetic pole and the rotor magnetic pole being 30 ° so that the motor model can be easily understood. The torque Tm decreases. By making the circumferential width of the stator magnetic pole and the rotor magnetic pole larger than 30 °, it is possible to reduce the torque drop at the torque transfer portion. Further, when the current of each phase is increased or decreased at a phase slightly earlier than the current phase of each phase shown in FIG. 8, the increase and decrease of each phase current can be accelerated due to the influence of the voltage induced in each winding. Further, although depending on the characteristics of the stator magnetic pole and the rotor magnetic pole, as shown by the broken line in FIG.

  The major features of the motor and its control device shown in FIG. 1 are that each winding current is a direct current, and that the current of each winding is also used in a configuration that can contribute to the torque generation of the adjacent stator magnetic poles. The four-quadrant operation of the clockwise direction CW and the counterclockwise direction CCW and the positive torque and the reverse torque is possible by increasing or decreasing the number of direct currents.

  In these characteristics, the motor configuration and the control device configuration are closely related, and the downsizing of the inverter can be realized. An example in the case of the control device of FIG. 2 will be described. Assume that the voltage of the DC power supply 53A is 200V and the current capacity of each transistor is 10A. Now, the rotor of the motor shown in FIG. 1 is rotating at a certain rotational speed ωr, and is the rotational position approaching the rotational position θr = 30 ° shown in FIG. 4A, and the A-phase windings A0D, A0G, It is assumed that a current of 10 A is passed through winding 561 and C-phase windings A0H and A0E, that is, windings 563, respectively. At this time, it is assumed that the magnetic flux density between the stator magnetic pole and the rotor magnetic pole facing the stator is about 2.0 T, which is magnetically saturated with a current of 10 A and is the saturation magnetic flux density. The voltage Vx of the A-phase winding and the C-phase winding is expressed by equation (12). Here, as shown in FIG. 4A, the interlinkage magnetic flux φ of the A-phase winding and the C-phase winding has the same value, and the rotational change rate dφ / dt of the interlinkage magnetic flux φ is also the same. It is the same value. Then, it is assumed that the voltage represented by the equation (12) is just 200V. At this time, the electric power P1, which is the output of the inverter and the input / output of the motor, is expressed by the following equation (17).

P1 = (200V) × (10A) × (2 windings) (17)
= 4000 [W]
On the other hand, the inverter shown in FIG. 119 is a conventional three-phase AC inverter and is usually used well. And about the thing which star-connected the three-phase AC motor connected to this inverter, the maximum output is verified. The voltage of the DC power supply 84D is 200V, and the current capacity of each transistor is 10A. For example, assuming that 200 V is applied from the U-phase winding 834 to the V-phase winding 835 and a maximum current of 10 A is applied, the output P2 at that time is expressed by the following equations (18) and (19).

P2 = (200V) × (10A) (18)
= 2000 [W] (19)
Note that the same amount of power is supplied when the U-phase winding 834 and the V-phase winding 835 and the W-phase winding 836 are energized half by half. That is, in the system of FIG. 119, when the induced voltage of the motor winding is a value close to the DC power supply 84D, the maximum current of the transistor being used and the peak current of the three-phase sine wave current are 3 When a phase sine wave current is applied, approximately the same motor output can be obtained regardless of the phase of the three-phase current.

  When the combination of the motor of FIG. 1 and the control device of FIG. 2 is compared with the normal three-phase AC motor of FIG. 119 and the inverter, the output of 4000 W is 3 transistors and the output of 2000 W is 6 transistors. When the output per one is compared, it becomes four times. Comparing under the same output conditions, the motor of FIG. 1 and the control device of FIG. 2 can be three transistors, which is half the number of transistors, and the current capacity of the transistors is half 5A, and can output 2000 W of the same output. Become.

  Note that in the configuration of FIG. 2, a DC-DC converter including the transistor 56A or the like is necessary, and it should be noted that the withstand voltages of the transistors 564, 565, and 566 need to be greater than 200V.

  In general, 50 to 100 or more motors for automobile auxiliary machines with a battery voltage of 12V are used. When the DC-DC converter of FIG. 2 is shared by a plurality of motors and the driving of each motor is driven by three transistors, the driving device can be greatly simplified.

  Moreover, in an electric vehicle, a hybrid vehicle, etc., two or more motors are often used for driving a vehicle. Usually, the fuel consumption in the city driving mode, that is, the driving efficiency is a problem. Although it depends on the type of automobile, it is often less than or equal to ½ of the maximum motor torque in the urban driving mode. Accordingly, a motor-side power generation capacity during regeneration, that is, a regeneration capacity, that is 1/2 or less of the maximum output capacity during rapid acceleration is sufficient. When an automobile must decelerate rapidly, it can be considered that it is sufficient to use a mechanical brake function together from the viewpoint of safety. From such a viewpoint, the DC-DC converter as shown in FIG. 2 can be shared by a plurality of motors, and a driving circuit for driving one motor has three transistors 564, 565, 566. And three diodes 567, 568 and 569. Therefore, the inverter shown by the broken line in FIG. 2 is a simple inverter that can drive one motor by three transistors and three diodes, and can reduce the cost. At the same time, the forward voltage drop of the current and the voltage drop at the diode during regeneration are about l / 2 compared to a normal three-phase AC inverter as shown in FIG. 119, which is efficient and generates little heat. In this respect, the inverter can be downsized.

  Next, the case of the control device shown in FIG. 3 will be described. This is a DC three-phase inverter. 84D is a DC voltage source. 87D is an A-phase winding and is driven by transistors 871 and 872. 87E is a B-phase winding and is driven by transistors 873 and 874. 87F is a C-phase winding and is driven by transistors 875 and 876. Regenerative diodes are attached before and after each winding. In the case of the control device of FIG. 3, the DC-DC converter as shown in FIG. 2 is not necessary.

  Now, when the voltage of the DC voltage source 84D is 200V and the current capacity of each transistor is 10A, when the maximum voltage and the maximum current are output to the A-phase winding and the C-phase winding, the maximum output P3 is expressed by the following formula ( 20).

P3 = (200V) × (10A) × (2 windings) (20)
= 4000 [W]
In the configuration in which the three-phase AC motor is connected to the three-phase AC inverter shown in FIG. 119 with a star connection, the maximum output P2 is 2000 W. Therefore, the motor shown in FIG. 1 and FIG. In the case of the same output, the same level of output is possible with a current capacity of 1/2, that is, a current capacity of 5A. As a motor system, the cost can be significantly reduced in this case as compared with the conventional motor system.

  A proposal of a motor having a part similar in appearance, such as a switched reluctance motor shown in FIG. 120, and its control device is known in a publicly known patent or conference presentation, but the current capacity of the inverter is reduced to 1/4 or 1 / There is no proposal other than the DC motor system, and the motor and control device of the present invention is a motor system that has not existed so far.

  Even in the case of a DC motor, four transistors are required for the variable speed four-quadrant operation. The current capacity of each transistor must be twice the current capacity of the transistor used in the motor of the present invention and the control device of FIG. 3, and the control device of the motor system of the present invention is relatively 6 / (4 × 2) = 6/8, and the cost can be reduced and the size can be reduced. Further, in the case of the DC motor and in the case of the combination of the motor of the present invention in FIG. 1 and the control device in FIG. 2, the current capacity becomes 3/8, and the cost can be further reduced.

  The motor system shown in FIGS. 1, 2, and 3 has been described as being able to reduce the cost and size of the inverter due to its motor characteristics. Other features will be described below. The motor of FIG. 1 is inexpensive because it does not use expensive rare earth magnets, and there is no problem of rare earth metal resource depletion and price increase. Compared with the switched reluctance motor of FIG. 120, the thickness of the winding in the slot can be doubled, so the winding resistance is small. However, the motor of FIG. 1 is disadvantageous in that it has a long coil end, and it is necessary to reduce the burden by increasing the number of poles in a motor having a small stator core axial thickness. Since the motor of FIG. 1 has a robust rotor, it is physically easy to use high-speed rotation, and high output is possible. The torque of the motor in FIG. 1 uses the attractive force generated between the stator magnetic pole and the rotor magnetic pole, and the principle of torque generation is simple, and it is easy to obtain a characteristic with a relatively small torque ripple. Vibration and low noise can be achieved. However, it should be noted that sudden fluctuations in the suction force in the radial direction cause vibration of the stator core. Since the motor shown in FIG. 1 is a reluctance motor, no permanent magnet is used, and no magnetic flux is present inside the motor when no current is applied. Will not occur. This occurs when a gasoline engine is used for a hybrid vehicle or the like and travels with a gasoline engine during high-speed traveling.

  In addition, although the example of the present invention has been described, various modifications and combinations are possible. Below, the example of various deformation | transformation is demonstrated. First, the shape of the motor illustrated in the present invention including the motor shown in FIG. 1 is an example, and various modifications are possible. The circumferential width Ht of the stator magnetic poles is illustrated as Ht = 360 ° / (6 × 2) = 30 ° for ease of understanding of the configuration when the number of stator magnetic poles M = 6, for example. By setting the Ht and the rotor magnetic pole width Hr to an angle larger than 30 °, the circumferential width in which the stator magnetic pole generates torque can be increased. Then, the rotor magnetic pole width Hr is set to 30 ° or more, and more continuous rotational torque can be generated at the current switching unit.

  Conversely, in order to facilitate the increase of the current of each phase winding, current is applied to the windings of the slots adjacent to each other in the circumferential direction of the slot at the rotational position θr where the stator magnetic pole and the rotor magnetic pole do not face each other. What is necessary is just to supply. For that purpose, it is advantageous that the stator magnetic pole width Ht and the rotor magnetic pole width Hr are smaller than 30 °. In conclusion, it is necessary to select the values of Ht and Hr by various driving methods, and both cases are included in the present invention.

  Further, the width of the opening of the slot can be reduced, and the slot can be driven like a synchronous motor. In addition, the rotor magnetic poles are arranged uniformly in the circumferential direction, and a model with a constant circumferential width is shown in FIG. 1, but in order to reduce torque ripple, the rotor magnetic poles are arranged unevenly or at low speed. As for the driving method in the above and the driving method at high speed rotation, a wide rotor magnetic pole and a narrow rotor magnetic pole can be arranged for generating torque or reducing vibration.

  The shape of the stator magnetic pole and rotor magnetic pole is also shown in FIG. 1 as a simple salient pole shape, but can be variously modified in the axial direction, the circumferential direction, and the radial direction. In particular, in the vicinity of both ends of each stator magnetic pole and rotor magnetic pole in the circumferential direction, the magnetic flux is rapidly increased when the rotor magnetic pole approaches the stator magnetic pole by deforming in the radial direction so that the air gap between the stator and the rotor is increased. Therefore, vibration and noise due to a sudden change in the radial suction force can be reduced.

  Also, the current values of the respective phases are shown as ON and OFF as basic algorithms in FIGS. 4 and 11 and the like, but since they can be driven by the difference in attractive force, for example, all windings It is also possible to apply a current to the wire and generate torque by the difference between the currents. That is, this is a method of superimposing and adding a superimposed current Iset to each of the A-phase, B-phase, and C-phase winding currents Ia, Ib, and Ic shown in FIGS. The superposed current Iset can be set to a constant current value or a current value at a constant ratio of the current amplitude value, or can be defined by various control conditions. The superimposed current Iset can also be considered as a field current that excites the magnetic flux of all the stator magnetic poles.

  This superimposed current Iset has advantages and disadvantages depending on the motor type. For example, one problem with the motor of the present invention is vibration and noise of the motor. One of the causes of this problem is a change accompanying the rotation of the suction force in the radial direction of the motor and a change in the suction force accompanying an increase / decrease in current. In this respect, by exciting each stator magnetic pole to some extent, it is possible to reduce fluctuations in the radial attractive force and reduce vibration and noise. In some cases, the response when the winding currents Ia, Ib, and Ic increase or decrease can be improved.

  Further, the motor torque T is a constant torque as shown in FIGS. 4 and 11 and the torque ripple does not become zero. Due to leakage magnetic flux, soft magnetic non-linearity, etc., the motor torque T is accurately a complex function and value depending on the current values Ia, Ib, Ic and the phase angle θre of the rotor electrical angle. . Therefore, in order for the motor to rotate while generating a constant torque T, a current obtained by adding a current for correcting the torque error to each phase current Ia, Ib, Ic as shown in FIGS. There is a need.

  In addition, the current value of each phase has been described in many cases when the number of stator magnetic poles is M = 6 and there are three phases of A phase, B phase, and C phase. It is necessary to expand each phase current according to the number.

  The motor is illustrated as a two-pole motor. However, when the outer diameter of the motor is larger than 100 mm, for example, it is often multipolar. The length of the coil end is reduced to about ¼, and the burden such as the rate of increase in winding resistance due to the coil end of each phase winding can be greatly reduced.

  Further, in FIG. 1 and the like, the arrangement of the coil ends is shown by one path from the slot. However, even if the coil ends are divided into two or three and wound around other slots of the same phase, they are electromagnetically the same. The winding path can be deformed.

  Further, in the motor shown in FIG. 1 and the like, the winding is a full-pitch winding, the winding pitch is 180 °, and concentrated winding is shown. However, some modifications are possible, and almost the same effect can be obtained. Those modified versions are also included in the present invention. One of the deformations is a slot shape, and the slot can be divided into two to form distributed winding. Although the winding is complicated, there is an effect that the torque fluctuation is smooth, and there is also an effect that the winding can be easily inserted when the amount of one winding is large.

  As another modification, the winding pitch is not limited to 180 °, and a similar effect can be obtained even if the winding is somewhat short. If the windings are wound so that the phase windings of slots adjacent to each other in the circumferential direction overlap each other, the windings are complicated, but when the phase current is switched, torque fluctuations are smoothed. It is also possible to obtain an effect.

  Next, FIG. 9 shows an example of the present invention in which the number of stator magnetic poles M = 6 and the number of rotor magnetic poles K = 2. For ease of explanation, the motor is a two-pole motor, and the motor model type is 6S2R.

  Each slot of the stator 11F is wound with full-pitch winding and concentrated winding. 111 and 114 are A-phase windings, 113 and 116 are B-phase windings, and 115 and 112 are C-phase windings. Winding. The teeth 117, 118, 119, 11A, 11B, and 11C sandwiched between the slots constitute salient poles. The width of the tip of the tooth is Ht, the width of the opening of the slot is Hs, and the sum of both widths (Hs + Ht) is 60 ° in electrical angle. The rotational position of the rotor is indicated by θr. Compared to the stator of FIG. 1, the circumferential width Ht of the stator magnetic pole is larger than the width Hs of the slot opening.

  The rotor 11E is a salient pole-shaped rotor made of a soft magnetic material. Most of the portion of 11D is a space, and it is possible to embed a nonmagnetic material for the purpose of reducing windage loss during rotation. The width of the rotor magnetic pole is Hm as shown.

  The motor of FIG. 9 shows an example of a two-pole motor whose operation is easy to explain. However, it can be multi-polarized like other motors of the present invention. Show. The slots of the stator 12T are 121, 122, 123, 124, 125, 126, 127, 128, 129, 12A, 12B, 12C, 12D, 12E, 12F, 12J, 12K, 12L, 12M, 12N, 12P, 12Q, 12R. , 12S, and the rotor 12U has eight salient poles 12V.

  Next, the operation of the motor of FIG. 9 will be described with reference to (a) to (f) of FIG. In this example, the width Hs of the opening of the slot is 20 °, and the width Hm of the rotor magnetic pole is 40 °. FIG. 12 shows the rotor rotation position θr expressed in electrical angle on the horizontal axis, the current of each phase, and the torque of each phase. The A-phase current is Ia, the B-phase current is Ib, and the C-phase current is Ic.

  The torque of the motor shown in FIGS. 9, 10, and 11 is such that the winding is full-pitch winding and concentrated winding, and the stator teeth 117, 118, 119, 11A, 11B, and 11C are arranged almost all around. Therefore, in order to generate torque, current is passed through at least two windings to generate torque. Then, a reluctance torque is obtained by generating an attractive force between the salient pole-like teeth of the stator and the salient pole of the rotor. The torque generated between the stator teeth 117 and 11A and the rotor magnetic pole 11E is Ta, the torque generated between the stator teeth 119 and 11C and the rotor magnetic pole 11E is Tb, and the stator teeth 11B and 118 are The torque generated between the rotor magnetic poles 11E is Tc. It should be noted that at this time, each attractive force generates the same attractive force and torque regardless of whether the direction of the magnetic flux is positive or negative.

  When the rotor is in the vicinity of the rotational position of θr = 30 ° shown in FIG. 11A, a positive current Ia is supplied to the A-phase winding 131 and a negative current − is supplied to the A-phase winding 134 on the opposite side. Run Ia. At the same time, a positive current Ic is supplied to the C-phase winding 135 and a negative current -Ic is supplied to the opposite C-phase winding 132. No current flows through the B-phase windings 133 and 136. The currents Ia, Ib, and Ic in each phase are the currents shown in (A), (C), and (E) of FIG. In this state, in accordance with Ampere's law, the magnetomotive forces of the A-phase current Ia and the C-phase current Ic act in the direction indicated by the thick arrows from the stator salient poles 11A to 117 in the direction indicated by the arrows. Magnetic flux is induced in the direction of 117. Then, torque Ta shown in FIG. 12B is generated in the counterclockwise direction CCW. Here, the permeability of the soft magnetic part of the stator and the rotor is sufficiently large, the permeability of the wide space between the stator and the rotor is sufficiently small, and the magnetic resistance of the narrow air gap between the stator and the rotor is sufficiently large In a simple model that is assumed to be small, the magnetic field strength [A / m] acting in the vicinity of the stator teeth 118, 119, 11B, and 11C is almost zero, and the magnetic flux passing through these teeth in the radial direction is almost zero. And the torque is almost zero.

  When the rotor rotates to CCW and rotates to the vicinity of the rotation position of θr = 50 ° shown in FIG. 11B, a positive current Ic flows through the C-phase winding 135 and a negative current − Ic is flowed. At the same time, a positive current Ib is supplied to the B-phase winding 133, and a negative current −Ib is supplied to the opposite B-phase winding 136. No current flows through the A-phase windings 131 and 134. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the bold arrows in the direction of the stator salient poles 118 to 11B, and the magnetic flux is induced in the direction indicated by the arrows. The Then, the rotor generates a torque Tc shown in FIG. However, the magnetic flux passes through the air portion in the vicinity of the slot openings of the windings 132 and 135, and since the magnetic resistance is large, the magnetic flux density is small and the torque Tc is not large. As the rotor magnetic pole 11E approaches the stator teeth 118 and 11B, the torque Tc increases rapidly.

  When the rotor rotates to CCW and rotates to near the rotation position of θr = 70 ° shown in FIG. 11C, the rotor further rotates with the torque Tc generated in CCW under the same current condition. Similarly, the torque Tc shown in FIG. 12F is generated from the rotor rotational position θr = 90 ° to θr = 110 ° in FIG.

  When the rotor rotates to the vicinity of the rotation position of θr = 110 ° shown in FIG. 11E, a positive current Ib flows through the B-phase winding 133 and a negative current −Ib flows through the B-phase winding 136. At the same time, a positive current Ia is supplied to the A-phase winding 131, and a negative current −Ia is supplied to the opposite A-phase winding 134. No current flows through the C-phase windings 131 and 134. In this state, in accordance with Ampere's law, magnetomotive forces of the A-phase current Ia and the B-phase current Ib act in the direction indicated by the thick arrows in the direction from the stator salient poles 11C to 119, and magnetic flux is induced in the direction indicated by the arrows. The Then, the rotor generates a torque Tb shown in FIG. When θr is from 110 ° to around 120 °, the magnetic flux passes through the air portions near the windings 133 and 136, so that the magnetic resistance is large and the generated torque Ta is small. The torque Tb increases rapidly in the vicinity of the rotor rotational position θr = 130 ° in FIG.

  Here, in FIGS. 11A and 11D, the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed. In FIGS. 11B and 11E, the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed. In FIGS. 11 (c) and 11 (f), the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed.

  As shown in FIGS. 11 and 12, the rotor can be rotated by changing the current that is sequentially applied depending on the rotor rotational position θr. The total torque Tm of the motor that has transferred the generated torques Ta, Tb, and Tc of each tooth is shown by a solid line in FIG. When the rotation direction end of the rotor magnetic pole 11E reaches the opening of the slot, the torque Tm decreases. There are many applications in which such a partial torque reduction is not a problem. In order to reduce this torque drop, there is a method of skewing the stator or the rotor, and other methods will be described later.

  In these operations shown in FIG. 11, the direction of the magnetic flux of each salient pole of the stator is the same direction, the direction of the magnetic flux of the rotor is reversed depending on the rotational position, and the A phase current Ia, B phase current Ib, C phase current Ic. These directions can be driven by a current in one direction. It is important that these three-phase currents are used in two current modes, and that each winding is also used. Also, by combining these three types of one-way currents, clockwise torque and counterclockwise torque can be generated continuously, and power running and regeneration are possible, so four-quadrant operation is possible. It can be said that there is. Therefore, it can be driven by the control device of FIG. 2 or the control device of FIG. 3, and the cost and size of the control device can be reduced.

  As for the magnitude of the current, the explanation has been made assuming that the magnitude of the current in each phase is the same, but it is also possible to change the current balance of each phase or to pass the current in all three phases. The rotor magnetic pole width Hm, the tooth tip width Ht, and the slot opening width Hs can take different values. A control method for reducing fluctuations in the radial attractive force of the motor when the rotor magnetic pole width Hm is 40 ° or more will be described later. As for the tip shape of the teeth of the stator, a simple salient pole shape has been illustrated and described, but various modifications such as a structure that narrows the opening of the slot and a structure that widens the air gap with the rotor at the circumferential end of the tooth Is possible. Similarly, various modifications can be made to the shape of the rotor magnetic poles.

  Moreover, although the said winding method of the winding of the motor shown to FIG. 1, FIG. 9 was demonstrated like lap winding, you may wind by a wave winding. Note that the full-pitch windings have an electrical angle of 360 ° pitch, and the pitch between the windings going back and forth in the rotor axial direction of each slot is 180 ° in electrical angle. 1 and 9, the difference between lap winding and wave winding is difficult to understand because it is a two-pole motor, but the difference between lap winding and wave winding appears when it is multipolarized like the eight-pole motor in FIG. 10. . A specific example in which the A-phase winding is a wave winding is shown in FIG. 121, 127, 12D and 12M are slots for winding the A-phase winding, and 124, 12A, 12J and 12Q are slots for winding the negative A-phase winding. 12X indicated by a solid line is a coil end of the A-phase winding on the front side of the paper surface, and 12Y indicated by a broken line is a coil end disposed on the back side of the paper surface. The wave winding is alternately arranged at both ends of the stator core in the rotor axial direction like the coil ends 12X and 12Y and winds one phase winding. The other B-phase and C-phase windings are similarly wound as wave windings of the respective phases.

  As described above, the connection relationship outside the slot has a degree of freedom, and may be lap winding, wave winding, or other winding methods. Electromagnetically the same action. However, these winding methods are deeply related to the structure of the winding machine for mass production and automation, and are selected according to the production method. The same applies to motors other than those shown in FIG. Further, not only concentrated winding in which one phase winding is wound in one slot but also distributed winding is possible. The spacing in the circumferential direction of the slots is not always uniform. For example, there is a method in which in-phase windings are distributed windings wound around two adjacent slots, and the circumferential width of teeth between these in-phase adjacent windings is made smaller than the others.

  Although the 6S4R motor of 6S4R in FIG. 1 and the 6S2R motor of 6S2R in FIG. 9 have been described as motor types, many motor types shown in FIG. 13 can be similarly realized. The horizontal axis in FIG. 13 is the number M of stator magnetic poles, and the vertical axis is the number K of rotor magnetic poles. FIG. 13 shows a combination of motor types up to a maximum value 14 of M and a maximum value 14 of K. Among them, combinations of integer multiples of a certain number of combinations of M and K include those in which a 2-pole model is multipolarized as 4-pole, 6-pole, 8-pole,. Even if the multi-pole type is omitted, many more motor types on the extension of FIG. 13 are possible. Those are also included in the present invention. Even if the number of M and K increases somewhat, the number of power transistors of the control device increases, but the total output capacity of the control device does not change just by dispersion, so the motor type with a small number of M and K Compared with the motor system, the sum of the voltage and current capacity of the power transistor is basically the same. In addition, if the number of M and K is large, the number of locations of the stator magnetic pole and the rotor magnetic pole where the force can be generated in the entire motor increases, and if the phases are different, a torque ripple canceling effect can be obtained and the merit is also improved. is there. However, it cannot be denied that the controller becomes complicated when the number of M and K increases, and it is not a good idea to increase the number of M and K extremely. Some combinations of the number M of the stator magnetic poles and the number K of the rotor magnetic poles will be described later.

  Next, a motor of the present invention having a different winding method will be described with reference to FIG. The motor of FIG. 14 is a motor in which the winding of the motor of FIG. In this motor, the number of stator magnetic poles M = 6 and the number of rotor magnetic poles K = 4. The windings A41 and A42 are A-phase windings, which are annularly wound and concentratedly wound to form a so-called toroidal winding. A47 and A48 are negative A-phase windings that are concentrated in the form of an annular winding. The windings A45 and A46 are B-phase windings that are concentrated in an annular manner. A4B and A4C are negative B-phase windings that are concentrated in an annular shape. The windings A49 and A4A are C-phase windings that are concentrated in the form of an annular winding. A43 and A44 are negative C-phase windings, which are annularly wound and wound intensively.

  The windings A42, A44, A46, A48, A4A, and A4C located outside the stator core can only constitute a closed magnetic circuit whose electromagnetic action passes through the outer space of the stator. The influence of the currents of these windings on the magnetic action inside the motor is negligibly small. Therefore, the motor of FIG. 14 exhibits almost the same magnetic characteristics as the motor of FIG.

  These six sets of windings can individually control the unidirectional current. The control device of FIG. 2 controls the current and voltage of the three windings, but the number of phases of this control device may be expanded to a six-phase control device using six power transistors. In the case of the control device in FIG. 3 as well, the number of elements may be doubled to obtain a six-phase control device. When the three-phase control device is a six-phase control device, if the voltage of each of the six windings is designed to be the same voltage as that of a three-phase motor, the number of power transistors is doubled. Since the current capacity can be halved, the output capacity of the control device is the same, and there is no significant difference in cost.

  If the motor output capacities are the same, the total value of the current capacities of the power transistors can be made the same in principle even if the number of phases is increased. Therefore, assuming that the price of the power transistor is proportional to the current capacity, the output capacity and cost of the power portion are the same regardless of the number of phases. However, in the case of a motor with a small output capacity of 100 W or less, for example, the ratio of the number of parts that affects the cost increases, and therefore a cost problem may occur when the number of phases increases.

  14 is equivalent to the A-phase winding of FIG. 1 if the windings A41 and A42 and A47 and A48 are connected in series so as to have the same current direction. It becomes. The windings A45 and A46 and A4B and A4C are equivalent to the B-phase winding of FIG. 1 if they are connected in series with a jumper so that the current directions are the same. The windings A49 and A4A and A43 and A44 are equivalent to the C-phase winding in FIG. When these windings are connected, the motor of FIG. 14 becomes a three-phase motor equivalent to the motor of FIG. 1, and is driven by the three-phase control device of FIG. 2 or the three-phase control device of FIG. Can do.

  In the motor shown in FIG. 14, the outer winding of the stator core is hardly useful electromagnetically. However, when the stator core has a small axial thickness in the rotor axis and the number of poles is small, the annular winding is the winding. In some cases, the length is short, and the windings are well organized, so that a practical motor structure can be obtained.

  The annular winding motor of FIG. 14 is an example of the motor type 6S4R, but can be realized for various motor types as shown in FIG. Further, FIG. 13 shows the case where the number M of the stator magnetic poles and the number K of the rotor magnetic poles are even numbers. However, even if each is an odd number, one winding or several windings are currents in both positive and negative directions. Can be energized, the other parts can be driven in the same way as when the number of M and K is an even number, and a motor with an odd number of M or K is also included in the present invention.

  Next, a motor of the present invention having a different configuration will be described with reference to FIG. It is a motor having a composite structure in which motors are incorporated on the outer diameter side and the inner diameter side. FIG. 15 shows an 8-pole motor. The motor shown in FIG. 9 has 8 poles, the rotor R1 is disposed on the outermost diameter side, the stator S1 corresponding to the rotor R1 is disposed on the inner side, and the inner diameter side of the stator S1 is disposed. A stator S2 is arranged, and a rotor R2 that acts on the stator S2 is arranged on the inner diameter side of the stator S2. That is, the motor type 6S2R is a multi-pole motor with 8 poles, and two motors are combined in the outer shape and the inner diameter. The rotor configuration shown in FIG. 1 or other motors can be similarly realized.

  In the configuration of FIG. 15, since the outer diameter side stator S1 and the inner diameter side stator S2 are arranged back to back, the currents in the back to back outer diameter side slot and the inner diameter side slot are just opposite currents. If designed in this way, the windings of each phase can be wound around the back-to-back slots, simplifying the windings and shortening the coil end length.

  The windings 46B, 46H, and 46Q are A-phase windings, and the direction of the current is the direction of the winding symbol, and a one-way current is applied. The windings 46E and 46L are negative A-phase windings. The windings 46D and 46K are B-phase windings. The windings 46G and 46N are negative B-phase windings. The windings 46F and 46M are C-phase windings. The windings 46C and 46J are negative C-phase windings. These windings can be treated as 6-phase windings. Therefore, the unidirectional current control device shown in FIGS. 2 and 3 can be controlled in six phases by connecting it to the motor shown in FIG. In addition, since the current capacity of the power transistor can be significantly reduced as compared with the normally used three-phase AC inverter, the cost and size can be reduced.

  Alternatively, the A-phase winding and the negative A-phase winding are connected in series so that the current direction is the same by a jumper, and similarly, the B-phase winding and the negative B-phase winding Can be driven by the three-phase one-way current control device shown in FIGS. 2 and 3, which is also low cost. And miniaturization are possible.

  FIG. 13 shows a list of many motor types. In particular, when the number M of the stator magnetic poles is increased, many windings intersect in the case of full-pitch winding, and the winding becomes complicated. The coil end portion tends to be large at the same time that the ease of manufacture becomes worse. In this regard, in the case of the configuration shown in FIG. 15, even if the number M of the stator magnetic poles is increased, each winding only has to be wound between the back-to-back slots. It won't fall and the windings won't be complicated. In the case of the annular winding, the winding does not become complicated even if the number M of the stator magnetic poles is increased.

  In addition, the outer diameter motor and the inner diameter motor have different diameters, so the electromagnetic conditions are different. When the two motors are electromagnetically optimized, the currents of the two motors have different values, and the above winding causes inconvenience. There is a problem to do. In order to solve this problem, it is possible to add a full-pitch winding as shown in the motor of FIG. 46R and 46U are A-phase windings, 46S and 46V are B-phase windings, and 46T and 46W are C-phase windings. These additional windings contribute to the electromagnetic action of the motor on the outer diameter side, and by optimizing both motors, higher output, smaller size, and lower cost can be realized.

  Next, an example of a composite motor having another configuration will be described with reference to FIG. Similarly to FIG. 15, the motor has a composite structure in which motors are incorporated on the outer diameter side and the inner diameter side, but the phases of the motor on the outer diameter side and the motor on the inner diameter side differ by 180 ° in electrical angle. FIG. 16 shows an 8-pole motor. The motor shown in FIG. 9 has 8 poles, the rotor R1 is disposed on the outermost diameter side, the stator S3 corresponding to the rotor R1 is disposed on the inner side, and the stator S3 is disposed on the inner diameter side of the stator S3. A stator S4 is disposed, and a rotor R2 that acts on the stator S4 is disposed on the inner diameter side of the stator S4. That is, the motor type 6S2R is a multi-pole motor with 8 poles, and two motors are combined in the outer shape and the inner diameter. The rotor configuration shown in FIG. 1 or other motors can be similarly realized.

  The rotor R1 and the rotor R2 are the same as the motor of FIG. Stator teeth G08, G09, G0A, G0B, G0C, G0D, G0E, G0F, G0G, G0H, G0J, G0K, G0L, G0M are the teeth of the outer diameter side stator S3 and the inner diameter side stator S4. It is integrated. The polarity of the teeth of the stator S3 on the outer diameter side and the stator S4 on the inner diameter side have opposite characteristics. For example, the magnetic flux passing through the rotor magnetic pole 461 of the rotor R1 on the outer diameter side passes through the teeth G08 or G09 of the stator. The rotor magnetic pole 466 of the rotor 2 on the side passes through. Therefore, the back yokes of the stator S3 and the stator S4 are not necessary, and the composite teeth G09 and the like are arranged on the circumference. In addition, it is necessary to fix these teeth from a rotor axial direction side surface etc. using some fixing means. Care must be taken with respect to the magnetic characteristics of the fixing means.

  The winding of the motor of FIG. 16 can be configured as a configuration in which the winding of the stator S3 on the outer diameter side and the winding of the stator S4 on the inner diameter side are integrated. The coil end portions of the windings are indicated by broken lines, G02 and G05 are A-phase windings, G03 and G06 are B-phase windings, and G01, G04 and G07 are C-phase windings. In either case, a one-way current represented by a winding symbol is applied. The three-phase current can be driven by the one-way current control device shown in FIGS. 2 and 3, and the cost and size can be reduced as described above. At this time, a unidirectional current is passed through each winding of the motors shown in FIGS. 15 and 16, and the unidirectional magnetic flux passes through each stator magnetic pole to act electromagnetically.

  In the case of the composite motor of FIG. 16, the amount of magnetic flux of the motor on the outer diameter side and the amount of magnetic flux of the motor on the inner diameter side need to be the same value, and the amount of magnetic flux of the motor on the inner diameter side tends to be slightly excessive. As this correspondence, an intermediate motor between the motors of FIGS. 15 and 16 can be realized. That is, the back yoke is arranged between the outer diameter side stator and the inner diameter side stator by the amount of magnetic flux imbalance. The windings in FIG. 16 are divided into two sets, and windings similar to the windings in FIG. 16 may be wound around the slots of the outer side stator and the inner diameter side stator, respectively.

  FIG. 15 shows a motor configuration in which two motors are effectively arranged in the outer diameter direction and the inner diameter direction. Two motors having an axial gap structure are arranged in the rotor axial direction, and both ends are respectively rotors. It is also possible to arrange the stators back to back at the center in the rotor axial direction. This combination is also a modification of the motor shown in FIG. 15 and is included in the present invention.

  As another form of the composite motor, the combination of the arrangement of the stator and the rotor is such that the stator S5 is arranged on the outermost side, the stator S6 is arranged on the innermost side, and the combined rotor R3 is placed between S5 and S6. And R4 can also be arranged. At this time, windings similar to FIG. 16 may be wound around the stators S5 and S6.

  Next, the present invention for improving the torque by devising the shape of the stator magnetic pole and the shape of the rotor magnetic pole will be described with reference to FIGS. The motor shown in FIG. 17 divides the tips of the teeth of each stator into two parts as compared with the motor shown in FIG. Specifically, A51 and A52, A53 and A54, A55 and A56, A57 and A58, A59 and A5A, and A5B and A5C. The number of rotor magnetic poles is 10 which is obtained by adding the increased number of teeth 6 to the number of stator magnetic poles 4 in FIG. 1, and the pitch of the rotor magnetic poles substantially matches the pitch of the divided teeth of the stator. A5K and coil ends A5S and A5N are A-phase windings. A5M and coil ends A5T and A5Q are B-phase windings. A5P and coil ends A5U and A5L are C-phase windings. The operation is similar to the operation shown in FIG. 4, and the A-phase current Ia, the B-phase current Ib, and the C-phase current Ic are the same current. However, the rotation angle of the rotor is reduced by the corresponding amount because the circumferential width of the rotor magnetic pole is reduced. As a result of the above, the motor of FIG. 17 is twice the number of teeth generating torque in comparison with the motor of FIG. 4, so the torque is doubled and the rotational speed is the tooth width. The motor is slowed by the ratio of

  Next, the motor of FIG. 18 divides the tips of the teeth of each stator into three parts as compared with the motor of FIG. Specifically, they are A61 and A62 and A63, A64 and A65 and A66, A67 and A68 and A69, A6A and A6B and A6C, A6D and A6E and A6F, A6G and A6H and A6J. The number of rotor magnetic poles is 16 which is obtained by adding the increased number of teeth 6 × 2 to the number of rotor magnetic poles 4 in FIG. 1, and the pitch of the rotor magnetic poles substantially matches the pitch of the divided teeth of the stator. A6R and coil ends A5S and A6U are A-phase windings. A6T and coil ends A5T and A6W are B-phase windings. A6V and coil ends A5U and A6S are C-phase windings. The operation is similar to the operation shown in FIG. 4, and the A-phase current Ia, the B-phase current Ib, and the C-phase current Ic are the same current. However, the rotation angle of the rotor is reduced by the corresponding amount because the circumferential width of the rotor magnetic pole is reduced. As a result, the motor shown in FIG. 18 has three times the number of teeth in principle in comparison with the motor shown in FIG. 4, so that the torque is tripled and the rotational speed is slower by the ratio of the tooth width. It is a motor.

  The motor shown in FIGS. 17 and 18 is a motor in which the number of teeth of each stator magnetic pole is increased twice or three times to increase the torque constant. Compared with the motor shown in FIG. 1, an increase in torque constant and torque can be expected, and the efficiency of the motor is improved. However, in many cases, the peak torque of the motor does not improve so much because magnetic flux is partially generated because leakage magnetic flux is generated around the stator magnetic pole and the rotor magnetic pole. As a countermeasure, the permanent magnet is arranged in the concave portion between the teeth in the direction opposite to the direction of the magnetic flux of the stator magnetic pole, thereby reducing the leakage magnetic flux between the teeth and improving the torque. Is possible. Further, the torque can be increased by disposing a magnet from the side surface of the tooth to the bottom of the recess to increase the magnitude of the magnetic flux passing to the tooth portion. Both approaches can also be used.

Next, a motor in which a field winding is added to the motor of FIG. 1 will be described with reference to FIG. Specifically, in the stator A0P1, the field windings A72 and A73 are wound around the stator magnetic pole A01 in a concentrated manner as indicated by broken lines, and the field windings A75 and A74 are wound around the stator magnetic pole A02. Winding in concentrated winding, winding field windings A76 and A77 around stator magnetic pole A03 in concentrated winding, winding field windings A79 and A78 around stator magnetic pole A04 in concentrated winding, and stator magnetic poles Field windings A7A and A7B are wound around A05 in a concentrated winding, and field windings A71 and A7C are wound around a stator magnetic pole A06 in a concentrated winding. The direction of the current is the direction of the symbol of each winding. Each field winding is connected in series with a jumper so that the direction of current is the same, and a field current If is supplied by a field current drive circuit as shown in FIG. A87 is a DC voltage source, A88 is a flywheel diode, A81, A82, A83, A84, A85, and A86 are each field winding shown in FIG. 19, A89 is a field current driving transistor, and A92 is a field current If. When the field current If, which is a current sensor to detect, is energized and the other phase currents Ia, Ib, and Ic are zero, the magnetic fluxes A7D and A7E indicated by the thick arrows are induced in the rotor A0K1, and the rotational torque Cancels and becomes zero. When the rotor rotates counterclockwise in this state, the magnetic flux interlinked with each field winding changes with rotation, but the total magnetic flux interlinked with the six field windings does not change. Therefore, as shown in FIG. 21, when the field windings are connected in series and the field current If is applied to rotate in the counterclockwise direction, the magnetic fluxes of the stator magnetic poles A01 and A04 increase, but A02 and A05. The magnetic flux decreases and the total magnetic flux does not change. At this time, the magnetic energy of each winding in FIG. 21 is transferred between the windings. Therefore, in principle, no power is transferred from the DC power source side to the motor field winding side.

  One problem in driving the motor of FIG. 1 at high speed is the supply of field magnetic flux and the burden of regeneration. For example, when rotating in the counterclockwise direction, it is necessary to supply a predetermined current to the windings A72 and A78 and the windings A79 and A73 as quickly as possible immediately before or immediately after the rotor magnetic pole A0K reaches the stator magnetic pole A01. Thereafter, a rotational torque is generated between the stator magnetic pole A01 and the rotor magnetic pole A0K to rotate, and the currents flowing through the windings A72 and A78 and the windings A79 and A73 immediately before the two magnetic poles face each other in front. Must be reduced to zero A as soon as possible. That is, one of the problems is to suddenly increase the current at a predetermined timing and then rapidly decrease the current at a predetermined timing.

  As shown in FIGS. 19 and 21, when the field current If is flowing and the field magnetic flux is established, the increase / decrease in the magnetic flux accompanying the increase / decrease in the currents of the windings A72, A78 and A79, A73 is increased. The current control is easy because the voltage is small and the burden of voltage is small. In addition, the total voltage variation of the six field windings shown in FIG. 21 is small, a relatively thin winding can be wound many times, and the field current can be controlled by a relatively small transistor A89. it can. As a result, the burden on the transistor of the control device of FIGS. 2 and 3 can be reduced, and the capacity of the control device can be reduced. It can also be seen that the power factor of the power supplied by the control device of FIGS. 2 and 3 is improved.

  In addition, when regenerating or using as a generator, the field winding configuration of FIGS. 19 and 21 is advantageous in that the power factor can be improved.

  Further, although the field winding shown in FIG. 19 is wound around each stator magnetic pole individually, it can be a full-pitch field winding as shown in FIG. H71 and H74 are A-phase field windings, H75 and H72 are B-phase field windings, and H73 and H76 are C-phase field windings. These three windings are wound in series with their current directions aligned, and a field current is applied. It is also possible to apply current to the three windings individually.

  Since the number of field windings of these full-pitch windings and concentrated windings is three, when applied to the control of FIG. 21, instead of six windings such as the field winding A81, three series windings are provided. Become a line. Further, the field winding may be formed of an annular winding as shown in FIG. Further, as shown in FIG. 15, a motor can be combined to form a field winding in which windings are wound around slots of both motors.

  The section and method for energizing the motor windings of FIG. 1 have already been described with reference to FIG. 4. When a torque in a certain rotational direction is generated, the rotor is rotating in the rotational direction and the rotor magnetic pole is approaching a certain stator magnetic pole. The currents in the opposite directions may be applied to the two windings in the slots on both sides in the circumferential direction of the stator magnetic pole. For this purpose, the current flowing through the windings in the slots in the circumferential direction is arranged such that positive currents and negative currents are alternately arranged in slots adjacent in the circumferential direction. At this time, a positive magnetic flux and a negative magnetic flux are alternately induced in the direction of the magnetic flux that is sequentially induced along with the rotation in each stator magnetic pole adjacent in the circumferential direction.

  The following conditions are important in order to exhibit these features in the motor and the control device of the present invention. In order to supply current and voltage more easily, the current in each winding must be unidirectional. Since each winding and its current can be used for two electromagnetic actions, the positive and negative directions of the winding current in the slots adjacent to each stator pole adjacent in the circumferential direction must be opposite. This is because the current in the winding of a slot can contribute to each of the stator poles adjacent to that slot, respectively, ie each winding can be used for two different functions, That is. By sequentially performing these operations as the rotor rotates, forward rotation, reverse rotation, positive torque, and negative torque of the rotor can be generated almost continuously with the winding current, that is, four-quadrant operation can be performed. is there. It is also important that the rotor magnetic poles are made of soft iron, the direction of the magnetic flux can be made in both directions, and the attractive force acts regardless of the direction of the magnetic flux.

  At this time, in order to be able to be used for different actions of the adjacent stator magnetic poles, it is necessary to connect the windings of the motors so that the current can be controlled independently. The simplest method that can control the current of each winding independently is a method in which voltage and current can be applied to each winding of the motor by an individual transistor, and specific examples are the connection of the windings of FIGS. The windings are arranged in parallel to the DC power source and current and voltage are individually applied, or the windings of the motor are separated and the current and voltage are individually applied. The fact that the winding can also be used not only means that the substantial resistance value of the motor winding can be reduced, but also means that the power transistor that drives the current of the winding can also be used.

  When these conditions are satisfied, the current to be driven is direct current, and each current can also be used for driving different stator poles of the motor, so that the power transistor can also be used, and the total current capacity of all the power transistors can be reduced, As a result, the control device can be reduced in size. Specifically, by combining the motor of the present invention shown in FIGS. 1 and 9 and the control device shown in FIGS. 2 and 3 or a control device in which these are multiphased as necessary, Cost reduction can be realized. Further, how much the current capacity of the control device can be reduced depends on the type of the motor of the present invention.

  In the case of the motor of FIG. 1, two of the three windings contribute to torque generation, and the path of the control device that supplies power to the two windings is another path, A characteristic is that power can be supplied simultaneously in each of the two paths.

  In contrast, in the case of the conventional three-phase AC inverter and three-phase AC motor shown in FIG. 119, the maximum output of the inverter is the product of the voltage of the DC voltage source and the current capacity of one transistor, and one power supply path It can be driven with only a minute output. That is, only the output for one power supply path can be supplied and driven by six transistors.

  In the motor and the control device of the present invention, when the number M of the stator magnetic poles is a large number, as described later, simultaneous power supply from three or more paths is possible.

  The connection method of each winding can simplify current control means such as a transistor for driving a current by connecting in-phase windings or anti-phase windings in series so that the current directions coincide with each other. For example, when the motor of FIG. 1 which is the motor of the present invention has four poles, two A-phase windings, B-phase windings and C-phase windings are arranged on the circumference, so that the windings of the respective phases are arranged. Two pieces can be wound in series. In the case of 6 poles, three in-phase windings can be wound in series. Of course, the currents to be supplied to the windings having different phases are separately supplied by different current control means and supplied with electric power.

  In the motor of the annular winding shown in FIG. 14, the windings A41 and A42 and the windings A47 and A48 are currents of opposite phases, and can be wound in series with a jumper in accordance with the energization direction.

  When the motor of FIG. 1 is rotated in the counterclockwise direction, when the stator magnetic pole A01 attracts the rotor magnetic pole A0K, a reverse current is simultaneously supplied to the windings A0D and A0E. Next, the stator magnetic pole A02 When attracting the rotor magnetic pole A0K, current is supplied to the winding A0E and the winding A0F, and current is not supplied to the winding A0D. In this case, the windings A0D and A0E may conduct the same current, but may have different current values. Such a current phase is neither the same phase nor a reverse phase. The A-phase current Ia energized in the winding A0D and the B-phase current Ib energized in the winding A0E are currents having different phases.

  The motor shown in FIG. 1, FIG. 9, FIG. 14, FIG. 15 and the like is controlled by supplying the A phase current Ia, the B phase current Ib, and the C phase current Ic with the control device as shown in FIG. The motor can be driven to rotate. At this time, each coil is energized with a current having at least two energization paths and supplying power simultaneously by two or more current control means.

  For example, when driven by the control device of FIG. 98, power can be supplied through two paths, so that the two current control means can supply power of (current capacity × power supply voltage × 2). The three-phase AC inverter shown in FIG. 119 can supply twice as much power as compared to (current capacity × power supply voltage × 1).

  In the control device as shown in FIG. 2 and FIG. 3, as described above, as a condition that power can be supplied simultaneously with two or more paths, that is, a condition that the current capacity of the control device can be reduced, the current of each winding is a one-way current. There is a connection of a motor that can control the current of each winding independently, and both adjacent stator poles adjacent to each other in the circumferential direction can be excited in two torque generation modes, and the winding can be used in both modes. It is that you are. The motor characteristics and the connection of each winding and the control device characteristics are all closely related.

  For example, the switched reluctance motor shown in FIG. 120 can be driven by a control device as shown in FIGS. 2 and 3, but cannot supply power through two paths simultaneously. Therefore, the current capacity of the transistor of the control device cannot be reduced as in the motor and the control device of the present invention. 120 differs from the motor shown in FIG. 120 in a winding method, a connecting method for connecting wires, a current application method, a control method, and the like. Both motor systems have large differences in motor efficiency, motor size and cost, control device efficiency, control device size and cost.

  As already explained, a field winding as shown in FIG. 19 can be applied. Then, as shown in FIG. 21, these field windings are connected in series with a jumper so that the current directions are the same, and a field current If is allowed to flow to constitute a relatively simple generator. it can. The windings A0D and A0G are A-phase windings and are the winding A9L of FIG. The windings A0F and A0J are B-phase windings, which are the winding A9M in FIG. The windings A0H and A0E are C-phase windings and are the winding A9N of FIG. A9H, A9J, and A9K are rectifying diodes that rectify three-phase voltages to obtain a DC voltage and a DC current. A9F is a choke coil for rectification, and A9P is a capacitor or a battery. A9Q is a load. A control element such as an IGBT can be used instead of the rectifier diodes A9H, A9J, and A9K. Further, a DC-DC converter can be configured by adding a control element such as IGBT between the choke coil A9F and the common line A9R and adding a diode between the choke coil A9F and the capacitor A9P. A DC-DC converter can also be arranged after the capacitor A9P. As shown in FIG. 21, a generator having a relatively simple configuration can be configured by supplying a field current If to the field winding.

  The power generation in the motor configuration in FIG. 19 assumes that the rotor magnetic pole A0K is now rotating in the counterclockwise direction CCW, faces the stator magnetic pole A01, faces directly in front, rotates from the rotor position to the CCW, and When the area facing the rotor magnetic pole decreases, a generated voltage is generated in a direction that prevents the magnetic flux from decreasing in the windings A0D and A0E and the windings A0H and A0E, and a generated current flows. The same applies to other stator magnetic poles and windings. Further, since the generated current flows through the two-phase winding of the three phases, the efficiency is good.

  An alternator, which is a generator for automobiles, is required to generate power of about 12V to 14V, which is a battery voltage, in a wide range of rotation speeds from 1,000 rpm to 10,000 rpm, for example. Moreover, the level of cost requirement is high and it is necessary to realize with a simple configuration. At present, most of the alternators used in automobiles around the world are said to be of the Landel type. A rotor pole with a claw pole structure is arranged on the rotor side, and the rotor shaft is wound around the rotor axis. A DC field winding is wound. Direct current is supplied from the stator side through a slip ring and a brush in a contact manner. Since an AC voltage is output to the three-phase AC winding on the stator side, the three-phase AC voltage is rectified with a diode to produce 12V DC. Current alternator issues include slip ring and brush life and reliability issues and rotor complexity. Another problem is that it is difficult to increase the output of the alternator due to the relationship between the winding of the rotor structure and the magnetic path. In addition, due to the claw pole structure of the rotor, there is a problem that the rotor is deformed at high speed rotation, and there is also a problem that the air gap between the stator and the rotor is enlarged.

  In FIG. 21, A93 is a voltage command, A9G is an output terminal of this generator, A9S is an output voltage signal, A9A is a voltage error by an adder, and the output is a field current command by a compensator A9B such as proportional or integral. . A9C is an adder which calculates a field current error from the difference between the field current command and the output of the field current detection means A92, and determines a field current voltage command by the compensators A9D and A9E such as proportional and integral. A so-called PWM signal is generated by a width modulator to generate a drive signal for the transistor A89. Thus, by controlling the field current If while monitoring the generated voltage, it is possible to generate a stable constant voltage in a wide range of rotation. The active power element is only the transistor A89 for field current, and the windings and rectifier of the generator can be realized at a low cost with a simple configuration. And since there is no brush and slip ring for field current, reliability is also improved. Although vibration and noise are important issues, the shape of each magnetic pole can be variously modified with respect to the detailed configuration of each part.

  An example of an alternator configured by the motor of the present invention is a configuration in which the motor structure with field winding shown in FIG. 19 is multipolarized to about 8 poles, and both the field winding and the power generation winding are stators. It is possible to provide a non-contact type highly reliable and simple alternator. The reason why the field winding can be arranged on the stator side and can be excited by direct current in the present invention is that the motor of the present invention functions with a unidirectional current, and the magnetic flux passing through each stator magnetic pole is also a unidirectional magnetic flux. Is attributed. This is a feature unique to the motor of the present invention. In the case of exciting a field from the stator side with a conventional AC generator, a three-phase AC field current is required, which is complicated and expensive. In addition, since the field winding and the power generation winding are on the stator side, it is relatively easy to increase the size and output. In addition, since the rotor of the motor of the present invention is a lump of iron, it can be constructed very robustly, and the amount of deformation of the rotor is small even at high speed rotation, and the air gap between the stator and the rotor can be reduced to the limit of machining and assembly accuracy. And speeding up is possible. As described above, the configuration of the alternator according to the present invention shown in FIGS. 19 and 21 can solve most of the problems of the conventional Landell type alternator.

  Note that the motor of the present invention can also generate power as a regenerative braking operation of the motor even when the field winding is not provided. That is, it is possible to generate electric power with the motor of FIG. 1 and the control device of FIG. 2 or FIG. The operation state is the same operation as in the case of FIGS. 7 and 8 in which the motor of FIG. 1 performs regenerative braking, and energy is regenerated from the motor side to the DC power source side. That is, it generates electricity. However, when power generation is performed using a motor drive device in this way, the number of transistors that are active power elements increases and is often somewhat expensive compared to the configuration of FIG. .

  Next, a motor in which a permanent magnet is added to the surface of the stator magnetic pole of the motor of the present invention will be described with reference to FIGS. The motor shown in FIG. 22 has the basic structure of the motor shown in FIG. 1, and has permanent magnets B07, B08, B09, B0A, B0B, and B0C arranged on the surface of the stator magnetic pole. B01, B02, B03, B04, B05, and B06 are the stator magnetic poles. A phase current Ia is supplied to A phase windings A0D and A0G, B phase current Ib is supplied to B phase windings A0F and A0J, and C phase current Ic is supplied to C phase windings A0H and A0E. The direction of current is the direction of the symbol shown in the figure, and a one-way current is applied.

  The direction of the magnetic pole of the permanent magnet is the direction shown in the figure, and the direction of the permanent magnet of a certain stator magnetic pole is applied when a current flows in the direction of the winding symbol shown in FIG. The direction of magnetomotive force. Since the stator side magnetic flux of the motor of the present invention is excited by a three-phase unidirectional current and the magnetic flux of each stator magnetic pole becomes a unidirectional magnetic flux, a motor configuration in which a permanent magnet is arranged on the stator side is possible.

  The magnetic flux in the motor is not simple because there are a plurality of magnetic paths and the magnetomotive force of the magnet and the magnetomotive force of the current act. Magnetic fluxes indicated by thick arrow lines H11, H12, H13, and H14 are induced at the rotor rotation position in FIG. 22, and each change as the rotor rotates. The torque of the motor is proportional to the rotational change rate of the interlinkage magnetic flux of the winding and the current of the winding, as shown by the equation (10). When the CCW torque is generated at the rotor rotational position of FIG. 22, the rotational change rate of the magnetic flux linked to the C-phase windings A0H and A0E is large, so that the CCW torque can be generated by energizing the C-phase current Ic. . Since the magnetic flux H12 does not change before and after this rotational position, it does not contribute to torque generation.

  At this time, assuming that the magnetic flux density in the portion where the permanent magnet and the rotor magnetic pole are opposed is a constant value Bx, and the magnetic flux in the portion where the rotor magnetic pole is not opposed is zero or does not affect the operation of the motor. Since the interlinkage magnetic flux of the A-phase windings A0D and A0G does not change before and after the rotor rotation position, torque generation due to the A-phase current Ia becomes zero. The operation is different from that of the motor shown in FIG.

  However, the actual magnetic circuit of the motor is in a state different from the above assumption, and by appropriately designing the magnetic circuit of the motor, it is possible to obtain characteristics that are convenient for the motor system. The convenient characteristic is that the motor torque is large and the current capacity of the power transistor of the control device can be reduced.

  FIG. 26 shows an example of the characteristics of the magnetic field strength H and the magnetic flux density B of the permanent magnet. The magnetic flux density Br at the operating point of Bh is the residual magnetic flux density, the operating point of Bn is the demagnetization limit point of the permanent magnet, and the magnetic flux density is B14. For example, if the operating point when the motor current is zero is Bk and the operating point is changed from Bm to Bj by the current of each winding, the magnetic flux density of the permanent magnet changes from B13 to B11. The torque generating action of the motor in FIG. 22 is also different from the above description. Specifically, when CCW torque is generated in FIG. 22, the A-phase windings A0D and A0G are energized with the A-phase current Ia, and the C-phase windings A0H and A0E are simultaneously supplied with the same C-phase current as Ia. When Ic is energized, the permanent magnets B07 and B0A exert a magnetomotive force in the direction of increasing the magnetic flux, the magnetic flux density increases at the operating point Bj in FIG. 26, and the magnetomotive force is exerted on the other permanent magnets B08, B09, B0B, B0C. Since it does not act, its operating point Bk does not change. At this time, the rotation change rate of the magnetic flux H12 increases in proportion to the increase value of the magnetic flux density at the operating point, the torque generated by the current Ic increases, and the torque is proportional to the difference between the magnetic fluxes B11 and B12 by the current Ia. Can be generated.

  Further, when the current Ic is larger than the current Ia, a magnetomotive force that weakens the magnetic fluxes of the permanent magnets B09 and B0C is generated in proportion to (Ic−Ia), and the magnetic flux density decreases to B13 at the Bm point. . And the torque by the electric current Ia increases by the magnetic flux fall of the permanent magnets B09 and B0C. At this time, the torque Tc generated by the C-phase currents Ac of the C-phase windings A0H and A0E is a permanent magnet characteristic and is proportional to (B11 + B13), and the torque Ta generated by the A-phase currents Ia of the A-phase windings A0D and A0G Is proportional to the difference (B11−B13) between the magnetic flux density B11 of the permanent magnets B07 and B0A and the magnetic flux density B13 of the permanent magnets B09 and B0C. In this way, with the motor of FIG. 22, electric power can be supplied to the motor from the A-phase winding and the C-phase winding, and the current capacity of the power transistor of the control device can be reduced.

  The motor shown in FIG. 22 can be driven by the control device shown in FIGS. 2 and 3 to reduce the current capacity of the power transistor. By adding a permanent magnet, the torque is increased from the characteristic Trm shown in FIG. 5 to the characteristic Tspm. It can be improved. In particular, in a small motor, the improvement in torque in a low current region is significant. Further, the A phase current Ia, the B phase current Ib, and the C phase current Ic correspond to so-called torque currents, and the inductance of each winding is reduced by the permanent magnet, so that the current response is remarkably improved, and the motor Can be controlled.

  At this time, if the current -Ib in the direction opposite to the winding symbol can be applied to the B-phase windings A0F and A0J, the magnetic flux H11 can be strengthened, so that the torque can be increased. The motor shown in FIG. 22 can be driven by a conventional three-phase AC inverter shown in FIG. 119 by changing the winding connection to a star connection.

  Next, the motor shown in FIG. 23 will be described. This motor has a configuration in which permanent magnets are arranged on the stator magnetic pole surface of the motor shown in FIG. The influence of the current of each winding on the characteristics of FIG. 26 of each permanent magnet is the same as that of the motor of FIG. When CCW torque is generated in the state of FIG. 23, the A phase current Ia is supplied to the A phase windings A0D and A0G, and the C phase current Ic is supplied to the C phase windings A0H and A0E. At this time, the magnetic flux H15 is strengthened, and the magnetic flux H16 does not change. 23, the relationship between the A-phase current and the C-phase current of the motor of FIG. 22 is reversed, and the torque Ta generated by the A-phase current Ia of the A-phase windings A0D and A0G is the permanent magnet F67, The torque Tc generated by the C-phase windings A0H and A0E and the C-phase current Ic in proportion to the sum of the magnetic flux density B11 of F6A and the magnetic flux density B13 of the permanent magnets F68 and F6B (B11 + B13) is a permanent magnet characteristic (B11-B13). ). The features of the motor of FIG. 23 are almost the same as the features of the motor of FIG.

  As described above, in the motor characteristics of FIG. 22 and the motor characteristics of FIG. 23, the torque generation ratio between the A-phase current Ia and the C-phase current Ic energized in the circumferentially adjacent slots of the stator magnetic pole that generates torque, Alternatively, since the ratio of the induced voltages is reversed, the balance between the A-phase current Ia and the C-phase current Ic is improved by superimposing these two types of motor configurations in the rotor axis direction. be able to. At this time, the three-phase windings penetrate the same winding in the rotor axial direction. By improving this balance, the current capacity of the power transistor of the control device can be reduced.

  At this time, if the current -Ib in the direction opposite to the winding symbol can be applied to the B-phase windings A0F and A0J, the magnetic flux H11 can be strengthened, so that the torque can be increased. The motor shown in FIG. 23 can be driven by a conventional three-phase AC inverter shown in FIG. 119 by changing the winding connection to a star connection.

  22 and 23 can have various characteristics by changing the shape of the permanent magnet on the surface of the stator magnetic pole. Specifically, the stator magnetic pole surface has a configuration in which a permanent magnet and a soft magnetic material are mixed, and the ratio of the permanent magnet is Rspm. When this ratio is Rspm = 1, the stator magnetic pole surfaces are all permanent magnets as shown in FIG. 22, and when the ratio is Rspm = 0.5, the permanent magnet and the soft magnetic material are 50% each, and the ratio is Rspm = 0. As shown in FIG. 1, the stator magnetic poles are all made of a soft magnetic material. By changing the ratio Rspm from 0 to 1, various intermediate characteristics from the motor characteristics of FIG. 1 to the motor characteristics of FIG. 22 can be obtained.

  Next, the motor of the present invention shown in FIG. 24 will be described. In this motor, the rotor magnetic poles in FIG. 22 are changed, the number K of the rotor magnetic poles is 2, and the circumferential width of the rotor magnetic poles is increased to 60 °. The stator windings and currents Ia, Ib, and Ic are the same as those in FIG. When torque is generated in the CCW in the state of FIG. 24, the A-phase current Ia is supplied to the A-phase windings A0D and A0G, and at the same time, the C-phase current Ic is supplied to the C-phase windings A0H and A0E. When the rotor magnetic pole rotates to CCW to the position facing the front of the permanent magnet B07, the B phase current Ib is supplied to the B phase windings A0F and A0J, and the C phase current Ic is supplied to the C phase windings A0H and A0E. To do. Similarly, it is possible to rotate the rotor magnetic pole by rotating the winding to change the winding and energize the current to attract the rotor magnetic pole and continuously generate torque. The direction of the magnetic flux of the rotor is reversed by 6 degrees during one rotation of the rotor. However, from the viewpoint of sequentially attracting and rotating the same rotor magnetic pole in the same direction, this is also a driving method similar to a synchronous motor.

  In addition, the current driving method for the windings can generate the rotational torque sequentially with the current of one phase of the three-phase windings, and while passing through the two windings as described above. Sequential rotational torque can be generated. If it is possible to drive a current in the direction opposite to the winding symbol shown in the figure, for example, if a current in the reverse direction is supplied to the B-phase winding at the rotational position in FIG. The phase current can also contribute to torque generation.

  The motor shown in FIG. 24 can be driven by the control device shown in FIGS. 2 and 3 and can reduce the current capacity of the power transistor. The addition of a permanent magnet increases the torque from the characteristic Trm shown in FIG. 5 to the characteristic Tspm. It can be improved. In particular, the improvement of torque in a low current region is remarkable in a small motor.

  Next, FIG. 25 shows another example of the present invention. This motor has a configuration in which six cylindrical permanent magnets shown in FIG. 24 are arranged separately. The electromagnetic operation is almost the same as that of the motor shown in FIG. Thus, the shape of the stator magnetic pole and the shape of the rotor magnetic pole can be modified into various shapes. Further, as described above, the permanent magnet on the surface of the stator magnetic pole can be mixed with the stator magnetic pole made of soft magnetic material. Further, as will be described later, it is possible to dispose the permanent magnet inside the soft magnetic material.

  As described above, the motor in which the permanent magnet is added to the surface of the stator magnetic pole can take various forms. The flexibility of each shape of each stator magnetic pole, each permanent magnet, and rotor magnetic pole is great, and various ideas can be made in terms of torque, cogging torque, and torque ripple. In terms of torque, characteristics can be greatly changed by changing the circumferential width of each of the magnetic poles of the stator magnetic pole and permanent magnet and the circumferential width of the rotor magnetic pole. For example, the circumferential width of the permanent magnet can be reduced to an electrical angle of 30 ° as in the motor of FIG. Further, in order to reduce cogging or the like at the boundary portion of the permanent magnet magnetic pole, it is possible to reduce the torque ripple by making the surface shape of the permanent magnet recessed in a circular arc shape with a so-called kamaboko shape. Both the stator and the rotor can be skewed. The surface shape of the rotor magnetic pole can also be a so-called kamaboko shape, for example. The deformation direction can be deformed in the radial direction, the circumferential direction, and the rotor axial direction. If the permanent magnet is damaged, the inner periphery of the stator can be protected by covering it with a material having a low magnetic permeability such as resin. The method of FIGS. 14 and 15 can also be adopted for the winding method of the winding.

  Also, the motor shown in FIGS. 22 and 25 can be multipolarized, and a motor having various values of the number M of stator magnetic poles and the number K of rotor magnetic poles as described later can be realized. Also, the characteristics required of the motor vary depending on the application. When different motor characteristics are required depending on the number of rotations, for example, the number of rotor magnetic poles is changed from 8 to 4 above a certain number of rotations. It is also possible to change the configuration with other power or electric actuators in conjunction with each other.

  The demagnetization of each permanent magnet is characterized in that it acts in the direction of strengthening the magnet when torque is normally generated. Since the rotor is robust, the air gap between the stator and the rotor can be reduced. Therefore, the permanent magnet can be used thinly, and the cost can be reduced. However, when incorrect motor current control is performed, or when the power transistor is broken, the magnet may be demagnetized, and it is necessary to consider how to deal with it. One countermeasure is to design a magnet that can be magnetized with a winding current of each phase, as will be described later. Since demagnetization does not occur when normally controlled, various unprecedented methods are possible. Further, as will be described later, it is possible to perform field weakening control and constant output control by changing the strength of the permanent magnet during rotation of the motor. It becomes possible to design a magnet with ease of demagnetization and magnetisation. In this case, not only a neodymium iron boron NdFeB-based magnet having a very large magnetic field strength H but also a so-called alnico magnet can be applied.

  Next, it is a motor which arrange | positions a part or all of a permanent magnet inside a stator core. Specific examples are shown in FIGS. 27, 28, 29, and 30 and will be described. The motor shown in FIG. 27 has a configuration in which a permanent magnet is disposed in the vicinity of the tip of the stator magnetic pole of the motor shown in FIG. B11, B12, B13, B14, B15, and B16 are permanent magnets. B17, B18, B19, B1A, B1B, and B1C are stator magnetic poles. The magnetic flux direction of the magnet is the direction shown in the figure, and is the magnetic flux direction of FIG. The circumferential width of the stator magnetic pole and each permanent magnet can be reduced, and conversely, it can be increased.

  Then, the magnetic characteristics of the stator can be realized between the characteristics of the permanent magnet and the characteristics of the soft magnetic material. The magnetic characteristics required for the stator magnetic pole vary depending on the application. For example, when high torque is required at low speed, the magnetic flux density of the stator magnetic pole is desired to be higher than the magnetic flux density of the rare earth magnet, that is, about 2.0 T near the saturation magnetic flux density of iron. If a high magnetic flux density material such as permendur is used, a magnetic flux density of about 2.5 T is possible. Conversely, when it is desired to perform high-speed rotation operation by performing field weakening or the like, the magnetic flux density on the surface of the stator magnetic pole when the winding current is zero is convenient, for example, about 0.5 T or less. Sometimes it is good. In the case of the structure in which the permanent magnet is arranged inside the core, various magnetic characteristics can be set and designed by the structures as shown in FIGS. 27, 28, 29, and 30.

  From the viewpoint of the control device, for example, when CCW torque is generated at the rotor rotational position θr in FIG. 27, the A-phase current Aa is supplied to the A-phase windings A0D and A0G, and simultaneously, the C-phase windings A0H and A0E are supplied. A C-phase current Ic is applied to increase the magnetic flux B1D, and a CCW torque is generated. At this time, conversely, the magnetic fluxes B1E and B1F in the other directions are desired to be as small as possible. This not only reduces the torque of the CW, but also increases the voltage of the C-phase winding. Since the voltage generated in the C-phase winding is mainly a value obtained by adding the time change rate of the magnetic flux B1D and the time change rate of the magnetic flux B1F, the magnetic flux B1F that generates a negative voltage during CCW is preferably a small value. . When the magnetic flux density of the magnetic flux B1F on the surface of the stator magnetic pole is lower, electric power can be supplied to the motor with a better balance between the A-phase winding and the C-phase winding. Improving the balance between the winding voltages leads to a reduction in the current capacity of the power transistor of the control device because there is a voltage limit between the DC power supply and the power transistor. From the viewpoint of voltage balance, it is desirable that the A-phase current Ia is larger than the C-phase current Ic because an effect of reducing the magnetic flux B1F can be generated.

  At this time, the generated torque can be increased by adding a current -Ic in the opposite direction to the current direction indicated by the current direction symbol of the winding to the C-phase windings A0F and A0J. explain. The current -Ic also has an effect of generating a magnetomotive force that decreases the magnetic fluxes B1E and B1F.

  FIGS. 28A, 28B, and 28C show an example in which only the stator magnetic poles are partially shown, and the magnets are divided and arranged. The amount of magnetic flux of the stator magnetic pole can be arbitrarily designed and selected. In FIG. 28B, the permanent magnet is divided into three parts. However, the number of divisions can be further increased, or the width of the soft iron portion between the permanent magnets can be increased. 29 (a), (b), (c) and (d) are examples of various magnets. The soft iron part can further obtain a direction of magnetic flux by adding a slit to each part. FIG. 30 shows an example in which permanent magnets are arranged in the back yoke portion. It is also possible to supply a magnetic flux to each stator magnetic pole by arranging a magnet in the back yoke portion.

  Next, a specific example using a permanent magnet is shown in FIG. 31 has a permanent magnet G71, G72, G73, G74, G75, G76 added to a part of the stator magnetic pole of the motor of FIG. 9, the rotor magnetic pole width Hm is about 75 °, and the circumference of the rotor magnetic pole. The shape of the direction end is rounded. Depending on the application, one of the problems of the motor of the present invention is to quickly increase or decrease the current of each phase to achieve high-speed rotation or secure torque. Another problem is to more smoothly change the suction force in the radial direction and reduce vibration and noise. The motor shown in FIG. 31 improves the characteristics shown in FIG. 12, which are the motor characteristics shown in FIG.

  Now, consider generating CCW torque and rotating to CCW. Structurally, CCW and CW have different characteristics. In an electric vehicle or a hybrid vehicle, CCW and CW are used differently. Air conditioner compressor motors use only one-way rotation. At the rotor rotational position θr in FIG. 31 (a), the A-phase currents Ia are supplied to the A-phase windings 131 and 134, and at the same time, the C-phase currents Ic are supplied to the C-phase windings 135 and 132 to generate the magnetic flux G7F. Energized to generate CCW torque. At this time, if the currents Ia and Ic have the same amplitude, magnetomotive force is not canceled between the stator magnetic poles G7D and G7A, but a magnetic flux component of G7F is generated between the permanent magnets G71 and G74. . This magnetic flux G7F does not generate torque.

  At the rotor rotational position θr in FIG. 31B, the magnetic flux G7F is continuously excited by the A-phase current Ia and the C-phase current Ic to generate CCW torque. Similarly, a magnetic flux component of G7F is generated at this time. This magnetic flux G7F does not generate torque.

  At the rotor rotational position θr in FIG. 31 (c), the magnetic flux G7F is continuously excited by the A-phase current Ia and the C-phase current Ic to generate CCW torque. However, when rotating at a high speed, it is necessary to reduce the current Ia around the rotor rotational position θr. At the rotor rotational position θr, a magnetic flux component of G7F is generated by the permanent magnets G71 and G74. Moreover, the magnetic flux component of G7G by the permanent magnets G75 and G72 starts to be generated.

  At the rotor rotational position θr in FIG. 31 (d), the A-phase current Ia decreases to zero, the B-phase windings 133 and 136 are energized, the magnetic flux G7H is excited, and the CCW torque is increased. appear. At this time, a magnetic flux component of G7F by the permanent magnets G71 and G74 and a magnetic flux component of G7G by the permanent magnets G75 and G72 are generated.

  From (c) of FIG. 31 to (d) of FIG. 31 described above, the A-phase current Ia rapidly decreases and the magnetic energy accompanying the magnetic flux G7E needs to be regenerated to the DC power source. However, the permanent magnets G72 and G75 are required. It is not necessary to regenerate the magnetic flux energy accompanying the magnetic flux component. In FIG. 11B, the regenerative energy is reduced by the magnetic flux components of the permanent magnets G72 and G75, compared with the necessity of regenerating all the magnetic energy of the magnetic flux passing through the stator magnetic pole 117, and the A-phase current is reduced. The decrease time of Ia can be shortened. That is, the burden of increasing or decreasing the excitation energy is reduced by the permanent magnets G72 and G75.

  At the same time, the radial attractive force between the stator and the rotor rapidly decreased between the A-phase current Ia and the C-phase current Ic from FIG. 31 (c) to FIG. 31 (d). However, the G7F magnetic flux component by the permanent magnets G71 and G74 and the G7G magnetic flux component by the permanent magnets G75 and G72 remain, and the attractive force does not become zero. In FIG. 11 (b), if the A-phase current Ia and the C-phase current Ic suddenly become zero, the radial attractive force becomes almost zero, compared with (c) in FIG. In the case of (d), fluctuations in the radial attractive force are alleviated by the permanent magnets G71, G74, G75, and G72. Further, also in FIG. 31D, the change in the A-phase current can be controlled gently to the extent that the variation in the radial attractive force is reduced without setting the A-phase current Ia to zero.

  Although the example of operation | movement of (a) to (d) of FIG. 31 was demonstrated, this 60 degree operation | movement is repeated similarly and a rotor can be rotated continuously.

  As for the shape of the permanent magnet, the thickness of the magnet may be a structure in which some magnets are thick and other magnets are thin, and the magnet characteristics are different. Also, two or more types of magnets such as rare earth magnets, cast magnets, ferrite magnets, and so-called bonded magnets can be used depending on the purpose. For example, a thick rare earth magnet always exhibits a high magnetic flux density characteristic, and the other portions can be used with the magnetic flux density varied by current. Further, as described above, it is also possible to magnetize and demagnetize a part of the magnet with the motor drive current while the motor is operating. Also, as for the magnet shape, many examples in which the magnet cross section is rectangular have been shown, but various shapes such as a triangular shape and a trapezoidal shape can be used. The permanent magnet arrangement in the vicinity of the stator magnetic pole is divided into the surface arrangement of the stator magnetic pole and the internal arrangement of the stator magnetic pole. However, if the soft iron part has a very thin shape as in FIGS. There is no big difference in the effects.

  Next, a motor configuration in which a permanent magnet is disposed on the surface of the rotor magnetic pole or the soft magnetic body of the rotor magnetic pole will be described. FIG. 32 shows a motor in which the number of stator magnetic poles M is 6, the number of rotor magnetic poles K is 10, and permanent magnets are arranged on the rotor surface. The stator magnetic pole and each winding of the stator are the same as the stator shown in FIG. The permanent magnet of the rotor G3M shown in G3D, G3E, G3Q, etc. indicates the direction of the magnetic flux with an arrow, and indicates the N pole or S pole which is the inner diameter side magnetic pole of the permanent magnet on the inner diameter side of the permanent magnet.

  (A), (b), (c), and (d) of FIG. 32 show a state in which the rotor G3M has advanced in the counterclockwise rotation direction CCW in that order. In FIG. 32 (a), in order for the rotor G3M to generate the CCW torque T, the G3D N-pole magnet is driven by the current Ia of the A-phase windings G37 and G38 and the current Ic of the C-phase windings G3B and G3C What is necessary is just to attract | suck to the direction of the magnetic pole G31. At the same time, the G3L S-pole magnet is attracted to the CCW, which is the direction of the stator magnetic pole G34, by the currents Ia and Ic. At the same time, the permanent magnets G3R and G3S repel the stator magnetic poles G31 and G34, respectively, to generate CCW torque. When torque is generated in the clockwise direction CW, a stator magnetic pole and a rotor magnetic pole having opposite relations are selected, and a current that generates a magnetomotive force is applied to the stator magnetic pole.

  Similarly, at the rotor rotational position in FIG. 32B, the B-phase current Ib is passed through the windings G39 and G3A, the current Ic is passed through, and the stator magnetic pole G32 and the S-pole permanent magnet G3E are attracted. The magnetic pole G35 and the N-pole permanent magnet G3F are attracted to generate CCW torque T.

  Similarly, at the rotor rotational position of FIG. 32C, the A-phase current Ia is supplied, the B-phase current Ib is supplied, the stator magnetic pole G33 and the N-pole permanent magnet G3H are attracted, and the stator magnetic poles G36 and S The permanent magnet G3G is attracted and a CCW torque T is generated. Thereafter, rotational torque can be generated similarly.

  (D) of FIG. 32 is an example of the motor which changed the permanent magnet of rotor G3X. The N-pole permanent magnet G3J and the like have the same configuration, and instead of the S-pole permanent magnet, a soft magnetic pole G3K is used. Since the magnetic flux such as the N-pole magnet G3J passes through the stator to the magnetic pole G3K and the like of this soft magnetic material, the magnetic flux similar to that when the S-pole permanent magnet is disposed passes and the generation of torque is also similar. is doing. This is a motor configuration example in which the number of magnets is reduced.

  Further, the permanent magnet disposed on the rotor can be disposed not only on the surface of the soft magnetic body of the rotor but also inside the soft magnetic body. The permanent magnet can be arranged inside the soft magnetic body by various methods. FIGS. 27, 28, 29, and 30 are examples of the stator magnetic pole, but the same applies to the rotor magnetic pole.

  Next, FIG. 33 shows a motor in which the number of stator magnetic poles M is 8, the number of rotor magnetic poles K is 12, and permanent magnets are arranged on the rotor surface. In this case, each winding of the stator cannot be the full-pitch winding shown in FIG. 1 due to the number of slots. The motor winding shown in FIG. 33 needs to be an annular winding shown in FIG. 14 or a motor combined as shown in FIG. 15 and wound around the slots of both motors. In FIG. 33, only the windings necessary to function electromagnetically are shown.

  (A), (b), (c), and (d) of FIG. 33 show a state in which the rotor G41 has advanced to CCW in that order. In the rotor rotation position of FIG. 33 (a), in order to generate the CCW torque, a current in the direction shown in the figure is supplied to the windings G43 and G44, and an attractive force is generated between the stator magnetic pole G4B and the magnet G42. . At the same time, a current in the direction shown in the figure is applied to the windings G43 and G4A, and an attractive force is generated between the stator magnetic pole G4Z and the magnet G4J. At the same time, a current in the direction shown in the figure is applied to the windings G47 and G46, and an attractive force is generated between the stator magnetic pole G4E and the magnet G4L. At the same time, the windings G47 and G48 are energized in the direction shown in the figure, and an attractive force is generated between the stator magnetic pole G4F and the magnet G4M. Here, since the currents to be supplied to the windings G43 and G48 overlap, it is necessary to supply a current twice as large as that of the other windings to these two windings. The total torque generated by the four stator magnetic poles G4B, G4Z, G4G, and G4F is the motor torque T. The permanent magnets arranged on the CCW side of the rotor magnetic pole magnets G42, G4A, G4U, and G4M also generate CCW torque by the repulsive force.

  Similarly, at the rotor rotational position shown in FIG. 33 (b), in order to generate a CCW torque, a current in the direction shown in the figure is applied to the windings G43 and G44, and an attractive force is generated between the stator magnetic pole G4B and the magnet G42. Is generated. At the same time, the windings G45 and G44 are energized in the direction shown in the figure, and an attractive force is generated between the stator magnetic pole G4C and the magnet G4V. At the same time, a current in the direction shown in the figure is applied to the windings G49 and G48 to generate an attractive force between the stator magnetic pole G4G and the magnet G4W. At the same time, the windings G47 and G48 are energized in the direction shown in the figure, and an attractive force is generated between the stator magnetic pole G4F and the magnet G4M.

  Similarly, at the rotor rotational position of FIG. 33 (c), in order to generate the CCW torque, a current in the direction shown in the figure is supplied to the windings G45 and G44, and the attractive force is generated between the stator magnetic pole G4C and the magnet G4Y. Is generated. At the same time, a current in the direction shown in the figure is applied to the windings G45 and G46, and an attractive force is generated between the stator magnetic pole G4D and the magnet G4K. At the same time, a current in the direction shown in the figure is applied to the windings G49 and G48 to generate an attractive force between the stator magnetic pole G4G and the magnet G4W. At the same time, a current in the direction shown in the figure is applied to the windings G49 and G4A, and an attractive force is generated between the stator magnetic pole G4H and the magnet G4P.

  Similarly, at the rotor rotational position of FIG. 33 (d), in order to generate the CCW torque, a current in the direction shown in the figure is applied to the windings G45 and G46, and the attractive force is generated between the stator magnetic pole G4D and the magnet G4K. Is generated. At the same time, a current in the direction shown in the figure is applied to the windings G47 and G46, and an attractive force is generated between the stator magnetic pole G4E and the magnet G4R. At the same time, a current in the direction shown in the figure is applied to the windings G49 and G4A, and an attractive force is generated between the stator magnetic pole G4H and the magnet G4P. At the same time, a current in the direction shown in the figure is applied to the windings G43 and G4A, and an attractive force is generated between the stator magnetic pole G4J and the magnet G4Q. Thereafter, rotational torque can be generated similarly.

  Next, FIG. 34 shows a motor in which the number of stator magnetic poles M is 10, the number of rotor magnetic poles K is 16, and permanent magnets are arranged on the rotor surface. Each winding of the stator of FIG. 34 can be the full-pitch winding shown in FIG. 1, but in the case of this rotor magnetic pole number 16, the direction of the rotor magnetic pole on the opposite side by 180 ° in electrical angle is reversed. The whole volume cannot be adopted. The winding of the motor shown in FIG. 34 needs to be an annular winding shown in FIG. 14, or a motor is combined as shown in FIG. 15 and wound around the slots of both motors. FIG. 34 shows only the windings necessary to function electromagnetically.

  (A), (b), (c), and (d) of FIG. 34 show a state in which the rotor GG1 has advanced to the CCW in that order. In the rotor rotation position of FIG. 34 (a), in order to generate the CCW torque, a current in the direction shown in the figure is supplied to the windings G51 and G52, and an attractive force is generated between the stator magnetic pole G5B and the magnet G5Z. . At the same time, a current in the direction shown in the figure is applied to the windings G59 and G58, and an attractive force is generated between the stator magnetic pole G5U and the magnet G5V. At the same time, a current in the direction shown in the figure is applied to the windings G57 and G58, and an attractive force is generated between the stator magnetic pole G5H and the magnet G5V. At the same time, a current in the direction shown in the figure is applied to the windings G55 and G54, and an attractive force is generated between the stator magnetic pole G5E and the magnet G5M. The total torque generated by the four stator magnetic poles G5B, G5U, G5H, and G5E is the motor torque T. The permanent magnets arranged on the CCW side of the rotor magnetic pole magnets G5Z, G5V, G5V, and G5M also generate CCW torque by the repulsive force.

  Similarly, at the rotor rotational position of FIG. 34 (b), in order to generate the CCW torque, a current in the direction shown in FIG. 34 is supplied to the windings G53 and G52, and an attractive force is generated between the stator magnetic pole G5C and the magnet G5L. Is generated. At the same time, a current in the direction shown in the figure is applied to the windings G59 and G5A, and an attractive force is generated between the stator magnetic pole G5J and the magnet G5R. At the same time, a current in the direction shown in the figure is supplied to the windings G59 and G58, and an attractive force is generated between the stator magnetic pole G5U and the magnet G5M. At the same time, the windings G55 and G56 are energized in the direction shown in the figure, and an attractive force is generated between the stator magnetic pole G5F and the magnet G5N.

  Similarly, at the rotor rotational position of FIG. 34 (c), in order to generate the CCW torque, a current in the direction shown in the figure is supplied to the windings G53 and G52, and an attractive force is generated between the stator magnetic pole G5C and the magnet G5T. Is generated. At the same time, a current in the direction shown in the figure is applied to the windings G59 and G5A, and an attractive force is generated between the stator magnetic pole G5J and the magnet G5R. At the same time, a current in the direction shown in the figure is applied to the windings G57 and G56, and an attractive force is generated between the stator magnetic pole G5G and the magnet G5P. At the same time, a current in the direction shown in the figure is supplied to the windings G53 and G52, and an attractive force is generated between the stator magnetic pole G5C and the magnet G5L.

  Similarly, at the rotor rotational position of FIG. 34 (d), in order to generate the CCW torque, a current in the direction shown in the figure is applied to the windings G5B and G5A, and the attractive force is generated between the stator magnetic pole G5K and the magnet G5S. Is generated. At the same time, a current in the direction shown in the figure is applied to the windings G57 and G58, and an attractive force is generated between the stator magnetic pole G5H and the magnet G5Q. At the same time, a current in the direction shown in the figure is applied to the windings G55 and G54, and an attractive force is generated between the stator magnetic pole G5E and the magnet G50. At the same time, a current in the direction shown in the figure is applied to the windings G53 and G54 to generate an attractive force between the stator magnetic pole G5D and the magnet G5T. Thereafter, rotational torque can be generated similarly.

  The example of the present invention in which the permanent magnet is arranged on the rotor has been described above. In addition to these examples, various combinations of the number M of the stator magnetic poles and the number K of the rotor magnetic poles are possible, and these configurations in which a permanent magnet is arranged on the rotor are also included in the present invention. 32, 33, and 34 show examples in which permanent magnets are arranged on the entire circumference of the rotor, but permanent magnets may be arranged separately in the circumferential direction of the rotor surface. Moreover, it can also be set as the structure which has arrange | positioned the permanent magnet part and the soft iron part in the circumferential direction. Various modifications can be made to the shape of each permanent magnet and the shape of the soft iron portion.

  Further, when the motor of the present invention using a permanent magnet for the stator magnetic pole or the rotor magnetic pole is used as a generator, it may be possible to eliminate the exciting current such as the field current If shown in FIG. In addition, when rectifying the power generation voltage of each winding, in order to utilize the voltage in both positive and negative directions, it is necessary to perform so-called full-wave rectification.

  It is possible to combine the motors having various configurations described so far. For example, in the case of an 8-pole motor, the surface of the stator magnetic pole is composed of permanent magnets for 4 poles, the other half is composed of the stator magnetic poles of soft iron, and the overall motor characteristics are intermediate between the two. be able to. Also, it can be combined with conventional motor technology. Various combinations of motors with various configurations can be realized.

  Next, the configuration of the linear motor will be described. It is possible to linearize the various motors. 35A is a longitudinal sectional view of the linear motor, and FIG. 35B is a transverse sectional view. 35 is a linear motor having a configuration in which the motor of FIG. 1 is composed of an outer diameter side and an inner diameter side as in the motor of FIG. 15, and the motor is linearized and finally cylindrical. is there. The schematic linear motor configuration is an example in which the center side is a stator and the outer peripheral side is a slider. B26, B25, B24, B23, B23, B22, and B21 are annular stator magnetic poles. B21 and B27 form an annular salient pole. B2T and B2C are annular windings that are A-phase windings, and B2Q and B2K are negative A-phase windings. B2R and B2L are annular windings and B-phase windings, and B2N and B2H are negative B-phase windings. B2P and B2J are annular windings and C-phase windings, and B2S and B2M are negative C-phase windings. B2G and B2B are annular slider magnetic poles. Similarly, B2F and B2A, B2E and B29, and B21 and B27 are annular slider magnetic poles, respectively. FIG. 35 (a) illustrates a part of the linear motor, and can be configured to have an arbitrary length. In FIG. 35B, B2U is a slider magnetic pole, B2V is an air gap between the stator and the slider, and B2H is a stator magnetic pole. The generation of force is the same electromagnetic action as the generation of torque of the motor in FIG. 1, and the movement is linearized from the circular motion. In addition, a sliding bearing or a rolling bearing is disposed between the stator and the slider to accurately maintain and guide the mutual gap. In addition, current can be energized by connecting in-phase windings and opposite-phase windings with crossovers so that the current directions are the same, and can be driven by a control device as shown in FIGS. A small system can be achieved at low cost. Since each 2/3 of the windings contribute to torque generation, a highly efficient linear motor can be obtained. The torque can be further increased by adding a permanent magnet to the stator magnetic pole. In addition, the linear motor of FIG. 35 can be configured such that the inner diameter side and the outer diameter side are reversed, and the winding side can be a movable portion, and various modifications are possible. Other motors with different motor types can be made into linear motors, or permanent magnets can be used.

  Next, various types of motors are shown. The case where the number M of stator magnetic poles is 6 and the number K of rotor magnetic poles is 2 or 4 is shown in FIGS. Even when the number M of the stator magnetic poles is 6 and the number K of the rotor magnetic poles is an odd number of 3 or 5, as shown in FIGS. 36 and 37, the motor of the present invention can be realized.

  FIG. 36 shows a motor of motor type 6S3R. Since this motor has a rotor magnetic pole number K of 3 and an asymmetric structure, full-pitch winding cannot be adopted. The winding of the motor shown in FIG. 36 needs to be an annular winding shown in FIG. 14 or a combination of the motors as shown in FIG. 15 and winding the windings around the slots of both motors. In FIG. 36, only the windings necessary to function electromagnetically are shown.

  In order to generate CCW torque at the rotor rotational position of FIG. 36 (a), a current in the direction shown in the figure is applied to the windings G11 and G16 to generate an attractive force between the stator magnetic pole A01 and the rotor magnetic pole G17. CCW torque can be obtained. In order to generate CCW torque at the rotor rotational position shown in FIG. 36 (b), a current in the direction shown in the figure is applied to the windings G13 and G12 to generate an attractive force between the stator magnetic pole A03 and the rotor magnetic pole G18. CCW torque can be obtained. Thus, a continuous torque can be obtained by sequentially changing the winding to be energized as the rotor rotates. This motor has six windings, and each winding can be used when two stator magnetic poles on both sides in the circumferential direction generate torque respectively, and each winding is also used.

  Next, FIG. 37 shows a motor of motor type 6S5R. Since this motor also has a rotor magnetic pole number K of 5 and an asymmetric structure, it is not possible to adopt full-pitch winding. The winding of the motor shown in FIG. 36 needs to be an annular winding shown in FIG. 14 or a combination of the motors as shown in FIG. 15 and winding the windings around the slots of both motors.

  In order to generate CCW torque at the rotor rotational position of FIG. 37 (a), a current in the direction shown in the figure is applied to the windings G11 and G14 to generate an attractive force between the stator magnetic pole A06 and the rotor magnetic pole G25. CCW torque can be obtained. Further, a current in the direction shown in the figure can be applied to the windings G15 and G14 to generate an attractive force between the stator magnetic pole A05 and the rotor magnetic pole G24, thereby obtaining CCW torque.

  When rotating to the CCW to the rotor rotation position of FIG. 37 (b), currents in the direction shown in the figure are applied to the windings G15 and G12, and an attractive force is generated between the stator magnetic pole A04 and the rotor magnetic pole G23, and the torque of the CCW is increased. Obtainable. Further, a current in the direction shown in the figure can be applied to the windings G15 and G14 to generate an attractive force between the stator magnetic pole A05 and the rotor magnetic pole G24, thereby obtaining CCW torque. Thus, a continuous torque can be obtained by sequentially changing the winding to be energized as the rotor rotates. This motor has six windings, and each winding can be used when two stator magnetic poles on both sides in the circumferential direction generate torque respectively, and each winding is also used.

  In the motors shown in FIGS. 36 and 37, the circumferential widths of the stator magnetic poles and the rotor magnetic poles can be changed as necessary. Further, the number of the stator magnetic poles and the rotor magnetic poles can be multiplied by an integer to make a modification corresponding to so-called multipolarization. In the basic form of FIGS. 36 and 37, a geometrically unbalanced structure can be multipolarized to improve the balance. Magnetically, it is possible to change the direction of the winding so that the total magnetic flux is balanced.

  Further, torque can be generated even when K is 6 or more, and there is a feature that operation with a small torque ripple is possible particularly in a low torque range. However, when a large torque is generated, the rotor magnetic poles are close to each other, which increases the leakage magnetic flux, which is disadvantageous.

  Although the case where the number K of rotor magnetic poles is an odd number has been shown, the motor of the present invention can be realized also when the number M of stator magnetic poles is an odd number. However, in that case, inconvenience occurs in the magnetomotive force generation of the stator magnetic poles in one place with respect to the direction of current. There are several ways to deal with this, as shown below. One method does not use inconvenient stator poles as a motor. Alternatively, the current direction of one winding is controlled so that current can flow in both directions. Alternatively, two sets of windings having opposite current directions are arranged in slots in which irregularities occur so that a unidirectional current can be applied to each of the windings. Since this problem is a problem of a part of the motor, there are various ways to deal with it with some cost burden. If it is desired to realize a specific relationship between the number of stator magnetic poles and the number of rotor magnetic poles, or if there are restrictions in realizing the motor, M and K can be set to odd numbers.

  Further, an example of a motor in which the shape of the stator magnetic pole, the shape of the slot, the shape of the rotor magnetic pole, etc. are modified will be described with reference to FIG. This motor has a motor type 6S4R and is a full-pitch winding. The broken line shows the arrangement of the coil ends. Windings 226 and 229 are A-phase windings, windings 228 and 22B are B-phase windings, and windings 22A and 227 are C-phase windings. The current direction of each winding is the direction of the illustrated symbol. For example, the current direction of the winding 226 is the direction from the front side of the page to the back side of the page. The current direction of the winding 229 is the direction from the back side of the page to the front side of the page. Regarding the winding method of the winding, it is possible to use the annular winding shown in FIG. 14 or to combine the motors and wind the windings around the slots of both motors as shown in FIG.

  Reference numerals 22C, 22D, 22E, 22F, 22G, and 22H denote stator magnetic poles. Various modifications can be made to the shape of the stator magnetic pole, and the circumferential width of the stator magnetic pole of the motor shown in FIG. 1 is approximately 30 ° in electrical angle. Compared to this value, the stator magnetic pole width Ht of FIG. Is rather close to 60 °. And the opening part of each slot which has arrange | positioned each phase coil | winding has the collar shape 222, and the circumferential direction width | variety Hs of a slot opening part is an example small enough to be able to insert a coil | winding.

  The rotor magnetic poles 221 have a wide rotor magnetic pole width Hm in the circumferential width and 225 have a narrow rotor magnetic pole width Hh. The aim of these shapes is that at low speed rotation, both stator poles are driven alternately according to the rotor rotation position, and as the rotation speed increases, driving with narrow rotor poles 225 is reduced, and with wide rotor poles. The purpose is to increase the drive and simplify the drive algorithm to reduce the increase / decrease in current of each winding, to reduce iron loss, to reduce vibration and noise.

  Slits 223 and 224 are provided in the rotor magnetic pole having a wide circumferential width. The purpose of this is to prevent the magnetic flux of the rotor magnetic pole from becoming excessive and to make the distribution of the magnetic flux of the rotor magnetic pole portion uniform. Various shapes of slits, holes and the like can be provided in the rotor magnetic pole. Furthermore, a conductor or conductor plate that forms a closed circuit in the slit can be arranged to induce a current in the conductor so as to prevent a change in magnetic flux across the slit, thereby restricting the distribution of the magnetic flux. In addition, although four slits are described as being separated into two rotor magnetic poles, these slits may be four slits connected from one rotor magnetic pole to the other rotor magnetic pole.

  Here, the rotation position of the rotor is indicated by θr as the angle from the center position of the A-phase winding to the corner of the rotor magnetic pole in the counterclockwise rotation direction. When CCW torque is generated at the rotational position θr in FIG. 38, the A phase current Ia and the current Ic of the same magnitude may be supplied to the A phase windings 226 and 229 and the C phase windings 22A and 227, respectively. . The magnitudes of the currents Ia and Ic can be roughly determined by the magnitude of the torque command Tc. To be precise, the rotor rotational position θr depends on the influence of leakage magnetic flux leaking into the space and the magnetic nonlinearity of the soft magnetic material. It is also a function of At this time, the current Ib of the B-phase windings 228 and 22B is set to zero. However, these current values can take various values depending on the purpose. For example, in high-speed rotation, it is effective to control each phase current by advancing the phase in consideration of the current response delay time. In addition, a current of a certain magnitude in each phase can be added as an offset value, the radial attractive force in the motor can be stabilized, and vibration and noise can be reduced in some cases. Similarly, there is a current control method for smoothly changing the magnetic flux in the motor. Further, as will be described later, at this rotational position θr, the motor torque can be increased by energizing the C-phase current Ic in the direction opposite to the direction of the C-phase current shown in FIG. It is also possible to improve. However, in that case, it is necessary to change and increase the control device, which is both a merits and demerits. As described above, since each phase current can drive a motor by various methods and values, the current energization method and current value for basic and static torque generation described in the present invention are modified. However, the present invention does not exclude them.

  Next, it is possible to multiply the motor shown in FIG. 38 by an integer multiple in the circumferential direction to make a so-called multi-pole. By making the multi-pole, the length of the coil end portion can be greatly reduced, and the thickness of the back yoke portion of the stator can be reduced. Therefore, it is assumed that the two-pole motor shown and described as an example of the present invention is rather multi-polar.

  These modifications of the motor shape shown here can be applied to other types of motors shown in the present invention, and motors in which the shape of each part of the other motors shown in the examples of the present invention are deformed are also included in the present invention. Is included.

  Next, the motor of the present invention when the number M of stator magnetic poles is 8 will be described. As shown in FIG. 39, the current directions of the windings in each slot when M is 8 can be arranged so that the current directions of the windings of adjacent slots are opposite to each other. However, the current directions of the windings B3H and B3M at the electrical angle of 180 ° are the same direction, and the windings cannot be wound with full-pitch winding like the stator of FIG. The annular winding as shown in FIG. 14 or the winding of the composite motor as shown in FIG. 15 can be manufactured because the windings of the individual slots can be wound.

  At this time, the number K of rotor magnetic poles can be an integer of 2 or more. In the case of the motor type 8S2R with K = 2, the rotor 11E shown in FIG. 9 is obtained, and can be driven by being sequentially attracted by each stator magnetic pole as in the motor shown in FIG. However, in this case, since the direction of the magnetic flux is the same direction, it is necessary to add a magnetic flux bypass path between the stator back yoke and the rotor back yoke. Alternatively, in FIG. 39A, the rotor magnetic poles B3B and B3D can be removed. The torque reduction portion can be improved by generating torque due to leakage magnetic flux, rotor skew, or stage skew described later.

  A feature of the motor of the motor type 8S2R is that the degree of freedom regarding the overlap with the torque of the adjacent stator magnetic pole is higher than that of the motor of 6S2R. In the case of a 6S2R motor, current is generated by superimposing currents that generate torque on the two stator magnetic poles in the vicinity of the rotational position where torque generation is changed between the two stator magnetic poles. There is a problem that negative torque is generated in the third stator magnetic pole.

  To solve this problem, in the motor of the motor type 8S2R, torque is generated by each of these two stator magnetic poles in the vicinity of the rotational position where the torque generation is changed by the adjacent first and second stator magnetic poles. When the current is superimposed and energized, the magnetomotive force to the third stator magnetic pole close to the rotor magnetic pole hardly acts, so that it is easy to realize good torque generation.

  Examples of the motor format 8S4R are shown in FIGS. 39 (a) and 39 (b). In this motor, the rotor is geometrically unbalanced, but can be driven in the same manner as 6S4R shown in FIG. Multipolarization can solve problems such as rotor balance. When generating a CCW torque, at the rotor position in FIG. 39 (a), current is supplied to the windings B3N and B3B in the direction of the winding symbol, and the stator magnetic pole B35 and the rotor magnetic pole B3B are attracted. CCW torque is generated. At the same time, a current is passed through the windings B3Q and B3H in the direction of the winding symbol, and the stator magnetic pole B38 and the rotor magnetic pole B3D are attracted to generate CCW torque.

  The rotor position in FIG. 39 (a) is slightly rotated from the rotor position to the CCW, and at the rotor position in FIG. 39 (b), a current is passed through the windings B3J and B3K in the direction of the winding symbol, so B3A is sucked and CCW torque is generated. At the same time, a current is supplied to the windings B3Q and B3P in the direction of the winding symbol, and the stator magnetic pole B37 and the rotor magnetic pole B3C are attracted to generate CCW torque. Similarly, the winding torque to be energized can be selected according to the rotor rotational position θr to obtain the rotational torque.

  39 (a) and 39 (b), it can be driven by the current of four of the eight windings. Further, each winding can be used when torque is generated by the stator magnetic poles on both sides in the circumferential direction of the winding, and each winding is also used.

  Next, the motor shown in FIG. 40 is an example in which the stator is the same as in FIG. 39 and the number K of rotor magnetic poles is six. The motor type is 8S6R. In FIG. 40, the stator is shown in a slightly specific manner, with an annular winding configuration, with the addition of a winding B3X on the back side of the back yoke and a path B3Y indicated by a broken line. When generating CCW torque, at the rotor position in FIG. 40 (a), current is supplied to the windings B3J and B3H in the direction of the winding symbol, and the stator magnetic pole B31 and the rotor magnetic pole B3R are attracted. CCW torque is generated. At the same time, a current is applied to the windings B3J and B3K in the direction of the winding symbol, and the stator magnetic pole B32 and the rotor magnetic pole B3S are attracted to generate CCW torque. At the same time, a current is supplied to the windings B3N and B3M in the direction of the winding symbol, and the stator magnetic pole B35 and the rotor magnetic pole B3U are attracted to generate CCW torque. At the same time, a current is supplied to the windings B3N and B3P in the direction of the winding symbol, and the stator magnetic pole B36 and the rotor magnetic pole B3V are attracted to generate CCW torque.

  The rotor position in FIG. 40 (a) is slightly rotated from the rotor position to the CCW, and at the rotor position in FIG. 40 (b), a current is passed through the windings B3Q and B3H in the direction of the winding symbol. B3W is sucked and CCW torque is generated. At the same time, a current is passed through the windings B3J and B3H in the direction of the winding symbol, and the stator magnetic pole B31 and the rotor magnetic pole B3R are attracted to generate CCW torque. At the same time, a current is supplied to the windings B3L and B3M in the direction of the winding symbol, and the stator magnetic pole B34 and the rotor magnetic pole B3T are attracted to generate CCW torque. At the same time, a current is supplied to the windings B3N and B3M in the direction of the winding symbol, and the stator magnetic pole B35 and the rotor magnetic pole B3U are attracted to generate CCW torque. Similarly, the winding torque to be energized can be selected according to the rotor rotational position θr to obtain the rotational torque.

  Next, the motor of the present invention when the number M of stator magnetic poles is 10 will be described. As shown in FIG. 41, the current direction of the windings of each slot when M is 10 can be arranged so that the current direction of the windings of adjacent slots is opposite. Then, the current direction of the winding at an electrical angle of 180 °, for example, B5M and B5S is opposite, and the winding can be wound with full-pitch winding. In the case of full-pitch winding, windings arranged in 10 slots are B5M and B5S for A phase winding, B5P and B5U for B phase winding, B5R and B5W for C phase winding, B5T and B5N for D phase winding. The E-phase winding is composed of B5V, B5Q and five sets of windings. Further, it can be wound with an annular winding as shown in FIG. 14 or a composite motor winding as shown in FIG.

  At this time, the number K of rotor magnetic poles can be an integer of 2 or more. In the case of the motor type 10S2R with K = 2, the rotor 11E shown in FIG. 9 is obtained, and can be driven by being sequentially attracted by each stator magnetic pole like the motor shown in FIG. The torque reduction portion can be improved by generating torque due to leakage magnetic flux, rotor skew, or stage skew described later.

  The feature of the motor 10S2R is that the degree of freedom regarding the overlap with the torque of the adjacent stator magnetic poles is higher than that of the 6S2R motor. In the case of a 6S2R motor, current is generated by superimposing currents that generate torque on the two stator magnetic poles in the vicinity of the rotational position where torque generation is changed between the two stator magnetic poles. There is a problem that negative torque is generated in the third stator magnetic pole.

  To solve this problem, in the motor of the motor type 10S2R, torque is generated by these two stator magnetic poles in the vicinity of the rotational position where the torque generation is changed by the adjacent first and second stator magnetic poles. When the current is superimposed and energized, the magnetomotive force to the third stator magnetic pole close to the rotor magnetic pole hardly acts, so that it is easy to realize good torque generation. Further, in the case of the motor type 8S2R, it is difficult to wind all the nodes, and it is necessary to devise the direction of the magnetic flux, but the motor type 10S2R is simple in that respect. However, since the motor is a five-phase motor, the utilization factor of the windings is lowered, and the control device is somewhat complicated.

  FIG. 41 shows a motor example of the motor type 10S4R. This motor can be driven to rotate by a method similar to the method shown in FIG. In accordance with the rotor rotational position θr, the rotor magnetic poles B5E, B56 and B5C, B5D can be excited by energizing the unidirectional currents indicated by symbols in the figure alternately to generate rotational torque. The control device of FIG. 2 is expanded to five phases, and the time ratio of generating torque by energizing each winding of FIG. 41 is 2/5, which is compared with the conventional example of FIG. The current capacity of the power transistor is (1/6) / (2/5) = (1 / 2.4) times. The current capacity of the control device can be reduced. That is, the size can be reduced.

  Next, FIG. 42A shows an example of a motor in which the circumferential width of the stator magnetic pole and the rotor magnetic pole in FIG. 41 is increased. G91, G92, G93, G94, G95, G96, G97, G98, G99, and G9A are ten stator magnetic poles. Each winding is the same as FIG. 41, and each of the A phase, B phase, C phase, D phase, and E phase windings is B5M and B5S, B5P and B5U, B5R and B5W, B5T and B5N, B5V and B5Q. Each current carries a one-way current Ia, Ib, Ic, Id, Ie in the direction of the symbol shown in the figure. Note that the description of the coil end portion of the whole volume winding is omitted.

  When generating CCW torque, the currents Ia and Id are supplied to the A-phase windings B5M and B5S and the D-phase windings B5T and B5N in the direction of the winding symbol at the rotor position in FIG. A magnetic flux G9F indicated by a solid line with a thick arrow is excited to attract the stator magnetic pole G91 and the rotor magnetic pole G9B to generate CCW torque. At the same time, the magnetic flux G9F is also excited between the stator magnetic pole G96 and the rotor magnetic pole G9D, an attractive force is generated, and a CCW torque is generated. On the other hand, at the same time, current is supplied to the B-phase windings B5P and B5U and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux G9G indicated by the broken line with thick arrows, and the stator magnetic pole G93 and the rotor magnetic pole G9C To generate a CCW torque. At the same time, the magnetic flux G9G is also excited between the stator magnetic pole G98 and the rotor magnetic pole G9E, an attractive force is generated, and a CCW torque is generated.

  At the rotor position of FIG. 42 (b) rotated slightly to CCW, current is supplied to the A-phase windings B5M, B5S and D-phase windings B5T, B5N in the direction of the winding symbol, and is indicated by a solid line G9H with a thick arrow. The magnetic flux is excited to attract the stator magnetic pole G91 and the rotor magnetic pole G9B, thereby generating CCW torque. At the same time, the magnetic flux G9H is also excited between the stator magnetic pole G96 and the rotor magnetic pole G9D, an attractive force is generated, and a CCW torque is generated. On the other hand, at the same time, current is supplied to the C-phase windings B5R and B5W and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux G9J indicated by the broken line with thick arrows, and the stator magnetic pole G94 and the rotor magnetic pole G9C To generate a CCW torque. At the same time, a magnetic flux indicated by a thick broken line with an arrow is also excited between the stator magnetic pole G99 and the rotor magnetic pole G9E, an attractive force is generated, and a CCW torque is generated.

  In the rotor position of FIG. 42 (c), a current is supplied to the B-phase windings B5P and B5U and the D-phase windings B5T and B5N in the direction of the winding symbol, and a magnetic flux G9K indicated by a solid line with a thick arrow is excited. The stator magnetic pole G92 and the rotor magnetic pole G9B are attracted to generate CCW torque. At the same time, a magnetic flux G97 indicated by a solid line with a thick arrow is also excited between the stator magnetic pole G97 and the rotor magnetic pole G9D, and an attractive force is generated to generate a CCW torque. On the other hand, at the same time, current is supplied to the C-phase windings B5R and B5W and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux G9L indicated by the broken line with thick arrows, and the stator magnetic pole G94 and the rotor magnetic pole G9C To generate a CCW torque. At the same time, a magnetic flux indicated by a thick broken line with an arrow is also excited between the stator magnetic pole G99 and the rotor magnetic pole G9E, an attractive force is generated, and a CCW torque is generated.

  In the rotor position of (d) of FIG. 42, a current is passed through the B-phase windings B5P and B5U and the D-phase windings B5T and B5N in the direction of the winding symbol, and the magnetic flux G9M indicated by a solid line with a thick arrow is excited. The stator magnetic pole G92 and the rotor magnetic pole G9B are attracted to generate CCW torque. At the same time, the magnetic flux G9M is also excited between the stator magnetic pole G97 and the rotor magnetic pole G9D, an attractive force is generated, and a CCW torque is generated. At the same time, current is supplied to the C-phase windings B5R and B5W and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux G9N indicated by the broken line with thick arrows, and the stator magnetic pole G94 and the rotor magnetic pole G9C. To generate a CCW torque. At the same time, the magnetic flux G9N indicated by the broken line is also excited between the stator magnetic pole G99 and the rotor magnetic pole G9E, and an attractive force is generated to generate a CCW torque. Similarly, rotational torque can be generated by controlling the current of each phase according to the rotor rotational position. By rotating in the CCW and CW directions, positive and negative torque can be controlled, and so-called four-quadrant operation is possible.

  As described above, the motor shown in FIG. 42 generates torque with four rotor magnetic poles, and the phase relation between the stator magnetic pole and the rotor magnetic pole is different for each of the two rotor magnetic poles, so that torque ripple is also reduced. Moreover, it can be said that the electric current of four of the five windings is involved in the generation of the rotational torque and is effectively generating the torque at most rotational positions.

  However, in the motor type 10S4R in FIG. 42, since two sets of increase / decrease in magnetic flux intersect with each other as compared with the motor type 6S2R in FIG. 11, energy exchange with the motor by the current of each winding is complicated. The voltage generated in each winding may not be a simple superposition of the example of the motor type 6S2R in FIG. When the magnetic flux indicated by the bold broken line increases or decreases while rotating at the rotational position of FIG. 42A, the induced voltages of the A-phase windings B5M and B5S and the D-phase windings B5T and B5N do not have the same value. Be careful.

  When the circumferential end of the rotor magnetic pole in the rotational direction approaches the slot opening of the stator as shown in FIG. 42 (c), the torque generated at that portion tends to decrease. As countermeasures for the problem of torque reduction, there are various methods such as rotor skew and rotor shape described later to generate torque due to leakage magnetic flux. In addition, when a five-phase full-pitch winding is configured in the shape of the motor shown in FIG. 42 or its multiphase, the windings of each phase intersect at the coil end portion, and the coil end portion tends to be large. With respect to this point as well, by forming a composite motor as shown in FIG. 15, each phase winding is wound between slots of the same phase with inner and outer diameters, and the winding is simplified even if the number of phases increases. There is a way. There is also a method of making an annular winding as shown in FIG.

  Also, the shape and position of the rotor magnetic pole can be modified to the shape shown in FIG.

  When the 6S4R motor type shown in FIG. 9 is quadrupled on the circumference like the motor of FIG. 10, the number of stator magnetic poles is 24, and the number of rotor magnetic poles is 8. Control is possible with the three-phase currents Ia, Ib, and Ic, and the control can be performed without crossing the two sets of magnetic fluxes, and the problem of voltage imbalance unlike the motor of FIG. 42 does not occur. That is, the relationship between the current of each phase winding and the interlinkage magnetic flux is not electromagnetically complicated by the multipolarization.

  Next, FIG. 45 shows a motor example of the motor type 10S6R. The stator of this motor has the same configuration as the stator of FIGS. The number K of the rotor magnetic poles is 6, which is an example in which the rotor magnetic poles are uniformly arranged on the circumference. When CCW torque is generated at the rotor rotational position of FIG. 45 (a), currents Ia and Id are passed through the A-phase windings B5M and B5S and the D-phase windings B5T and B5N in the direction of the winding symbol, A magnetic flux B6A indicated by a solid line with an arrow is excited to attract the stator magnetic pole B51 and the rotor magnetic pole B61 to generate CCW torque. At the same time, the magnetic flux B6A is also excited between the stator magnetic pole B56 and the rotor magnetic pole B64, an attractive force is generated, and a CCW torque is generated.

  On the other hand, at the same time, current is supplied to the C-phase windings B5R and B5W and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux B6B indicated by the broken line with thick arrows, and the stator magnetic pole B54 and the rotor magnetic pole B63. To generate a CCW torque. At the same time, the magnetic flux B6B is also excited between the stator magnetic pole B59 and the rotor magnetic pole B66, an attractive force is generated, and a CCW torque is generated.

  In the rotor position of FIG. 45 (b) rotated slightly to CCW, currents Ia and Id are supplied to the A-phase windings B5M and B5S and the D-phase windings B5T and B5N in the direction of the winding symbol, and solid lines with thick arrows Is excited, the stator magnetic pole B51 and the rotor magnetic pole B61 are attracted, and CCW torque is generated. At the same time, the magnetic flux B6A is also excited between the stator magnetic pole B56 and the rotor magnetic pole B64, an attractive force is generated, and a CCW torque is generated.

  On the other hand, at the same time, a current is supplied to the B-phase windings B5P and B5U and the E-phase windings B5V and B5Q in the direction of the winding symbol to excite the magnetic flux B6C indicated by the broken line with a thick arrow. To generate a CCW torque. At the same time, the magnetic flux B6C is also excited between the stator magnetic pole B58 and the rotor magnetic pole B65, an attractive force is generated, and a CCW torque is generated. Similarly, rotational torque can be generated by controlling the current of each phase according to the rotor rotational position.

  FIG. 46 shows a motor example of the motor type 10S8R. The stator of this motor has the same configuration as the stator of FIGS. The number K of the rotor magnetic poles is 8, which is an example in which the rotor magnetic poles are uniformly arranged on the circumference. When CCW torque is generated at the rotor rotational position of FIG. 46 (a), currents Ia and Id are applied to the A-phase windings B5M and B5S and the D-phase windings B5T and B5N in the direction of the winding symbol, and are thick. A magnetic flux B7A indicated by a solid line with an arrow is excited to attract the stator magnetic pole B51 and the rotor magnetic pole B71, thereby generating CCW torque. At the same time, the magnetic flux B7A is also excited between the stator magnetic pole B56 and the rotor magnetic pole B75, an attractive force is generated, and a CCW torque is generated.

  At the same time, currents Ib and Id are passed through the B-phase windings B5P and B5U and the D-phase windings B5T and B5N in the direction of the winding symbol, and the magnetic flux B7B indicated by the broken line with a thick arrow is excited, The rotor magnetic pole B72 is attracted and CCW torque is generated. At the same time, the magnetic flux B58 is also excited between the stator magnetic pole B57 and the rotor magnetic pole B76, an attractive force is generated, and a CCW torque is generated.

  In the rotor position of FIG. 45 (b) rotated slightly to CCW, currents Ia and Id are supplied to the A-phase windings B5M and B5S and the D-phase windings B5T and B5N in the direction of the winding symbol, and solid lines with thick arrows Is excited to attract the stator magnetic pole B51 and the rotor magnetic pole B71 to generate CCW torque. At the same time, the magnetic flux B7A is also excited between the stator magnetic pole B56 and the rotor magnetic pole B75, an attractive force is generated, and a CCW torque is generated.

  At the same time, the currents Ia and Ic are supplied to the A-phase windings B5M and B5S and the C-phase windings B5R and B5W in the direction of the winding symbol to excite the magnetic flux B7C indicated by the broken line with thick arrows, The rotor magnetic pole B78 is attracted and CCW torque is generated. At the same time, the magnetic flux B7C is also excited between the stator magnetic pole B55 and the rotor magnetic pole B74, an attractive force is generated, and a CCW torque is generated. Similarly, rotational torque can be generated by controlling the current of each phase according to the rotor rotational position.

  Next, the motor of the present invention when the number M of the stator magnetic poles is 12 will be described. One of the motors is a motor type 12S4R in which the motor type 6S2R shown in FIGS. 9 and 11 is multipolarized to four poles, but the motor characteristics are 6S2R characteristics. The other is a motor in which the 6S4R motor of FIG. 1 is multipolarized to 4 poles, and the motor characteristics are the characteristics of 6S4R of FIG. These motors are three-phase motors. However, a part of the rotor magnetic pole is shifted in the circumferential direction by an electrical angle of about 0 ° to 30 °, or the circumferential magnetic pole width of a part of the stator magnetic pole and the rotor magnetic pole is expanded or reduced by about 0 ° to 30 °. It is possible to modify such as. It is possible to reduce torque ripple, vibration and noise.

  As another example of the motor when M is 12, an example of the motor type 12S10R is shown in FIG. The stator magnetic poles are arranged at equal intervals, and are B81, B82, B83, B84, B85, B86, B87, B88, B89, B8A, B8B, B8C. In the case of a motor of this shape, in the case of an AC motor, it can be considered as a six-phase AC motor. However, if there is a restriction that current flows in one direction to an adjacent slot, the current direction of the slot on the opposite side of 180 ° Since the currents have the same polarity, it is impossible to wind the full-pitch winding. However, the winding in the current direction shown in FIG. 47 can be modelly wound by the annular winding as shown in FIG. 14 and the winding between the slots of the mutual motor in the composite motor as shown in FIG. . B8R is an A-phase winding and conducts an A-phase current Ia. B8S is a B-phase winding and conducts a B-phase current Ib. B8T is a C-phase winding and supplies a C-phase current Ic. B8U is a D-phase winding and supplies a D-phase current Id. B8V energizes E phase current Ie with E phase winding. B8W is an F-phase winding and energizes an F-phase current If. B8X is a G-phase winding and energizes a G-phase current Ig. B8Y is an H-phase winding and energizes an H-phase current Ih. B8Z is a J-phase winding and conducts a J-phase current Ij. B8D is a K-phase winding and energizes a K-phase current Ik. B8E is an M-phase winding and energizes an M-phase current Im. B8F is an N-phase winding and energizes an N-phase current In.

  When generating the CCW torque, at the rotor position in FIG. 47, currents Ia and If are applied to the A-phase winding B8R and F-phase winding B8F in the direction of the winding symbol to excite the magnetic flux B8G, and the stator The magnetic pole B8C and the rotor magnetic pole B8K are attracted to generate CCW torque. At the same time, in parallel, currents Ia and Ib are applied to the A-phase winding B8R and B-phase winding B8S in the direction of the winding symbol, the magnetic flux B8H is excited, and the stator magnetic pole B81 and the rotor magnetic pole B8L are attracted. CCW torque is generated. At the same time, in parallel, currents Ic and Ib are supplied to the C-phase winding B8T and B-phase winding B8S in the direction of the winding symbol, the magnetic flux B8J is excited, and the stator magnetic pole B82 and the rotor magnetic pole B8M are attracted. CCW torque is generated. At the same time, in parallel, currents Ig and If are passed through the G-phase winding B8X and F-phase winding B8F in the direction of the winding symbol, the magnetic flux J11 is excited, and the stator magnetic pole B86 and the rotor magnetic pole B8N are attracted. CCW torque is generated. At the same time, currents Ig and Ih are applied to the G-phase winding B8X and the H-phase winding B8Y in the direction of the winding symbol to excite the magnetic flux J12 and attract the stator magnetic pole B87 and the rotor magnetic pole B8P. CCW torque is generated. At the same time, in parallel, currents Ij and Ih are supplied to the J-phase winding B8Z and H-phase winding B8Y in the direction of the winding symbol, the magnetic flux J13 is excited, and the stator magnetic pole B88 and the rotor magnetic pole B8Q are attracted. CCW torque is generated. Similarly, rotational torque can be generated by controlling the current of each phase according to the rotor rotational position.

  In FIG. 47, the sum of the radial magnetic fluxes of the magnetic fluxes B8G, B8H, B8J, J11, J12, and J13 may not be zero. Can loop. Further, it is not always necessary to energize the current for exciting the magnetic flux, and the magnitude of each current can be selected. Therefore, when a certain torque is generated, there are many combinations of the current values of the respective windings of the motor, and it can be said that there are various driving methods. It can be selected for convenience such as reducing the burden on the inverter, reducing iron loss, reducing copper loss, and reducing winding voltage.

  In the motor shown in FIG. 47, two sets of windings are inserted into each slot to operate as a 12-phase motor, or a part of the windings can be configured to pass a bidirectional current. It can also be a whole volume.

  Next, the motor of the present invention when the number M of stator magnetic poles is 14 will be described. The winding may be a full-pitch winding, an annular winding as shown in FIG. 14, or a composite motor winding as shown in FIG. However, it becomes a 7-phase motor, and in the case of full-pitch winding, the windings of each phase intersect in a complicated manner, making it difficult to manufacture. In that respect, the winding of each phase does not interfere with the annular winding shown in FIG. 14 and the winding of the composite motor shown in FIG. It is easy and there is no problem that the coil end portion of the motor becomes large.

  In the case of full-pitch winding, as shown in FIG. 48, the A-phase winding is energized with current Ia at B9V and B9H. The B-phase winding is energized with current Ib at B9X and B9K. The C-phase winding passes current Ic at B9Z and B9M. The D-phase winding is energized with current Id by B9G and B9P. The E-phase winding is energized with current Ie at B9J and B9W. The F-phase winding passes current If through B9L and B9Y. The G-phase winding passes current Ig with B9N and B9F. B91, B92, B93, B94, B95, B96, B97, B98, B9A, B9B, B9C, B9D, and B9E are stator magnetic poles.

  FIG. 48 shows an example of the motor type 14S4R. This motor can be driven by passing a current in the same manner as the motor of FIG. However, the difference from the motor of FIG. 4 is that there is a lot of free space on the outer periphery of the rotor, and that space can be used for some other purpose. It can be used when the rotor has shape constraints.

  FIG. 49 shows an example of the motor format 14S6R. This motor has six rotor magnetic poles and can overlap the torque generation section of each rotor magnetic pole. Therefore, the motor shown in FIG. 48 has a feature that a larger torque can be obtained and torque ripple can be reduced. .

  FIG. 50 shows an example of the motor type 14S8R. This motor has 8 rotor magnetic poles, and when the rotor rotates, 4 rotor magnetic poles can be used for torque generation, and torque can be generated by the current of 8 windings out of 14 windings. it can. The motor torque is large, the current capacity of the power transistor of the control device can be reduced, and the cost and size can be reduced.

  When the CCW torque is generated at the rotor rotation position of FIG. 50, the current Ia is supplied to the A-phase windings B9V and B9H, and at the same time, the current Ie is supplied to the E-phase windings at B9J and B9W. Energized to attract the stator magnetic pole B91 and the rotor magnetic pole C11 to generate CCW torque. At the same time, the stator magnetic pole B98 and the rotor magnetic pole C14 are attracted by the magnetic flux C17 to generate CCW torque.

  At the same time, the current Ib is supplied to the B-phase windings B9V and B9H, and the F-phase winding is supplied with the current If at B9L and B9Y at the same time, thereby exciting the magnetic flux C18, and the stator magnetic pole B93 and the rotor magnetic pole C12. To generate CCW torque. At the same time, the magnetic flux C18 attracts the stator magnetic pole B9A and the rotor magnetic pole C15 to generate CCW torque.

  At the same time, the current Ic is supplied to the C-phase windings B9Z and B9M, and at the same time, the current Ig is supplied to the G-phase windings at B9N and B9F, thereby exciting the magnetic flux C19 and the stator magnetic pole B95 and the rotor magnetic pole. C13 is sucked and CCW torque is generated. At the same time, the magnetic flux C19 attracts the stator magnetic pole B9C and the rotor magnetic pole C16 to generate CCW torque. In the following, the rotor can be rotated sequentially by exciting the stator magnetic poles that can generate a desired torque in accordance with the rotational position of the rotor. A so-called four-quadrant operation is possible by a combination of forward rotation, reverse rotation, power running torque, and regenerative torque.

  As described above, in the motor shown in FIG. 50, torque can be generated simultaneously by four stator magnetic poles or six stator magnetic poles, and the torque can be increased. Conversely, by limiting the stator magnetic poles that generate torque, the generated voltage of the winding can be reduced, and higher speed rotation can be realized within the constraints of the DC power supply voltage. Also, in terms of the control device, each winding is a unidirectional current, and each winding is also used to excite the stator poles on both sides. The current capacity of the transistor can be reduced, and the cost and size of the control device can be reduced.

  Next, FIG. 51 shows an example of the motor type 14S10R. This motor is the same stator as the motor shown in FIG. 50, and the rotor has ten stator magnetic poles. For example, when CCW torque is generated in the state of the rotor rotational position of FIG. 51, the magnetic flux indicated by the arrow is excited in the same manner as in the case of the motor of FIG. be able to. The motor characteristics are similar to the motor of FIG.

  Next, FIG. 52 shows an example of the motor type 14S12R. This motor is the same stator as the motor shown in FIG. 50, and the rotor has ten stator magnetic poles. For example, when the CCW torque is generated in the state of the rotor rotational position of FIG. 51, the magnetic force indicated by the arrow can be excited to generate an attractive force to generate the CCW torque. However, the distribution of magnetic flux in FIG. 52 is very different from that in FIGS. This is such a magnetic flux distribution because the number of stator magnetic poles M = 14 and the number of rotor magnetic poles K = 12 are close, and the magnetic flux distribution is similar to a so-called vernier motor. There is a point. However, in the motor of FIG. 52, the current direction of each winding is opposite to the current direction of the winding of the adjacent slot, the direction of the magnetic flux is different, and each rotor magnetic pole is individually excited by each stator magnetic pole. Is different. As a result, the difference is that the back yoke of the stator can be made thinner and the control device can be made smaller. The torque characteristics and the like of the motor shown in FIG. 52 are similar to the motor characteristics shown in FIGS. 50 and 51, so that a large torque can be generated, and the torque generator can be selectively controlled.

  As described above, when the number M of stator magnetic poles and the number K of rotor magnetic poles are large values as shown in FIGS. 50, 51, 52, etc., torque can be generated with many stator magnetic poles. In many cases, torque can be achieved.

  At this time, each winding current excites a soft magnetic material to induce a magnetic flux, but each magnetic flux is often linked to a plurality of windings and has a large mutual inductance. As a result, the voltage Vz of each winding is affected by the total magnetic flux φa interlinked, and becomes the product of the number of turns Nw of the winding and the time change rate of the total magnetism. The voltage of the winding is larger than the voltage when only the magnetic flux of one specific stator magnetic pole is generated. Therefore, when the motor is driven within the constraints of the DC power supply voltage, the drivable rotational speed ωr is restricted by the DC power supply voltage.

  As a countermeasure, when operating at a high rotational speed ωr, the stator magnetic pole that generates torque is limited to reduce the generated voltage of the winding and realize higher speed rotation within the constraints of the DC power supply voltage. You can also

  Also, in terms of the control device, the current passed through each winding is a one-way current, and each winding is also used to excite both adjacent stator poles, so power is simultaneously supplied to the motor from multiple winding paths. Therefore, the current capacity of the power transistor can be reduced, and the control device can be reduced in cost and size.

  In the above specific motor examples, many cases have been described in which the number M of stator magnetic poles and the number K of rotor magnetic poles are even. However, even when the number M of the stator magnetic poles or the number K of the rotor magnetic poles is an odd number, the motor of the present invention and its control can be controlled by a partial change such that the partial windings can be driven by a bidirectional current. Some of the features shown in the device examples can be obtained consistently and are included in the present invention.

  In FIG. 39 and the like, the circumferential width of the stator magnetic pole and the rotor magnetic pole is shown as 360 ° / (8 × 2) = 22.5 °, but the circumferential width is increased or decreased depending on the required characteristics of the motor. You can also. Each pitch can also be moved in the circumferential direction for the purpose of reducing torque ripple, and can be arranged at unequal intervals. Further, the shape of each magnetic pole and the shape of the slot can also be deformed in the circumferential direction, the rotor axial direction, and the radial direction. It is also possible to perform skewing or delete a part of the magnetic pole. Of course, multipolarization is also possible. In addition, when the number of poles is increased, the configuration of the electrical angle of 360 ° is not made exactly the same, but can be modified such as being slightly shifted in the circumferential direction. It is also possible to arrange a plurality of motors having different configurations in the circumferential direction. Further, it is possible to realize a configuration in which the number K of rotor magnetic poles is larger than the number M of stator magnetic poles.

  Next, FIG. 53 shows an example in which the circumferential width Ht expressed by the angle of the stator magnetic pole is larger than 360 ° / (2M). This motor is a motor type 6S4R. The stator magnetic pole width Ht and rotor magnetic pole width Hm of the 6S4R motor shown in FIG. 1 are 30 °, whereas the stator magnetic pole width of FIG. 53 is 40 ° and the rotor magnetic pole width is also about 40 °. In the motor of FIG. 1, the width at which a certain stator magnetic pole attracts the rotor magnetic pole to generate torque is 30 °, whereas in the motor of FIG. 53, it is widened to 40 °. Therefore, in the motor of FIG. 53, when the generation of torque is transferred between two stator magnetic poles, it is possible to perform control with a margin so that the generated torque on one side is reduced while the generated torque on the other side is increased. However, the slot area is reduced and the winding resistance is increased.

  Next, another motor example of the present invention is shown in FIG. A rotor magnetic pole 162 having a small circumferential magnetic pole width is added to the 6S2R motor of FIG. Others are the same as the motor shown in FIG. In the motor of FIG. 54, the rotor magnetic pole 161 having a large circumferential magnetic pole width is called a main salient pole, and the rotor magnetic pole 162 having a small circumferential magnetic pole width is called an auxiliary salient pole. The main salient pole magnetic pole 161 has a circumferential magnetic pole width Hm of about 40 °, and the auxiliary salient pole magnetic pole 162 has a circumferential magnetic pole width Hh of about 20 °. In this example, the four salient poles of the rotor have an electrical angle difference of 90 degrees from each other and are arranged at equal intervals. The motor shown in FIG. 54 is also an example of a motor in which the rotor magnetic pole width has two different values compared to the 6S4R motor shown in FIG.

  In FIG. 54, a motor having a two-pole configuration is shown for ease of explanation. However, in reality, it is used in a multipolar form such as 4 poles or 8 poles. By reducing the number of coils, the motor can be reduced in size, for example, by shortening the coil end portion or reducing the thickness of the back yoke portion of the stator. FIG. 55 shows a configuration in which the two-pole motor shown in FIG. Compared to the motor of FIG. 10, an auxiliary salient pole magnetic pole 173 is added to the rotor 171. Reference numeral 172 denotes a main salient pole magnetic pole. Between the electrical angles of 360 °, the four salient poles of the rotor are arranged at equal intervals with an electrical angle difference of 90 degrees between them.

  The purpose of the motor configuration in FIG. 54 is to improve the torque reduction portion shown in FIG. 12G, which is a characteristic of the motor in FIG. The operation of the motor of FIG. 54 will be described with reference to FIGS. 56 (a) to (f). In this example, the width of the opening of the slot is 20 °, the width Hm in the circumferential direction of the main salient pole magnetic pole 161 of the rotor is 40 °, and the width in the circumferential direction of the auxiliary salient pole magnetic pole 162 is 20 °. FIG. 57 shows the rotor rotational position θr expressed in electrical angle on the horizontal axis, the current of each phase, and the torque of each phase. The A-phase current is Ia, the B-phase current is Ib, and the C-phase current is Ic.

  The torque of the motor shown in FIG. 54, FIG. 55, and FIG. Therefore, in order to generate torque, current is passed through at least two windings to generate torque. Then, a reluctance torque is obtained by generating an attractive force between the salient pole-like teeth of the stator and the salient pole of the rotor. The current direction of each winding is the current direction indicated by the winding symbol. Therefore, the direction of the magnetic flux of each stator magnetic pole is also a one-way magnetic flux.

  The stator magnetic poles 11C and 119 generate torque generated between the rotor main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162, and the stator magnetic poles 118 and 11B contact the rotor main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162. The torque generated between the stator magnetic poles 117 and 11A and the rotor main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 is Tc. At this time, it is a characteristic point of the reluctance torque that each attractive force generates the same attractive force and torque regardless of whether the direction of the magnetic flux is positive or negative.

  When the rotor is in the vicinity of the rotational position of θr = 30 ° shown in FIG. 56 (a), a positive current Ia is supplied to the A-phase winding 111 and a negative current − is supplied to the A-phase winding 114 on the opposite side. Run Ia. At the same time, a positive current Ic is supplied to the C-phase winding 115, and a negative current −Ic is supplied to the opposite C-phase winding 112. No current flows through the B-phase windings 113 and 116. The currents Ia, Ib, and Ic of the respective phases are the currents shown in (A), (C), and (E) of FIG. In this state, in accordance with Ampere's law, magnetomotive forces of the A-phase current Ia and the C-phase current Ic act in the directions indicated by the thick arrows in the direction from the stator magnetic poles 11A to 117, and the direction indicated by the arrows, from the stator magnetic poles 11A to the stator magnetic poles. Magnetic flux is induced in the direction of 117. Then, torque Tc shown in FIG. 57 (F) is generated in the counterclockwise direction CCW. Here, the permeability of the soft magnetic part of the stator and the rotor is sufficiently large, the permeability of the wide space between the stator and the rotor is sufficiently small, and the magnetic resistance of the narrow air gap between the stator and the rotor is sufficiently large In a simple model that is assumed to be small, the magnetic field strength H [A / m] acting in the vicinity of the stator magnetic poles 118, 119, 11B, and 11C is almost zero, and these teeth that are the stator magnetic poles pass through the teeth in the radial direction. The magnetic flux is almost zero and the torque is almost zero.

  When the rotor rotates to CCW and rotates to the vicinity of the rotational position of θr = 50 ° shown in FIG. 56B, a positive current Ia flows through the A-phase winding 111 and a negative current − Run Ia. At the same time, a positive current Ib is supplied to the B-phase winding 113, and a negative current −Ib is supplied to the opposite B-phase winding 116. No current flows through the C-phase windings 115 and 112. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the thick arrows in the direction of the stator salient poles 118 to 11B. Magnetic flux is induced in the direction of 119. Then, the rotor generates torque Ta shown in FIG. In this motor model, the circumferential width of the auxiliary salient pole magnetic pole 162 is as narrow as 20 °, so the torque width near θr = 60 ° shown in FIG.

  When the rotor rotates to CCW and rotates to the vicinity of the rotational position of θr = 70 ° shown in FIG. 56C, a positive current Ic flows through the C-phase winding 115 and a negative current −Ic flows through the C-phase winding 112. Shed. At the same time, a positive current Ib is supplied to the B-phase winding 113, and a negative current −Ib is supplied to the opposite B-phase winding 116. No current flows through the A-phase windings 111 and 114. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the thick arrows in the direction of the stator salient poles 118 to 11B. Magnetic flux is induced in the direction of 11B. Then, the rotor generates a torque Tb shown in FIG. And it rotates to CCW and rotates to the rotation position of (theta) r = 90 degrees shown to (d) of FIG.

  The motor shown in FIG. 54 has a periodicity of 60 ° in electrical angle, and a similar drive can be performed at a cycle of 60 °. The relative relationship between the operation at θr = 30 ° in FIG. 56 (a) and the operation at θr = 90 ° in FIG. 56 (d) is opposite in the direction of current and the direction of magnetic flux. However, both torques T have the same magnitude in the CCW direction. Thus, when the rotational position θr is between 90 ° and 150 °, torque can be generated and rotated in a similar manner to the motor operation when the rotational position θr is between 30 ° and 90 °. Similarly, rotation can be performed in a similar manner between 150 ° and 210 °, between 210 ° and 270 °, between 270 ° and 330 °, and between 330 ° and 30 °. Specifically, as shown in FIG. 57, torques Ta, Tb, Tc are obtained by changing the currents Ia, Ib, Ic sequentially energized according to the rotor rotational position θr, and the rotor is rotated.

  A solid line in FIG. 57G shows the motor torque Tm obtained by transferring the generated torques Ta, Tb, and Tc of each tooth. The torque Tm in (G) of FIG. 57 has a characteristic in which the torque slightly decreases when the current of each phase winding is switched. The torque Tm has improved torque drop compared to the torque Tm shown in FIG. Further, in order to reduce the partial torque drop of the torque Tm shown in FIG. 57 (G), the circumferential widths Hm and Hh of the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 of the rotor are slightly widened. By setting to, further improvement can be achieved.

  Further, when the rotor rotational speed is large, various devices such as advancing the current phase or changing the current waveform as shown by the broken line in FIG. 6 are possible.

  Next, the relationship between the circumferential width Hs of the opening of the slot, the circumferential width Hm of the main salient pole magnetic pole 161 of the rotor, the circumferential width Hh of the auxiliary salient pole magnetic pole 162, and the output torque T of the motor will be described. To do. First, consider the conditions under which a continuous torque can be generated. Consider the conditions under which the motor torque Tm can be generated continuously in the CCW direction at the rotor rotational position of FIG. Now, consider the case where the main salient pole magnetic pole 161 is rotating to CCW and reaches the tooth 117, and the rotation position θr is such that the left end of the main salient pole magnetic pole 161 coincides with the left end of the tooth 117. The rotational position θr in FIG. 58 is just the rotational position at which the main salient pole 161 can no longer generate torque in the CCW direction. Consider the conditions under which the auxiliary salient pole 162 can generate torque in the CCW direction at this rotational position θr. The condition is that the sum Hg of the width of the gap between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 and the width of the auxiliary salient pole magnetic pole 162 is as shown in the following equations (21) and (22): It is larger than the width Hf obtained by adding 60 ° to the circumferential width Hs of the opening of the slot.

Hg> Hf (21)
(360 ° − (Hm + Hh) × 2) / 4 + Hh> 60 ° + Hs (22)
Next, as shown in FIG. 59, a condition is considered in which the main salient pole magnetic pole 161 does not generate torque in the clockwise direction CW when the auxiliary salient pole magnetic pole 162 generates torque in the CCW direction. The condition is that the width Hb of the gap between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 is larger than the width Ht of the stator teeth as shown in the following equations (23) and (24).

Hb> Ht (23)
(360 ° − (Hm + Hh) × 2) / 4> 60 ° −Hs (24)
For example, assuming the circumferential width Hs of the opening of the slot and the circumferential width Hm of the main salient pole magnetic pole 161, and obtaining the circumferential width Hh of the auxiliary salient pole magnetic pole 162 that meets the conditions, (22) The width Hh can be expressed by the following equation from the equation and the equation (24).

Hm + 2Hs−60 ° <Hh <−Hm + 2Hs + 60 ° (25)
In order to simplify the conditions, the circumferential width Hm of the main salient pole magnetic pole 161 is larger than the circumferential width Hh of the auxiliary salient pole magnetic pole 162 as shown in the following equation (26).

Hm> Hh (26)
Further, since the rotation angle of at least 60 ° is driven by the main salient pole magnetic pole and the auxiliary salient pole magnetic pole, the condition of the following equation (27) is satisfied.

Hm + Hh> 60 ° (27)
The table of FIG. 60 shows examples of specific widths that satisfy the conditions of the expressions (25), (26), and (27). The horizontal axis is the circumferential width Hs of the opening of the slot, the vertical axis is the circumferential width Hm of the main salient pole magnetic pole 161, and the minimum value that satisfies the condition of the circumferential width Hh of the auxiliary salient pole magnetic pole 162 in the table. It is indicated by Min and the maximum value Max. For example, when the circumferential width Hs = 15 ° of the opening of the slot and the circumferential width Hm = 50 ° of the main salient pole magnetic pole 161, the circumferential width Hh of the auxiliary salient pole magnetic pole 162 satisfying the condition is the minimum value. The maximum value is 20 ° from 20 °.

  In FIG. 60, a thick double line frame is shown, but the outer thick line frame is a range in which the correlation between the Hs, Hm, and Hh has a margin. The inner thick line frame is a range in which the degree of freedom of selection of each value is further large.

  In addition, the circumferential width Hh of the auxiliary salient pole magnetic pole 162 in FIG. 58 may be practically used even if the range between the minimum value and the maximum value is slightly different. For example, the auxiliary salient pole magnetic pole 162 and the teeth of the stator may generate attractive force and torque even if the auxiliary salient pole magnetic pole 162 and the tooth 119 are not opposed to each other in the radial direction in FIG. it can. Therefore, continuous torque can be generated even with a value of Hh that is slightly smaller than Hh represented by the equation (22) simply created as a model. Furthermore, since there are many applications that can be used even with somewhat intermittent motor torque, the present invention can be put into practical use even if the expressions (22) and (24) are slightly deviated.

  As described above, in the motor configuration of FIG. 54, the function of the auxiliary salient pole magnetic pole 162 is the generation of torque at the torque reduction portion in the motor of FIG. As a result, torque can be generated over the entire circumference of the motor, particularly at low speed rotation, and free motor operation at low speed rotation is possible. As one of the motor operation methods of FIG. 54, when the rotation becomes high-speed rotation, torque generation at the main salient pole magnetic pole 161 is mainly performed, torque generation at the auxiliary salient pole magnetic pole 162 is reduced, and sufficiently high-speed rotation is performed. Then, the torque can be generated only by the main salient pole magnetic pole 161, and the torque can be positively not generated by the auxiliary salient pole magnetic pole 162. By performing the operation only with the main salient pole 161, it is possible to reduce the complexity of the control and make the control simpler. Moreover, since the change of the magnetic flux in the auxiliary salient pole magnetic pole 162 and the stator magnetic pole can be reduced, there is an effect of reducing iron loss.

  However, at this time, when control is performed in which the three-phase unidirectional current is sequentially switched with rotation, a phenomenon in which magnetic flux is induced in the auxiliary salient pole magnetic pole 162 occurs due to a transient unbalance of the two currents. . In other words, a negative effect due to the presence of the auxiliary salient pole magnetic pole 162 also occurs. Therefore, when the required specification of the motor shown in FIG. 9 is only required to improve the torque reduction part of FIG. 12G to some extent, the auxiliary salient pole 162 of the motor of FIG. 54 is partially deleted. Is also possible. Specifically, the auxiliary salient pole magnetic pole 162 shown in FIG. 54 does not exist from the end in the rotor axial direction to the other end, but can be reduced to, for example, about half. As shown in FIG. 55, in the case of a multi-pole motor, eight auxiliary salient poles 173 are arranged on the circumference, but the number can be reduced to, for example, four. As described above, since the auxiliary salient pole has advantages and disadvantages, the amount of the auxiliary salient pole can be varied depending on the application.

  Further, as a method of changing the electromagnetic action of the motor between the low speed rotation and the high speed rotation, a rotor capable of generating a high torque at a low speed rotation like the motor shown in FIG. 54, and a slight torque ripple as shown in FIGS. Although it tends to be large, a large average torque can be obtained at high speed rotation, and a rotor having a quiet characteristic is arranged in parallel on the same axis, and the motor is moved by moving the stator and the rotor relatively in the axial direction of the rotor. The electromagnetic characteristics can also be changed.

  When the motors shown in FIGS. 54 and 55 are driven at a low speed, the main salient poles 161 and the auxiliary salient poles 162 can be alternately attracted to obtain a continuous torque. However, in high-speed rotation, inertial rotation is possible for a short time, and it is simpler to obtain rotational torque with only the main salient pole magnetic pole 161, and iron loss is small, and vibration and noise are also small. it can.

  In order to operate more effectively during this high speed rotation, it is preferable to eliminate the auxiliary salient pole magnetic pole 162 and increase the circumferential width of the main salient pole magnetic pole 162. When the circumferential width of the main salient pole magnetic pole 162 is large, it is possible to provide a margin for the time to increase or decrease the current that generates torque. In addition, it is preferable in terms of control that the circumferential width Ht of the stator magnetic pole is large so that the leakage magnetic flux between the teeth does not cause a problem.

  In order to meet these requirements, it is only necessary to change the shape of the rotor magnetic pole during rotation. For example, a mechanism KS capable of moving in the rotor axial direction is provided on the stator side, and a mechanism KR that is rotatable but interlocked with the mechanism KS in the rotor axial direction is provided on the rotor side. If the structure is interlocked with the radial movement of the magnetic pole 162, the auxiliary salient magnetic pole 162 can be moved in the radial direction by driving the mechanism KS on the stator side with a small servo motor or the like. And the presence or absence of the auxiliary salient pole magnetic pole 162 can be controlled during rotation from the stator side. FIG. 61A shows an image of the auxiliary salient pole magnetic pole 162 moving in the radial direction by an arrow line F41 and a broken line.

  Alternatively, if the rotor side mechanism KR is a mechanism capable of moving the auxiliary salient pole magnetic pole 162 in the circumferential direction, the auxiliary salient pole magnetic pole 162 is driven by driving the stator side mechanism KS by a small servo motor or the like. Is adjacent to the main salient pole magnetic pole 161 as indicated by an arrow F42, and is made a part of the main salient pole magnetic pole 161, substantially widening the main salient pole magnetic pole 161, and at the same time eliminating the auxiliary salient pole magnetic pole 162. become.

  Further, if the main salient pole 161 is divided into, for example, SS1 and SS2, and the salient pole SS1 indicated by the broken line moves in the direction indicated by the arrow F41 in FIG. The circumferential width of the magnetic pole 161 can be increased. In normal high-speed rotation, since the power supply voltage of the control device is finite, the amount of magnetic flux is reduced by field weakening or the like so that the voltage on the motor side does not increase. As described above, when the main salient pole magnetic pole is divided into two in the circumferential direction at high speed rotation, the amount of magnetic flux per unit width in the circumferential direction of the rotor magnetic pole 161 is reduced, which is preferable in terms of voltage.

  Further, there are various methods for driving a part of the rotor side from the stator side. For example, a small motor is built in the rotor side, and the rotation of the motor is controlled from the stator side to control the rotor shape and rotor. It is also possible to change the position of the magnetic pole.

  Next, FIG. 62 shows another example in which the motor characteristics can be changed during rotation. F43 is a rotor shaft, and F4B is a bearing. F44 is a stator core and F45 is a coil end. F46, F47, F48, F49, and F4A are rotor units. By changing the relative positions of the stator and the rotor in the axial direction of the rotor as indicated by arrows, the electromagnetic characteristics of the motor can be changed so as to obtain the desired characteristics. The target characteristics are, for example, characteristics having the four rotor salient poles shown in FIG. 1 in the range of an electrical angle of 360 ° at low speed rotation, and at an electrical angle of 360 ° as shown in FIGS. 9 and 10 at medium speed. It has a simple characteristic with two rotor salient poles in the range, and the electromagnetic characteristic in which the rotor axial length of the rotor of FIGS. 9 and 10 is reduced so that the entire field magnetic flux of the motor is reduced at high speed rotation. It is a characteristic that it has. If such a characteristic is obtained, torque ripple is small and large torque can be obtained at low speed, high efficiency operation with low iron loss can be realized at medium speed, and the field flux at high speed can be reduced to reduce motor flux. The constant voltage is maintained appropriately so that the induced voltage does not become excessive, and excellent constant output characteristics can be obtained. Also, good characteristics from low speed to high speed can be obtained with a control device having a relatively small current capacity.

  Next, consider the relationship between the circumferential width Ht of the stator magnetic poles and the circumferential widths Hm and Ht of the rotor magnetic poles. For example, at the rotor position in FIG. 54, the rotor magnetic pole 162 is in the circumferential range of the stator magnetic pole 118, so that the current Ib of the B-phase windings 113 and 116 and the C-phase windings 115 and 112 are in this state. Even if the current Ic is increased or decreased, the rotor magnetic pole 162 does not generate torque. Even when the rotor is rotating, it is possible to control without adversely affecting other parts by changing Ib and Ic within the range of the stator magnetic pole 118. In this way, the circumferential width Ht of the stator magnetic poles and the circumferential widths Hm, Ht of the rotor magnetic poles can be set to different values to reduce adverse effects such as unintended negative torque.

  Next, a countermeasure for the problem that the torque decreases when the motor shown in FIG. 9 is rotated to the CCW and reaches the rotational position shown in FIG. As shown in FIG. 63, by adding soft magnetic projections 151, 152, 153, 154 to the inner diameter side of the rotor radius R at the circumferential end of the rotor 13E, torque is generated in the vicinity of this rotational position. Can be increased and improved. The distance between the stator teeth 118 and the protrusions 152 is shortened, and the magnetic resistance therebetween is reduced. The relationship between the stator teeth 11B and the protrusions 153 is the same. For example, in the case of a motor that only needs torque in the CCW direction, only the soft magnetic bodies 152 and 153 need be added.

  In addition, as in the motor shown in FIG. 47, when the sum of the radial magnetic fluxes between the stator and the rotor does not become zero, the magnetic flux may not be allowed to pass by the unbalance. In such an unbalanced relationship, by providing some magnetic bypass between the stator and the rotor, the amount of magnetic flux passing between the stator magnetic pole and the rotor magnetic pole can be increased, and torque can be generated more effectively. It can be configured. In addition, as another method, two motors in which the magnetic flux amount relationship between the stator and the rotor is opposite to each other are arranged in parallel, and the stator and the rotor are magnetically connected to each other. Torque can be generated.

  Next, FIG. 64 shows an example in which the circumferential positions of the rotor magnetic poles are arranged asymmetrically. The main salient poles 161 are directed upward and downward in the drawing, while the auxiliary salient poles 162 are inclined upward to the water line on the left and right sides of the drawing. In a motor or the like used for one-way rotation, the torque characteristics of one-way rotation can be improved by arranging the asymmetric rotor poles in this way.

  65 is a motor in which the 6S4R two-pole motor shown in FIG. 1 is converted into four poles, and the number of stator magnetic poles and rotor magnetic poles is doubled. However, the arrangement of the eight rotor magnetic poles is not equidistant, and the four rotor magnetic poles C78, C79, C7A, C7B are mechanical angles in the clockwise direction CW with respect to the other stator magnetic poles C74, C75, C76, C77. In this example, the electrical angle is shifted by 15 ° and 30 °. Thus, torque ripple can be reduced by shifting the positions of some of the magnetic poles in the circumferential direction. However, it is necessary to pay attention to the problem of torque imbalance.

  Next, the shape of the mutually opposing surfaces of the stator magnetic pole or the rotor magnetic pole will be described. Since it is a relative shape between the stator and the rotor, either shape may be used. Further, the shape of both the stator and the rotor may be a shape obtained by deforming a rectangle, and a shape that can obtain the effect of the shape relatively shown in FIG. 66 may be used. FIG. 66F is a diagram showing the outer peripheral surface shape of the rotor magnetic pole A0K of the motor shown in FIG. It has an almost rectangular shape. The horizontal axis represents about a half circumference from 0 ° to 180 ° at the rotor rotational position θr. The vertical direction is the rotor axial direction. 66E shows a shape obtained by skewing the shape of FIG. Skew generally has the effect of reducing torque ripple. However, there is a tendency for the torque to decrease slightly, and attention must be paid to this point. In the case of the motor of the present invention, the effect of reducing torque ripple and the effect of expanding the torque generation range of the rotor can be expected. FIG. 66D shows a magnetic pole shape with fine irregularities. The component of the magnetic flux in the magnetic pole in the rotor axial direction is small, and the eddy current in the magnetic steel sheet is small. Further, since the structure is symmetric in the rotor axial direction, the force acting in the rotor axial direction is also small. Although it is excellent in terms of electromagnetics, it has a fine structure, so it must be devised for its production. (C) in FIG. 66 is a so-called step skew, which has an effect similar to that in (E) of FIG. 66, and is used when inconvenience occurs when the skew is performed. In the case of FIG. 66 (C), the rotor axial direction vibration may occur because of the asymmetry in the rotor axial direction. FIG. 66B shows a configuration in which step skew is realized symmetrically in the rotor axis direction.

  FIG. 66A shows a shape in which there is a protrusion having a length Sb on the left side in the circumferential direction and a protrusion having a length Sc on the right side in the circumferential direction. The portion of the original magnetic pole shape in FIG. 66 (F) which is the same position in the circumferential direction as the original original shape has a rotor axial length twice as large as Sa, and has a rotor axial width Sd in the circumferential direction. There are protrusions. Since Sa × 2 is sufficiently larger than Sd, the characteristics are similar to those of the magnetic pole shown in FIG. 66 (F). However, since the circumferential protrusion acts, the torque generation range in the circumferential direction of the rotor Has the effect of extending the length by Sb or Sc. Since the number of protrusions is small, the torque that can be generated by the protrusions is not large, but it is effective when it is necessary to expand the torque generation range. For example, the motor of FIG. 9 has several places where it is difficult to generate torque. In such a case, a structure such as a step skew is suitable.

  Also, the shape may be symmetrical, such as each magnetic pole shape in the vicinity of the rotor rotational position θr of 180 ° in FIG. This can make the characteristics of CCW and CW more symmetrical. However, there is a manufacturing burden such as an increase in the types of molds when creating the shape of the electromagnetic steel sheet.

  FIG. 66 shows an example in which the rotor magnetic pole width is 30 °, but the electrical angle can be set to a value of about 20 ° to 60 °. 9 has a configuration in which two rotor magnetic poles are arranged at an electrical angle of 360 °, and is different from the motor model of the four rotor magnetic poles shown in FIG. Therefore, in the case of the motor model of FIG. 9, the rotor magnetic pole width Hm should be set from 40 ° to 90 ° in electrical angle, and the various shapes in FIG. 66 double the angle of the rotor rotational position θr on the horizontal axis. It is necessary to look at the degree. Specifically, for example, a point whose horizontal axis is 30 ° is replaced with about 40 ° to 90 °, the 90 ° position is 180 °, and the 180 ° position is 360 °. Therefore, for example, when the rotor magnetic pole width Hm is 60 ° in electrical angle, the space between the rotor magnetic poles is 120 ° in electrical angle, and when Hm is 80 °, the space between the rotor magnetic poles is 100 °. It is.

  Next, a motor having six stator magnetic poles with an electrical angle of 360 ° as shown in FIGS. 9 and 11 and two rotor magnetic poles, for example, continuously torque without interruption in the CCW direction. A structure that generates the above will be described. For example, in FIG. 11B, since the slot opening width Hs is 20 °, the magnetic flux 11X indicated by the arrow is smaller than the air gap length between the stator and the rotor, and the generated torque is not zero but small. There is. In order to cover the range of 360 ° with the six stator magnetic poles, it is necessary for one stator magnetic pole to generate torque in the range of 60 ° or more.

  As shown in FIGS. 9 and 11, since the shape of the stator magnetic pole facing the rotor is almost rectangular and the circumferential width Ht is 40 ° in electrical angle, the rotor magnetic pole is oriented toward the stator. Consider the rotor magnetic pole shape shown in FIG.

  When the width Sf in FIG. 66 is 40 ° in electrical angle, the following equation is obtained.

Sb + (Se−Sf) / 2 + Sf ≧ 60 °
∴ Sb + Se / 2 ≧ 40 °
When the width Sf in FIG. 66 is larger than 40 ° in electrical angle, the following equation is obtained.

Sb + (Se−Sf) / 2 + 40 ° ≧ 60 °
Ｓ Sb + (Se−Sf) / 2 ≧ 20 °
At this time, the magnitude of torque differs depending on the shape of each magnetic pole. In addition, since the rotation is performed while inheriting the generation of torque between the stator magnetic poles adjacent in the circumferential direction, 60 ° in the conditional expression may be set to a slightly large value when it is desired to have a slight margin. There is design freedom.

  In the above example, it is assumed that the stator magnetic pole is rectangular and the rotor magnetic pole shape is as shown in FIG. 66, but both magnetic pole shapes may have various shapes. For example, a shape including a curved shape may be used, or a structure in which the radial direction, that is, the air gap changes, may be used. The structure in which the circumferential force and the radial force do not change suddenly with rotation is important for reducing the vibration of the motor, and can be improved by devising each magnetic pole shape.

  In addition, the structure that continuously generates torque without interruption by the motor as shown in FIGS. 9 and 11 is a multi-pole motor, and in the case of a multi-pole motor, half the number of rotor magnetic poles is set to an electrical angle of 20 in the circumferential direction. A continuous torque can be generated even if the method is shifted. For example, in the case of an 8-pole motor, the rotor poles for 2 poles are in the initial state, the rotor poles for the next 2 poles are shifted to CCW by 20 °, and the rotor poles for the next 2 poles are If the rotor magnetic poles of the next two poles are shifted to CCW by 20 °, torque can be continuously generated. In addition, this structure has a point-symmetric rotor shape with respect to the rotor center point, and is excellent in terms of mechanical balance. Further, the method of shifting the rotor magnetic pole in the circumferential direction and the method of FIG. 66 may be combined.

  Next, regarding the example of the motor of the present invention shown in FIG. 9, the relationship between the shape of the stator magnetic pole facing the rotor, the shape of the rotor magnetic pole facing the stator, and the torque that can be generated between them is shown in FIGS. explain. The motor torque of the motor type 6S2R shown in FIG. 9 is different from the motor of the motor type 6S4R shown in FIG. On the other hand, when the circumferential width of the slot is reduced, there is a problem that the winding space is reduced. Even if only the slot opening is narrowed, there is a problem of leakage magnetic flux between the stator magnetic poles and a problem of magnetic saturation of the stator magnetic poles. Therefore, in the case of the motor as shown in FIG. 9, the circumferential width of the stator magnetic pole and the size of the slot opening have a trade-off relationship with respect to torque and leakage flux, and it is difficult to achieve both of them. is there.

  67 (S1) is a diagram in which the shape of the stator magnetic poles 117, 118, 119, 11A, 11B, and 11C in FIG. The horizontal axis is the angle of the circumferential position of the stator. This horizontal axis is indicated by an electrical angle in the case of the 8-pole motor of FIG. The vertical axis is the rotor axial direction. The shape of the stator magnetic pole of the motor shown in FIG. 9 facing the rotor is shown by six rectangles, the circumferential width is 40 °, and the circumferential width of the slot opening between the stator magnetic poles is 20 °.

  (R1) in FIG. 67 is a diagram in which the shape of the rotor magnetic pole 11E in FIG. Here, the horizontal axis and the vertical axis are the same as (S1). The circumferential width of the rotor magnetic pole is 60 °, and the rotor axial width is the same as the rotor axial width of the stator magnetic pole.

  67 (T1) is the torque of the motor of FIG. 9, and the torque when the rotor 11E of the rotor magnetic pole shape (R1) is rotated to CCW when the stator magnetic pole shape (S1) is the same is shown in FIG. Torque expressed as a model. The simple modeling assumes that the air gap length between the stator and the rotor is almost zero, and that there is almost no leakage flux in the space near the stator and the rotor. The torque (T1) has a characteristic that an effective torque can be generated during 40 ° and the torque is zero during 20 °.

  67 (S2) is a result of skewing the stator magnetic pole (S1) by 20 °, and the torque (T2) at that time has a trapezoidal torque characteristic as shown in the figure, and the portion where the torque is zero is only a specific point. It is said. However, as shown in the figure, the torque reduction portion is in a wide range, for example, in the range of 40 ° to 80 °. In (S2) and (R2), the torque characteristic (T2) is the same even if the stator magnetic pole (S2) is rectangular and the rotor magnetic pole (R2) is skewed by 20 °.

  FIG. 67 (S3) shows that the stator magnetic pole (S1) has a width of about 60 °, and the surface shape of the stator magnetic pole is expanded to the full width in the circumferential direction. The rotor magnetic pole (R3) has the same shape. In this case, since the stator magnetic poles are adjacent to each other in the circumferential direction, the magnetic path of the stator magnetic poles is the rotor magnetic pole 11E. The magnetic saturation occurs depending on the rotational position of. For this reason, as shown in the figure, the torque characteristic (T3) is such that a large torque is generated in a range where the facing area between the stator magnetic pole and the rotor magnetic pole is small, and the torque gradually decreases due to magnetic saturation as the facing area increases.

  The stator magnetic pole (S4) in FIG. 68 has the shape shown at the right end of FIG. 66 (C), and is a modified shape in which the stator magnetic poles (S1) and (S3) are combined. The stator magnetic poles (S1) and (S3) are combined in half. The stator magnetic poles adjacent to each other in the circumferential direction have a symmetrical shape in the rotor axial direction and are simultaneously shifted in the rotor axial direction. As a result, the rotor pole direction width of the stator magnetic pole and the rotor axis direction width of the rotor are about 1.5 times larger in the diagrams of (S4) and (R4). As a result, the torque characteristic (T4) is considerably improved in both the average torque and the torque reduction portion. And the clearance gap between stator magnetic poles can be enlarged, and the leakage magnetic flux between stator magnetic poles can be reduced significantly compared with said (S3).

  The stator magnetic pole (S5) in FIG. 68 is an example in which the above (S4) is trapezoidal. The rotor magnetic pole (R5) is the same as (R4). These torque characteristics (T5) are slightly improved. Furthermore, by increasing the circumferential width of the stator magnetic poles and separating the relative rotor axial positions of the stator magnetic poles adjacent to each other in the circumferential direction, both an increase in torque and a reduction in leakage magnetic flux between the stator magnetic poles can be achieved. Can do. That is, in the stator magnetic pole (S1), the torque increase and the leakage magnetic flux reduction between the stator magnetic poles, which are in a trade-off relationship, can be made compatible by adopting the shape of the stator magnetic pole (S5).

  The problem here is that the shape of the stator magnetic pole (S5) is larger in the rotor axial direction than the stator magnetic pole (S1). However, the length of the three-phase winding can be shortened as shown by adding the path of the three-phase winding to the stator magnetic pole (S5), and a concave shape in the rotor axial direction of each stator magnetic pole is provided. Thus, the amount of protrusion of the coil end portion in the rotor axial direction can be reduced. In addition, the space where the tip side of the trapezoidal stator magnetic pole can be utilized at the intersection of the three-phase windings is also effective in manufacturing the motor in terms of winding arrangement and processing. As a result, the motors indicated by (S5), (R5), and (T5) are superior to the motors indicated by (S1), (R1), and (T1) in terms of torque characteristics, motor size, cost, and the like. It is also possible to use a different motor.

  In addition, when manufacturing the stator magnetic pole (S5) by the structure which laminated | stacked the electromagnetic steel plate, it is necessary to manufacture the electromagnetic steel plate from which the circumferential direction width | variety of a tooth | gear differs in a lamination direction one sheet at a time. Therefore, it is necessary to devise a mold for producing electromagnetic steel sheets by press punching. In order to realize the laminated structure of electrical steel sheets such as skew with existing technology, in the normal phase mold for electrical steel sheet processing, the rotation direction position of the mold is slightly changed for each press processing using a stepping motor etc. There is a technology that realizes a laminated core that is skewed by rotating each time. Such a rotatable mold is arranged at two stations of the normal phase mold, and a rotatable mold for pressing one side of the slot and a rotatable mold for pressing the opposite side of the slot are provided. By arranging and processing while rotating both molds in opposite directions, sheets of electromagnetic steel sheets with different tooth widths can be manufactured and laminated one by one. In this case, the joining between the electromagnetic steel sheets can adopt a caulking structure called a so-called dowel.

  As another method, the pressing of one side of the slot can be rotated, and the other slot processing is set to a fixed position, and sheets of electromagnetic steel sheets having different tooth widths are manufactured. Then, the sheet | seat of the electromagnetic steel plate to laminate | stack is performed, and the laminated body of an electromagnetic steel plate is made. In this case, the above caulking structure cannot be adopted. However, also in this case, the above-described caulking structure can be adopted if an alignment mechanism in some rotational direction is provided somewhere in the normal phase mold so that the caulking structure can be adopted.

  The motor shown in FIG. 68 is effective with a relatively thin motor. In addition to the circumferential direction and radial direction of the magnetic flux, a rotor axial direction component is also generated, so care must be taken to increase iron loss due to fluctuations in the magnetic flux rotor axial direction component, and countermeasures should be taken as necessary. There is a need. For example, a magnetic path in the axial direction of the rotor may be provided in a part of the back yoke. A material capable of increasing or decreasing the magnetic flux in the three-dimensional direction, such as a dust core, can also be used.

  Further, FIG. 68 shows the case where the circumferential width of the rotor magnetic pole is 60 °. However, in order to allow the torques of the respective phases to be overlapped, it is of course possible to have a size of 60 ° or more. is there.

  Next, one of the countermeasures concerning the leakage magnetic flux of the motor of the present invention will be described with reference to FIG. There are many uses for motors that require a relatively large torque at a high rotational speed. The limit of the peak torque of the motor of the present invention is often limited in that a part of a magnetic path through which a magnetic flux contributing to torque generation passes is magnetically saturated. This magnetic saturation is often caused by an increase in the total magnetic flux φs due to a leakage magnetic flux in each part due to a large current in the winding. Therefore, in order to obtain a large peak torque, it is effective to secure a sufficient magnetic path cross-sectional area of each part of the motor and at the same time reduce the leakage flux of each part of the motor. Further, when a large torque is required at a high rotational speed, the power factor can be improved by reducing the leakage magnetic flux, and the motor efficiency is improved.

  69A is an enlarged cross-sectional view of a part of the motor cross-section of FIG. 9, and the winding 114 of FIG. 9 is the winding C92 of FIG. C91 is a stator core. C93 is a leakage magnetic flux generated at the opening of the slot when a large current is applied to the winding C92 from the back side to the front side. The leakage magnetic flux may cause the teeth that are the stator magnetic poles to be magnetically saturated, making it difficult to increase the torque.

  As a countermeasure against this, as shown in FIG. 69 (b), if a rectangular conductor C94 is used, an eddy current is induced in the rectangular conductor C94 due to an increase or decrease of the leakage flux C95, and the leakage flux C95 is reduced. appear. When the motor generates a static torque or rotates at a low speed, there is almost no effect, but when a large current is passed through the winding at a high speed rotation, the frequency of the leakage magnetic flux C95 increases, and the current value increases. The effect of reducing the leakage magnetic flux due to the eddy current in the conducting wire C94 appears. As described above, the motor power factor and the efficiency can be improved by using the flat wire C94.

  At this time, the leakage magnetic flux C95 is the largest in the vicinity of the opening of the slot. If it is 4 or more, the effect can be expected.

  Next, another method for reducing leakage magnetic flux will be described. FIG. 70 shows a configuration in which a conductor or a conductor plate constituting a closed circuit indicated by 505, 506, 507, and 508 is arranged in the circumferential direction of the rotor magnetic pole. With these conductors, the leakage magnetic flux component in the direction passing through the conductor is reduced by the current induced in the conductor. In the motor of the present invention, since the magnetic flux on the rotor side alternates with the rotation of the rotor, there is an effect of reducing the leakage magnetic flux. It can be applied to various motors of the present invention other than those shown in FIG. In addition, since the magnetic flux leakage from the rotor is also generated from the rotor axial end of the rotor magnetic pole, the magnetic flux leakage from the rotor axial direction can be reduced by changing the electromagnetic steel plate at the rotor axial end of the rotor to a conductor plate. it can.

  As shown in FIG. 71, the conductor may be provided with slits 509, 50B, 50D, and 50F in the rotor 501, and the conductors 50A, 50C, 50E, and 50G may be embedded in the slits. The conductor can be easily fixed and can be applied to high-speed rotation.

  Next, a method for improving the maximum torque of the motor will be described. The torque characteristics of the motor of the present invention shown in FIG. 54 are, for example, characteristics as shown in FIG. In the region from 0 to A1 where the current I is small, the torque characteristic shown by the equation (10) is shown, and a characteristic like a square function of the current is shown. In the current transition region from A1 to A2 in FIG. 5 where the magnetic flux density of a part of the motor approaches the saturation magnetic flux density, the torque increases from T1 to T2 as the current increases, as shown by equation (15). The torque characteristics that are linear functions of the current are shown. Further, when the current value is larger than A2, a portion where the motor magnetic circuit part is magnetically saturated is generated at a place other than the air gap portion between the stator magnetic pole and the rotor magnetic pole in the motor. The torque increase characteristic decreases, and torque saturation characteristics such as torques T2 to T3 at currents A2 to A3 in FIG. 5 are shown.

  The improvement of the maximum torque shown here is to improve the torque of T3 to T4 as shown by a one-dot chain line in the torque characteristics shown in FIG. Technically, the magnetic saturation inside the motor is reduced, and the air gap between the stator magnetic pole and the rotor magnetic pole is improved so that a larger magnetic energy can be given.

  First, the motor example in FIG. 54 will be described as to which part of the motor is likely to cause magnetic saturation. FIG. 54 shows a state where the main salient pole 161 faces the stator teeth 117 and 11A and the stator teeth are magnetically saturated. The magnetic flux passing through the tooth 117 leaks from the tooth 11A through the main salient pole magnetic pole 161 of the rotor and the magnetic flux component passing through the tooth 117 and into the stator magnetic pole 117 from the starter magnetic pole 11C through the vicinity of the opening of the slot due to the current of the winding 111. There are a magnetic flux component and a leakage magnetic flux component that enters the stator magnetic pole 117 from the stator magnetic pole 118 through the vicinity of the opening of the slot due to the current of the winding 112. Since a leakage magnetic flux component that does not contribute to torque generation is added from both adjacent stator magnetic poles, more magnetic flux passes through the teeth 117 than magnetic flux that passes through the main salient pole magnetic pole 161 of the rotor, and the stator magnetic pole 117 is more likely to be magnetically saturated. .

  On the other hand, when the auxiliary salient pole magnetic pole 162 generates torque, the circumferential width thereof is small, so that magnetic saturation is easily caused by the leakage magnetic flux from the surrounding stator magnetic pole. As explained, the magnetic saturation points are the stator teeth and the auxiliary salient poles 162 of the rotor. Hereinafter, six methods for improving the maximum torque of the motor will be described.

  FIG. 72 shows a first method for improving the maximum torque. FIG. 72A is an example of a cross section of the motor of FIG. 262 is a tooth portion of the stator core, 263 is a coil end of the winding, and 261 is a rotor core. FIG. 72B shows a case where a soft magnetic body 264 is added in the rotor axial direction in order to reduce the magnetic saturation of the teeth of the stator core. The tooth magnetic path becomes thicker and magnetic saturation is reduced. At this time, the flow of magnetic flux passes not only in the radial direction and the circumferential direction but also in the rotor axial direction. There is a problem that increases. Therefore, in the case of the configuration of FIG. 72 (b), it is necessary to devise such as providing slits or making cuts in the teeth of the electromagnetic steel sheet. Alternatively, an eddy current reduction measure is effective, such as using a powder magnetic core compressed by applying an electrical insulating film to soft magnetic powder.

  FIG. 73 shows a second method for improving the maximum torque. In FIG. 73 (a), a soft magnetic body 271 and a permanent magnet shown in the broken line are added. This broken line portion is enlarged and shown in FIG. 272 is a permanent magnet, and 271 is a soft magnetic material. One of the motor control methods shown in FIG. 73 (b) is to control the current in each slot with a direct current and direct current control. In that case, the direction of the magnetic flux passing through each tooth is one direction, which is characteristic. Then, the magnetic flux 274 is obtained by configuring the permanent magnet 272 in the direction opposite to the direction of the magnetic flux 275 passing through the teeth 262.

  FIG. 74 shows the magnetic characteristics of the teeth 262 at this time. The horizontal axis is the magnetic field strength H [A / m], and the vertical axis is the magnetic flux density B [T]. First, when a large current is applied to the motor without the soft magnetic body 273 and the permanent magnet 272, the magnetic flux density of the teeth 262 changes from zero to the Ba point, and the magnetic flux density becomes B1.

  Next, in the state where the soft magnetic body 273 and the permanent magnet 272 are added, the magnetic flux in the reverse direction passes through the teeth 262, so that it is the Bb point or the Bc point. When a large current is applied in this state, the magnetic flux density of the teeth 262 changes from zero to the Ba point, and the magnetic flux density is almost the same value as B1. However, the change in magnetic flux density is B3 or B4 shown in FIG. 74, which is significantly increased from B1. For example, if B1 and B2 are substantially the same value, double the magnetic flux can be passed through the tooth 262. Expressed as an increase in torque, this corresponds to increasing the torque at T3 to T4 in FIG. In the case of the configuration shown in FIG. 73 (b), it can be said that the time change rate of the rotor axial component of the magnetic flux passing through the teeth 262 is small in principle, but cannot be ignored. Eddy current reduction measures are also effective.

  FIG. 75 shows a third method for improving the maximum torque. In FIG. 75A, a soft magnetic body 293 indicated by a broken line portion and an excitation winding are added in place of the soft magnetic body 273 and the permanent magnet 272 shown in FIG. 73B. This broken line portion is enlarged and shown in FIG. Reference numeral 292 denotes a soft magnetic material, and reference numeral 291 denotes an excitation winding. The electromagnetic action is the same as that of the soft magnetic body 273 and the permanent magnet 272 of FIG. 73, and a magnetic flux 294 in the direction opposite to the magnetic flux 275 passing through the rotor can be created. And the maximum torque of a motor can be improved. One feature of the method of FIG. 75 is that the magnetic operating point can be easily changed by changing the magnitude of the current. For example, in the case of a light load, the current can be made zero, unnecessary control is not performed, and a current necessary for the winding 291 can be supplied only when a large torque is required. 73, the magnitude of the magnetic flux 274 can be controlled by inserting and removing the permanent magnet 272 with another actuator, and the direction of the magnet can be reversed. 73 and 75 can be variously modified, such as a modification in which the positions of the soft magnetic material, the permanent magnet, and the winding are combined with teeth adjacent in the circumferential direction. In addition, regarding the inverter that drives the motor, various inverters can be used, including not only an inverter that allows only one-way current to flow to each winding, but also an inverter that allows a free current to flow. In FIGS. 73 and 75, the magnetic fluxes 274 and 294 only pass through a soft magnetic material having a large relative permeability. The induced magnet 272 thickness may be small, and the exciting current of the winding 291 is also a relatively small value.

  FIG. 76 shows a fourth method for improving the maximum torque. In this method, permanent magnets 301, 302, 303, 304, 305, and 306 are arranged in the openings of the slots. As will be noted, the magnet directions N and S are opposite to the magnetomotive force due to the current in each slot to cancel and reduce the leakage magnetic flux in the vicinity of the slot opening. As a result, similarly to the effect of the structure shown in FIG. 73, the magnetic flux density of each tooth can be reduced, and the maximum torque of the motor can be increased. In particular, this method can be applied even when the length Wt of the motor in the rotor axial direction is increased. Therefore, the method of FIG. 76 is a technique that can be easily adopted even by a motor with a large output. On the other hand, the methods of FIGS. 72, 73, and 75 tend to gradually decrease the effect ratio as the rotor axial length Wt of the motor increases. is there. Note that the six permanent magnets 301, 302, 303, 304, 305, and 306 in FIG. 76 can be reduced to three for cost reasons. Further, even if the permanent magnets shown in FIG. 76 are arranged only in a part in the rotor axial direction, there is an equivalent effect. Further, since similar effects can be obtained if they can be arranged between the stator magnetic poles, they can also be arranged at the rotor axial end of the stator magnetic poles.

  As can be seen from the slot shape and the arrangement of the permanent magnets in FIG. 76, these permanent magnets can also be used as members for fixing the coils in the slots. Further, it is possible to change the shape such as the vertical and horizontal dimensions of the permanent magnet. Note that changes in characteristics due to the temperature rise of the permanent magnets are required. For example, a member having a large thermal resistance or a liquid cooling pipe for cooling can be disposed between the permanent magnet and the winding in each slot.

  FIG. 77 shows a fifth method for improving the maximum torque. 77A shows a shape in which a soft magnetic body 311 for reducing magnetic saturation is added to the cross-sectional view taken along the line AD-AD in FIG. Reference numeral 313 denotes a rotor core, and both ends of the rotor core 313 are auxiliary salient poles 162. FIG. 77 (b) is a side view of FIG. 77 (a). Like the shape of the auxiliary salient pole magnetic pole 162 in FIG. 77 (b), the auxiliary salient pole magnetic pole 162 has a trapezoidal shape from the tip to the rotor center, and the magnetic path is thickened. By thickening the base of the auxiliary salient pole 162, magnetic saturation is reduced and the maximum torque of the motor is improved.

  FIG. 77 shows a sixth method for improving the maximum torque. A soft magnetic material 311 for reducing magnetic saturation is added to the side surface of the auxiliary salient pole magnetic pole 162. The magnetic path of the auxiliary salient pole magnetic pole 162 becomes thick, magnetic saturation is reduced, and the maximum torque of the motor can be improved. The soft magnetic body 311 is disposed in the vicinity of the auxiliary salient pole magnetic pole 162. However, when rotating at high speed, a large centrifugal force is applied and the strength is required. As shown, a soft magnetic material connected in a ring shape in the circumferential direction may be used. It should be noted that the soft magnetic material in the vicinity of the auxiliary salient pole magnetic pole 162 is effective for measures for reducing eddy currents such as the breaks and the dust core. When a soft magnetic material is added, there is a degree of freedom in the direction of the magnetic flux, and it can be driven by a current control method in which the magnetic flux in the magnetic path portion alternates. Although various methods for improving the maximum torque of the motor have been shown, it is possible to reduce the size of the motor and reduce the cost by increasing the maximum torque.

  Next, the technique regarding the air gap between the stator and rotor of the motor of the present invention will be described. In particular, the performance of a reluctance motor that does not use a permanent magnet varies greatly depending on the air gap. In particular, the smaller the motor, the greater the proportion of the excitation load in the air gap portion, and the size of the air gap is important. 78A is a partially enlarged view of the motor, in which D51 is a stator magnetic pole and D52 is a rotor magnetic pole. The horizontal axis of the paper surface of FIG. 78 is the rotor axial direction, the vertical direction is the radial direction of the motor, and the front side to the back side of the paper surface is the circumferential direction. Both show a model of a configuration in which eight electromagnetic steel sheets are laminated. D53 is an air gap. In this example, the thickness of the electrical steel sheet is 0.35 mm and the air gap is 0.5 mm.

  78 (b) is a partially enlarged view of the stator magnetic pole D54, and the shape facing the air gap portion D56 is such that each electromagnetic steel sheet has a thickness in the rotor axial direction of the stator magnetic pole D51 of FIG. 78 (a). The shape is approximately doubled. Similarly, the rotor magnetic pole D55 has a shape expanded about twice as much as the rotor magnetic pole D52. D57, D58, D59, and D5A are soft magnetic materials for avoiding magnetic saturation of each magnetic path. 78 (b), the magnetic resistance of the air gap portion between the stator magnetic pole D54 and the rotor magnetic pole D55 can be reduced. Although the shape is orderly in FIG. 78, the magnetic resistance of the air gap portion can be reduced to a value close to the minimum ½ by further devising the shape. This is equivalent to the fact that the size of the air gap D53 can be reduced to nearly ½, and improvement in torque, improvement in power factor, and improvement in efficiency can be expected by devising each magnetic pole shape. At this time, the volume of FIG. 78 (b) is larger than that of FIG. 78 (a), but the increase in weight is slight and the increase in material cost is also slight.

  Next, a method for increasing the torque of the motor of the present invention will be described. FIG. 79 shows an example in which the stator is the same as that shown in FIG. 1 and the rotor magnetic poles are wound. Windings D61 and D62 for generating magnetomotive force of the rotor magnetic pole of D69 are wound, and windings D65 and D66 for generating magnetomotive force of the rotor magnetic pole of D6A are wound. Windings D63 and D64 for generating magnetomotive force of the rotor magnetic pole of D6B are wound, and windings D67 and D68 for generating magnetomotive force of the rotor magnetic pole of D6C are wound. The current Ir energized to these windings is supplied from the brushes D6F and D6G to the slip rings D6D and D6E attached to the rotor shaft D6H. In the case of this motor, the magnetic flux direction of the rotor magnetic poles is reversed six times for each rotation, so that it is necessary to give a current that changes in accordance with the magnetic flux direction.

  With such a motor configuration, since a magnetomotive force can be generated on the stator side and the rotor side, a large peak torque can be generated. Considering the case where the application is an electric vehicle, a large motor torque is required when going up a steep slope. However, there are many cases where such high torque operation is rare, and in normal operation, the life of the slip ring and the brush can be secured by floating the brushes D6F and D6G from the slip ring. As a result, it is a method that copes with the relatively low cost by adding a winding, a slip ring, and a brush to the rotor in addition to a high torque application that is used very rarely.

  Next, a method for rotating the motor of the present invention over a wide range by switching windings and a method for expanding the constant output range will be described. FIG. 80 shows an example of winding switching means. D71, D72, and D73 are three-phase windings such as the motor shown in FIG. D74, D75, and D76 are one end terminals of the respective windings. Each of the windings D71, D72, and D73 has a structure in which windings can be taken out from three locations as shown in the figure, and has a configuration in which a winding tap can be selected by the winding selection means D7A. D77, D78, and D79 are terminals of each winding connected to each selected winding tap. By adopting such a configuration, the number of turns of each winding can be selected from three ways, so that torque can be output in a wider rotation speed range. Since the winding voltage is proportional to the number of winding turns, all winding turns are used for low-speed rotation, and taps with a small number of turns are used for high-speed rotation. When constant output control is required, the constant output range can be expanded by this winding selection. In addition, an electromagnetic contactor etc. can be used for the switching of the said specific winding tap. If the current can be reliably interrupted when switching the windings, the contact can be made relatively small.

  Further, as a method of changing the voltage of the motor winding, the length of the air gap between the stator rotors can be changed. Specifically, for example, the rotor axial length Lr is twice as long as the stator axial length Ls, and the inner diameter of the stator is tapered by 10 mm at one end and the other end. The diameter of one end of the rotor is smaller than the inner diameter of the stator by twice the air gap length, and the taper of the rotor is tapered at the same angle. And it is set as the structure which can move the rotor axial direction position of the stator with respect to a rotor relatively to the axial direction length Ls of a stator. With such a configuration, when the position of the stator in the rotor axial direction relative to the rotor is moved by Ls, the amount of the air gap can be increased by 10 mm in diameter. The air gap length on one side can be varied by 5 mm. Thus, by varying the air gap length, the magnetic flux between the stator and the rotor can be varied, the induced voltage of the motor can be varied, and the variable range of the motor rotation speed can be expanded.

  In addition, the material of each part of the stator can be properly used depending on the characteristics, and the winding diameter can be devised. An example is shown in FIG. Reference numeral 351 denotes a back yoke portion of the stator. Now, it is assumed that a large current is passed through the winding 354 from the front side to the back side of the paper, and at the same time, a current of the same magnitude is passed through the winding 355 from the back side of the paper to the front side. The magnetic flux indicated by 356 is excited by the both currents to the teeth 352, and the magnetic fluxes of 357 and 358 pass through the teeth 352 from the adjacent teeth through the vicinity of the slot opening. In this manner, the stator teeth 352 acting to attract the rotor magnetic poles tend to gather magnetic flux and become magnetically saturated. In order to reduce the leakage magnetic fluxes 357 and 358, it is better that the width WSB of the opening of the slot is large, and the slot shape is as shown in FIG. At this time, the tip of the tooth becomes thinner and becomes more and more likely to be magnetically saturated. Therefore, it is convenient to use a permendule known as a high magnetic flux density material as the material of the tip 359 of the tooth. The maximum magnetic flux density of permendur composed of about 50% of iron and cobalt is about 2.5T, which is convenient but expensive. If only the tip of the tooth is used, there is an application that can be used even if it is somewhat expensive. Further, the part structure of the permendur can be shaped like 35A to reduce the magnetic saturation of the entire tooth.

  As for the thickness of the winding in the slot, the leakage flux 357 and 358 can be reduced by making the winding on the opening side of the slot relatively thin and increasing the magnetic flux density near the opening of the slot. You can also.

  In the motor of the present invention, it is also possible to change the shape of the stator and the rotor according to the operating conditions. For example, specifically, it has the shape shown in FIG. 54 for low-speed rotation, and can generate a substantially constant torque all around, and the auxiliary salient pole magnetic pole 162 is moved toward the inner diameter side at a certain number of rotations or more. The motor can be equivalent to this motor. Although the motor of FIG. 9 generates torque ripple, it has a simple configuration and is relatively simple to control. As a method of moving the auxiliary salient pole magnetic pole 162, for example, a lever on the rotor shaft can be moved in the rotor shaft direction within a certain range from the stator side. Then, the auxiliary salient pole magnetic pole 162 may be connected to the lever, and the auxiliary salient pole magnetic pole 162 may be configured to move in and out in the radial direction by moving the lever.

  By using such a motor, a motor with a small torque ripple can be realized at low speed, and a motor with low iron loss, power transistor switching loss, etc. can be realized at high speed rotation. At this time, it is also possible to change the magnetic pole width of the main salient pole salient pole magnetic pole 161 in conjunction with it. These mechanisms such as movement or deformation can be performed by various other methods.

  Further, the above-mentioned various measures for reducing magnetic saturation can be used in combination. Naturally, each effect can be obtained.

  Next, explanations relating to iron loss of the motor will be given. Since the motor of the present invention performs unidirectional current control, the magnetic flux of each part of the stator is also unidirectional magnetic flux. Therefore, the hysteresis loss of the special stator is small, and a normally used electrical steel sheet can be used. However, in the case of the motor shown in FIG. 1, the direction of the magnetic flux of the rotor changes six times in one rotation. In order to reduce the iron loss of the motor of the present invention, it is effective to reduce the iron loss by making the rotor material amorphous metal. Soft magnetic materials with low iron loss include magnetic steel sheets with high silicon-Si content, directional silicon steel sheets, and powder magnetic cores that have been pressed at high pressure by applying a film with high electrical resistance on the surface of iron powder. It can also be used.

  Next, the problem of winding will be verified. After that, a method for solving the winding problem will be described. 116, 117, and 118 will be described as conventional examples. The problem is that the arrangement of each winding is complicated and difficult to manufacture, the problem that the space factor of the winding is reduced, and the coil end portion due to the intersection of windings. The problem of the lengthening in the axial direction was shown. FIG. 82 shows an example of a stator of the motor of the present invention in which the two-pole motor shown in FIG. 54 is made into four poles, and is a diagram modeling the shape when viewed from the rotor axial direction. 116 and 118 is a configuration in which the winding is wound by the prior art. The A-phase windings are a winding H81 wound from the slot 371 to the slot 374 and a winding H84 wound from the slot 377 to the slot 37A. The B-phase windings are a winding H82 wound from the slot 373 to the slot 376 and a winding H86 wound from the slot 379 to the slot 37C. The C-phase winding is a winding H83 wound from the slot 375 to the slot 378 and a winding H86 wound from the slot 37B to the slot 372.

  83 is a longitudinal sectional view of a section AG-AG of the motor of FIG. Reference numeral 381 denotes a stator core in which electromagnetic steel plates are laminated in the rotor axial direction. 384 is a winding in the slot, 385 and 389 are coil ends, and LCE1 is a coil end length which is an amount of protrusion from the stator core 381. Reference numerals 386 and 387 denote insulating papers for maintaining insulation between the stator core 381 and the windings in the vicinity of the slots. The insulating paper 387 normally protrudes about 10 mm from the end of the stator core 381 in order to ensure insulation. Reference numeral 382 denotes a rotor, and reference numeral 383 denotes an air gap portion between the stator core 381 and the rotor 382. The alternate long and short dash line is the rotation center of the rotor. With such a configuration, a 2-pole motor and a 6-pole motor can be wound in the same manner, and the windings at each part of the coil end are almost uniformly distributed over the entire circumference. However, a plurality of windings overlap the coil end portion, resulting in a complicated arrangement, and there is a problem of productivity. There is also a problem that the winding space factor in the slot is lowered due to difficulty in winding. The length of the coil end in the rotor axial direction can be shortened to some extent by molding with a mold or the like because the windings of the coil end portion are dispersed. However, there is a limit to molding, and there is a problem that the coil end is long.

  Next, a winding winding method and a winding arrangement method for reducing these problems will be described with reference to FIG. FIG. 84 is a diagram showing a model of the slot position of the winding and the winding arrangement of the coil end portion as seen from the rotor axial direction of the stator example of the motor of the present invention having 4 poles and 12 slots. A positive A-phase winding is wound around the slot 371 and a negative A-phase winding is wound around the slot 374 with the winding 37D. Then, a positive A-phase winding is wound around the slot 377 and a negative A-phase winding is wound around the slot 37A with the winding 37G. Similarly, a positive B-phase winding is wound around the slot 373 and a negative B-phase winding is wound around the slot 376 with the winding 37E. Then, a positive B-phase winding is wound around the slot 379 and a negative B-phase winding is wound around the slot 37C with the winding 37G. Similarly, a positive C-phase winding is wound around the slot 375 and a negative C-phase winding is wound around the slot 378 with the winding 37F. Then, a positive C-phase winding is wound around the slot 37B and a negative C-phase winding is wound around the slot 372 with the winding 37J.

  84, the windings 37D, 37F, and 37H arranged on the outer diameter side are wound before the windings 37E, 37G, and 37J. The reason is, for example, that the winding 37E covers the end surfaces of the slots 374 and 375 in the rotor axial direction, and thus the windings 37D and 37F need to be wound around the winding 37E. Therefore, for example, if the windings 37D and 37F are wound, the winding 37E can be wound before the winding 37H.

  As shown in FIG. 84, the number of poles of a motor that can be wound by sorting concentrated windings on the outer diameter side and the inner diameter side is an integer multiple of four. In this case, since the number of windings is small and there is little physical interference with each other, the windings can be wound efficiently and with high space.

  FIG. 85 shows an example of a stator of a 6-pole motor in which the number of poles of the motor is not an integer multiple of 4. The A-phase windings are the winding 73K wound from the slots 731 to 734, the winding 73N wound from the slots 737 to 73A, and the winding 73R wound from the slots 73D to 73G. The B-phase windings are a winding 73L wound from slots 733 to 736, a winding 73P wound from slots 739 to 73C, and a winding 73S wound from slots 73F to 73J. The C-phase windings are a winding 73M wound from slots 735 to 738, a winding 73Q wound from slots 73B to 73E, and a winding 73T wound from slots 73H to 732. Among these windings, the winding 73T has an irregular shape and arrangement compared to other windings. Thus, if it is made partially irregular, similar windings can be obtained with 6 poles, 10 poles, 14 poles, 18 poles, and the like.

  Next, the winding shape and the core shape in the vicinity of the coil end portion of the motor shown in FIG. 84 will be described with reference to FIGS. 86, 87, 88, and 89. FIG. 86 is a cross-sectional view of the cross-section AF-AF of FIG. 84 on the slot 375 side. For comparison, the coil end 385 of FIG. 83 is indicated by a broken line. 391 is a stator core, 396, 397, and 398 are insulating papers, 394 is a winding wound around the slot 37B, and 395 is a coil end portion thereof. From the middle of the slot toward the rotor axial end surface of the stator core, it is gradually bent toward the outer diameter side and connected to the coil end 395. Reference numeral 399 denotes a winding 37E, which is located in the circumferential direction. As a result, it can be seen that the length LCE2 of the coil end in FIG. 86 is significantly shortened compared to the length LCE1 in the rotor axial direction of the conventional coil end.

  FIG. 87 is a sectional view of the slot 376 side of the section AE-AE in FIG. 40B is a stator core, 403, 404 and 405 are insulating papers, 406 is a winding wound around a slot 376, and 402 is a coil end portion thereof. The shape of the windings 406 and 402 is shown in FIG. 88 in which an arc shape is linearly expanded when viewed from the stator side. The horizontal axis in FIG. 88 is the circumferential direction, and the vertical direction is the rotor axial direction. Reference numerals 411, 412, 413, 414, and 415 denote inner peripheral side shapes of the respective teeth. Reference numerals 416 and 417 denote the winding 37E of FIG. 84 and the windings 406 and 402 of FIG. When the winding 416 reaches the portion 418 from the middle of the slot, it is bent in the circumferential direction and connected to the coil end portion. The same applies to the bent portion 419. Since the arc is bent without difficulty, the amount of protrusion of the coil end 402 in the rotor axial direction shown in FIG. 87 is shortened to keep the size of LCE2. Thus, by devising the slot shape of the stator core near the coil end, the length of the coil end in the rotor axial direction can be shortened. Note that the winding 401 is the winding 37F of FIG. 84 and is located in the circumferential direction.

  FIG. 89 is a diagram showing the shape of the stator core in the same portion as FIG. Reference numeral 724 denotes a stator core, and the coil end portion 37D in FIG. 84 is the coil end portion 723 in FIG. Reference numeral 721 indicated by a broken line indicates the shape of the slot 371 in FIG. A thick line 722 is the shape of the end of the slot shape 39B in FIG. 86 in the rotor axial direction. Therefore, a curved surface shape is formed from a broken line 721 to a thick line 722 in FIG. And it is set as the shape which winding can wind more easily along the said curved surface shape of a stator core toward the rotor axial direction side surface of a stator core from the slot inside.

  Note that the shape 39B of the stator core where the winding 394 in FIG. 86 is bent is shown by a smooth curve or an arc, but the same effect can be obtained even with a two-step, multi-step shape. I can do it. For example, it has a three-stage shape such as a slot shape 407 indicated by a broken line of the stator core 40B in FIG. The stator core is often composed of laminated electromagnetic steel sheets, and the smooth shape change of the slot increases the cost burden of a die for punching the electromagnetic steel sheets. If it is a shape like the slot shape 407, since it can manufacture by laminating | stacking three types of electromagnetic steel plates, it is possible to reduce the cost of production equipment.

  Further, in FIG. 87, since the circumferential magnetic path of the back yoke portion in the vicinity indicated by the broken line 39A is reduced, an annular soft magnetic body 408 is added to increase the circumferential magnetic path cross-sectional area. is doing. The soft magnetic body 408 can secure the circumferential magnetic path of the back yoke portion without increasing the length in the rotor axial direction of the motor. Further, as shown by the broken line, it is preferable that the soft magnetic body 408 be formed by bending electromagnetic steel plates laminated in the rotor axial direction so that the magnetic flux can easily pass through the portion 408.

  Next, improvement of the insulating paper 387 and 386 shown in FIG. 83 will be described. In the case of FIG. 83, the cross-sectional shape of the slot is the same from the one end of the stator core 381 to the other end in the rotor axial direction. Therefore, the terminal treatment of the insulating paper 386 and 387 often protrudes about 10 mm from the end of the stator core 381 in the rotor axial direction. The reason for this is that if the end of the stator core 381 is extended along the end surface of the rotor core along the end surface, the insulation distance between the insulating paper cut and the corner 388 of the stator core is shortened, or If the insulating paper is divided in the vicinity, there is a problem that the effective sectional area of the slot decreases and the winding space factor decreases. In the configuration of FIG. 86, as the slot shape approaches the end in the rotor axial direction, the slot cross-sectional area increases, and the insulating papers 398 and 397 are overlapped so as to secure an insulation distance from the stator core 391 while ensuring both insulation. Paper can be connected. Since the insulating paper 397 can be disposed along the slot shape 39B of the stator core 391, the length LCE2 of the coil end 395 in the rotor axial direction can be shortened.

  Similarly, in the configuration of FIG. 87, the slot cross-sectional area is increased, and both insulating papers are overlapped along the slot shape 407 while being overlapped like the insulating papers 405 and 404 to ensure an insulating distance from the stator core 40B. It can be connected.

  86 and 87, the coil end is fixed and radiated by placing the flange part of the motor in close contact with the coil end so that the coil end is fixed by the flange part and at the same time the heat of the coil end part is Can be effectively done by transmitting to the flange part.

  The winding method shown in FIG. 84 is high in productivity because each winding has little physical interference with other windings and winding is easy, and the space factor of the winding is high. It is also possible to improve. As shown in FIGS. 86 and 87, the arrangement and configuration of the windings can be facilitated by improving the slot shape, and the protrusion amount LCE2 of the coil end portions 395, 399, 401, 402 in the rotor axial direction is shortened. The effective utilization rate in the vicinity of the coil end, which has been a problem with conventional motors, can be greatly improved. As a result, it can be said that the motor has been downsized.

  Next, a configuration obtained by dividing the stator core of the motor of the present invention will be described with reference to FIG. FIG. 90 is a diagram in which each winding is modeled when the stator of the 8-pole motor of the present invention is viewed from the rotor axial direction. Reference numeral 171 denotes an 8-pole rotor, which is the same as the rotor of the motor shown in FIG. It is composed of a stator core joined at four points of division points 42E, 42F, 42G, and 42H. Although the shape of these dividing points is indicated by double lines, it can be modified into various shapes. Since these divided stators are divided into 360 degrees in electrical angle, the windings of each phase can be wound around the same divided core, and the respective divided stators can be manufactured independently. The A-phase winding is wound from the slot 421 to the slot 424. The B-phase winding is wound from slot 423 to slot 426. The C-phase winding is wound from slot 425 to slot 423.

  For example, when winding a winding around one of the four divided stator cores, since the arc shape is 90 °, the opening side of the slot is open, which is significantly easier than winding a winding around an annular stator core. There is a degree of freedom. Therefore, the winding can be directly wound by a winding machine, and the winding space factor in the slot can be improved.

  Further, the same effect can be obtained by dividing the stator core by an integer multiple of an electrical angle such as 360 ° or 720 ° in electrical angle. Also, for example, in the case of a stator core divided at 720 °, the shape of the stator or rotor can be deformed at a cycle of 720 °, and two in-phase rotor magnetic poles are slightly different from 360 in order to reduce torque ripple. It is possible to adopt a method of canceling torque ripple as a different pitch.

  In addition, the mating surfaces of the dividing points 42E, 42F, 42G, and 42H can be combined with each other on the uneven surface every other sheet or every several sheets of magnetic steel sheets so that mutual magnetic resistance is not hindered. Can be reduced. And the part which the magnetic steel sheet of the adjacent division | segmentation core mutually staggered on the unevenness | corrugation can also be fixed firmly with sufficient precision by laser welding etc. In addition, various methods such as pressure welding with bolts, adhesion fixation with an adhesive, gripping with a casing, welding fixation with laser welding, or the like are possible for mechanically fixing each divided core.

  Next, the mechanical support structure of the motor shown in FIGS. 15 and 16 in which the two motors are combined and incorporated will be described. Since there are rotors on the outer diameter side and the inner diameter side, and the stator is disposed between them, a specific configuration to be realized with high accuracy is not simple. As described above, the air gap length between the stator magnetic pole and the rotor magnetic pole needs to be as small as possible in order to ensure motor characteristics.

  FIG. 91 shows an example of a longitudinal sectional view of the composite motor shown in FIGS. E05 is an output shaft of the rotor, E01 is a rotor R1 arranged on the outer diameter side, E02 is a rotor R2 arranged on the inner side, E03 and E04 are a stator S1 and a stator S2 integrated back to back. E0A indicates the coil end of the stator winding. E06 and E07 are bearings that support the rotor. E08 is a fixed portion, which connects the stators S1 and S2, and can fix the entire motor by fixing the portion of E09. The alternate long and short dash line is the rotation center line of the rotor.

  On the other hand, when high-speed rotation is used, it is known that the value of the product D × N of the bearing diameter D and the rotational speed N greatly affects the life and reliability of the bearing. That is, when the rotational speed N of the motor is large, it is preferable that the bearing has a small diameter. From this viewpoint, the diameter of the bearing E0C is reduced. Further, when the diameter of the bearing is increased, there is a problem that the cost increases.

  91 has a structure in which the rotor R1 of E01 and the stators S1 and S2 are supported from one side, and therefore errors such as eccentricity are likely to occur. Therefore, the axial thickness of the stator and the rotor cannot be increased. In the case of a flat motor, the accuracy can be maintained as such.

  The motor shown in FIG. 92 is an example of a motor when the axial length of the stator and the rotor is about twice as long as the motor of FIG. 92. In order to keep the air gap between the stator and the rotor of FIG. 92 small, both ends in the axial direction of the stator and the rotor can be accurately supported on the rotation center line indicated by broken lines. E15 is an output shaft of the rotor, E11 is a rotor R1 disposed on the outer diameter side, E12 is a rotor R2 disposed on the inner side, and E13 and E14 are a stator S1 and a stator S2 integrated back to back. E1A indicates the coil end of the stator winding. E16 and E1C are bearings that support the rotor. The other end of the stator is also supported by the bearing E17 toward the center of the rotor shaft. E18 is a fixed portion, which connects the stators S1 and S2, and can fix the entire motor by fixing the portion E19.

  However, since the inner diameter of the bearing E1C is small, there is a problem that the support rigidity of the entire motor is restricted. Regarding this point, the rigidity of the entire motor may be raised by increasing the inner diameter of the bearing E1C to an allowable level. In the motor of FIG. 92, attention is required because the outer periphery E11 is the rotor R1 and the rotating part is exposed.

  Next, FIG. 93 shows a motor configuration in which a motor case and a bearing are added to the outer periphery of the motor shown in FIG. E1D is a motor case, and E1B is a bearing that supports the rotor E15. E18 is a fixing portion, and the entire motor can be fixed by fixing the portion E19. In the case of the motor shown in FIG. 93, the E21 portion or the front flange portion E22 portion can be fixed. When the part E21 or E22 is fixed, it can be handled in substantially the same manner as the conventional motor support structure shown in FIG. In the case of the motor configuration of FIG. 93, the position of each part of the motor can be accurately maintained with respect to the rotor rotation center line indicated by the alternate long and short dash line, so that the air gap between the stator and the rotor can be kept small. Performance can be ensured. At this time, the diameter of each of the bearings E1C, E16, E17, E1B used can be designed to be an appropriate diameter without being excessively large, so a long-life and highly reliable motor can be designed. Is possible.

  Next, a motor capable of simplifying the configuration of the inverter will be described with reference to FIGS. 94 and 95. FIG. In the motor of FIG. 94, the winding of the motor of FIG. 54 is changed to a double winding so-called bifilar winding. 54 is changed to windings 521 and 522 in FIG. 94, and winding 114 is changed to windings 527 and 528 in FIG. The winding 521 is wound in series with the winding 527, and the winding 522 is wound in series with the winding 528. Since it is desired to make the magnetic fluxes linked to the windings 521 and 522 as common as possible, it is electromagnetically preferable to wind two electric wires in parallel. Similarly, for the other windings, winding 113 in FIG. 54 is windings 525 and 526 in FIG. 94, winding 116 is windings 52B and 52C, winding 115 is windings 529 and 52A, and winding 112 is winding. Lines 523 and 524 are replaced. The winding method of FIG. 94 can be driven by the inverter shown in FIG. 95, so that the inverter is simplified and the cost can be reduced. However, if the bifilar windings have the same thickness, the size of the slot cross-sectional area is limited, so the thickness of each winding is reduced to about half the thickness, and the resistance value is increased. Loss increases and motor efficiency decreases slightly. In addition, since the current of the winding through which the regenerative current flows is relatively small, the thickness of the two windings of the bifilar winding can be made different to minimize the reduction in the motor efficiency.

  The inverter shown in FIG. 95 and each winding shown in FIG. 94 will be described. 53A is a direct current power source, which indicates a direct current power source obtained by converting a commercial alternating current power source of 50 Hz or 60 Hz into direct current, or a direct current power source such as a battery. VM indicates the positive side of the DC power supply 53A, and VL indicates the negative side. The winding 531 in FIG. 95 is a winding constituted by 521 and 527 in FIG. 94, and the winding 532 is a winding constituted by 522 and 528. At this time, the windings 531 and 532 are wound in parallel at a close distance so that most of the magnetic fluxes linked to both windings can be made to be the magnetic fluxes linked in common. Further, the windings 531 and 532 are arranged so that the polarities are reversed, as indicated by the dots of the windings. When the windings 531 and 532 are spatially separated in the slot as shown in FIG. 94, the windings 522 and 528 arranged at the back of the slot are the regenerative winding 532 in FIG. It is preferable that more magnetic flux is interlinked with the winding at the back of the slot. Similarly, the winding 533 is windings 525 and 52B, and the winding 534 is windings 526 and 52C. Winding 535 is windings 529 and 523, and winding 536 is windings 52A and 524.

  The method of driving the motor shown in FIG. 94 by the inverter shown in FIG. 95 can be driven by the same method as the method shown in FIG. 56 when driving to the CCW. However, the method for regenerating the magnetic energy of the motor is different. Specifically, in the state of FIG. 56A, a current is supplied to the winding 531 by the transistor 537, and at the same time, a current is supplied to the winding 535 by the transistor 539. 56 (b), the transistor 539 is turned off, and the magnetic energy of the magnetic flux linked to the winding 535 is regenerated to the power source 53A using the winding 536 and the diode 53D. The windings 535 and 536 have a large mutual inductance, and most of the magnetic fluxes are linked to both windings in common, so that energy can be supplied to each other. Before and after this time, a current is passed through the winding 533 by the transistor 538, and a magnetic flux indicated by an arrow in FIG. 56B is generated to generate a CCW torque. In the state of FIG. 56C, the transistor 537 is turned off, and the magnetic energy of the magnetic flux linked to the winding 531 is regenerated to the power source 53A using the winding 532 and the diode 53B. Before and after this time, a current is passed through the winding 535 by the transistor 539, and a magnetic flux indicated by an arrow in FIG. 56C is generated to generate a CCW torque. When rotating to the state of FIG. 56 (d), the rotational position is a rotational position advanced by 60 ° in electrical angle from the state of FIG. 56 (a) to the CCW direction, and the subsequent rotation in the CCW direction is the same. It will be driven by the algorithm. The inverter shown in FIG. 95 is a simple inverter composed of three transistors and three diodes, and can be reduced in cost. In the inverter shown in FIG. 95, the number of transistors is reduced by half, and the current capacity of the transistors is also reduced by half, so that the total current capacity of the inverter of FIG. 95 can be reduced to about 1/4. At the same time, the forward voltage drop of the current and the voltage drop at the diode during regeneration are halved compared to a normal three-phase AC inverter as shown in FIG. 119, which is more efficient and generates less heat. The inverter can be downsized.

  FIG. 96 shows an example of reducing the instantaneous overvoltage applied to the transistor of FIG. When the current flows through the transistor 537, the magnetic energy of the magnetic flux linked to the winding 531 is regenerated to the power source 53A by the winding 532 and the diode 53B in a simple principle. However, regarding the leakage magnetic flux component of the winding 531 and the like, the winding 532 has a magnetic flux component that cannot be regenerated because the magnetic flux component is not linked. These leakage magnetic flux components have a problem in that when the transistor 537 is turned off, an instantaneous excessive voltage is generated in the winding 531 and the transistor 537 is damaged. The diodes 541, 542, and 543 in FIG. 96 can rectify and collect excessive voltages generated in the windings 531, 533, and 535, and can absorb them to the resistor 546, the capacitor 544, and the Zener diode 545 which are overvoltage absorption circuits. Since the total amount of magnetic energy such as the leakage magnetic flux component is small, it can be solved by a relatively small overvoltage absorption circuit. The overvoltage absorption circuit can be variously modified.

  In FIGS. 94, 95, and 96, the case where there are three windings has been described. However, the control circuit of FIGS. 95 and 96 can be expanded for multi-phase such as four-phase and five-phase. Can be realized similarly.

  Next, in the case where the motor shown in FIGS. 1, 9, 54, etc. is a small capacity motor of several W, an example of the configuration of the inverter is shown in FIG. 97 and will be described. Reference numerals 551, 552, and 553 denote three-phase windings, 554, 555, and 556 denote current-carrying transistors, 557, 558, and 559 denote diodes, and 55A, 55B, and 55C denote resistances that absorb the magnetic energy of the motor. In this way, when the output capacity of the motor is small, the circuit can be simplified by absorbing the magnetic energy of the motor with a simple circuit and resistance. Here, the resistors 55A, 55B, and 55C can be combined with a capacitor, a Zener diode, and the like.

  Next, an example of the configuration of the inverter will be described with reference to FIG. 1, FIG. 9, FIG. Reference numerals 561, 562, and 563 denote three-phase windings, 564, 565, and 566 denote transistors that conduct current, and 567, 568, and 569 denote diodes. The regenerative current collected by the diodes 567, 568, and 569 is charged into the capacitor 56C, and is converted into a voltage by the transistor 56A that is a DC-DC converter, the choke coil Ldcc, and the diode 56B, and is charged into the DC power supply 53A. This DC-DC converter has a common configuration that is often used, and electrification charged in the capacitor 56C is applied to the choke coil Ldcc by the transistor 56A, and in this state, the transistor 56A is turned off. At this time, the magnetic energy accumulated in the choke coil Ldcc can be charged to the DC power supply 53A in the form of a current Irc using the diode 56B. In this way, the kinetic energy of the motor and the magnetic energy of the motor are regenerated and efficiently regenerated to the DC power source 53A. The regenerative voltage VH in FIG. 2 can be set according to the required regenerative characteristics. Further, the DC-DC converter can be modified.

  In an electric vehicle, a hybrid vehicle, etc., two or more motors are often used for driving a vehicle. Usually, the fuel consumption in the city driving mode, that is, the driving efficiency is a problem. Although it depends on the type of automobile, it is often less than or equal to ½ of the maximum motor torque in the urban driving mode. Accordingly, a motor-side power generation capacity during regeneration, that is, a regeneration capacity, that is 1/2 or less of the maximum output capacity during rapid acceleration is sufficient. When an automobile must decelerate rapidly, it can be considered that it is sufficient to use a mechanical brake function together from the viewpoint of safety. From such a viewpoint, the DC-DC converter as shown in FIG. 2 can be shared by a plurality of motors, and a driving circuit for driving one motor has three transistors 564, 565, 566. And three diodes 567, 568 and 569. Therefore, the inverter shown by the broken line in FIG. 2 is a simple inverter that can drive one motor by three transistors and three diodes, and can reduce the cost. At the same time, the forward voltage drop of the current and the voltage drop at the diode during regeneration are about l / 2 compared to a normal three-phase AC inverter as shown in FIG. 119, which is efficient and generates little heat. In this respect, the inverter can be downsized.

  Next, another circuit for driving the motor shown in FIGS. 1, 9, 54, etc. will be described with reference to FIG. Compared to FIG. 2, a transistor 571 and a diode 572 are added, and the negative connection point of the capacitor 573 is a common line VL. By adding the transistor 571, the regenerative voltage of the windings 561, 562, and 563 can be set to a large value of about (VH−VL). Regeneration time can be shortened.

  Next, a method of driving the motor shown in FIGS. 1, 9, 54, etc. with the inverter shown in FIG. 3 will be described. The three-phase winding A winding, B winding, and C winding in FIG. 54 correspond to windings 87D, 87E, and 87F in FIG. For example, when a current is passed through the winding 87D, it is possible to control the transistors 871 and 872 to apply a DC voltage, regenerate, and control the flywheel. However, only one-way current can be passed through the winding 87D. The same applies to the winding currents of other phases. The relationship between the rotational position θr of the motor and each phase current may be energized according to the relationships of FIG. 11, FIG. 12, FIG. 56, FIG. As described above, when the motor of the present invention is driven by the inverter of FIG. 3, the capacity can be reduced to about ½ compared to the total current capacity of the conventional inverter, so that the control device can be reduced in size and cost. Is possible.

  Next, a method for making the motor of the present invention have a high torque and a high-speed rotation drive will be described. First, as a method of realizing high torque, the method shown in FIG. 73 (b) and the method shown in FIG. 76 show a method of passing more magnetic flux from the rotor side to the soft magnetic part of the stator magnetic pole. It was. As a result, it was shown that the maximum torque can be increased from the operating point of torque T2 in FIG. 5 to the operating point of torque T4. However, since the magnetic flux increases at this time, the magnetic flux and voltage linked to each winding also increase. Therefore, these methods are rather disadvantageous from the viewpoint of driving the motor speed to high speed.

  As a countermeasure, a method for reducing the time change rate of the magnetic flux linked to each winding, reducing the induced voltage of the winding, and facilitating the driving up to high speed rotation will be described. In FIG. 99, the direction of the rotor magnetic flux 681 is reversed as compared with FIG. 73 (b). Since the rotor magnetic flux 681 is excited by the current of each winding, the current direction of each winding of the motor is reversed and the direction of the magnetic flux 681 is reversed. The magnetic flux 274 and the magnetic flux 681 excited by the permanent magnet 272 are oriented in the same direction at the tooth 262, and the tooth is likely to be magnetically saturated. In the above description of increasing the torque, the method of obtaining a large magnetic flux change B4 with the magnetic flux density difference of B4 using the magnet 272 as the initial operating point Bc shown in FIG. If the current direction is reversed, if the initial operating point is Bc and the operating point when the reverse current is applied to each winding is Bb, the difference in magnetic flux density is B5, which is a small value. Can do. Since the induced voltage of the winding is proportional to the change in magnetic flux, high-speed rotation becomes easier.

  FIG. 100 shows another method for reducing the change in the magnetic flux passing through the stator magnetic pole. In this configuration, the current direction of the winding is reversed compared to FIG. Also in this case, in FIG. 74, if the magnetic flux density operating point of the stator magnetic pole is Bc and the operating point when the current is applied is Bb, the change in the magnetic flux density of the stator magnetic pole is B5, and the induced voltage of the winding is Therefore, high-speed rotation becomes easier.

  When acting as shown in FIG. 73 (b) or FIG. 76, a large torque can be generated due to the large magnetic flux density difference B4, while the three-phase current directions are all switched in the opposite direction and the magnetic flux density High speed rotation can be realized by reducing the induced voltage with the difference as B5. Examples of control devices capable of such control are shown in FIGS. Both control devices can energize both positive and negative current directions. The motor is a case of a three-phase motor shown in FIG. 1, FIG. 10, FIG. In the case of a motor with four or more phases, the circuit may be expanded according to the number of phases.

  In the control device shown in FIG. 101, each winding has a star connection, and is connected from the center K of the star connection to an intermediate point between the transistors E5D and E5F. Therefore, a positive current and a negative current can be supplied to the A-phase winding E5J by the transistors E51 and E52. The same applies to the B-phase winding E5K and the C-phase winding E5L. In such a control device, when a large torque is required at medium speed or less, a current as shown in FIG. 6 is applied to each phase winding, and the configuration in the vicinity of the stator magnetic pole is shown in FIG. 73 (b) or FIG. Thus, it is possible to output a large torque. On the other hand, when high-speed rotation is performed with the same motor, the directions of the A-phase current Ia, B-phase current Ib, and C-phase current are reversed, and the current is applied as shown in FIG. 99 or FIG. The amount of change can be reduced, the induced voltage of each winding can be reduced, and it can be easily driven to high speed rotation.

  In the control device of FIG. 102, each of the windings 561, 562, and 563 has a so-called H-bridge configuration with four transistors, so that positive and negative currents can be freely supplied to each winding. Similarly, in this case, the current direction is reversed between the energization at the time of high torque and the energization at the time of high speed rotation, so that the operation in both operation modes is possible. In both the above-described operation modes, the motor current control at that time is unidirectional current control. Since the motor is a reluctance motor in the case of a motor configuration that does not use a permanent magnet as shown in FIGS. 1, 10, and 54, the motor characteristics are the same in any of the above operating modes.

  Further, in the case of a control device that can freely energize currents in both directions as shown in FIGS. 101 and 102, the motor torque can be further increased. Since torque can be generated by supplying current to more windings simultaneously, the continuous output torque of the motor can be improved. In addition, since the power can be supplied through a plurality of paths at the same time as the control device, the number of elements of the power transistor increases, but the total current capacity can be reduced.

  For example, in FIG. 4A, torque is increased if a B-phase current in the opposite direction is supplied to the B-phase windings A0F and A0J. Since the rotor rotational position θr in FIGS. 4A and 4D is just at the boundary position, the torque does not increase even when current is supplied to the remaining one-phase winding. At the rotor rotational position θr in FIG. 4C, the torque increases if a reverse C-phase current is applied to the C-phase windings A0H and A0E. However, there is a problem that the cost of the control device increases.

  Further, in a motor having a configuration in which a permanent magnet is added as shown in FIGS. 22 to 34, the average torque is particularly effective. For example, in the case of the motor shown in FIG. 23, when CCW torque is generated at the rotor rotational position θr, the A phase current Ia is supplied to the A phase windings A0D and A0G, and the C phase windings A0H and A0E are supplied with the C phase. The current Ic is energized to generate torque for this motor. Further, the torque can be increased by passing the current -Ic in the direction opposite to the current direction Ic indicated by the current symbol to the remaining B-phase windings A0F and A0J. In this state, current is passed through the three windings to generate CCW torque, so that the joule loss of the motor can be reduced and the continuous output torque of the motor can be increased overall.

  In addition, in the control device shown in FIGS. 101 and 102, when a current in any direction is applied to three windings, the number of power transistors increases, but power can be supplied through a plurality of paths simultaneously. As a result, the number of power transistor elements increases, but the total current capacity can be reduced. Therefore, there is a possibility of cost reduction and size reduction.

  Further, it is possible to further increase the torque of the motor shown in FIG. 73 in which the torque is increased or the motor shown in FIG. Since a magnetomotive force that generates torque is generated in all windings, the joule loss of the motor can be reduced, and the continuous output torque of the motor can be increased. These high torques are particularly suitable for applications where the installation space of the motor is limited.

  In the motor configuration shown in FIG. 99 or 100, no magnetic flux is induced in the rotor when no motor current flows, and no iron loss occurs even when the rotor is in a rotating state. That is, so-called drag torque does not occur and no loss occurs. This is an important characteristic when operating at light loads. In a hybrid vehicle, when high-speed running is performed with an engine of an internal combustion engine, it is an important characteristic that the motor is also rotated in a state of zero current and that the motor loss is zero at this time.

  Next, applied technology in the type of the present invention to which a permanent magnet is added will be described. A motor using a permanent magnet is shown in FIGS. These motors can achieve high torque and high efficiency by using permanent magnets. Since the motor of the present invention can be controlled with a unidirectional current, it is shown that when torque is generated, the permanent magnet can be driven while applying a magnetomotive force in the direction of increasing the permanent magnet. That is, it is possible to drive such that no current is supplied in the direction in which the permanent magnet is demagnetized. On the other hand, in the case of a conventional permanent magnet application motor, a magnetomotive force in the direction of demagnetizing the permanent magnet also acts during acceleration / deceleration, so use a sufficiently thick magnet to prevent demagnetization or an expensive magnet that is difficult to demagnetize There is a cost problem.

  In the present invention, since it can be controlled only by the current in the direction not demagnetized, it is possible to use an inexpensive permanent magnet having a small demagnetization resistance, to make the magnet thinner, and to reduce the cost. However, the motor according to the present invention does not prevent the current from flowing in the direction in which the permanent magnet is demagnetized. Therefore, when the control device malfunctions, there is a problem that the permanent magnet is demagnetized. As another requirement of motor characteristics, there is also a requirement that high-speed rotation control is performed by demagnetizing the magnet so that the magnetic flux density of the magnet can be reduced.

  As a countermeasure, if the control device can be provided with a demagnetizing means and a magnetizing means for the permanent magnet, the above problems can be solved and other requirements can be met. Specifically, the permanent magnet of the motor is a magnet with the minimum necessary thickness, and the demagnetization magnetization is relatively easy. Then, as a demagnetizing means and a magnetizing means for the permanent magnet, a power transistor used for driving the motor is used. Since the motor can reduce the amount of permanent magnets and does not require additional hardware as magnetizing means and demagnetizing means, the motor is low in cost.

  On the other hand, various types of permanent magnets can be used as the permanent magnets. For example, so-called alnico magnets containing aluminum, nickel, and cobalt have a high magnetic flux density and a small coercive force, and are suitable for this purpose. For example, when the permanent magnets F68 and F6B are magnetized in the state shown in FIG. 23, the B-phase currents Ib are supplied to the B-phase windings A0F and A0J and simultaneously the C-phase windings A0H and A0E are magnetized. What is necessary is just to energize the C-phase current Ic. At this time, the magnitudes required for the currents Ib and Ic are determined by the magnet characteristics of the motor. Further, when the permanent magnets F68 and F6B are demagnetized in the state of FIG. 23, the current Ia may be supplied to the A-phase windings A0D and A0G. The magnetizing current can be performed by two windings, and the demagnetization is performed by one current. Therefore, the demagnetizing ability is likely to be insufficient.

  Among some permanent magnets, some permanent magnets are not demagnetized by motor current, and some other permanent magnets are easy to demagnetize and increase magnetism. It is also possible to facilitate the process. Note that this magnetizing and demagnetizing can be realized by increasing or decreasing the strength of all magnets to the same extent. However, the purpose of magnetizing and demagnetizing is adjustment of the voltage between the motor terminals, so that the object can be achieved by varying the strength of some magnets. If the magnet whose strength is variable is a specific magnet, the motor design can be made easier, and the operation of magnetizing and demagnetizing can be simplified.

For example, the motor control device can perform demagnetization and magnetization in FIG. 2, and then the rotation control of the motor can also be performed by this control device. If the demagnetization current is insufficient, the current capacities of the three power transistors 564, 565, and 566 may be increased. In addition, since the current required for magnetizing and demagnetizing is an instantaneous current, it is also possible to add another power element such as a magnetic contactor and thyristor to configure and add a functional unit dedicated to magnetizing and demagnetizing. It is. It will be arranged in parallel with the motor control inverter. In any case, the magnet magnetizing and demagnetizing method of the motor of the present invention is specifically determined by positioning at the rotor rotation position θr convenient for magnetizing and demagnetizing, and the size necessary for magnetizing and demagnetizing. This current can be realized by energizing at least the necessary minimum time. Also, when operating at high speed rotation, when demagnetizing so that the magnetic flux density of the magnet can be reduced, it is reduced when the corresponding permanent magnet comes to a rotational position where it can be demagnetized during rotation before entering high speed rotation. It can be demagnetized by applying a magnetic current. The same applies to the magnetization. When the rotation starts from the high speed rotation to the low speed rotation, the magnetization can be increased by the magnetizing current in the same manner.

  Next, an example of a method for driving the motor of the present invention will be described. As described above, in the case of the motor as shown in FIGS. 9 and 10, the motor can be rotationally driven by applying each phase current at each rotor rotational position θr as shown in FIG. 11. Each stator magnetic pole sequentially attracts the rotor magnetic poles to generate rotational torque. At this time, the direction of the current applied to each of the windings wound around the circumferential slots of the stator magnetic pole is reversed, and the magnetism of the stator magnetic pole adjacent in the circumferential direction is N pole and S pole. Alternating. Therefore, as the rotor rotates, the stator magnetic poles adjacent to each other in the circumferential direction are sequentially excited while continuously reversing the direction of the magnetic flux to be excited, and the rotor magnetic poles are continuously attracted. This is a control method in which rotational torque is generated and driven. At this time, the magnetic flux of the rotor magnetic pole is configured to reverse the direction of the magnetic flux as it rotates. The currents Ia, Ib, Ic of each phase and the torques Ta, Tb, Tc generated by the stator magnetic poles at this time are as shown in FIG.

  In such a driving method of the motor of the present invention, the method of passing current to the windings of the slots in the circumferential direction in synchronization with the rotor is a driving method similar to that of a normal synchronous motor. However, in the present invention, each phase current is a unidirectional current.

  The control of current and torque as shown in FIG. 12 can be realized by the control device shown and described in FIG. And the data regarding the electric current and torque of FIG. 12 is implement | achieved by storing the information in DATA.

  FIG. 107 shows an example of a specific speed control device that realizes the current control and torque control of the motor of the present invention. Reference numeral 591 denotes a motor according to the present invention, which represents a three-phase motor shown in FIG. 1, FIG. 9, FIG. An example of control of such a three-phase motor will be shown. Since the motor of the present invention is a motor that has a sequential and discontinuous control surface that is controlled by energizing the unidirectional currents Ia, Ib, and Ic, and is driven using a magnetically nonlinear region, the conventional control is performed. There are differences from the device. Reference numeral 592 denotes an A-phase winding, 593 denotes a B-phase winding, and 594 denotes a C-phase winding. Reference numeral 595 denotes a position detector that detects the rotational position θr of the rotor. Reference numeral 596 denotes an interface of the position detector 595 that outputs the rotor rotational position θr and the rotational speed ωr of the rotor. Reference numeral 599 denotes a speed command signal, and a speed error 59B, which is a difference from the rotor rotational speed ωr, is obtained by the adder 59A and is output to the compensator 59C. The compensator 59C performs proportional calculation, integral calculation, differential calculation, and the like, and outputs the added value as a torque command signal TC to the current-voltage calculator 59G.

  The current-voltage calculator 59G receives the torque command TC, inputs the rotor rotational position θr and the rotor rotational speed ωr, and uses the database DATA described later, to use the A-phase, B-phase, and C-phase current commands Ica, Icb. , Icc and three-phase voltage commands Vfa, Vfb, Vfc are obtained and output.

  The A-phase current command Ica calculates the difference from the A-phase current detection value Isa with an adder, the compensation calculation is performed by the compensator J31, and the voltage command Vea is added to the A-phase voltage prediction command value Vfa to add the A-phase voltage. Command value Vca is output to voltage width modulation amplifier PWM. The B phase current command Icb is calculated by the adder to calculate the difference from the B phase current detection value Isb, the compensation calculation is performed by the compensator J32 to be the voltage command Veb, and the B phase voltage prediction command value Vfb is added to the B phase voltage. Command value Vcb is output to voltage width modulation amplifier PWM. The C phase current command Icc calculates the difference with the C phase current detection value Isc by the adder, the compensation calculation by the compensator J33 is performed as the voltage command Vec, and is added to the C phase voltage prediction command value Vfc to obtain the C phase voltage. Command value Vcc is output to voltage width modulation amplifier PWM.

  The function of the pulse voltage width modulation amplifier PWM is to control the voltage time width in proportion to the input voltage command values Vca, Vcb, Vcc, and to perform pulse width modulation to amplify the average voltage proportional to the input voltage command value. This is a function that combines a so-called PWM modulator and a function of power amplification by a power transistor as shown in FIG. The three-phase voltage and current are supplied to the windings of each phase of the motor in accordance with the three-phase voltage inputs Vca, Vcb, and Vcc. There are various modulation amplifiers for reducing the switching loss of the power transistor or improving the response, and the amplification method is not particularly limited.

  The motor current of each phase is detected by a current detector to obtain detected currents Isa, Isb, Isc, and the difference is obtained by the current commands Ica, Icb, Icc and the adders of the A phase, B phase, C phase, The A phase compensator J31, the B phase compensator J32, and the C phase compensator J33 respectively calculate voltage components for correcting the current error of each phase.

  The voltage Vac to be applied to the A-phase winding is obtained by adding the output voltage of the compensator J31 and the voltage command Vfa to the voltage width modulation amplifier PWM, amplifying the power, and supplying the current and voltage to the A-phase winding. give. The voltage Vcb to be applied to the B-phase winding is obtained by adding the output voltage of the compensator J31 and the voltage command Vfb to the voltage-width modulation amplifier PWM and amplifying the power to supply current and voltage to the B-phase winding. . The voltage Vac to be applied to the C-phase winding is obtained by adding the output voltage of the compensator J31 and the voltage command Vfc to the voltage width modulation amplifier PWM, amplifying the power, and supplying current and voltage to the C-phase winding. . As a result, if the current value command values Ica, Icb, Icc of each winding and the voltage values Vfa, Vfb, Vfc necessary to energize the currents can be given, accurate and highly responsive current and voltage control can be achieved. It becomes possible.

  The contents of the database DATA are for calculating and obtaining A-phase, B-phase, and C-phase current commands Ica, Icb, and Icc when a torque command TC, a rotor rotational position θr, and a rotor rotational speed ωr are given. Stores data. At this time, it is preferable that the three-phase voltage commands Vfa, Vfb, and Vfc are also obtained by calculation. The contents of the database DATA may be mathematical expressions or numerical values. Output information for all combinations of inputs may be recorded. In this case, however, the amount of data increases and there is a problem of cost.

  Next, a specific example in which the database DATA for controlling the motor of the present invention is obtained in advance will be described. For example, for the motor shown in FIG. 4A having a specific shape, the current values Ia = Ic = Ix, Ib = 0, and the rotational position θr = 30 ° of the A-phase and C-phase, and the nonlinear finite element method The magnetic field analysis is performed to obtain the interlinkage magnetic flux φx interlinked with each phase winding. At the same time, the torque Tx is also obtained. Under this condition, the magnetic fluxes linked to the A-phase winding and the C-phase winding are equal, and the magnetic fluxes linked to the B-phase winding are almost zero. In this way, as shown in the table of FIG. 109, the flux linkage number Ψ and torque T under all conditions used in motor operation are obtained. Rows indicate current conditions. For example, In means a certain combination of three-phase currents Ia, Ib, and Ic. The column indicates the rotor rotational position θr, and θm indicates a certain value. Ψmn represents the number of flux linkages Ψmna, Ψmnb, Ψmnc and torque Tmn of the A-phase winding, B-phase winding, and C-phase winding. Since the flux linkage number Ψ = Nw × φ is the product of the number of winding turns Nw and the linkage flux φ, it may be expressed by the linkage flux φmna, φmnb, φmnc of each phase.

  Here, as for the combination conditions of the three-phase currents Ia, Ib, and Ic, if 10 discrete representative values are determined for each, 1000 combinations can be made. However, combinations of current values that are not used in normal operation can be omitted. The rotor rotational position θr ranges from 0 ° to 360 °, but the same data is repeated because the three-phase magnetic circuit and the winding are symmetrical, and is omitted to 1/2, 1/3, or 1/6. can do. In this way, the motor model is subjected to magnetic field analysis by the nonlinear finite element method, and all the data in the table of FIG. 109 is obtained.

  The method of obtaining the interlinkage magnetic flux of each winding under the actual motor use conditions using the data of FIG. 109 is based on data values in the vicinity of the obtained condition since the data to be stored is discrete sample values. Find by interpolation calculation. In most cases, sufficient accuracy can be obtained by proportionally calculating from data values in the vicinity of the desired condition. More specifically, the data in the table of FIG. 109 does not store the flux linkage data or the like when the variable combination Ia, Ib, Ic, θr to be calculated has the same value. Approximation of magnetic flux linkage number Ψmna, Ψmnb, Ψmnc and torque Tmn corresponding to the condition of an arbitrary motor operating point by interpolating from the data of matrix Ψmn of the closest combination of variables and the upper, lower, left and right data of the matrix Ψmn A value can be obtained.

  Next, database conversion will be described. In the table data of FIG. 109, assuming that the flux linkage number and torque data in the vicinity of all the operating conditions of the motor are discretely secured, the current voltage calculator 59G will be obtained from the table of FIG. In principle, the phase current command values Ica, Icb, Icc and the phase voltage command values Vfa, Vfb, Vfc under the conditions of the arbitrary rotational position θr and the torque command value TC are obtained. However, it is difficult to find and calculate such data in real time when operating the motor. In order to solve this problem, if the database is converted so that the data required by the current-voltage calculator 59G can be easily obtained, a necessary value can be obtained by a simple calculation during motor control.

  In one example, the table in FIG. 109 is converted into the table in FIG. 110 before the control in FIG. 107 is performed. The rows in the table of FIG. 110 are the conditions for the torque command value TC, and the columns are the conditions for the rotor rotational position θr. Each table in the table of FIG. 110 stores data for calculating the corresponding torque command value TC, each phase current value and each phase voltage value at the rotor rotational position θr. In the table of FIG. 110, the corresponding data Pmn obtained from the torque command value TC and the rotational position θr, which are the inputs of the current / voltage calculator 59G, are searched for, and each of them is interpolated from the upper, lower, left and right data. Approximate values of the phase current command values Ica, Icb, Icc and the phase voltage command values Vfa, Vfb, Vfc can be obtained.

  The data Pmn in FIG. 110 can be generated in various forms. The information on each phase current command value Ica, Icb, Icc is each phase current at which the torque Tn is obtained at the rotational position θm. Further, in order to calculate each phase voltage command value Vfa, Vfb, Vfc, information on the rotational speed ωr is required from the equation (4). One method is a method of storing the value of the rotational change rate (dΨ / dθr) of the flux linkage number Ψ from the equation (4). In that case, the voltage command values Vfa, Vfb, and Vfc of each phase can be obtained by performing approximate calculation from the rotational change rate of the magnetic flux linkage in the front, rear, left, and right matrices according to the equation (4) and multiplying by the rotational speed ωr. . Note that the flux linkage number ψ is the product of the linkage flux φ and the number of turns Nw, and ψ = Nw × φ, and therefore the rotation change rate data (dφ / dθr) of the linkage flux φ may be used.

  Even when the data of (dΨ / dθr) is not stored in the data of FIG. 107, the rotational change rate of the linkage flux is calculated from the flux linkage number data before and after the corresponding matrix data of FIG. It is also possible to calculate each time. In another method, each phase voltage command value at each rotation speed is stored as a direct value in each data column in FIG. 110, and each phase voltage command value Vfa, Vfb, Vfc at that motor rotation speed is stored. Is obtained by interpolation calculation. In this way, not only the data format of FIG. 110 but also various data tables can be created.

  The table data in FIG. 110 is an example of the DATA in FIG. Using this table data, approximation of each phase current command value Ica, Icb, Icc and each phase voltage command value Vfa, Vfb, Vfc according to torque command value TC, rotation position θr, rotation speed dθr / dt = ωr The value can be determined.

  Note that the torque Tx obtained by the nonlinear finite element method of magnetic field analysis matches well with the equation (10) or (15), and may be calculated from the flux linkage number ψ. When the current and voltage are obtained by the method of the present invention, the error is small even if the motor soft magnetic material is in a magnetic saturation nonlinear region, and even if a permanent magnet is included in the motor, it can be handled without being aware of the permanent magnet. . This is because the physical quantity to be handled is calculated only with the basic physical quantities of voltage, current, magnetic flux, and rotation speed, and therefore there are few error elements.

  For a long time, induction motors, transformers and the like have been expressed by inductance L. In fact, these old instruments have been used in a magnetically linear region near 1.6T. Even if the value of the inductance L as a proportional coefficient of current and magnetic flux is treated as a constant value, there are few problems. However, in recent years, variable frequency control is performed by an inverter, and due to demands for downsizing and cost reduction, devices are frequently used in a magnetically non-linear region near magnetic saturation at continuous rated torque. And it is common sense that many motor manufacturers seek motor characteristics by nonlinear finite element method and pursue performance. In the case of a motor that uses a magnetically non-linear region, it is used in a region where the value of inductance L, which is a proportional constant between current and magnetic flux, changes greatly, and there is a contradiction in the voltage equation of the motor. Then, it is assumed that the inductance L is a value that changes depending on conditions.

  In a simple case, the voltage V of the motor winding, the current I, the number of turns Nw, and the magnetic flux φ are represented by the following equations (28) and (29).

V = L × (dI / dt) = Nw × (dφ / dt) (28)
L × I = Nw × φ = Ψ (29)
The motor can be expressed as a model expression as shown in FIG. The input of the motor is a voltage V and a current I, the inside of the motor is a voltage V, a current I, a flux linkage number ψ, and a rotational speed ωr, and the motor output is a torque T and a rotational speed ωr. Here, there is an inductance L as a coefficient between the current I and the flux linkage number ψ as in the equation (29), but if it is non-linear, there is almost no meaning as a proportional constant.

  The data as shown in FIG. 109 can be obtained relatively easily by the nonlinear finite element method, and can be converted into the table data of FIG. 110. Thus, the control method of the present invention enables consistent development from motor development to motor control. Is possible. The validity of the control method of the present invention using the number of magnetic flux linkages is described in IEEJ Transactions D, Journal of Industrial Applications, IEEE Trans. IA, Vol. 127, no. On pages 2,2007 and 158 to page 166, the inventors have reported, "Proposal of inductance calculation method and modeling using the number of flux linkages of a synchronous reluctance motor". Although it is an example of a conventional synchronous motor, accurate torque can be calculated from the number of magnetic flux linkages even in a non-linear region of the magnetic saturation region, and even a motor with a permanent magnet built in the rotor is aware of the permanent magnet. This shows that an accurate torque can be calculated using the number of magnetic flux linkages, and that the voltage of each winding can be calculated from the number of magnetic flux linkages.

  Next, a method for controlling the motor of the present invention to reduce vibration and noise will be described. The motor shown in FIG. 103 is a motor in which the circumferential width Hm of the rotor magnetic pole of the rotor J61 is increased to 75 ° as compared with the motor shown in FIG. The circumferential width Ht of the stator magnetic pole is 40 °, and the circumferential width Hs of the slot opening is 20 °. Now, CCW torque is generated in the rotor, rotated to CCW, and each current Ia, Ib, Ic and torque T when rotating in the order of (a), (b), (c), (d) in FIG. And the radial attractive force FSR acting between the stator side and the rotor side will be described. For ease of understanding, a two-pole motor model is illustrated.

  In the state of the rotor rotational position θr = 30 ° in FIG. 103 (a), the A phase current Ia is supplied to the A phase windings 131 and 134, and at the same time, the C phase current Ic is supplied to the C phase windings 135 and 132. Then, the magnetic flux J62 indicated by the broken line is excited to generate the CCW torque Ta. At this time, a radial attractive force FSR acts between the stator magnetic poles 117 and 11A and the rotor. In the direction of the stator magnetic poles 11C and 119, since the magnetomotive forces of the currents Ia and Ic cancel each other, no magnetic flux is generated. FIG. 104 shows the current of each phase at each rotor rotational position θr shown in FIG. 103 and the generated torque.

  When rotating to the rotor rotational position θr = 55 ° in FIG. 103 (b), the C-phase current Ic2 is energized, and at the same time, the B-phase current Ib2 is energized to the B-phase windings 133 and 136 to excite the magnetic flux J62. Torque Ta is generated. At this time, if the components of the currents Ia and Ic that excite the magnetic flux J62 are zero, the magnetic flux J62 changes abruptly, the radial attractive force FSR decreases rapidly, and vibrations are generated in the stator and the rotor. .

  As a countermeasure, at the rotor rotational position θr = 55 ° in FIG. 103 (b), the stator magnetic poles 117 and 11A are almost opposed to the convex portions of the rotor, so that the magnetic flux J62 does not generate CCW or CW torque. . The rotation region where the magnetic flux J62 does not generate torque continues to the rotor rotation position shown in FIGS. 103 (c) and 103 (d). Utilizing this, the A-phase current Ia1 and the C-phase current Ic1 that maintain the magnetic flux J62 to such an extent that the radial attractive force FSR does not rapidly decrease are supplied. After all, each phase current is expressed by the following equations (30) to (32).

Ia = Ia1 (30)
Ib = Ib2 (31)
Ic = Ic1 + Ic2 (32)
Thus, both the magnetic flux J63 and the magnetic flux J62 in FIG. 103 are excited. At this time, the generation of torque by the magnetic flux J63 and the maintenance of the minimum amount of the magnetic flux J62 are compatible. At this time, the magnetic flux in the direction of the stator magnetic poles 11C and 119 is zero. This is because the excitation current in that direction is expressed by the following equation (33).

Ia + Ib-Ic = Ia1 + Ib2- (Ic1 + Ic2) (33)
= 0
This is because Ia1 and Ic1 are equal, and Ib2 and Ic2 are equal.

  At the rotor rotational position θr = 70 ° in FIG. 103 (c), the condition is the same as in the state of FIG. 103 (b), and the magnetic flux J62 is decreased to such an extent that the radial attractive force FSR does not decrease rapidly. The current components Ia1 and Ic1 that excite the magnetic flux J62 are gradually reduced. At this time, the components of the currents Ic2 and Ib2 that generate torque energize the current value necessary for generating the torque Tc.

  At the rotor rotational position θr = 90 ° in FIG. 103 (d), the magnetic flux J62 begins to generate CW torque, so that the current components Ia1 and Ic1 are set to zero. The components of currents Ic2 and Ib2 that generate torque energize a current value necessary for generating torque Tc.

  In the state of FIG. 103 (d), it is only necessary that the torque of the CCW is the desired torque T3. Therefore, the current components Ia1 and Ic1 are not zero but generate some CW torque T4, T5 = T3 + T4 may be used as the torque T5 obtained by increasing the CCW torque from T3 so that the components of the currents Ic2 and Ib2 that generate the CCW torque supplement the CW torque.

  Further, the currents Ia, Ib, and Ic of the three phases between (b) to (c) and (d) of FIG. 103 are not necessarily values of the expressions (30), (31), and (32). It shows the basic idea. Accordingly, the value may be a slightly different value as long as the desired torque T3 is obtained and the radial attractive force does not fluctuate rapidly.

  For example, at the rotor rotational position θr in FIGS. 103B and 103C, the magnetic flux J62 can be maintained by making the C-phase current Ic larger than the B-phase current Ib. That is, this method can also suppress the generation of torque and the rapid fluctuation of the radial attractive force FSR. In the case of this method, the A-phase current can be made zero in the meantime. As described above, the value of the three-phase current that achieves both the generation of torque and the suppression of the rapid fluctuation of the radial attractive force FSR is naturally limited, but can take many kinds of values.

  In addition, vibration and noise are manifested in relation to the resonance frequency of the motor and peripheral parts, and often cause problems. Therefore, it is necessary to control the increase / decrease in the radial attractive force as a function of time, and each current waveform with the horizontal axis as the rotor rotational position θr varies depending on the rotational speed of the motor.

  The state of FIG. 103 (d) is a position rotated by 60 ° from the state of FIG. 103 (a) to the CCW, and the above operation can be repeated and continuously rotated. And it can control so that radial direction attractive force FSR may not fluctuate rapidly, and can reduce vibration and noise. At this time, unnecessary current from the viewpoint of torque generation is also energized, so that each phase current of the motor is selected based on a balance between efficiency, vibration and noise.

  In FIGS. 103 and 104, a large torque ripple is generated in the torque Tm shown in FIG. 104 (G) generated by this motor. However, the torque ripple can be reduced by the skew shown in FIG. It is.

  Next, the motor shown in FIG. 105 is a motor example in which the circumferential width of the rotor magnetic pole of the motor shown in FIG. 1 is increased from 30 ° to 45 ° in electrical angle. As described with reference to FIGS. 103 and 104, a driving method for reducing a rapid change in the axial suction force FSR will be described with reference to FIGS. 105 and 106. FIG. This is a method for controlling the vibration and noise of the motor of the present invention to be small. Now, CCW torque is generated in the rotor, rotated to CCW, and each current Ia, Ib, Ic and torque T when rotating in the order of (a), (b), (c), (d) in FIG. And the radial attractive force FSR acting between the stator side and the rotor side will be described. For ease of understanding, a two-pole motor model is illustrated.

  In the state of the rotor rotational position θr = 30 ° in FIG. 105 (a), the A-phase current Ia is supplied to the A-phase windings A0D and A0G, and at the same time, the C-phase current Ic is supplied to the C-phase windings A0H and A0E. The magnetic flux K12 indicated by the broken line is excited to generate a CCW torque Ta. At this time, a radial attractive force FSR is acting between the stator magnetic poles A01 and A04 and the rotor. In the direction of the stator magnetic poles A06 and A03, no magnetic flux is generated because the magnetomotive forces of the currents Ia and Ic cancel each other. FIG. 106 shows the current of each phase at each rotor rotational position θr shown in FIG. 105 and the generated torque.

  When rotating to the rotor rotational position θr = 45 ° in FIG. 105 (b), the A-phase current Ia3 is energized, and simultaneously the B-phase current Ib3 is energized to the B-phase windings A0F and A0J to excite the magnetic flux K13. Torque Tb is generated. At this time, if the components of the currents Ia and Ic for exciting the magnetic flux K13 are zero, the magnetic flux K13 changes abruptly, the radial attractive force FSR decreases rapidly, and vibrations are generated in the stator and the rotor. .

  As a countermeasure, at the rotor rotational position θr = 45 ° in FIG. 105 (b), the stator magnetic poles A01 and A04 are almost opposed to the rotor projections, so that the magnetic flux K12 does not generate a CCW or CW torque. . The rotation region where the magnetic flux K12 does not generate torque continues until the rotor rotation position θr = 60 ° in FIG. 105 (c). Utilizing this, the A-phase current Ia4 and the C-phase current Ic4 that maintain the magnetic flux K12 to such an extent that the radial attractive force FSR does not rapidly decrease are supplied. After all, each phase current becomes the following formula.

Ia = Ia3 + Ia4
Ib = Ib3
Ic = Ic4
Thus, both the magnetic flux K12 and the magnetic flux K13 in FIG. 105 are excited. At this time, the generation of torque by the magnetic flux K13 and the maintenance of the minimum amount of the magnetic flux K12 are compatible. At this time, the magnetic flux in the direction of the stator magnetic poles A02 and A05 is zero. The excitation current in that direction is expressed by the following equation.

Ia-Ib-Ic = Ia3 + Ia4-Ib3-Ic4
= 0
This is because Ia3 and Ib3 are equal, and Ia4 and Ic4 are equal.

  At the rotor rotational position θr = 60 ° in FIG. 105 (c), the A-phase current Ia4 and the C-phase current Ic4 are set to zero, and the magnetic flux K12 is set to zero. The A-phase current Ia4 and the C-phase current Ic4 are gradually decreased from the state shown in FIG. 105 (b) until the rotation to CCW by 15 °. At this time, the current components of the currents Ia3 and Ib3 that generate torque energize the current value necessary for generating the torque Tb.

  At the rotor rotational position θr = 75 ° in FIG. 105 (d), the C-phase current Ic and the B-phase current Ib are energized, the magnetic flux K11 is excited, and the torque Tc is generated. On the other hand, the magnetic flux K13 is a position where CCW torque cannot be generated, and the exciting current components Ia3 and Ib3 of the magnetic flux K13 are reduced to such an extent that the radial attractive force FSR does not rapidly decrease. Zero until the rotational position θr = 90 °.

  In the following rotation in FIG. 105, the currents Ia, Ib, and Ic of the respective phases can be similarly controlled and can be continuously rotated by the same operation. As for the magnitude of each phase current, the torque current component is determined according to the magnitude of the load torque, and the total radial direction attractive force FSR does not change abruptly for the current component that generates the radial direction attractive force FSR. In addition, it is necessary to determine so as not to resonate with peripheral components. Further, the driving method shown in FIGS. 105 and 106 is an example, and various modifications can be made.

  Next, a method of driving a motor that obtains continuous torque by alternately selecting rotor magnetic poles that generate torque will be described. The motor is, for example, the motor shown in FIG. 1 and has the relationship between the rotor rotational position θr, current, and torque shown in FIGS. 4 and 6. A method for realizing such relationship control can be realized by the control device of FIG. The current / voltage calculator 59G stores the control algorithm of the current and voltage of the motor to be controlled as shown in FIG. 6, and the magnitude of each current and the magnitude of each voltage depend on the torque command value TC and the rotor rotational position θr. And the rotor speed ωr are input, and the motor phase current command values Ica, Icb, Icc and the phase voltage command values Vfa, Vfb, Vfc are calculated. The motor current control algorithm shown in FIG. 1 is characterized in that the rotor magnetic poles that generate the drive torque change with the rotation of the rotor, and generate torque alternately. That is, a combination in which the rotor magnetic pole and the stator magnetic pole approach each other with the rotation of the rotor is selected from all the stator magnetic poles, and a current in a specific current direction is applied to the windings in the circumferential direction of the selected stator magnetic pole. . Although the torque generated in each rotor magnetic pole is intermittent with respect to the rotor rotation angle, the torque generated by a plurality of rotor magnetic poles as a whole rotor can be a substantially continuous torque.

  Next, at low speed rotation, the rotor magnetic poles are alternately selected and rotated by a motor driving method that obtains continuous torque. As the rotor rotational speed increases, the torque generated at a specific rotor magnetic pole increases, and the other rotor A method for reducing torque generation at the magnetic pole will be described. In the motor shown in FIG. 54, a rotor is composed of a main salient pole magnetic pole 161 having a wide circumferential magnetic pole width and an auxiliary salient pole magnetic pole 162 having a narrow circumferential magnetic pole width. In order for this motor to continuously generate torque, the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 need to generate alternating torque. FIG. 57 shows the relationship between the phase currents Ia, Ib, and Ic that generate the CCW torque and the torques Ta, Tb, and Tc. In the low-speed rotation, the rotor magnetic pole approaches as the rotor magnetic pole rotates in this way, and the opposing stator magnetic pole is selected from among all the stator magnetic poles. Current in the current direction is applied, and the torque generated in each rotor magnetic pole is intermittent, but the rotor as a whole generates almost continuous rotational torque. In high-speed rotation, as the rotor magnetic pole rotates, the stator magnetic poles adjacent to each other in the circumferential direction are sequentially excited while reversing the direction of the magnetic flux to be excited in order to attract the rotor magnetic poles almost continuously. Rotational torque can be generated and driven. The rotor magnetic pole used at high speed rotation is the main salient pole magnetic pole 161. This driving at high speed is to control the motor of FIG. 9 like the current and torque shown in FIG. In FIG. 112, when the rotor rotational position θr is from 0 ° to 210 °, torque is generated by alternately using the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162. After 210 °, torque is generated only by the main salient pole magnetic pole 161. An example of current and torque when it occurs is shown.

  The current of each phase can gradually shift from the control shown in FIG. 57 to the control of FIG. 12 from low speed rotation to high speed rotation. Alternatively, the control as shown in FIG. 57 may be performed up to a certain number of revolutions, and the control may be switched to the control of FIG. 12 above the number of revolutions. In high-speed rotation, even if the generated torque is intermittent, the time interval for generating the torque is small, so that it rotates with inertia during zero torque, and there are few practical problems. FIG. 113 shows an example of the drive ratio of the main salient pole magnetic pole 161 and the drive ratio of the auxiliary salient pole magnetic pole 162 with the horizontal axis as the rotation speed. The drive ratio of the main salient pole 161 increases to R0, R1, R2, and R3 along with the rotation speeds 0, N1, N2, and N3, and the drive ratio of the auxiliary salient pole 162 decreases to R4, R5, R6, and R7. The weights are weighted so that the average torque obtained by adding both torques is approximately the same.

  The control shown in FIG. 57 can generate a continuous torque, but there is a large fluctuation in the attractive force between the stator magnetic pole and the rotor magnetic pole, and a radial force is alternately generated in different parts. It tends to cause vibration of the stator core with low mechanical rigidity. In this respect, since the attractive force between the stator magnetic pole and the rotor magnetic pole in the case of FIG. 12 becomes a radial attractive force like an AC synchronous motor, fluctuations in the radial attractive force are relatively small, and vibration and noise can be reduced. Suitable for operation at high speed. Further, in the control of FIG. 57, since the current in the radial direction increases and decreases, there are problems such as an increase in iron loss, switching loss of the control device, and reactive power.

  Next, a motor control method using the linkage flux φ or flux linkage number ψ shown in FIGS. 107, 108, 109, 110, etc. will be described. As described in the control device of FIG. 107, and as shown by the equation (2), the voltage of each winding is mainly proportional to the time change rate (dφ / dt) of the linkage flux φ. As shown in the equation (4), the winding voltage is proportional to the product of the rotational change rate (dφ / dθ) of the flux linkage φ and the rotational speed ωr. On the other hand, the flux linkage φ under each current condition of each winding can be accurately obtained by performing a magnetic field analysis on the motor using a nonlinear finite element method. Using this analysis data, the rotational change rate (dφ / dθ) of the interlinkage magnetic flux of each winding can be calculated. As a result, the rotational speed detector ωr is obtained by using the rotational position detector, and the equation (4) is obtained by multiplying the rotational change rate (dφ / dθ) of the interlinkage magnetic flux of the winding by the number of turns Nw of the winding. Can be used to accurately determine the voltage of that winding. However, this calculation is for the case where the current of each winding is constant, and when the current value changes, it is also necessary to calculate the voltage component accompanying the change in current.

  The linkage flux φ and the flux linkage number ψ of each winding are functions of the rotor rotational position θr and the phase currents Ia, Ib, and Ic. Therefore, in order to indicate a function, it can be expressed as φ (θr, Ia, Ib, Ic), ψ (θr, Ia, Ib, Ic). Now, when controlling the motor, if the rotor rotation position and each phase current change from (θ1, A1, B1, C1) to (θ2, A2, B2, C2) in a short time of Δt The voltage Vx of the winding can be written as the following equation (34).

Vx = Nw × dφ / dt
≒ Nw x {φ (θ2, A2, B2, C2)
−φ (θ1, A1, B1, C1)} / Δt (34)
Here, for θ2, (Δt × ωr) may be added to θ1. Each current value is a target current value after time Δt, and can be obtained by interpolation calculation from the data table shown in FIG.

  At this time, the current target value cannot be selected arbitrarily, and the maximum value of the current change rate is limited by the maximum value of the voltage that can be applied to the winding as shown in the equation (28). Since there is a limit on the maximum value that can be changed from the current value (A1, B1, C1), the current value (A2, B2, C2) is limited to that range.

  The voltage Vx represented by the equation (34) is roughly divided into the following three voltage components. The first voltage component VX1 is a component in which each current is constant and the interlinkage magnetic flux φ changes with rotation, and is a value represented by equations (1) to (15).

  The second voltage component VX2 is a voltage component based on a change in linkage flux when the current value of each winding changes from (A1, B1, C1) to a current value (A2, B2, C2). This voltage component is expressed by equation (28), and the inductance L needs to be obtained by an incremental inductance Linc at the operating point where the current values are (A1, B1, C1).

  The third voltage component VX3 is a voltage component based on the time change rate of the magnetic flux φt between the stator magnetic pole A02 and the stator magnetic pole A05 in the case of the rotational position θr in FIG. When the torque is generated in the CCW in the state of FIG. 1, the magnetic flux indicated by the thick line is induced to generate the CCW torque, but the stator magnetic pole A02 and the stator magnetic pole A05 are caused by an error between the A phase current Ia and the C phase current Ic. Is generated, the third voltage VX3 is generated.

  In summary, the voltage of each winding can be calculated by the above equation (34) if the interlinkage magnetic flux in the current state used for motor control at each rotor rotational position is known. For the voltage component of the second voltage component VX2, the voltage component of the second voltage component VX2 is calculated by the equation (28) if the incremental inductance at the current operating point at that time is known. be able to.

  Further, the current value and voltage value of each winding at each operation mode of the motor, that is, each torque command value TC, rotor rotation position θr, and rotor rotation speed ωr, are referred to as current information IJ of each phase and voltage information VJ of each phase. It is also possible to drive the motor by storing it in a memory and interpolating and using the current value and voltage value of the motor operating state from these discrete data. This is because the relationship between current information, magnetic flux linkage number information, torque information, etc. of each winding is expressed as a voltage equation for each winding, so these relationships are related to the driving conditions such as motor torque, rotor rotation position, etc. It means that the current information IJ of the winding and the voltage information VJ of each phase winding can be replaced and stored in the memory.

  Although the description has been given using the relationship between the linkage flux and the voltage, the linkage flux φ or the flux linkage number ψ is calculated using the value of the nonlinear inductance L instead of the linkage flux from the relationship of the equation (29). It can also be expressed and is equivalent. However, the inductance L is a value that changes with the current value of each phase, and handling is often complicated.

  Next, FIG. 111 shows a control device in which torque feedback control is added to the control device shown in FIG. In the case of the control device of FIG. 107, the motor current value and voltage value are accurately given to control the motor at high speed and with high accuracy. However, when the power supply voltage of the control device fluctuates, or when an error in information such as motor parameters or fluctuates due to temperature or the like, there is a problem that the motor control error increases.

  In the control device shown in FIG. 111, the torque detection device J3C inputs the current detection values Isa, Isb, Isc and the rotor rotational position θr for each of the three phases, and utilizes information DATA that stores the interlinkage magnetic flux of each winding of the motor. Then, the estimated value Ts of the motor torque is detected. The torque estimate value Ts is subtracted from the torque command value TC by the adder J3E and fed back, and compensation operation such as proportionality and integration is performed by the compensator J3B, and a torque error correction value Tcm is output to the current voltage calculator J3D. . The current voltage calculator J3D converts the torque error correction value Tcm into a three-phase current command Ica, Icb, Icc or a three-phase voltage command Vfa, Vfb, Vfc or both. In this way, when an error occurs in the motor torque due to a power supply voltage fluctuation or a motor parameter error, the control accuracy of the motor control device can be improved by adopting a configuration that can feed back the torque error, Higher accuracy and higher response control can be realized.

  Although various methods have been shown for the current control method of the motor of the present invention, in order to obtain both torques of the two stator magnetic poles that overlap in a specific section, a current obtained by adding two current components is applied. You can also. In addition, it is possible to superimpose a current of another component on a current for generating a basic torque in order to reduce vibration and noise. In the motor of FIG. 9, FIG. 10, etc., offset current can be superimposed on each three-phase current, and control can be performed so that the change in the attractive force in the radial direction becomes small. There is also an advantageous current control method for shortening the current increase / decrease time. Like these various current control methods, there are current control methods obtained by modifying the current control method shown in the present invention, and these current control methods are also included in the present invention.

  Next, a method for detecting the rotor rotational position θr from the inductance, winding current and winding voltage of the motor, not the position detector, will be described. This is so-called sensorless position detection or sensorless speed detection. FIG. 114 shows the characteristics of a motor model similar in shape to the motors shown in FIGS. 54 and 55, analyzed by a magnetic field using the nonlinear finite element method. The rotor rotational position θr [°] where the horizontal axis represents the electrical angle, the vertical axis is the interlinkage magnetic flux [mWb] of the A phase winding and the C phase winding, and the current value Ia = Ic of the A phase and the C phase Is a parameter. The number of poles is 14, the circumferential width Hm expressed by the electrical angle of the main salient pole magnetic pole 172 is 40 °, the circumferential width Hh of the auxiliary salient pole magnetic pole 173 is 20 °, and the circumferential width of the slot opening. In this example, Hs is 20 ° and the B-phase current Ib = 0.

  114 and the two-pole model of FIG. 54, the interlinkage magnetic flux φ of the main salient pole magnetic pole 161 excited by the A-phase and C-phase windings when the rotor rotational position θr is 0 ° to 50 °. Can be increased with the rotation of the rotor to generate CCW torque, and from 50 ° to 100 °, the interlinkage magnetic flux φ of the A-phase and C-phase windings decreases with the rotation of the rotor to generate CW torque. Can do. When the rotor rotational position θr is at 100 °, the auxiliary salient pole 162 comes near the stator magnetic poles 117 and 11A sandwiched between the A-phase winding and the C-phase winding. The interlinkage magnetic flux φ of the A-phase and C-phase windings increases with the rotation of the rotor to generate a CCW torque, and the interlinkage magnetic flux φ of the A-phase and C-phase windings varies between 140 ° and 180 °. Along with this, the torque of CW can be generated.

  At this time, the relationship between the current value I and the torque T is the relationship as shown in FIG. 5 and the relationship between the equation (10) and the equation (15). In FIG. Proportional. In the region where the current values Ia and Ic as parameters in FIG. 114 are small and magnetically linear, the characteristic gradient increases with the current value.

  In this motor, the characteristic corresponding to FIG. 114 when the C-phase current Ic is supplied to the C-phase windings 115 and 112 and at the same time the B-phase current Ib is supplied to the B-phase windings 113 and 116 is shown in FIG. Compared with the electrical angle, the phase is delayed by 120 °. The characteristics corresponding to FIG. 114 when the B-phase current Ib is supplied to the B-phase windings 113 and 116 and at the same time the A-phase current Ia is supplied to the A-phase windings 111 and 114 are compared with those in FIG. Thus, the phase is delayed by 240 ° in electrical angle. Therefore, by supplying the same current to two of the three phases, three types of characteristics, that is, the characteristic shown in FIG. 114, the characteristic shifted by 120 ° and the characteristic shifted by 240 ° can be obtained. .

  In addition, the inductances La and Lc when energizing the A-phase and C-phase windings are roughly similar to those in FIG. 114, and have the AC characteristics in FIG. The horizontal axis is the rotor rotational position θr. The inductances Lc and Lb when the C-phase and B-phase windings are energized generally have the CB phase characteristics of FIG. The inductances Lb and La when the B-phase and A-phase windings are energized generally have the B-A characteristic of FIG. Each has a characteristic having a relative phase difference of 60 °. Since the period of each characteristic is reluctance, the period is 180 degrees in electrical angle.

  Each of the interlinkage magnetic flux characteristics in FIG. 114 and the inductance characteristics in FIG. 115 is a characteristic when current of the same magnitude is applied to two windings, and is only one of the representative characteristics. In addition to this, there are many other characteristics such as a characteristic when only one winding is energized and a characteristic when a current is energized through three windings. Both characteristics can be used as information for estimating the rotor rotational position θr from the values of the winding voltage and current. In FIG. 115, the rotation change rate (dL / dθr) of each inductance value, that is, the gradient of each characteristic in FIG. 115 can also be used for specifying the rotor rotational position θr.

  One specific method for estimating the rotor rotational position θr from the inductance characteristics of FIG. 115 is to measure the inductance of the three windings at an arbitrary rotor rotational position θr by the method described above. 115, the rotor rotation position θr from 0 ° to 180 ° can be identified. However, at very specific rotation positions, the same inductance may be obtained at two rotation positions. When it is necessary to detect such a specific rotor rotational position, the rotational change rate (dL / dθr) of the inductance value or the inductance under other conditions can be added as position detection information.

  Conversely, various simplified methods such as estimating the rotor rotational position θr from the two inductance characteristics are also conceivable. Further, for example, considering an application used between 500 rpm and 2000 rpm, even if the rotor rotational position θr cannot be detected at a specific rotational position, the rotor rotational position θr is determined from the past history. Since it can be estimated, there is almost no problem in motor speed control and current / voltage control.

  54. Various motors according to the present invention, not limited to the motor according to the present invention shown in FIG. 54, exhibit magnetic characteristics peculiar to each. Therefore, the rotor rotation position can be determined by measuring the voltage and current of the motor winding by each method. Can do. In the case of the motor shown in FIG. 9, the characteristics corresponding to those in FIGS. 114 and 115 are further simplified. The motor of the present invention is mainly composed of a soft magnetic material, has a salient pole shape, and the motor itself has characteristics similar to a magnetoresistive change type resolver position detector. Various position detection techniques that have been used in the instrument can be applied and used.

  In addition, a specific method for measuring the inductance is a so-called disturbance injection method. In addition to the control current and voltage of the motor, a pulsed voltage or current for measurement is added, and the current and voltage values at that time are added. 115 can be obtained from FIG. 115, and the rotor rotational position θr can be detected. Alternatively, in order to detect the rotor rotational position θr, a current or voltage having a frequency fx different from the control frequency of the motor current is superimposed on the current and voltage for driving the motor, and a signal having the frequency fx is selected from the motor winding. Thus, the rotor rotational position θr can be detected by detecting the characteristic as shown in FIG. Alternatively, the rotor rotational position θr can be measured using the voltage and current values for generating the motor torque.

  The method of estimating the rotational speed ωr of the rotor can be calculated from the time change rate (dωr / dt) of the rotational position obtained by the above method. As another method, as shown in FIG. 114, when two currents are constant, the rate of change in rotation (dφ / dθr) in a section where the flux linkage changes at a certain gradient at a certain winding current value. Therefore, the rotor rotational speed ωr can be detected using the winding voltage Vs in this section. That is, the value ks of the flux linkage rotation rate change rate (dφ / dθr) when the winding current is a certain current value Is is unique to the motor and is known, so that the following equations (35) and (36) The rotor rotational speed ωr can be detected.

Vs = Nw × (dφ / dt)
= Nw × (dφ / dθr) × (dθr / dt) (35)
∴ ωr = dθr / dt
= Vs / Nw / (dφ / dθr)
= Vs / (Nw × ks) (36)
In the motors shown in FIGS. 9 and 10, a specific method for detecting the rotational position and rotational speed of the rotor is to apply a constant current that provides a certain magnetic flux density to all three-phase windings. This is a method of detecting by rotating the rotor and measuring the voltage of each phase winding at that time. At this time, since a substantially rectangular voltage is generated in each winding, the speed and position of the rotor can be measured by measuring the timing at which the magnitude and polarity of the voltage are switched. Since the magnitude of the voltage is proportional to the magnetic flux density and the rotational speed, it can be used for detecting the rotational speed. Since the voltage switching point is a position where both ends of the rotor magnetic pole in the circumferential direction are opposed to both ends of the stator magnetic pole in the circumferential direction, it can be estimated at which position the rotor is located at the time of switching of the voltage polarity.

  Using these relationships, the rotor rotational position and the rotor rotational speed can be detected during operation of the motor by detecting the signal component in a state where the motor drive current and the position detection current are superimposed. Can do. In addition, the rotor rotational position and voltage characteristics when current is applied to one line of the three-phase windings, and the rotor rotational position and voltage characteristics when current is applied to the two lines also have unique characteristics. So it can be used.

  As described above, using the rotational position θr of the rotor and the speed information obtained from the voltage and current of the motor winding, torque control or speed control of the motor of the present invention is performed by the control device as shown in FIGS. Alternatively, position control or the like can be performed. Since the position detector is not used, the position detection sensor, the wiring for the position detector, the connector and the like are unnecessary, and the reliability of the system can be improved. The cost can be reduced and the size can often be reduced.

  As mentioned above, although the example of the various form regarding this invention was demonstrated, this invention can be variously deformed and is included in this invention. For example, there is no restriction on the number of poles of the motor, and the present invention can be applied to rotors having various structures and various shapes. Various torque ripple reduction techniques can also be applied to the motor of the present invention. For example, a method of smoothing the shape of the stator magnetic pole and rotor magnetic pole in the circumferential direction, a method of smoothing in the radial direction, a method of moving some rotor magnetic poles in the circumferential direction and canceling torque ripple components, etc. There is.

  Various forms of motors are possible, expressed as an air gap shape between the stator and the rotor, and an inner rotor type motor, outer rotor type motor, or air gap shape whose air gap shape is cylindrical. It can also be transformed into a disk-shaped axial gap type motor or the like. It can also be transformed into a linear motor. In addition, a motor shape in which the air gap shape is a slightly tapered shape from the cylindrical shape is also possible. In addition, a plurality of motors including the motor of the present invention can be combined and manufactured. Moreover, the structure which abbreviate | omitted and a part of this invention motor was also possible.

  In addition to using a normal silicon steel plate as the soft magnetic material, it is possible to use an amorphous magnetic steel plate, a powder magnetic core formed by compression molding powdered soft iron, and the like. In particular, in a small motor, a three-dimensional shaped part can be formed by punching, bending, and forging a magnetic steel sheet to form a part of the above-described motor of the present invention.

  In the motor of the present invention, the rotor is particularly robust, and high output using high-speed rotation is possible. With regard to this speeding up, windage loss of the motor may be a problem, and measures for reducing windage loss such as making the outer periphery of the rotor close to a circle or making the inner periphery on the stator side circular are effective. Further, it is also effective to eliminate the rotor axial space of the stator and rotor. There is also an effect of noise reduction.

  As for the control device of the present invention, the device mainly controlling the three-phase unidirectional current has been described. However, in the case of a motor having four or more phases, the control device having four or more phases should be technically expanded. Can be realized. As for the power control element, the case of a conventional power transistor has been described. However, a power element such as a thyristor or a power element to which a new technology such as SiC or GaN is applied can be used. As for the current waveform to be applied to the motor, an example in which the current of each phase has a specific shape has been described, but it is also possible to control with a current of various waveforms such as control with a sine wave current. Even for these variously modified motors, the modified technology intended for the motor of the present invention is included in the present invention.

  The motor system comprising the motor of the present invention and its driving device can be reduced in size, weight, and efficiency at low cost. Therefore, it can be effectively used for devices that cope with environmental issues and resource issues such as electric vehicles and hybrid vehicles. Similarly, application to industrial equipment and home appliances is also possible.

  Specifically, a wide range of motors of the present invention can be driven without using expensive rare earth permanent magnets, are low in cost, and do not have resource problems such as rare earth metals, which have been a problem in recent years. The control device can simultaneously supply power through a plurality of paths by combining with the motor of the present invention, and the cost, weight, etc. per unit output of the control device can be greatly reduced.

A01, A02, A03, A04, A05, A06 Stator magnetic pole A07, A08, A09, A0A, A0B, A0C Slot A0D, A0G A phase winding A0F, A0J B phase winding A0H, A0E C phase winding A0L A phase winding Coil end of wire A0M Coil end of B phase winding A0N Coil end of C phase winding A0P Magnetic flux excited by A phase current and C phase current A0K Rotor magnetic pole A0Z Rotor shaft θr Rotor rotation position 117, 118, 119, 11A , 11B, 11C Stator magnetic pole 117, 114 A phase winding 113, 116 B phase winding 115, 112 C phase winding 11D Space 11E Rotor magnetic pole

Claims (15)

Arbitrarily defined M stator poles of the stator between 360 ° in electrical angle;
Winding of each phase wound at an electrical angle of approximately 180 ° pitch in each slot between the magnetic poles of the stator,
K rotor magnetic poles that are arbitrarily determined by different number from the M rotors between 360 ° in electrical angle;
A DC power source PS1 for supplying a DC voltage VM;
DC power supply PS2 that supplies a regenerative voltage VH that is a voltage greater than the voltage VM;
Current control means for each phase for energizing the windings of each phase by increasing the current in one direction using the DC power source PS1 and decreasing the current in one direction using the DC power source PS2.
One end of each phase winding is connected to the DC power supply PS1, the other end of the winding is connected to the anode of a diode, the cathode of the diode is connected to the DC power supply PS2,
The current energization path from the DC power supplies PS1 and PS2 to the windings of each phase has three or more parallel energization paths, and two or more current energization paths are simultaneously supplied in parallel by the current control means. A motor characterized by that. Here, M is an integer of 6 or more, and K is an integer of 2 or more. M first group of stator poles arbitrarily defined in the stator between 360 ° in electrical angle;
A second group of M stator poles arranged on the inner diameter side of the first group of stator poles, and having the same phase stator poles arranged at the same angle in the circumferential direction;
The first group of windings and the second group of windings are so-called toroidal windings so as to go around the first group of back yokes and the second group of back yokes, Windings of each phase of the first group wound around each slot between the magnetic poles of the first group of stators at an electrical angle of approximately 180 °;
Windings of each phase of the second group wound around each slot between the magnetic poles of the stator of the second group at an electrical angle of approximately 180 °;
K first group of rotors which are arranged on the outer diameter side opposite to the stator magnetic poles of the first group, and are arbitrarily determined with an electric angle of 360 ° different from the M number of rotors. Magnetic poles,
The rotor magnetic poles are arranged on the outer diameter side so as to face the second group of stator magnetic poles, and include K second group rotor magnetic poles of the rotor at an electrical angle of 360 °. The motor described in.   In the magnetic pole shape of each stator, the shape of the pair of stator magnetic poles facing the rotor is an uneven shape, and there are two or more convex portions facing the rotor. The motor according to claim 1, wherein the pitch is substantially the same as the pitch of the uneven shape of the pair of stator magnetic poles.   The motor according to claim 1, wherein a field winding is added to each stator magnetic pole.   The motor according to claim 1, further comprising a permanent magnet on a surface of each stator magnetic pole facing the rotor.   The motor according to claim 1, further comprising a permanent magnet inside a soft magnetic body of each stator magnetic pole.   The value of M, which is the number of magnetic poles of each stator magnetic pole between electrical angles of 360 °, is 6, and the value of K, which is the number of magnetic poles of each rotor magnetic pole, is 2. motor.   The value of M, which is the number of magnetic poles of each stator magnetic pole between electrical angles of 360 °, is 6, and the value of K, which is the number of magnetic poles of each rotor magnetic pole, is 4. motor. In the shape of each stator magnetic pole facing the rotor, one of the rotor axial directions is large in the circumferential direction and the other end of the rotor axial direction is in the circumferential direction so that the rotor axial shape of each stator magnetic pole is trapezoidal. With a small shape DIJ,
The shape of the stator magnetic pole adjacent to the circumferential direction in the rotor axial direction is a shape DIK obtained by inverting the shape DIJ in the rotor axial direction.
In the circumferential direction, the stator magnetic pole shapes DIJ and DIK are arranged alternately,
The size of the slot opening between the stator pole of the shape DIJ and the stator pole of the shape DIK can be increased by relatively separating the position in the rotor axial direction between the area center of the shape DIJ and the area center of the shape DIK. The motor according to claim 1, wherein the motor is configured.   The motor according to claim 1, wherein a soft magnetic material and a permanent magnet are added to the end of each stator magnetic pole in the rotor axial direction.   The motor according to claim 1, wherein an excitation winding for exciting a soft magnetic material and a magnetic flux is added to each stator magnetic pole or each rotor magnetic pole end in the rotor axial direction.   The motor according to claim 1, wherein a permanent magnet is disposed in the vicinity of a rotor side end of each stator slot.   The motor according to claim 1, wherein the motor has a structure in which an intermediate tap of the total number of turns can be taken out on both sides of the winding of each stator, and includes means for switching a connection destination of the winding.   2. The motor according to claim 1, further comprising: a demagnetizing unit that reduces a magnetized state of the permanent magnet of the motor; and a magnetizing unit that increases the magnetized state of the permanent magnet of the motor. 2. The motor according to claim 1, wherein, in the current control of each winding, a voltage signal is generated and the current is controlled using the flux linkage information of each winding determined by the current flowing through each winding and the rotor rotational position.

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