JP2011061956A - Device for controlling electric motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve controllability by suppressing a current detection error by avoiding an effect of switching noise. <P>SOLUTION: A phase current of an electric motor and a rotation angle of a rotor of the electric motor are acquired at the timing of a prescribed phase of a PWM carrier signal, and a voltage command is operated on the basis of a difference between a phase current command and the phase current, and the rotation angle. Then, a duty ratio of PWM control is operated on the basis of the voltage command output at the timing of the prescribed phase in a succeeding carrier frequency of the PWM carrier signal, and the PWM carrier signal. Here, a phase of the acquisition timing of the phase current and the rotation angle is set so as to be variable on the basis of the timing of the switching operation of an inverter. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a control device for an electric motor.

  2. Description of the Related Art Conventionally, there has been known a motor control device that controls an output torque of an electric motor by controlling a switching operation of an inverter in accordance with a duty command for PWM control. By the way, in the PWM control, switching noise of the semiconductor switch is superimposed on the current. Therefore, Patent Document 1 discloses a motor control device that improves controllability by performing current detection while avoiding the influence of switching noise. Is disclosed. Specifically, the control device performs current control with a carrier signal generator that outputs a triangular carrier signal, a current detection trigger generator that outputs a current detection trigger, a current detector that detects a motor current, and the like. A current controller and a PWM controller that outputs a PWM signal are provided. Here, the current detection trigger generator outputs the current detection trigger from the peaks and valleys of the carrier signal by the current detection and current control processing time.

JP 2008-131712 A

  However, according to the technique disclosed in Patent Document 1, since the current detection is performed only at a predetermined timing, the influence of switching noise may not always be avoided.

  The present invention has been made in view of such circumstances, and an object thereof is to improve the controllability by suppressing the current detection error by avoiding the influence of switching noise.

  In order to solve this problem, the present invention acquires a phase current of the motor and a rotation angle of the rotor of the motor at a predetermined phase timing of the PWM carrier signal, and a phase current command corresponding to the torque command of the motor. And a voltage command is calculated based on the difference between the acquired phase current and the acquired rotation angle. Then, a duty command for PWM control is calculated based on a voltage command and a PWM carrier signal output at a predetermined phase timing in the next carrier cycle of the PWM carrier signal. Here, the phase of the acquisition timing of the phase current and the rotation angle is variably set based on the timing of the switching operation of the inverter.

  According to the present invention, by making the phase of the acquisition timing of the phase current and the rotation angle variable, current detection can be performed at a timing that avoids the influence of switching noise, thereby suppressing a current detection error. Thus, controllability can be improved.

Explanatory drawing which shows typically the whole structure of the motor control system concerning 1st Embodiment. Illustration of noise remaining time Tn Explanatory diagram showing three-phase voltage commands vu * to vw * corresponding to the rotation angle Explanatory drawing which shows three-phase voltage command vu * -vw * and the PWM pulse waveform of each phase in the timing A shown in FIG. Explanatory drawing which shows the ON / OFF timing of the PWM pulse of each phase with respect to the rotation angle θ The flowchart which shows the determination procedure of the acquisition timing Tg concerning a 1st method. Explanatory drawing which shows the change of acquisition timing Tg Explanatory drawing which shows the ON / OFF timing of the PWM pulse of each phase with respect to the rotation angle θ The flowchart which shows the determination procedure of the acquisition timing Tg concerning a 2nd method. Explanatory drawing which shows the change of acquisition timing Tg Explanatory drawing which shows typically the whole structure of the motor control system concerning 2nd Embodiment. Explanatory drawing which shows acquisition timing Tg on condition that the calculation time is maximum for each modulation factor Explanatory drawing which shows acquisition timing Tg on condition of the maximum calculation time for every voltage phase Explanatory drawing which shows the modification of the motor control system concerning 2nd Embodiment. Explanatory drawing which shows acquisition timing Tg on condition of the calculation time maximum

(First embodiment)
FIG. 1 is an explanatory diagram schematically showing the overall configuration of the motor control system according to the first embodiment. The motor control system according to the present embodiment is a motor control system that controls a drive motor of an electric vehicle (for example, an electric vehicle). This motor control system is mainly composed of a motor 10, an inverter 20, and a control unit 30.

  The motor 10 is mainly composed of a rotor and a stator, and has a plurality of phase windings (in this embodiment, a U-phase winding, a V-phase winding, and a W-phase) that are star-connected around a neutral point. This is a permanent magnet synchronous motor in which three phase windings comprising windings are wound around a stator. The motor 10 is driven by an interaction between a magnetic field generated by supplying three-phase AC power to each phase winding from an inverter 20 described later, and a magnetic field generated by a permanent magnet of the rotor. The rotor and the output shaft connected to the rotor rotate. The output shaft of the motor 10 is connected to an automatic transmission of an electric vehicle, for example.

  The inverter 20 is connected to a power source (not shown), and is power conversion means that converts DC power from the power source into AC power and supplies the AC power to the motor 10. The AC power is generated corresponding to each phase of the motor 10, and the AC power of each phase generated by the inverter 20 is supplied to the motor 10.

  The inverter 20 is a circuit in which an upper arm connected to the bus on the positive side of the power source and a lower arm connected to the bus on the negative side of the power source are connected in series to each phase of the U phase, the V phase, and the W phase. In correspondence with. Each arm of the circuit corresponding to each phase is mainly composed of a semiconductor switch capable of controlling conduction in one direction (for example, a self-extinguishing switching element such as a transistor such as an IGBT). The reflux diodes are connected in reverse parallel.

  The on / off state of each arm, that is, the on / off state (switching operation) of the semiconductor switch is controlled through a PWM command (a duty command for PWM control) output from the control unit 30. The semiconductor switches constituting the individual arms are turned on when turned on by a PWM command from the control unit 30 and turned off (cut off) when turned off.

  The control unit 30 is a motor control device that controls the output torque of the motor 10 by controlling the switching operation of the inverter 20. As the control unit 30, a microcomputer mainly composed of CPU, ROM, RAM, and I / O interface can be used. The control unit 30 performs a calculation for controlling the inverter 20 in accordance with a control program stored in the ROM. Then, the control unit 30 outputs a control signal (PWM command) calculated by this calculation to the inverter 20.

  The control unit 30 can acquire (read) information detected by various sensors. The position sensor (for example, resolver) is attached to the motor 10 and detects position information indicating the rotor position of the motor 10, specifically, the rotor phase (electrical angle), that is, the rotation angle θ of the rotor. The current sensor detects actual currents iu, iv, iw of each phase in the motor 10. Information acquisition from each sensor by the control unit 30 is performed at a timing according to a calculation result of an acquisition timing calculation unit 38, which will be described later, serving as one function of the control unit 30.

  The control unit 30 drives the inverter 20 by a driving method such as PWM wave voltage driving. The PWM wave voltage drive generates a PWM pulse from a DC voltage by PWM control and applies it to the motor 10. Specifically, an equivalent sine wave AC voltage is calculated by calculating a duty command based on a PWM carrier signal. Is applied to the motor 10.

  When the control unit 30 grasps this functionally, the current calculation unit 31, the voltage calculation unit 32, the three-phase conversion unit 33, the PWM conversion unit 34, the dead time compensation unit 35, and the dq axis conversion unit. 36, a differential calculation unit 37, and an acquisition timing calculation unit 38.

  The current calculation unit 31 calculates the d-axis and q-axis current commands id * and iq * corresponding to the torque command Te * based on the torque command Te * of the motor 10 given from the outside and the motor rotation speed ω, respectively. To do. By taking into account the characteristics of the motor 10 and the like, the relationship between the torque command value Te * and the motor rotational speed ω and the d-axis and q-axis current commands id * and iq * is obtained in advance through experiments and simulations. The current calculation unit 31 holds a map that defines this relationship. The current calculation unit 31 calculates the d-axis and q-axis current commands id * and iq * with reference to the map. Here, the differential calculation unit 37 calculates the rotor angular velocity (electrical angle), that is, the motor rotation speed ω, by time-differentiating the rotation angle θ obtained from the position sensor. As the motor rotation speed ω necessary for the calculation of the d-axis and q-axis current commands id * and iq *, the calculation result of the differential calculation unit 37 can be used.

The d-axis and q-axis current commands id * and iq * output from the current calculation unit 31 are subtracted from the d-axis and q-axis currents id and iq corresponding to the actual current of the motor 10, respectively. q-axis current deviations are respectively calculated. The d-axis and q-axis current deviations are output to the voltage calculation unit 32. Here, the d-axis and q-axis currents corresponding to the actual current of the motor 10 are calculated by the dq-axis conversion unit 36 converting the coordinates of the three-phase actual currents iu to iw based on the rotation angle θ. (See Formula 1).

For example, the voltage calculation unit 32 calculates the d-axis and q-axis voltage commands vd * and vq * such that the current deviations of the d-axis and the q-axis are each 0 using PI control (see Formula 2). .

  Here, Kpd is a d-axis proportional gain, Kid is a d-axis integral gain, Kpq is a q-axis proportional gain, and Kiq is a q-axis integral gain. S is a Laplace operator. The calculated d-axis and q-axis voltage commands vd * and vq * are output to the three-phase converter 33, respectively.

The three-phase conversion unit 33 converts the d-axis and q-axis voltage commands vd *, vq * into the voltage commands vu *, vv *, vw * corresponding to the three phases after referring to the rotation angle θ ( (See Formula 3). The voltage commands vu * to vw * for each phase are output to the PWM converter 34, respectively.

  As described above, in the present embodiment, the current control calculation for calculating the three-phase voltage commands vu * to vw * includes the current calculation unit 31, the voltage calculation unit 32, the three-phase conversion unit 33, and the dq axis. This is performed by the conversion unit 36 and the differential calculation unit 37 (first calculation means). That is, the phase currents iu to iw of the motor 10 are acquired by the dq axis conversion unit 36, and the rotation angle θ of the rotor is acquired by the three-phase conversion unit 33 and the differential calculation unit 37, respectively. The phase current command corresponding to the motor torque command Te * (corresponding to the d-axis and q-axis current commands id * and iq * in this embodiment) and the acquired phase currents iu to iw (in this embodiment). , Corresponding to the d-axis and q-axis currents id, iq), and the voltage commands vu * to vw * are calculated based on the obtained rotation angle θ.

Next, the PWM converter 34 calculates the PWM commands tu, tv, tw corresponding to the respective phases based on the voltage commands vu * to vw * of the respective phases and the voltage vdc of the power supply (see Formula 4). ). The PWM commands tu to tw are duty commands for PWM control corresponding to the three-phase voltage commands vu * to vw *.

  Here, T0 is the period (s) of the PWM carrier signal. Also, tu is a u-phase pulse width (s) that is a u-phase PWM command, tv is a v-phase pulse width (s) that is a v-phase PWM command, and tw is a w-phase pulse width that is a w-phase PWM command ( s). The PWM commands tu to tw corresponding to each phase are output to the dead time compensation unit 35.

  The dead time compensator 35 calculates a compensation voltage determined based on the three-phase actual currents iu to iw, and adds the calculated compensation voltage to the PWM commands tu to tw corresponding to the respective phases. Each phase PWM command tu ', tv', tw 'is calculated. Here, by acquiring the relationship between the three-phase actual currents iu to iw and the compensation value in advance through experiments and simulations, the dead time compensation unit 35 holds a map that defines this relationship. The dead time compensation unit 35 calculates a compensation value with reference to the map. This dead time compensation prevents the upper arm and the lower arm from being simultaneously turned on and short-circuited in each phase, so that the upper arm and the lower arm are turned off at the same time when switching on / off. (So-called dead time) is provided.

  The final PWM command tu ′ to tw ′ of each phase that is calculated is output to the inverter 20, and the inverter 20 is driven in accordance with the final PWM command tu ′ to tw ′ of each phase. The three-phase voltages vu, vv, vw are applied to the motor 10 through the motor 10. As described above, in the present embodiment, based on the voltage commands vu * to vw * and the PWM carrier signal output by the PWM conversion unit 34 and the dead time compensation unit 35, the duty command of the PWM control, that is, the PWM command Tu to tw (corresponding to PWM commands tu ′ to tw ′ in this embodiment) are calculated (second calculating means).

  Further, the PWM commands tu ′ to tw ′ for each phase output from the dead time compensation unit 35 are also output to the acquisition timing calculation unit 38. The acquisition timing calculation unit 38 calculates a timing (hereinafter referred to as “acquisition timing”) Tg for acquiring a sensor output based on a noise remaining time Tn, which will be described later, and PWM commands tu ′ to tw ′ of each phase. 1 computing means). Sensor outputs from the position sensor and the current sensor are acquired according to the acquisition timing Tg calculated by the acquisition timing calculation unit 38. One of the features of this embodiment is a method for determining the acquisition timing Tg by the acquisition timing calculation unit 38, and the contents thereof will be described below.

  FIG. 2 is an explanatory diagram of the noise remaining time Tn. In the figure, (a) shows the transition of a certain phase PWM pulse (PWM pulse waveform), and (b) shows the output of the current sensor. As shown in the figure, noise resulting from the switching operation (hereinafter referred to as “switching noise”) is superimposed on the sensor output of the current sensor by the switching operation according to the PWM command. The noise remaining time Tn is a time that is affected by switching noise in the sensor output of the current sensor. The noise remaining time Tn is obtained in an optimum value by considering the switching characteristics of the inverter 20 through experiments and simulations.

  FIG. 3 is an explanatory diagram showing three-phase voltage commands vu * to vw * (V) corresponding to the rotation angle (electrical angle (°)) θ. FIG. 4 is an explanatory diagram showing three-phase voltage commands vu * to vw * at timing A shown in FIG. 3 (see (a)) and PWM pulse waveforms for each phase (see (b)). . In the figure, the hatched area indicates the noise remaining time Tn (hereinafter, the same applies to the drawings described later). FIG. 5 is a diagram corresponding to FIG. 4 and is an explanatory diagram showing the on / off timing of the PWM pulse of each phase with respect to the rotation angle θ. In the figure, various thick lines indicate the PWM pulse on / off timing from the peak timing or valley timing of the PWM carrier signal.

  In FIG. 5, a region corresponding to non-hatching indicates timing at which sensor output can be acquired without being affected by switching noise. As can be seen from FIG. 5, there are various timings for acquiring the sensor output without being affected by the switching noise at each rotation angle θ. In the present embodiment, an acquisition timing determination method will be described by exemplifying two methods.

  First, as a first method, the acquisition timing Tg is determined so as to maximize the control calculation time. In general, the control unit 30 acquires a sensor output and performs a current control calculation (a series of calculations up to the voltage commands vu * to vw * described above) at the return timing of the PWM carrier signal, and performs the next corresponding return. Voltage commands vu * to vw * are output at the timing (turnback timing in the next carrier cycle). That is, if the calculation is not completed before the next turn-back timing, the time from when the sensor output is acquired to when the voltage commands vu * to vw * are output is further doubled. Therefore, it is possible to lengthen the control calculation time by setting the sensor output acquisition timing as early as possible.

  FIG. 6 is a flowchart showing a procedure for determining the acquisition timing Tg according to the first method. The process shown in this flowchart is called at a predetermined cycle and executed by the acquisition timing calculation unit 38.

  First, in step 10 (S10), the acquisition timing calculation unit 38 compares the on / off timings of the PWM pulses corresponding to the voltage commands vu * to vw * output last time, and the next PWM carrier signal folding timing (first timing). The phase whose on / off timing is closest to the timing T1) is selected. Here, the on / off timing of each phase is calculated from the PWM command (pulse width) tu ′ to tw ′ of each phase, and the first timing T1 is set as the reference timing of the time axis.

  In step 11 (S11), the acquisition timing calculation unit 38 determines whether or not the value obtained by adding the remaining noise time Tn to the previous on / off timing Tonoff (p) related to the selected phase is smaller than zero. If an affirmative determination is made in step 11, the process proceeds to step 12 (S12). On the other hand, if a negative determination is made in step 11, the process proceeds to step 13 (S13).

  In step 12, the acquisition timing calculation unit 38 determines the first timing T1 as the acquisition timing Tg.

  In step 13, the acquisition timing calculation unit 38 compares the PWM pulse on / off timing corresponding to the voltage commands vu * to vw * output this time, and turns on / off the next PWM carrier signal return timing (first timing T1). Select the phase with the closest timing.

  In step 14 (S14), the acquisition timing calculation unit 38 obtains a value obtained by subtracting the previous on / off timing Tonoff (p) and the remaining noise time Tn from the current on / off timing Tonoff (t) related to the selected phase. To determine whether it is greater than zero. If an affirmative determination is made in step 14, the process proceeds to step 15 (S15). On the other hand, if a negative determination is made in step 14, the process proceeds to step 16 (S16).

  In step 16, the acquisition timing calculation unit 38 determines a timing obtained by adding the remaining noise time Tn to the previous on / off timing Tonoff (p) as the acquisition timing Tg.

  On the other hand, in step 17, the acquisition timing calculation unit 38 determines a timing obtained by adding the remaining noise time Tn to the current on / off timing Tonoff (t) as the acquisition timing Tg.

  According to such a method, the phase of the acquisition timing of the phase currents iu to iw and the rotation angle θ is variably set based on the timing of the switching operation of the inverter 20. Specifically, the phase of the acquisition timing Tg is set based on the three-phase voltage commands vu * to vw * applied to the motor 10 and the noise remaining time Tn. In particular, according to this method, from the viewpoint of maximizing the control calculation time, the phase of the acquisition timing Tg is advanced so as to approach the return timing of the PWM carrier signal.

  FIG. 7 is an explanatory diagram showing a change in the acquisition timing Tg, in which the horizontal axis is the rotation angle (electrical angle (°)) and the vertical axis is the time (μs) from the valley timing. In this method, the maximum control calculation time can be secured under the condition that the return timing of the PWM carrier signal is the acquisition timing Tg. As shown in FIG. 5, when the remaining noise time Tg overlaps with the turnaround timing, the sensor output can be acquired without being affected by the switching noise by shifting the timing by the remaining noise time Tn. . This is to improve the controllability by suppressing the current detection error. In addition, as shown in FIG. 8, when the voltage amplitude (modulation rate) is large, the on / off timing approaches with the turnaround timing interposed therebetween. For this reason, as described above, if only the noise remaining time Tn is shifted, it may be influenced by subsequent switching noise in terms of timing. Therefore, as shown in FIG. 7B, the acquisition timing Tg is set at a timing at which the influence of noise is suppressed by changing the acquisition timing Tg of the sensor output according to the on / off timing after the turn-back timing. be able to. This is to improve the controllability by suppressing the current detection error.

  Next, the second method is a method of determining the acquisition timing Tg so that the controllability is improved, that is, the dead time is minimized. Contrary to the first method, this method reduces the dead time of control by performing from the sensor output acquisition timing Tg to the output of the voltage commands vu * to vw * in the shortest time. This will improve the performance.

  FIG. 9 is a flowchart showing a procedure for determining the acquisition timing Tg according to the second method. The process shown in this flowchart is called at a predetermined cycle and executed by the acquisition timing calculation unit 38.

  First, in step 20 (S20), the acquisition timing calculation unit 38 compares the on / off timings of the PWM pulses corresponding to the voltage commands vu * to vw * output this time. Thereby, the acquisition timing calculation unit 38 selects the phase with the closest on / off timing to the timing (second timing T2) obtained by subtracting the control calculation time (predetermined time required for the current control calculation) from the return timing of the PWM carrier signal. To do. Here, the on / off timing of each phase is calculated from the PWM command (pulse width) tu ′ to tw ′ of each phase, and the second timing T2 is set as the reference timing of the time axis.

  In step 21 (S21), the acquisition timing calculation unit 38 determines whether or not the value obtained by subtracting the second timing T2 from the current on / off timing Tonoff (t) related to the selected phase is smaller than zero. If an affirmative determination is made in step 21, the process proceeds to step 23 (S23). On the other hand, if a positive determination is made in step 21, the process proceeds to step 24 (S24).

  In step 23, the acquisition timing calculation unit 38 selects the next phase closest to the second timing T2, and then performs the process of step 21.

  In step 24, the acquisition timing calculation unit 38 determines whether or not a value obtained by subtracting the second timing T2 from the current on / off timing Tonoff (t) related to the selected phase is greater than the noise remaining time Tn. If an affirmative determination is made in step 24, the process proceeds to step 25 (S25). On the other hand, if an affirmative determination is made in step 24, the process proceeds to step 26 (S26).

  In step 25, the acquisition timing calculation unit 38 determines the second timing T2 as the acquisition timing Tg.

  In step 26, the acquisition timing calculation unit 38 determines the current on / off timing Tonoff (t) related to the selected phase as the acquisition timing Tg.

  According to such a method, the phase of the acquisition timing of the phase currents iu to iw and the rotation angle θ is variably set based on the timing of the switching operation of the inverter 20. Specifically, the phase of the acquisition timing Tg is set based on the three-phase voltage commands vu * to vw * applied to the motor 10 and the noise remaining time Tn. In particular, according to this method, from the viewpoint of improving controllability, the phase of the acquisition timing Tg is delayed so as to approach the output timing of the voltage command in the next carrier cycle.

  FIG. 10 is an explanatory diagram showing changes in the acquisition timing Tg, in which the horizontal axis represents the rotation angle (electrical angle (°)), and the vertical axis represents the time (μs) from the valley timing. According to the second method, it is possible to reduce the influence of the error of the sensor output due to the switching noise while improving the controllability.

  Note that the two methods described above may be executed independently or may be executed while switching according to system requirements. In addition to the above-described conditions, the sensor output acquisition timing Tg may be determined under conditions that are not affected by switching noise. In the above description, it is assumed that the calculated PWM pulse and the on / off timing of the switching operation coincide with each other. However, there are cases where the timings of the two coincide with each other due to various factors. In this case, it is preferable to calculate the sensor output acquisition timing Tg so as to avoid the influence of switching noise in consideration of the deviation.

(Second Embodiment)
FIG. 11 is an explanatory diagram schematically showing the overall configuration of the motor control system according to the second embodiment. The motor control system according to the present embodiment is different from that of the first embodiment in the calculation method of the acquisition timing calculation unit 38 configuring the control unit 30. Note that description of points that are the same as in the first embodiment will be omitted, and hereinafter, differences will be mainly described.

In the present embodiment, the acquisition timing calculation unit 38 acquires the sensor output acquisition timing based on the d-axis and q-axis voltage commands vd * and vq *, the rotation angle (electrical angle) θ, and the remaining noise time Tn. Tg is calculated. As shown in FIG. 7 and the like, the sensor output detection timing Tg periodically changes with respect to the rotation angle θ. FIG. 12 is a plot of the acquisition timing Tg when the maximum calculation time is used for each modulation factor (voltage amplitude Va (see Formula 5)). In FIG. 12, the horizontal axis indicates the rotation angle ( Electrical angle (°)), and the vertical axis represents time (μs) from the valley timing.

Further, FIG. 13 is a plot of the acquisition timing Tg when the maximum calculation time is used for each voltage phase α (see Equation 6). In FIG. 13, the horizontal axis represents the rotation angle (electrical angle (° )), The vertical axis represents time (μs) from the valley timing. As can be seen from the figure, the sensor output acquisition timing Tg changes according to the voltage amplitude Va and the voltage phase α.

  Therefore, in consideration of the remaining noise time Tn, the relationship between the acquisition timing Tg with respect to the d-axis and q-axis voltage commands vd *, vq * (voltage amplitude Va and voltage phase α) and the rotation angle θ is tested and simulated. Get through. And the acquisition timing calculating part 38 hold | maintains the numerical formula or map which reflected the said relationship, and calculates the acquisition timing Tg from the said numerical formula or map.

  Thus, according to the present embodiment, the d-axis and q-axis voltage commands vd * and vq * for calculating the three-phase voltage commands vu * to vw * to be applied to the motor 10, the rotation angle θ of the rotor, and Based on the remaining noise time Tn, the phase of the acquisition timing Tg is set. As a result, when the d-axis and q-axis voltage commands vd * and vq * do not change, the sensor output acquisition timing Tg can be calculated only by the rotation angle θ, thereby avoiding the influence of switching noise with a small calculation. can do.

  As shown in FIG. 14, the motor control system of the second embodiment may further include a correction unit 39. The correction unit 39 adds a value corresponding to the compensation voltage from the dead time compensation unit 35 to the d-axis and q-axis voltage commands vd * and vq * input to the acquisition timing calculation unit 38. Thus, the acquisition timing Tg can be calculated using the d-axis and q-axis voltage commands vd * and vq * taking into account the value corresponding to the compensation voltage by the dead time compensation unit 35.

  Even in the configuration shown in FIG. 11, when calculating the mathematical expression or map used by the acquisition timing calculation unit 38, the ON / OFF timing of the switching operation and the remaining noise time Tn are calculated by increasing or decreasing the compensation voltage. Also good. Thus, the acquisition timing Tg is calculated using the d-axis and q-axis voltage commands vd * and vq * taking into account the value corresponding to the compensation voltage by the dead time compensation unit 35 without adding the correction unit 39. Can do.

  Further, as can be seen from FIG. 12 and the like, the acquisition timing Tg considering the switching noise changes in one-third cycle with respect to the rotation angle θ. There is no need to have. For example, as shown in FIG. 15, this period is 6 minutes of the rotation angle θ when using a calculation to improve the voltage utilization rate by superimposing the third harmonic on the three-phase voltage commands vu * to vw *. It is sufficient to hold a mathematical expression or map corresponding to one period of

DESCRIPTION OF SYMBOLS 10 ... Motor 20 ... Inverter 30 ... Control unit 31 ... Current calculation part 32 ... Voltage calculation part 33 ... Three phase conversion part 34 ... PWM conversion part 35 ... Dead time compensation part 36 ... dq axis conversion part 37 ... Differentiation calculation part 38 ... Acquisition timing calculation unit 39... Correction unit

Claims (5)

In the motor control device for controlling the output torque of the motor by controlling the switching operation of the inverter according to the duty command of the PWM control,
At the timing of the predetermined phase of the PWM carrier signal, the phase current of the motor and the rotation angle of the rotor of the motor are respectively acquired, and the phase current command corresponding to the torque command of the motor and the acquired phase current First computing means for computing a voltage command based on the difference and the obtained rotation angle;
Based on the voltage command and the PWM carrier signal output by the first calculation means at the timing of the predetermined phase of the PWM carrier signal in the next carrier cycle of the carrier cycle in which the phase current and the rotation angle are acquired. , Second calculating means for calculating a duty command of the PWM control,
The motor control device according to claim 1, wherein the first calculation means variably sets the phase of the phase current and the rotation angle acquisition timing based on a switching operation timing of the inverter.   The first calculation means sets the phase of the acquisition timing based on a three-phase voltage command applied to the electric motor and a remaining time of noise caused by the switching operation of the inverter. The motor control device according to claim 1.   The first calculation means includes d-axis and q-axis voltage commands for calculating a three-phase voltage command to be applied to the electric motor, a rotation angle of the rotor, and noise remaining due to the switching operation of the inverter. The motor control device according to claim 1, wherein the phase of the acquisition timing is set based on time.   4. The motor control device according to claim 2, wherein the first calculation unit advances the phase of the acquisition timing so as to approach the return timing of the PWM carrier signal. 5.   4. The motor control device according to claim 2, wherein the first calculation unit delays the phase of the acquisition timing so as to approach the output timing of the voltage command in the next carrier cycle. 5. .

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