JP2010243295A - Pulse compression device and target detection device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To achieve a pulse compression device capable of suppressing a side lobe even when using a nonlinear FM chirp transmission signal. <P>SOLUTION: A Fourier transform section 51 of a pulse compression section 15 transforms a reception signal Sr(t) on a time axis into a reception signal Sr(ω) on a frequency axis. A matched filter 52 multiplies a matched filter coefficient Ks(ω) formed of a complex conjugate of a pulse like transmission signal to the reception signal Sr(ω) and outputs a signal Sd(ω). A smoothing processing section 53 multiplies the signal Sd(ω) by a smoothing coefficient Ks(ω) for smoothing a frequency spectrum of an output of the matched filter and outputs a signal Sds(ω) after the processing of smoothing. An inverse Fourier transform section 54 transforms the signal Sds(ω) after the processing of smoothing on the frequency axis into a signal Sds(t) after the processing of smoothing on the time axis and outputs it. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a pulse compression device that performs pulse compression on a modulated pulse signal and a target detection device including the pulse compression device.

  Conventionally, there is a radar device that detects a target from a reflected signal by transmitting a pulse signal to a detection target region. Among such radar apparatuses, there is a radar apparatus using pulse compression as shown in Patent Document 1.

  In a radar apparatus using pulse compression, as shown in Patent Document 1, a pulse-shaped transmission signal using a linear FM chirp signal is transmitted, and a reception signal obtained by reflection on a target is converted into a pulse consisting of a matched filter. Pulse compression is performed by the compression circuit. As a result, the pulse-compressed signal has a main lobe with a peak shorter than the pulse width of the transmission signal on the time axis and higher than the level of the reception signal. Thereby, the distance resolution is improved while suppressing the transmission power.

JP-A-11-142507

  When pulse compression is performed using the transmission signal of the linear FM chirp signal as described above, a range (time axis) side lobe always occurs together with the main lobe. Therefore, in Patent Document 1, a delay circuit different from the pulse compression circuit is provided to cancel the range side lobe, but a circuit for canceling the range side lobe is required separately.

  Further, as a method of suppressing the range side lobe after pulse compression, there is a method of transmitting a pulsed transmission signal applied with a predetermined window function such as a Gaussian window or a Hamming window.

  However, if a window function is applied to a pulsed transmission signal in the time domain, the amplitude at both ends of the pulse waveform is smaller than when a rectangular pulsed signal is transmitted, so the total transmission power is weakened. End up. In particular, in a semiconductor radar device using a semiconductor as an oscillation element, a large power like a magnetron cannot be obtained, and thus a decrease in detection capability due to the use of a window function is a serious problem.

  In order to solve this problem, there is a method of using, as a transmission signal, a nonlinear FM chirp signal that performs window function processing on the frequency spectrum of the transmission signal.

  6A is a diagram illustrating frequency transition within the pulse width Tw of the linear FM chirp transmission signal, and FIG. 6B is a diagram illustrating frequency transition within the pulse width Tw of the nonlinear FM chirp transmission signal. is there.

As shown in FIG. 6, even if the transition frequency width Δf within the pulse width Tw is the same, the non-linear FM chirp signal is included in one pulse waveform of the transmission signal like the above-described linear FM chirp signal. The frequency transition in is not linear (FIG. 6A), and the frequency transition in the pulse waveform is non-linear (FIG. 6B). Specifically, as shown in FIG. 6B, the nonlinear FM chirp signal has a frequency component near the center frequency (f _C ) in the transition frequency width Δf included in one pulse waveform (pulse width Tw). It has a frequency characteristic in which the time ratio is high and the time ratio of frequency components in the vicinity of both end frequencies (f _L , f _H ) is low.

  An object of the present invention is to realize a pulse compression device capable of sufficiently suppressing side lobes and a target detection device including the pulse compression device when this nonlinear FM chirp signal is used.

  The present invention relates to a pulse compression apparatus that performs pulse compression using a nonlinear FM chirped pulse-shaped transmission signal whose frequency spectrum is defined by a predetermined window function. The pulse compression device according to the present invention includes a detection unit, a smoothing processing unit, a smoothing coefficient generation unit, and a pulse compression unit. A detection part calculates | requires the amplitude of the frequency spectrum of the signal obtained by autocorrelating a pulse-shaped transmission signal. The smoothing processing unit smoothes the detection signal output from the detection unit. The smoothing coefficient generation unit divides the output of the smoothing processing unit by the frequency spectrum of the signal obtained by autocorrelating the pulsed transmission signal to generate a smoothing coefficient. The pulse compression unit performs pulse compression using a reception signal with respect to the transmission signal, a matched filter coefficient based on the pulse-like transmission signal, and a smoothing coefficient.

  FIG. 7 is a diagram illustrating a frequency spectrum of a nonlinear FM chirp transmission signal having a constant amplitude. As shown in FIG. 7, when a pulse-shaped transmission signal having a constant amplitude using a non-linear FM chirp signal (hereinafter referred to as a non-linear FM chirp transmission signal) is used, the frequency spectrum of the transmission signal and the reception signal has smooth characteristics. In other words, the entire band has a characteristic with minute unevenness. This minute unevenness is considered to occur when a pulse-like transmission signal having a constant amplitude is generated by window function processing. Since such minute irregularities exist, even if the pulse compression processing is performed using the same replica as the transmission signal, the sufficient range sidelobe suppression effect expected in the window function processing cannot be obtained.

  Therefore, the present invention performs pulse compression using a smoothing coefficient for canceling out the fine irregularities of the frequency spectrum included in the transmission signal together with the matched filter coefficient. This smoothing coefficient is generated by smoothing the amplitude of the spectrum obtained by autocorrelation of the pulsed transmission signal and dividing the smoothed signal by the frequency spectrum obtained by autocorrelation of the pulsed transmission signal. To do. Thereby, the above-mentioned minute unevenness is suppressed, and a sufficient range side lobe suppression effect is realized. At this time, since the pulse compression is not a pure matched filter, the main lobe is also slightly attenuated. However, as shown in the drawing of the embodiment described later, since the attenuation amount of the main lobe is very small compared to the suppression amount of the range side lobe, the range side lobe is relatively significantly suppressed. .

  Further, the pulse compression apparatus according to the present invention performs smoothing processing by moving average processing the amplitude information of the detection signal, that is, the autocorrelation signal of the pulsed transmission signal.

  In addition, the smoothing processing unit of the pulse compression device according to the present invention performs smoothing processing so that the detection signal approaches the frequency spectrum of a predetermined window function.

  These configurations show specific examples of the spectrum smoothing process. By using these methods, the minute irregularities on the frequency spectrum as described above are suppressed.

  Further, the smoothing processing unit of the pulse compression device according to the present invention performs smoothing processing on a part of the frequency domain of the detection signal.

  In this configuration, the frequency region to which the smoothing process is applied can be set as appropriate, and the side lobe suppression amount and the main lobe attenuation amount can be optimized.

  The target detection device of the present invention includes the above-described pulse compression device, a transmission unit that generates a transmission signal, and radiates the transmission signal into the detection region, and receives a reflection signal of the transmission signal from the target. Then, an antenna that outputs a reception signal and a target detection unit that detects a target based on an output signal from the pulse compression unit are provided.

  In this configuration, by providing the above-described pulse compression device, the range side lobe of the detection signal by the pulse compression process is suppressed, so that the main lobe is easily detected and the target detection performance is improved.

  According to the present invention, side lobes can be effectively suppressed even when pulse compression is performed using a nonlinear FM chirp transmission signal. And detection capability can be improved by using such a pulse compression process.

2 is a block diagram showing a main circuit configuration of a radar apparatus 11, a block diagram showing a main circuit configuration of a pulse compression unit 15, and a block diagram of a smoothing coefficient generation unit. FIG. It is a flowchart which shows the setting method of smoothing coefficient Ks ((omega)). It is a figure which shows frequency spectrum SpTL of the autocorrelation signal of a transmission signal, and frequency spectrum SpGF of the autocorrelation signal smoothed. It is a figure which shows the frequency spectrum of signal Sds ((omega)) after a smoothing process. It is a figure which shows the simulation result which compared the pulse compression at the time of using the matched filter process and smoothing process of this invention, and the pulse compression result using only the matched filter process of a prior art example. It is a figure which shows the frequency transition within the pulse width Tw of a linear FM chirp transmission signal, and a figure which shows the frequency transition within the pulse width Tw of a non-linear FM chirp transmission signal. It is a figure which shows the frequency spectrum of the nonlinear FM chirp transmission signal with constant amplitude.

A target detection apparatus including a pulse compression apparatus according to an embodiment of the present invention will be described with reference to the drawings. Note that the target detection device shown in the present embodiment is, for example, the radar device 11 and may be another device as long as it detects a target using a nonlinear FM chirp transmission signal having a constant amplitude.
FIG. 1A is a block diagram showing a main circuit configuration of the radar apparatus 11, and FIG. 1B is a block diagram showing a main circuit configuration of the pulse compression unit 15. FIG. 1C is a block diagram illustrating a main configuration of the smoothing coefficient generation unit 500.

  The radar apparatus 11 includes a transmission unit 12, a DPX (duplexer) 13, an antenna 14, a pulse compression unit 15, and a target detection unit 16.

The transmission unit 12 includes an oscillation element such as a semiconductor or a magnetron, and transmits a non-linear FM chirp transmission signal having a pulse-like waveform at a predetermined transmission timing. Here, the non-linear FM chirp transmission signal is near the center frequency (f _C ) in the transition frequency width Δf included in one pulse waveform (pulse width Tw) as shown in FIG. The frequency component has a high time ratio, and the frequency components of the frequency components at both ends (f _L , f _H ) have a low time ratio. Such a nonlinear FM chirp transmission signal can be obtained by multiplying the frequency spectrum of the transmission signal by a window function. Examples of window functions include a Gaussian window, a Hann window, and a Hamming window.

  The non-linear FM chirp transmission signal output from the transmission unit 12 is output to the DPX 13.

  The DPX 13 transmits the nonlinear FM chirp transmission signal from the transmission unit 12 to the antenna 14. The antenna 14 radiates the supplied non-linear FM chirp transmission signal to the outside while continuing to rotate at a constant rotation speed at a predetermined rotation speed, and during a period in which the non-linear FM chirp transmission signal is not supplied, The reflected signal is received and the received signal is output to DPX 13. The DPX 13 outputs a reception signal from the antenna 14 to the pulse compression unit 15.

  As shown in FIG. 1B, the pulse compression unit 15 includes a Fourier transform unit 51, a matched filter 52, a smoothing processing unit 53, and an inverse Fourier transform unit 54. In addition, the pulse compression unit 15 includes a smoothing coefficient generation unit 500. The smoothing coefficient generation unit 500 includes an autocorrelation processing unit 501, a detection unit 502, a smoothing processing unit 503, and a smoothing coefficient calculation unit 504. The smoothing coefficient generation unit 500 may be in the pulse compression unit 15 or may be different from the pulse compression unit 15.

  The Fourier transform unit 51 sequentially acquires and stores the reception signal Sr (t) on the time axis input from the antenna 14 via the DPX 13 at every predetermined sampling timing, sequentially Fourier transforms every predetermined number of samplings, and the frequency A reception signal Sr (ω) on the axis is generated.

  The matched filter 52 performs pulse compression processing on the frequency axis by multiplying the received signal Sr (ω) by the matched filter coefficient Km (ω). Here, the matched filter coefficient Km (ω) is preset by the complex conjugate signal of the nonlinear FM chirp transmission signal, and the phase on the time axis of each frequency component of the reception signal Sr (ω) matches. It is a coefficient to be made. The signal Sd (ω) that is the reception signal Sr (ω) multiplied by the matched filter coefficient Km (ω) is output to the smoothing processing unit 53.

  The smoothing unit 53 applies a smoothing coefficient Ks (ω) set in advance to the output signal Sd (ω) from the matched filter 52 by using the nonlinear FM chirp transmission signal and the Gauss function that are regulated to have a constant amplitude. ) To smooth the signal Sd (ω). The smoothing processing unit 53 outputs the smoothed signal Sds (ω) multiplied by the smoothing coefficient Ks (ω) to the IFFT processing unit 54.

The smoothing coefficient Ks (ω) is set by the smoothing coefficient generation unit 500 by the following method. The smoothing coefficient Kw (ω) generation process may be performed offline in advance or may be performed in real time.
FIG. 2 is a flowchart showing a method for setting the smoothing coefficient Ks (ω). FIG. 3 is a diagram for explaining the setting concept of the smoothing coefficient Ks (ω), and shows the frequency spectrum SpTL of the autocorrelation signal of the transmission signal and the frequency spectrum SpGF of the autocorrelation signal after smoothing processing.

  When the smoothing coefficient Ks (ω) is set, first, the nonlinear FM chirp transmission signal regulated to have a constant amplitude is converted into a signal on the frequency axis using Fourier transform (FIG. 2: S101). In this embodiment, a case where a transmission signal is used is shown, but a reception signal obtained by transmitting a nonlinear FM chirp transmission signal to a target set in advance at a predetermined position may be used.

  Next, the autocorrelation processing unit 501 multiplies the transmission signal on the frequency axis by the matched filter coefficient Km (ω) obtained from the complex conjugate of the transmission signal, and acquires the frequency spectrum SpTL of the autocorrelation signal of the transmission signal (FIG. 2: S102). Here, since the transmission signal is regulated to have a constant amplitude as described above, the frequency spectrum SpTL of the autocorrelation signal has a waveform having minute irregularities, as shown in FIG.

  Next, the detection unit 502 acquires only the amplitude information of each spectrum component of the frequency spectrum SpTL of the autocorrelation signal. Specifically, amplitude information is acquired by performing a square root process on the sum of the square of the real part and the square of the imaginary part of each spectral component. That is, the amplitude Am (ω) of each spectral component is calculated by the following equation. Here, the real part of the autocorrelation signal is Re (ω) and the imaginary part is Im (ω).

  The smoothing processing unit 503 smoothes the amplitude information of each spectral component of the autocorrelation function of the transmission signal obtained in this way (FIG. 2: S103). As the smoothing process, a fitting process to a predetermined window function or a moving average process may be used.

  Next, the smoothing coefficient calculation unit 504 calculates the smoothing coefficient Ks (ω) for each spectral component by dividing the frequency spectrum SpGF of the autocorrelation signal after smoothing processing thus obtained by the autocorrelation signal. To do.

Note that the low frequency range from the lowest frequency (f _L ) of the frequency spectrum SpTL of the transmission signal to the lower threshold frequency f _THL and the high frequency range from the highest frequency f _H to the upper threshold frequency f _THH are smooth. Do not process. That is, smoothing processing is performed only for a predetermined frequency section FP of the autocorrelation signal. This was obtained by the inventor through simulation and experiment. Even if the low frequency range and the high frequency range are changed, the suppression amount of the side lobe hardly changes, whereas the main lobe is changed. This is based on an increase in attenuation.

  Thus, if the low frequency range and the high frequency range are appropriately set, the side lobe suppression amount and the main lobe attenuation amount can be optimized.

In the above description, the smoothing process is performed on the spectral components from the lower limit threshold frequency f _THL to the upper limit threshold frequency f _THH of the frequency spectrum SpTL of the autocorrelation signal of the transmission signal regardless of the amplitude level. The smoothing process may be performed only on the spectral components having a predetermined amplitude level or higher.

  By using the smoothing coefficient Ks (ω) calculated in this way, the signal Sds (ω) after the smoothing process becomes a frequency spectrum SpD as shown in FIG. FIG. 4 is a diagram showing the frequency spectrum SpD of the signal Sds (ω) after the smoothing process.

  As shown in FIG. 4, the signal Sds (ω) after the smoothing process has a smooth frequency spectrum without minute irregularities in the fitting section FP excluding the predetermined frequency range at both ends of the entire frequency band.

  The inverse Fourier transform unit 54 performs inverse Fourier transform on the signal Sds (ω) after smoothing on the frequency axis, and generates a signal Sds (t) after smoothing on the time axis.

  In this way, in addition to multiplying the received signal by the matched filter coefficient Km (ω) by the matched filter 52, the pulse compression unit 15 multiplies the smoothing unit 53 by the smoothing coefficient Ks (ω). Thereby, the processing in the pulse compression unit 15 is not a matched filter, and the main lobe of the signal output from the pulse compression unit 15 is attenuated. However, the loss due to multiplication by the smoothing coefficient Ks (ω) is very small. In exchange for this, the side lobe level can be reduced by windowing the frequency spectrum with the original predetermined window function.

  FIG. 5 shows a time-axis waveform of a signal obtained by performing a pulse compression process and a smoothing process on a signal Sds (t), that is, a received signal Sr (t) after a smoothing process on the time axis, and a smoothing process by performing only the conventional pulse compression The time axis waveform of the signal Sds ′ (t) that is not shown. Here, FIG. 5B is a partially enlarged view of the time (−1.0 ≦ t [μs] ≦ + 1.0) in FIG.

  As shown in FIG. 5, by performing pulse compression and smoothing as in this embodiment, the attenuation amount of the main lobe can be minimized, and the range side lobe can be significantly suppressed.

  The smoothed signal Sds (t) output from the IFFT processing unit 54 of the pulse compression unit 15 is input to the target detection unit 16 as an output signal from the pulse compression unit 15.

  The target detection unit 16 compares the output signal from the pulse compression unit 15 with a predetermined threshold value, and detects that there is a reflection signal from the target if it is equal to or higher than the predetermined threshold value. At this time, even if the output signal from the pulse compressor 15 is pulse-compressed, as shown in FIG. 5, the range side lobe is suppressed with almost no attenuation of the main lobe. In addition, the target can be detected with high accuracy.

In the above description, the example in which the smoothing coefficient Ks (ω) is multiplied after the multiplication of the matched filter coefficient Km (ω) is shown. However, the matched filter coefficient Km (ω) is You may make it multiply.
Furthermore, a coefficient obtained by multiplying the smoothing coefficient Ks (ω) and the matched filter coefficient Km (ω) in advance may be stored in advance, and this coefficient may be multiplied by the matched filter. If a coefficient obtained by multiplying the smoothing coefficient Ks (ω) and the matched filter coefficient Km (ω) in advance is used, a real-time multiplication process for the received signal can be performed only once, so that the processing load is reduced. Can do.

  In the above description, an example in which correlation processing or smoothing processing is performed on the frequency axis has been described. However, correlation processing or smoothing processing may be performed on the time axis.

  In the above description, the Fourier transform has been described as an example, but the Fourier transform method may be any method such as DFT or FFT, and further, a time axis signal can be converted into a frequency axis signal. For example, other conversion methods such as wavelet conversion may be used.

  In the above description, the radar apparatus has been described as an example. However, if the device transmits a pulse transmission signal of a non-linear chirp method, such as a fish detector or sonar, the above configuration and processing can be applied. it can.

11-radar apparatus, 12-transmission unit, 13-DPX, 14-antenna, 15-pulse compression unit, 16-target detection unit, 51-Fourier transform unit, 52-matched filter, 53-smoothing processing unit, 54- Inverse Fourier transform unit, 500-smooth coefficient generation unit, 501-autocorrelation processing unit, 502-detection unit, 503-smoothing processing unit, 504-smooth coefficient calculation unit

Claims (5)

A pulse compression device that performs pulse compression using a non-linear FM chirped pulse-shaped transmission signal in which a frequency spectrum is defined by a predetermined window function,
A detector for obtaining an amplitude of a frequency spectrum of a signal obtained by autocorrelating the pulsed transmission signal;
A smoothing processing unit that smoothes the detection signal output from the detection unit;
A smoothing coefficient generating section that generates a smoothing coefficient by dividing the output of the smoothing processing section by a frequency spectrum of a signal obtained by autocorrelating the pulsed transmission signal;
A pulse compression unit that performs the pulse compression using a reception signal for the transmission signal, a matched filter coefficient based on the pulsed transmission signal, and the smoothing coefficient;
A pulse compression device comprising:   The pulse compression device according to claim 1, wherein the smoothing process performs a smoothing process by moving and averaging the detection signals.   The pulse compression device according to claim 1, wherein the smoothing processing unit performs a smoothing process so that the detection signal approaches a frequency spectrum of the predetermined window function.   The pulse compression device according to any one of claims 1 to 3, wherein the smoothing unit performs a smoothing process on a part of a frequency region of the detection signal. While comprising the pulse compression device according to any one of claims 1 to 4,
A transmission unit for generating the transmission signal;
An antenna that radiates the transmission signal into a detection area, receives a reflection signal of the transmission signal by a target, and outputs the reception signal;
A target detection unit for detecting the target based on an output signal from the pulse compression unit;
Target detection device with

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