JP2010051011A - High dynamic range variable gain amplifier - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a high dynamic range variable gain amplifier. <P>SOLUTION: A multi-stage, low-power, high dynamic range variable gain amplifier (100) has an input stage (120) cascaded with one or more current amplifier stages (160A, 160B) whereby the gain of each stage (120) may be independently controlled. The input stage (120) may be composed of a variable transconductance amplifier (227) using variable emitter degeneration. The current amplifier (160A, 160B) may be composed of a differential Darlington amplifier (510) coupled to a differential cascade amplifier (520). The transconductance amplifier (227) converts an input voltage signal to a current signal. The variable gain amplifier (100) is designed for efficient low power operation. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a variable gain amplifier (VGA), and more particularly to a VGA used in a communication device.

  In a wireless communication environment, a wireless communication receiver receives signals that vary rapidly and widely in signal power. Receivers such as those used in wideband digital code division multiple access (CDMA) mobile stations are required to control the power of the demodulated signal for proper signal processing. Furthermore, it is necessary to control the transmission power in a transmitter as used in a CDMA mobile station in order to avoid excessive interference to other mobile stations. These same power control concepts apply to narrowband analog frequency modulation (FM) radio communication receivers and transmitters.

  There are dual mode CDMA / FM radio communication devices required to control the transmitted and received signal power of digital CDMA and analog FM modulation. In these dual-mode mobile stations, the control process is complicated by the different dynamic ranges and industry regulation standards associated with CDMA and FM signals. That is, the amplitude of the received CDMA signal can vary over a range of approximately 80 decibels, but the amplitude of the received FM signal can vary over the same range as 100 decibels. Providing separate automatic gain control (AGC) circuits for CDMA and FM signals complicates and increases the cost of this type of dual mode mobile station. It is therefore desirable to provide an AGC circuit that can operate on both CDMA and FM signals.

  1A and 1B show an exemplary environment for a VGA performing AGC functions. Figures 1A and 1B are commonly referred to, for example, as the telecommunications industry standard "Mobile Station-Base Station Compatibility Standard for Dual Mode Broadband Spread Spectrum Cellular Systems" (TIA / EIA / IS-95), generally referred to simply as IS-95. 2) is a block diagram of a dual mode CDMA / FM mobile phone 900 designed in accordance with FIG. VGA is used for receive and transmit AGC amplifiers 902 and 904, respectively, of cellular telephone 900. The front end receiver portion of the cellular telephone 900 includes an antenna 906, a duplexer circuit 908, a low noise amplifier (LNA) and mixer circuit 910, and a filter 930. As the cellular telephone 900 moves throughout the coverage area of the CDMA system, the signal level at the antenna 906 changes from approximately -110 dBm to -30 dBm. Note that each of these front-end elements provides the same gain as any signal level applied to it over its operating range. The operating range is such that the dynamic range of the signal applied to receiving AGC amplifier 902 is the same as the dynamic range of the signal at antenna 906 (approximately 80 dB). Similarly, the signal level at the antenna changes by approximately 100 dB when the cellular phone 900 moves throughout the coverage area of the FM system.

  The output of the receive AGC amplifier 902 is provided to a baseband analog application specific integrated circuit (BAASIC) 912 that converts the analog signal to a digital signal. The analog-to-digital signal conversion process works best if the signal level applied to the analog-to-digital converter is unchanged. The receive AGC amplifier 902 performs the function of compensating for variable input power so as to keep the output power of the receive AGC amplifier 902, and thus the input to the analog to digital converter, constant.

  The mobile station modem ASIC 914 performs demodulation on CDMA and FM signals, as well as various digital and power control functions associated with CDMA operation.

This type of function is a known technique and is not important to the present invention. Therefore, it is not further described here. User interface 916 provides an interface to a human operator. This type of user interface 916 is also known in the art.

  The mobile station modem ASIC 914 also provides the BAASIC 912 with a baseband modulated digital representation of the CDMA waveform or a modulated analog representation of the FM waveform. BAASIC 912 converts the baseband signal representation to an analog intermediate frequency (IF) form at a constant signal level and provides it to transmit AGC amplifier 904. Transmitter AGC amplifier 904 performs power control on the signal and provides it to upconverter 918, power amplifier and driver circuit 920, isolator 922, transmission / reception switching circuit 908 and antenna 906. As cellular phone 900 moves throughout the coverage area of the cellular system, the transmitted signal level at antenna 906 varies in the opposite direction from the received power, and the transmitted level is close to maximum when the received power is at a minimum value. The transmission power level is changed by the AGC amplifier 904. Note that the input power to the AGC amplifier 904 is typically fixed, and the gain of the power amplifier 920 is also fixed.

  More information about automatic gain control loops in wireless communication systems, and more information about power control in general, can be found in US Pat. No. 5,283,536 (named “High Dynamic” filed on Feb. 1, 1994). Range Closed Loop Automatic Gain Control Circuit (HIGH DYNAMIC RANGE CLOSED LOOP AUTOMATIC GAIN CONTROL CIRCUIT)), US Patent No.5,107,225 DYNAMIC RANGE CLOSED LOOP AUTOMATIC GAIN CONTROL CIRCUIT ”and US Pat. No. 5,267,262 (named“ TRANSMITTER POWER CONTROL SYSTEM ”filed on November 30, 1993). Name applied for on November 12, "Method and APPARATUS FOR AUTOMATIC GAIN CONTROL IN A DIGITAL RECEIVER" Patent Nos. 5,469,115 and U.S. Pat.No. 5,283,536, each of which is entitled `` HIGH DYNAMIC RANGE CLOSED LOOP AUTOMATIC GAIN CONTROL CIRCUIT '' And is incorporated herein by reference.

Like the mobile communication receivers and transmitters described above, the mobile communication receivers and transmitters have a high compression point, low noise injection.
And designed to have low power consumption. A receiver with a high compression point and low noise injection has a high dynamic range and can detect signals over a wide range of power levels. A transmitter with a high compression point and low noise injection has a high dynamic range and can transmit signals over a wide range of power levels. Low power receivers and transmitters extend battery life. Therefore, these characteristics are important when designing variable gain amplifiers for communication systems in which signals are transmitted and received over a large range of power levels.

  The receiver must be able to detect information from a strong signal broadcast by a nearby powerful transmitter and from a weak signal broadcast by a distant low power transmitter.

  The range in which the receiver can detect from a weak signal to a strong signal is called the dynamic range of the receiver. Similarly, the transmitter must be able to transmit low power signals to nearby receivers and high power signals to remote receivers. The dynamic range of the receiver is determined by its detectable minimum and maximum signal levels. The minimum detectable signal level of the receiver is determined by the noise figure of the receiver. Similarly, if the signal level falls near or below the noise floor, the minimum transmittable power is set by the transmitter noise figure. The noise figure of VGA is partly a function of noise injection characteristics and VGA gain. In general, the higher the receiver gain, the better the noise will be. That is, if the index is good, a very weak signal can be detected even in the presence of noise.

  The maximum detectable signal level of the receiver can be established by the intermodulation distortion (IMD) operation of the receiver. When multiple signals pass through any device, mixing operations occur between the signals due to device non-linearity. For example, in regions where CDMA and analog FM systems coexist, third order IM products from analog FM systems generally fall into the CDMA passband. This IM product acts as a “jammer” that contributes to the IMD and interferes with the detection and demodulation of the desired signal in the receiver. VGA IMD operation is in part a function of its linearity and gain. In general, the lower the receiver gain, the better the IMD operation. As noted above, this is in contrast to noise figure requirements. Thus, VGA design for a receiver with a large dynamic range involves a difficult trade-off between IMD operation and noise figure.

  In general, multiple transmitter VGAs are designed to receive a relatively constant input power level and provide different ranges of output power levels, while multiple receiver VGAs have different ranges of output power levels. Similar design considerations are relevant for multiple transmitter VGAs, with the difference that they are designed to provide a relatively constant output power level relative to the input power level.

  In addition, mobile receivers are designed to be compact, lightweight and have a long operational life. Mobile receivers are powered by a minimum number of battery cells to reduce their size and weight and increase their portability. Since battery voltage is proportional to the number of battery cells, AGC circuits (including variable gain amplifiers (VGA)) must operate at low supply voltages. It is also desirable to increase battery life in order to increase the period between battery replacement or recharge. Therefore, AGC circuits (including VGA) must have low DC current and power consumption.

  This requirement for low DC power consumption includes design trade-offs similar to those already mentioned. More DC power is required for high gain amplifiers with good noise figure. However, a low gain amplifier with good IMD operation requires less DC power. Existing VGA designs are inefficient and cannot save DC power at a sufficiently low gain level.

  What is needed is a VGA with high dynamic range, good noise figure and IMD operation, as well as low DC power consumption.

  According to the present invention, a VGA with high dynamic range, good noise figure and IMD operation and with minimum DC power consumption is provided. The VGA can be used in automatic gain control (AGC) amplifiers for receivers and transmitter chains in cellular telephones. The VGA achieves power gain by converting an input voltage signal into a current signal and amplifying the current signal. The amplified current signal can be converted to a voltage signal by terminating a VGA with appropriate impedance.

  A VGA consists of at least two cascaded stages, an input stage and a current amplifier. The input stage can be further divided into a CDMA input stage and an FM input stage. The outputs of both input stages are coupled to the input of the current amplifier and can be selected by a CDMA / FM mode signal. In an embodiment, the FM input stage is single-ended and the CDMA input is balanced. The gain of the VGA can be increased by cascading two or more current amplifier stages. The transconductance gain of the input stage can be controlled by a control signal.

  High dynamic range and low power consumption VGA is created by combining multiple technologies. In the first embodiment, well suited for dual mode receive AGC amplifiers such as amplifier 902 of FIG. 1, the CDMA input stage comprises a variable conductance amplifier cascaded with a Gilbert cell attenuator. A variable transconductance amplifier converts a changing voltage signal into an output current signal with a transconductance controlled by a FET transistor operating as a variable emitter degeneration resistor. Emitter degeneration provides variable local series feedback, which allows the CDMA input stage to handle a wide dynamic range of the input signal, while providing good noise figure and IMD. Provide behavior. When there is a low level input signal, the channel resistance of the FET transistor can be changed to increase the gain of the input stage, which improves the receiver's ability to detect noise figure and weak signals Is done. On the other hand, when there is a high level input signal, the channel resistance of the FET transistor is changed, reducing the gain of the input stage. This improves the IMD operation of the receiver. The Gilbert cell attenuator attenuates the additional current so that no subsequent current amplification stage is overdriven to their non-linear range when a large input signal is applied.

  In this first embodiment, the FM input stage is an emitter degeneration bipolar differential amplifier followed by a Gilbert cell attenuator. The differential pair converts the input voltage into a current, feeds it to the Gilbert cell attenuator, and further attenuates the current input to the next stage current amplifier. Unlike the CDMA input stage, the FM input stage uses a fixed gain transconductance stage rather than variable emitter degeneration. This is because the amplifier can quickly become non-linear and saturate, and the industry standard (IS-95) linearity requirements for FM signals are much relaxed than that for CDMA signals.

In a second embodiment, well suited for a transmit AGC amplifier such as amplifier 904 of FIG. 1, FM and CDMA signals are shunted at the input, connected to the transconductor and the Gilbert cell attenuator. It can be handled by a fixed-gain transconductance input stage with a differential pair with shunt-series feedback. Shunt-series feedback at the input allows for accurate and linear input impedance without using brute force matching. The output of the differential pair can be AC connected to the transconductor by a set of capacitors. The transconductor converts the voltage output of the differential pair into current using an emitter-degenerated differential amplifier. The current is then applied to a Gilbert cell attenuator, which further attenuates the current flowing through the next stage of the current amplifier. Since the input level to the transmit AGC amplifier 904 is generally constant, a variable gain input stage is not necessary.

  In a first embodiment suitable for use as a receiving AGC amplifier 902, each current amplifier consists of two sections, a differential Darlington amplifier and a differential cascade amplifier. These current amplifiers are translinear circuits that can be controlled by changing the ratio of “tail currents” in which the current gain biases the translinear loop. . The current gain of each current amplifier stage can be independently controlled by one or more control signals.

  In a second embodiment, which is convenient for use with the transmit AGC amplifier 904, each of the current amplifiers consists of two sections, a differential Darlington amplifier and a simple differential pair. This current amplifier is a hybrid of a feedback current amplifier and a translinear loop.

  In each of the above embodiments, the gain of the variable-gain stage is controlled by a gain control circuit. The gain control circuit changes the gain of the current amplifier according to the applied AGC control voltage (RX gain control or TX gain control in FIG. 1). The gain control circuit includes an exponential generator that ensures VGA linearity (in decibels) over a wide dynamic range.

  It is therefore an advantage of the present invention to provide a VGA with a high dynamic range on CDMA and FM signals. Mobile receivers using this type of VGA can detect signals over a wider input power range. A further advantage is that the VGA consumes minimal DC power. Therefore, the VGA can be used in the mobile communication device, and can advantageously protect the operating life of the battery. An additional advantage is that the gain of the VGA can be changed approximately linearly in decibels by linearly adjusting the DC control voltage.

1 is a schematic diagram of an exemplary CDMA / FM dual mode communication device that can be used with the present invention. FIG. 1 is a schematic diagram of an exemplary CDMA / FM dual mode communication device that can be used with the present invention. FIG. 1 is a block diagram of an exemplary three-stage variable gain amplifier of the present invention. FIG. 3 is a schematic diagram showing the CDMA input stage of FIG. FIG. 3 is a schematic diagram showing the transconductance amplifier bias control circuit of FIG. FIG. 5 is a schematic diagram showing the exponential function generator of FIG. FIG. 4 is a schematic diagram illustrating a combination of partial parts in FIGS. 2 and 3 configured to illustrate the beneficial properties of the present invention. FIG. 3 is a schematic diagram showing the current amplifier of FIG. FIG. 8 is a schematic diagram showing the tail current source of FIG.

The function, purpose and advantages of the present invention will become more apparent from the following detailed description with reference to the drawings, in which the same reference numerals are used for the same items throughout.
The present invention is directed to a monolithic integrated circuit variable gain amplifier (VGA). VGA provides a gain proportional to the control voltage. The VGA provides an exponential voltage gain as a function that increases linearly within the applied control voltage. This control voltage provides a linear power gain in decibels (DB) that is directly proportional to the linear increase within the applied control voltage. VGA can provide linear power gain over a large dynamic range (ie, a factor of l00,000,000) over 80 decibels. VGA provides a linear power gain that can withstand process changes that occur during VGA configuration.

  VGA can be used for many applications, including in receivers and transmitters. When the VGA is operating at the receiver, the input typically varies over a large dynamic range while the VGA output is relatively constant.

When the signal level input to the VGA operating at the receiver is small, the gain of the VGA must be relatively large. When the signal level input to the VGA operating at the receiver is large, the VGA gain must be relatively small. It must typically have good noise characteristics when providing a relatively high gain and must have good intermodulation behavior when providing a relatively low gain.

  When the VGA is operating at the transmitter, the input can typically be constant, but the output of the VGA varies over a wide dynamic range. When the VGA signal level output needs to be large, the VGA gain must be relatively large and the intermodulation operation must support the resulting large signal level. When the signal level output from the VGA operating at the transmitter needs to be small, the gain of the VGA must be relatively small and the noise characteristics of the VGA can be important.

  FIG. 2 is a block diagram of one embodiment of a variable gain amplifier (VGA) 100 that adjusts the power level of an input signal over a wide dynamic range. The embodiment of FIG. 2 is suitable for use as the receiving AGC amplifier 902 of FIG. VGA 100 comprises three stages: an input stage 120 and two cascaded current amplifier stages 160A and 160B. A plurality of current amplifier stages 160 are cascaded following the input stage 120 to increase the dynamic range of the VGA 100. In the first embodiment, the input stage 120 comprises: a separate FM input stage 121 and a CDMA input stage 122 with respective input ports 171 and 170. FM input stage 121 and CDMA input stage 122 are alternately connected to current amplifier 160A by switch 123. This switch 123 is controlled by a CDMA / FM mode selection signal. When the communication device is in CDMA mode, switch 123 connects CDMA input stage 122 to current amplifier 160A and disconnects FM input stage 121. Conversely, when the communication device is in the FM mode, the switch 123 connects the FM input stage 121 to the current amplifier 160A and disconnects the CDMA input stage 122.

  FIG. 2 also shows bias ports 110, 130, 150A, 150B for control voltages applied to VGA 100. FIG. The gain of each stage is controlled, for example, by a control voltage that can be formed by a receiver detection circuit that determines the signal strength. Each stage consists of various components including active elements such as transistors.

  The VGA input signal provided to the input port 170 of the CDMA input stage l22 is split into two signal paths that are balanced, i.e., transmitting signals that are 180 degrees out of phase with each other. The VGA input signal is applied through the VGA input port 170. However, the VGA input signal provided to the input port 171 of the FM input stage 121 is one end ground. The output of input stage 120 and the input of current amplifier 160A are connected by port 190.

  Since it operates at a low power supply voltage of about 3.6 volts, the input stage 120 converts the input voltage signal to a current signal, preventing the VGA active devices from operating and distorting the input signal in their non-linear regions. The low power supply voltage of VGA 100 also reduces the power consumption of VGA 100.

  FIG. 3 shows one embodiment of the CDMA input stage 122. The balanced signal is applied to the input port 170 of the VGA. The CDMA input stage 122 includes a variable transconductance amplifier 227 coupled to a Gilbert cell attenuator 226 and provides four functions. First, the variable transconductance amplifier 227 converts the input voltage signal into a current signal. Second, the combination of variable transconductance amplifier 227 and Gilbert cell attenuator 226 allows variable amplification of the signal, which is exponentially (by adjusting the control voltage linearly at bias port 110 ( Can be varied linearly (in decibels). Third, the increased emitter degeneration in variable transconductance amplifier 227 reduces the IMD of VGA 100 when the input signal voltage is large and when the IMD is most prominent. The transconductance of the input stage 120, and thus the IMD, is reduced so that the emitter degeneration in the variable transconductance amplifier 227 is increased. Finally, the reduced emitter degeneration within the variable transconductor amplifier 227 improves the VGA 100 noise figure when the input signal voltage is small and noise operation is most critical. As emitter degeneration in variable transconductance amplifier 227 is reduced, the transconductance of input stage 120 increases, improving the noise figure of the receiver.

  The variable transconductance amplifier 227 comprises two bipolar junction transistors (BJTS) 235 and 236, two current sources 238 and 239, and a field effect transistor (FET) 237. Current sources 238 and 239 are connected to the emitters of BJTS 235 and 236 in series. The source connection 228 and drain connection 229 of FET 237 are connected to the emitters of BJTS 235 and 236, respectively. The balanced signal at VGA input port 170 is applied to the bases of BJTS 235 and 2S6. The balanced current of variable transconductance amplifier 227 is output from the collectors of BJTS 235 and 236.

  The transconductance of the variable transconductance amplifier 227 can be adjusted by changing the emitter degeneration of the BJTS 235 and 236. As a result, the gain of the VGA 100 can be changed. The emitter degeneration of BJTS 235 and 236 is formed by changing the channel resistance of FET 237. FET 237 behaves like a variable resistor in its ohmic region and provides variable emitter degeneration for both BJTS 235 and 236. The drain-source bias voltage of FET 237 must therefore be less than the knee voltage of FET 237. By changing the voltage applied to the bias port 290, the bias between the gate and source junction of the FET 237 can be adjusted and this adjustment can change the channel resistance. The transconductance of the variable transconductance amplifier 227 can be increased by reducing the channel resistance of the FET 237. Thus, by providing a variable channel resistance through FET 237, the present invention can reconcile competing design considerations between noise figure and IMD operation. Furthermore, the DC efficiency of VGA 100 is improved. That is, when there is a need to amplify the low level input signal, the CDMA input stage 122 draws enough DC current, but for the high level input signal it reduces its transconductance to This is because the DC current consumption is reduced.

  The differential output current of the variable transconductance amplifier 227 is connected to the Gilbert cell attenuator 226. A Gilbert cell attenuator 226 changes the current amplitude of the signal applied to its input. Gilbert cell attenuator 226 comprises a first set of BJTa 231 and 234 and a second set of BJTa 232 and 233. The attenuation level of the Gilbert cell attenuator 226 is established by a control voltage applied to the bias port 110. When the first set of BJTs 231 and 234 is biased by a control voltage applied to bias port 110, Gilbert cell attenuator 226 attenuates the output current of variable transconductance amplifier 227 to make it variable. The transconductance amplifier output current component flows through the first set of BJTs 231 and 234 rather than the second set of BJTs 232 and 233. Therefore, the balanced current at port 190 of Gilbert cell attenuator 226 is reduced. Variable transconductance amplifier 227 and Gilbert cell attenuator 226 are biased by a common power supply 230.

  The preferred embodiment of FM input stage 121 is similar to CDMA input stage 122 except that its FET 237 is replaced with a fixed resistor. As previously mentioned, the fixed resistance of the FM input stage 121 provides a fixed transconductance. That's because industry standards (eg, IS-95) allow compression of the input signal (ie, allow the VGA to move non-linearly) at input levels much lower than that of the CDMA input signal. Alternatively, input stage 120 may comprise only a single fixed transconductance stage similar to that of FM input stage 121. This alternative embodiment is particularly well suited for use as the transmit AGC amplifier 904 of FIG.

  As described above, one aspect of the design is that the control voltage applied to the bias port 130 of the transconductance bias control circuit 140 is linearly adjusted and the transconductance of its variable transconductance amplifier 227 varies exponentially. Is to do. To implement this result, the control voltage at the bias port 130 of the transconductance bias control circuit 140 is linearly adjusted and the channel resistance of the FET 237 also varies exponentially. FIG. 4 illustrates one embodiment of a transconductance bias control circuit 140 that facilitates this result. The transconductance bias control circuit 140 includes an exponential function generator 360, first and second operational amplifier circuits 353 and 354, a low pass filter 352, and a current source 341.

  The exponential function generator 360 converts the control voltage into two output currents output from the output terminal 358 of the exponential function generator 360 to the first operational amplifier circuit 353. The ratio of these current magnitudes is exponentially proportional to the control voltage.

In the exemplary embodiment of FIG. 1, the control voltage is either RX gain control or TX gain control, or a variant that is scaled or temperature compensated. The generation of this control voltage is outside the scope of the present invention, which is described, for example, in US Pat. No. 5,469,115, which is hereby incorporated by reference.

  FIG. 5 shows one embodiment of the exponential function generator 360. Exponential function generator 360 includes a differential amplifier 465 that outputs a set of FET current mirrors 474. Differential amplifier 465 comprises a parallel set of BJTs 461 and 462 connected to current source 472. The set of FET current mirrors 474 comprises four FETs 464, 466, 468 and 470. Because of the exponential input voltage-output current relationship of BITs 461 and 462, their collector-current ratio is proportional to the difference in base voltage between BJTs 461 and 462 as determined by the control voltage signal. Thus, a linear voltage difference change across the baice port 130 is converted to an exponentially related (linear in decibel) current at the output 358. Current mirror 474 simply receives the exponentially related current generated by bipolar differential pair 461 and 462 and provides it for use through the amplifier. Exponential function generator 360 is biased by power supply 400.

  Referring again to FIG. 4, the first and second operational amplifier circuits 353 and 354 operate in cooperation with the exponential function generator 360 to control the channel resistance of the FET 237 of FIG. The first operational amplifier circuit 353 comprises a master FET 344, which is preferably identical to the FET 237, the reference resistor 346 and the differential operational amplifier 348. The output current from exponential function generator 360 is connected to master FET 344 and reference resistor 346. A differential operational amplifier 348 applies the voltage across the drain and source terminals of the master FET 344 and the reference resistor 346, which is made equal by changing the bias voltage applied to the gate of the master FET 344. The The bias voltages applied to the gates of FET 237 and master FET 344 are generally equal. However, the gate bias voltage applied to FET 237 through bias port 122 is low pass filtered to prevent thermal noise from transconductance bias control circuit 140 from being input to FET 237. Low-pass filtering is completed by a low-pass filter 352 formed by a series resistor 350 and a shunt capacitor 351.

  The second operational amplifier circuit 354 ensures that the master FET 344 and FET 237 have the same power supply voltage. The second operational amplifier comprises a non-inverting, unity gain operational amplifier 349 and resistors 345 and 347 to detect the drain-source voltage across FET 237 via source connection 228 and drain connection 229.

  The exponent function generator 360, and the current source 341 connected around the master FET 344 and the reference resistor 346, provide a voltage drop across the reference resistor 346, and hence a voltage drop across the drain-source of the master FET 344. Is designed to be less than the knee voltage of the FET. As a result, the operation of operational amplifier circuits 353 and 354 forces FET 237 and master FET 344 to operate at similar quiescent points within their ohmic region. Thus, the channel resistances of FET 237 and master FET 344 are generally the same and vary exponentially with a linearly regulated control voltage applied to bias port 130.

  FIG. 6 is a combination of the elements in FIGS. 2 and 3 configured to illustrate the beneficial properties of the present invention. One of the problems overcome by the device shown in FIG. 6 is the process variation of μcCox and hence the channel resistance of FET 237 as a function of the voltage applied to its gate.

As previously mentioned with respect to FIG. 3, FET 237 controls the transconductance of variable transconductance amplifier 227. The variable emitter degeneration provided by FET 237 allows input stage 120 to handle a wide range of signals.

  Since the attenuation due to the input stage 120 is critical to the operation of the circuit and the characteristics of the stage are set by the FET 237, it is very important to set the resistance value of the FET 237 accurately. The channel resistance as a function of the applied gate voltage is difficult to control one by one in the manufacturing process, so an outer control loop is used to achieve one inertia. FIG. 6 shows a control loop used to immunize the operation of the CDMA input stage 122 against FET 237 process variations.

  Resistor 346 is an on-chip resistor. This resistor is made large to minimize process variations. Resistor 346 is used as a reference resistance for the control loop.

  Note that the total current from output 358 of exponential function generator 360 is set by current source 341. Thus, if the current through one of the balanced outputs of output 358 increases, the current through the other one of the balanced outputs of output 358 decreases. Also note that the voltage drop across resistor 346 is the same as the voltage drop across master FET 344. Since each voltage is one of the inputs to operational amplifier 348, the voltage drop is the same. The output of operational amplifier 348 controls the resistance of master FET 344 so that the voltage drop across it is the same as the value provided by the current through resistor 346 and the value of resistor 346. Thus, the voltage drop across resistor 346 increases as the current through resistor 346 increases and the current through current master FET 344 decreases. In response, the channel resistance of the master FET 344 must also increase so that the voltage drop remains the same. The same output voltage of operational amplifier 348 applied to the gate of master FET 344 is also applied to the gate of FET 237. Resistor 350 and capacitor 351 provide a low pass filter between the output of operational amplifier 348 and the gate voltage of FET 237, but the DC voltage applied to the gates of master FET 344 and FET 237 is the same. is there.

  In the preferred embodiment, master FET 344 and FET 237 are in close proximity to each other on a common substrate. Thus, even if process variation from VGA component to VGA component is significant, in one VGA component, the channel resistance characteristics vs. gate voltage of master FET 344 and FET 237 closely follow each other. Thus, the resistance of FET 237 is set equal to the resistance of master FET 344. As the channel resistance of FET 237 decreases, the current through transistors 235 and 236 increases. Thus, the present invention provides a method for accurately performing variable emitter degeneration of CDMA input stage 122.

  FIG. 7 shows an embodiment of the current amplifiers 160A and 160B shown in FIG. The input of the current amplifier 160 shown in FIG. 7 can be connected to the output of the input stage 120 or the output of another current amplifier 160. The current amplifier 160 includes a Darlington differential amplifier 510, a cascaded differential amplifier 520, and a tail current source 570. Current amplifier 160 is biased by power supplies 508 and 506 and current sources 596 and 598. The Darlington differential amplifier 510 has BJTs 580, 586, 588 and 594 and the resistor 582 in topology shown in FIG. 7 so that the Darlington differential amplifier 510 has shunt-series feedback of electrical resistance. 584, 590, 592 and provides enhanced current gain and process variation insensitivity.

  In the present invention, the shunt-series feedback of electrical resistance provided by the resistors 582, 584, 590, 592 of the present invention makes the feedback current through the resistor equal to the input current through the input port 190. Special attention must be paid. Thus, they also provide a current divider, so they increase the current gain of the differential Darlington amplifier 510 by the ratio of the feedback resistors.

  Cascade differential amplifier 520 provides a translinear loop that provides variable current amplification according to the ratio of tail current 512 formed by tail current source 570. The cascaded differential amplifier comprises BJTs 500, 502, 504 and 506 in a differential current mirror (translinear loop) topology, the differential current mirror changing the tail current 512 to increase the gain of the current amplifier. It is possible to change.

  The gain of current amplifier 160 is controlled by tail current source 570. The tail current source 570 is connected to the Darlington differential amplifier 510 and the cascaded differential amplifier 520 via the differential port 512. The current amplification of each current amplifier 160 can be changed exponentially by using the control current formed by the exponential function generator 360 of FIGS. 4 and 5 applied to the control port 150. is there. The tail current source 570 is biased by the power source 509.

  FIG. 8 shows one embodiment of the tail current source 570. Tail current source 570 comprises: exponential function generator 861; this is similar to exponential function generator 360 output 358 or similar to exponential function generator 360 (FIGS. 4 and 5) producing the same output 859 Or the same thing. The exponential function generator 861 is connected to a set of bipolar current mirrors 860. In FIG. 8, both circuits are connected to a power source 509, but they can also be connected to different power sources. A set of bipolar current mirrors 860 includes first BJT groups 822, 824 and 830, and second BJT groups 832, 834 and 840, and first resistor groups 826, 828 and 844, and second. Resistor groups 836, 838 and 842. The purpose of the set of bipolar current mirrors is to receive the control current provided by the exponential function generator 861 and convert it to a tail current 512.

  In an embodiment of the invention, exponential function generators 360 and 861 are the same element and, like current amplifiers 160A and 160B, advantageously provide a single control current that can be projected to CDMA input stage 122. . This embodiment reduces the current gain of current amplifiers 160A and 160B (and thus the DC current drain on the battery), at the same time and ratio as the transconductance of CDMA input stage 122 is reduced. , Providing even more DC efficiency. Furthermore, this arrangement ensures that all current amplifications in all stages are exponentially (decibel and linear) related to the control voltage of the AGC amplifier.

  Thus, the present invention provides a VGA with high dynamic range on CDMA and FM signals that has maximum element sharing within CDMA and FM modes. Mobile receivers using this type of VGA can detect signals over a wider input power range. VGA also consumes minimal DC power. Thus, VGA can be used in mobile communication devices and can advantageously protect the operating life of the battery. Finally, the VGA gain can be varied linearly in decibels by linearly adjusting the DC control voltage.

  The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without using the inventive capabilities. Thus, the present invention is not limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and features disclosed herein.

Claims (32)

A transconductance input stage comprising at least one transistor having variable emitter degeneration, having a pair of differential voltage inputs for receiving the amplified signal, and having a pair of differential current outputs;
At least one current amplifier connected to the differential current output to amplify the signal to be amplified; and the transconductance input to apply a control signal to the transconductance input stage and the at least one current amplifier. A control circuit connected to the stage and the at least one current amplifier, wherein the control signal exponentially increases the gain of the transconductance input stage and the at least one current amplifier in response to a linear change in control voltage. Changing the control circuit;
A variable gain amplifier comprising: The transconductance input stage comprises:
An analog modulation input stage having a fixed transconductance; and
A digital modulation input stage having a variable transconductance, wherein the variable transconductance is varied by the control signal;
The variable gain amplifier of claim 1 comprising:   3. The apparatus according to claim 2, further comprising at least one mode selection switch for alternately connecting the analog modulation input stage and the digital modulation input stage to the at least one current amplifier in response to a mode selection signal. Variable gain amplifier. The transconductance input stage comprises:
A first bipolar junction transistor having a base connected to a first input of the differential voltage input;
A second bipolar junction transistor having a base connected to a second input of the differential voltage input; and a source connected to an emitter of the first bipolar junction transistor; and an emitter of the second bipolar junction transistor. A slave field effect transistor having a drain connected to and a gate connected to the control circuit for receiving the control signal;
The variable gain amplifier of claim 1, wherein the control signal changes a channel resistance of the dependent field effect transistor, thereby changing the variable emitter degeneration.   The variable gain amplifier of claim 4, wherein the transconductance input stage further comprises an attenuator for limiting the differential current output. The control circuit includes:
An exponential function generator for converting a linear change in the control voltage into an exponential change in the control current;
A first operational amplifier circuit connected to the exponential function generator and receiving the control current, the first operational amplifier circuit controlling the channel resistance of the dependent field effect transistor;
A second operational amplifier circuit for controlling a drain-source voltage of the subordinate field effect transistor; and a current mirror for providing the signal to the first operational amplifier circuit and the at least one current amplifier;
The variable gain amplifier of claim 4 comprising: The first operational amplifier circuit includes:
A master field effect transistor connected in parallel with a reference resistor; and
An operational amplifier for making the channel resistance of the subordinate field effect transistor the same as the channel resistance of the master field effect transistor;
The variable gain amplifier of claim 7, comprising: The at least one current amplifier comprises:
Electrical resistance shunt-differential Darlington amplifier with series feedback,
A differential cascade amplifier connected as a translinear loop to the differential Darlington amplifier; and a tail current source connected to the control circuit, the differential Darlington amplifier, and the differential cascade amplifier, The tail current source forming a differential pair of currents, and
The gain of the current amplifier is proportional to the ratio of the tail current of the differential pair,
The variable gain amplifier according to claim 1. The differential Darlington amplifier is:
A first bipolar junction transistor having a base connected to one of the differential current outputs of the transconductance input stage;
A second bipolar junction transistor having a base connected to the other one of the differential current outputs of the transconductance stage;
A first current divider connected to the collector of the first bipolar junction transistor at a first end, and connected to the base of the first bipolar junction transistor at a second end; and the second A second current divider connected at a first end to a collector of a bipolar junction transistor and connected at a second end to the base of the second bipolar junction transistor;
The variable gain amplifier of claim 8, wherein a current gain of the differential Darlington amplifier increases with a ratio of resistances in the first and second current dividers. An input stage including a transconductance amplifier;
A current amplifier connected to the input stage, and means for applying a linearly regulated control voltage to the current amplifier to exponentially change the gain of the amplifier as a function of the applied control voltage. An amplifier that processes the input signal provided.   The amplifier of claim 10, wherein the transconductance amplifier has a variable transconductance.   The amplifier of claim 11, wherein the input stage further comprises an attenuator connected to the transconductance amplifier.   The amplifier of claim 12, wherein the attenuator is a Gilbert cell attenuator.   The amplifier of claim 11, further comprising a transconductance amplifier bias control circuit connected to the transconductance amplifier. The input signal includes two balanced signals, and the transconductance amplifier is further an active element, each balanced signal being applied to a respective input of the active element;
A current source connected to each of the active elements; and a variable resistor connected to the active element and the current source;
15. The amplifier of claim 14 comprising: The attenuator further comprises a second active element and a third active element,
The amplifier of claim 15, wherein the second active element and the third active element are connected to a first active element. The transconductance bias control circuit further includes an exponential function generator,
A first operational amplifier circuit connected to the exponential function generator;
A second operational amplifier circuit connected to the first operational amplifier circuit; and
A current source connected to the first operational amplifier circuit;
15. The amplifier of claim 14 comprising: The transconductance amplifier bias control circuit further includes a low-pass filter connected to the first operational amplifier circuit;
18. The amplifier of claim 17 comprising: The exponential generator is
A pair of active elements,
A current source connected to the active element; and a pair of current mirrors respectively connected to the active element;
18. The amplifier of claim 17 comprising: The first operational amplifier circuit includes:
Master active element,
A reference resistor connected to the master active element; and a differential amplifier having first and second inputs and an output;
The amplifier of claim 17, wherein the master active device is connected to the first input and the output of the differential amplifier, and the reference resistor is connected to the second input of the differential amplifier. The second operational amplifier circuit further comprises a non-inverting unity gain amplifier having first and second inputs;
A first input resistance connected to the first input of the non-inverting single gain amplifier; and a second input resistance connected to the second input of the non-inverting single gain amplifier. The amplifier of claim 17. The current amplifier stage comprises:
Darlington differential amplifier,
12. The amplifier of claim 11, comprising: a cascaded differential amplifier connected to the Darlington differential amplifier amplifier; and a tail current generator connected to the Darlington differential amplifier and the cascaded differential amplifier. The Darlington differential amplifier is
A pair of first active elements,
A pair of second active elements respectively connected to the first active elements;
A pair of first resistors respectively connected to the first active element and the second active element;
A pair of second resistors respectively connected to the first active element and the second active element; and a pair of current sources respectively connected to the first active element and the second active element; 23. The amplifier of claim 22 comprising: The cascade differential amplifier is:
A pair of first active elements; and a pair of second active elements respectively connected to the first active elements;
23. The amplifier of claim 22 comprising: The tail current generator is
23. The amplifier of claim 22, comprising: an exponential function generator; and a pair of current mirrors connected to the exponential function generator. The exponential generator is
A pair of active elements,
26. The amplifier of claim 25, comprising: a current source connected to the active element; and a pair of current mirrors respectively connected to the active element. The pair of current mirrors includes:
A pair of first active elements,
A pair of second active elements respectively connected to the first active elements;
A pair of third active elements respectively connected to the first and second active elements;
A pair of first resistors respectively connected to the first active element; and a pair of second resistors respectively connected to the first, second, and third active elements. 26. The amplifier of claim 25. The amplifier includes a transconductance amplifier connected to the current amplifier, the transconductance amplifier applying an input signal to the transconductance amplifier, and modifying the current amplitude of the input signal;
A method of processing an input signal in an amplifier comprising:   29. The method of claim 28, further comprising the step of applying a linearly variable control voltage to the amplifier to produce a corresponding exponent change within the current amplitude of the input signal.   30. The method of claim 28, comprising the step of forming a pair of currents, wherein the amplitude ratio varies exponentially with the control voltage so as to change the current amplitude of the input signal.   29. The method of claim 28, further comprising modifying degenerative feedback in the transconductance amplifier to change the current amplitude of the input signal. Applying a control voltage to the mutual conductance amplifier;
Converting the control voltage into a pair of currents, the amplitude ratio being exponentially proportional to the control voltage;
Converting the current pair into an internal voltage; and a reduction in the mutual-conductance amplifier at the internal voltage such that a linear change in the amplitude ratio of the pair of currents changes the current amplitude of the signal into a linear form. The process of correcting sexual feedback,
30. The method of claim 28, further comprising modifying the current amplitude of the input signal.

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